High-dynamic stable tracking method for non-spread-spectrum continuous wave signals

Abstract

The invention belongs to the technical field of wireless communication, and particularly discloses a high-dynamic stable tracking method for non-spread-spectrum continuous wave signals. The method comprises the following steps: obtaining a digital signal based on ADC sampling, and carrying out digital frequency conversion on all sampling points of a current symbol according to a current estimated frequency result to obtain a digital baseband signal; a digital baseband signal under 4 times of oversampling is obtained through multi-stage downsampling, matched filtering processing is carried out on the digital baseband signal under 4 times of oversampling, and an output result of a preset position in matched filtering output is determined as an optimal sampling point to serve as output of a tracking loop; and performing timing error estimation according to an output result of the matched filtering, adjusting the sampling deviation of the next section of data by using a timing error estimation result, solving a phase discrimination value according to the optimal sampling point, and adjusting the digital frequency conversion of the next section of data by using the phase discrimination value. According to the scheme, the technical problem that the non-spread-spectrum continuous wave signal cannot be continuously and stably tracked in a high-dynamic scene is solved.

Description

Technical Field

[0001] The invention belongs to the technical field of wireless communications, and in particular relates to a high dynamic stability tracking method for a non-spread spectrum continuous wave signal. Background Art

[0002] The relative motion between the two communicating parties will bring about the Doppler effect. As the relative motion speed and acceleration between the transmitter and the receiver increase, the Doppler frequency deviation and frequency deviation acceleration of the received signal increase, and the code deviation of the transmitter and the receiver will also change. In addition to being related to relative motion, the code deviation is also related to the inconsistency of the crystal oscillators of the transmitter and the receiver. If the receiver cannot accurately and stably track the changes in the frequency deviation and code deviation of the received signal, it will affect the demodulation of the receiver and may even cause communication failure or interruption.

[0003] At present, most common high-dynamic tracking methods are based on the spread spectrum system. The core idea is that the carrier loop assists the code loop, and the NCO estimated by the carrier loop is fed back to the code loop to assist the code loop in eliminating the impact of high dynamics. The carrier loop includes a phase-locked loop, a frequency-locked loop, etc. However, the non-spread spectrum system cannot use the code tracking loop to restore the original information, so the above algorithm cannot be used.

[0004] Common carrier synchronization algorithms for non-spread spectrum systems mostly assume that the frequency offset remains unchanged over a period of time, which is applicable to situations where the residual frequency offset is less than the symbol rate. For high-dynamic scenarios, the received signal frequency offset is greater than the symbol rate, and the current algorithm cannot guarantee stable tracking of non-spread spectrum continuous wave signals in high-dynamic scenarios. Summary of the invention

[0005] The purpose of the present invention is to provide a high dynamic and stable tracking method for a non-spread spectrum continuous wave signal, so as to solve the technical problem that a non-spread spectrum continuous wave signal cannot be continuously and stably tracked in a high dynamic scenario.

[0006] To achieve the above object, the present invention adopts the following technical solution:

[0007] A high-dynamic and stable tracking method for a non-spread spectrum continuous wave signal comprises: obtaining a digital signal based on ADC sampling, and performing digital frequency conversion on all sampling points of a current symbol according to a currently estimated frequency result to obtain a digital baseband signal; obtaining a digital baseband signal under 4 times oversampling through multiple levels of downsampling, and performing matched filtering on the digital baseband signal under 4 times oversampling, determining an output result of a preset position in the matched filtering output as an optimal sampling point to serve as an output of a tracking loop; performing timing error estimation according to the output result of the matched filtering, and adjusting the sampling deviation of the next segment of data using the timing error estimation result, and obtaining a phase-locked value according to the optimal sampling point, and adjusting the digital frequency conversion of the next segment of data using the phase-locked value.

[0008] A high dynamic and stable tracking system for non-spread spectrum continuous wave signals comprises: a digital frequency conversion module, which is used to obtain a digital signal based on ADC sampling, and to perform digital frequency conversion on all sampling points of a current symbol to obtain a digital baseband signal according to a currently estimated frequency result; a matched filter module, which is used to obtain a digital baseband signal under 4 times oversampling through multi-stage downsampling, and to perform matched filtering on the digital baseband signal under 4 times oversampling, and to determine the output result of a preset position in the matched filter output as an optimal sampling point as the output of a tracking loop; a timing error estimation module, which is used to perform timing error estimation according to the output result of the matched filter, and to adjust the sampling deviation of the next segment of data using the timing error estimation result; and a phase detector module, which is used to obtain a phase detection value according to the optimal sampling point, and to adjust the digital frequency conversion of the next segment of data using the phase detection value.

[0009] A computer-readable storage medium stores a computer program, wherein the computer program is configured to execute the above-mentioned high dynamic stability tracking method for non-spread spectrum continuous wave signals when running.

[0010] An electronic device comprises a memory and a processor, wherein the memory stores a computer program, and the processor is configured to execute the high dynamic stability tracking method of the non-spread spectrum continuous wave signal through the computer program.

[0011] In the present invention, a digital signal is obtained based on ADC sampling, and according to the current estimated frequency result, all sampling points of the current symbol are digitally converted to obtain a digital baseband signal; after multi-stage downsampling, a digital baseband signal with 4 times oversampling is obtained, and the digital baseband signal with 4 times oversampling is processed by matched filtering, and the output result of a preset position in the matched filtering output is determined as the optimal sampling point to serve as the output of the tracking loop; a timing error is estimated according to the output result of the matched filtering, and the sampling deviation of the next segment of data is adjusted using the timing error estimation result, and a phase detection value is obtained according to the optimal sampling point, and the phase detection value is used to adjust The digital frequency conversion of the next segment of data gradually improves the signal-to-noise ratio of the received signal through multi-stage filtering and extraction, improves the accuracy of timing deviation estimation, adjusts the sampling deviation through the timing deviation estimation result, adjusts the number of sampling points of the symbol under high oversampling multiples, and reduces the phase error caused by adjusting the number of sampling points; at the same time, a combination of code deviation tracking loop and frequency deviation tracking loop is adopted to realize stable tracking and compensation of code deviation and frequency deviation in high dynamic scenarios of non-spread spectrum continuous wave signals, solves the technical problem that non-spread spectrum continuous wave signals cannot be continuously and stably tracked in high dynamic scenarios, and achieves the technical effect of continuous and stable tracking of non-spread spectrum continuous wave signals in high dynamic scenarios. BRIEF DESCRIPTION OF THE DRAWINGS

[0012] Figure 1It is a flow chart of a high dynamic stability tracking method for a non-spread spectrum continuous wave signal in an embodiment of the present invention;

[0013] Figure 2 It is a schematic diagram of an implementation flow of a high dynamic stability tracking method for a non-spread spectrum continuous wave signal in an embodiment of the present invention;

[0014] Figure 3 is a schematic diagram of a simulation result of a constellation diagram output by matched filtering in an embodiment of the present invention;

[0015] Figure 4 is a schematic diagram of a flow chart of timing error estimation in an embodiment of the present invention;

[0016] Figure 5 is a schematic diagram of the number of sampling points of each symbol after the timing deviation is adjusted in an embodiment of the present invention;

[0017] Figure 6 1 is a flow chart of a phase-locked loop phase detector according to an embodiment of the present invention;

[0018] Figure 7 Schematic diagram of output result of a phase-locked loop phase detector in an embodiment of the present invention;

[0019] Figure 8 It is a structural schematic diagram of a high dynamic stability tracking system for a non-spread spectrum continuous wave signal in an embodiment of the present invention. DETAILED DESCRIPTION

[0020] The present invention is further described in detail below in conjunction with the accompanying drawings and specific embodiments. The advantages and features of the present invention will become more apparent from the following description and claims. It should be noted that the drawings are all in very simplified form and are not in precise proportions, and are only used to conveniently and clearly assist in explaining the purpose of the embodiments of the present invention.

[0021] It should be noted that, in order to clearly explain the content of the present invention, the present invention specifically cites multiple embodiments to further illustrate different implementations of the present invention, wherein the multiple embodiments are enumerated rather than exhaustive. In addition, for the sake of brevity of explanation, the contents mentioned in the previous embodiments are often omitted in the subsequent embodiments. Therefore, the contents not mentioned in the subsequent embodiments can refer to the previous embodiments accordingly.

[0022] Example 1

[0023] A high dynamic and stable tracking method for non-spread spectrum continuous wave signals, such as Figure 1 As shown, the method includes:

[0024] S102, obtaining a digital signal based on ADC sampling, and performing digital frequency conversion on all sampling points of the current symbol according to the currently estimated frequency result to obtain a digital baseband signal;

[0025] S104, obtaining a 4-fold oversampled digital baseband signal through multi-stage downsampling, performing matched filtering on the 4-fold oversampled digital baseband signal, and determining an output result at a preset position in the matched filtering output as an optimal sampling point as an output of a tracking loop;

[0026] S106, perform timing error estimation based on the output result of the matched filter, and use the timing error estimation result to adjust the sampling deviation of the next segment of data, and at the same time obtain the phase detection value based on the optimal sampling point, and use the phase detection value to adjust the digital frequency conversion of the next segment of data.

[0027] As an optional implementation, after obtaining a digital signal based on ADC sampling, it also includes: caching the digital signal obtained by ADC sampling in RAM, wherein each time a sampling point is input, the address of the RAM storage number is increased by one; and reading the corresponding sampling point from RAM.

[0028] Specifically, the high dynamic stability tracking method of the non-spread spectrum continuous wave signal is not limited to including the following steps:

[0029] S1, cache the digital signal sampled by the ADC in the RAM, and read a certain sampling point from the RAM when needed.

[0030] S2, based on the current estimated frequency result, digital frequency conversion is performed on all sampling points of the current symbol to achieve frequency offset compensation, eliminate the frequency offset effect caused by high dynamics, and obtain a digital baseband signal.

[0031] S3, after multiple levels of downsampling, gradually reduces the sampling rate of the digital baseband signal, and finally reduces it to 4 times oversampling.

[0032] S4, performing matched filtering on the digital baseband signal under 4 times oversampling, and not limited to determining the 0th, 4th, 8th, 12th, 16th, ... output results in the matched filtering output results as the best sampling points. The best sampling points are given to subsequent decoding modules as the output of the tracking loop.

[0033] S5, make a timing error estimate based on the output result of the matched filter, convert the timing error estimation result into the number of sampling points to be adjusted and feed it back to step S1 to adjust the sampling deviation of the next segment of data, so as to achieve stable tracking of the high dynamic BPSK continuous signal code deviation.

[0034] S6: The residual phase deviation is estimated according to the optimal sampling point of the matched filter output in step S4 to obtain the phase detection value, and the phase detection value is fed back to the digital frequency conversion in step S2 after passing through the loop filter to construct a phase-locked loop feedback loop to achieve stable tracking of the frequency deviation of the high dynamic BPSK continuous signal.

[0035] As an optional implementation, reading the corresponding sampling point from the RAM includes: determining the starting address of the first sampling point of the first symbol, and determining a preset number of continuous sampling points as one symbol based on the starting address; updating the address information of the current sampling point according to the address of the sampling point of the previous symbol and the timing error estimation result, and reading the corresponding sampling point from the RAM.

[0036] The address of the sampling point read from the RAM is specifically not limited to determining the starting address of the first sampling point of the first symbol according to the capture result, starting from the address, continuously reading OSR sampling points as one symbol, OSR represents the oversampling multiple of the received signal. According to the address of the sampling point of the previous symbol and the address adjustment amount obtained by the timing error estimation module, the address information of the current sampling point is updated, and the corresponding sampling point is read from the RAM.

[0037] Furthermore, it is not limited to first determining the starting address and the number of sampling points of the current symbol and then reading it, and specifically includes:

[0038] The starting address of the current symbol is updated according to the starting address of the previous symbol and the address adjustment amount obtained by the timing error estimation module.

[0039] Update the number of sampling points of the current symbol according to the starting position of the current symbol and the starting position of the previous symbol.

[0040] According to the starting address of the current symbol and the number of sampling points of the current symbol, all sampling points of the current symbol can be read from the RAM.

[0041] As an optional implementation, after multiple levels of downsampling, a digital baseband signal with 4 times oversampling is obtained, including: reducing the sampling rate of the digital baseband signal by a preset multiple step by step, and after multiple levels of downsampling by the preset multiples, reducing the sampling rate to 4 times the symbol rate, to obtain a digital baseband signal with 4 times oversampling.

[0042] As an optional implementation, the sampling rate of the digital baseband signal is reduced step by step by a preset multiple, including: according to the number of sampling points of the current symbol, obtaining the first filtering result after the current symbol is input with a preset number of sampling points; performing a convolution operation when outputting the filtering result at the current sampling point; starting from the second segment of data of the current symbol, a filtering result is output each time a preset number of sampling points are input.

[0043] Downsampling includes multi-level filtering and extraction, in which the first level of filtering and extraction needs to consider the impact of the number of sampling points not being equal to the oversampling multiple OSR. Specifically, it includes the following steps:

[0044] After downsampling by z times, the sampling rate is reduced to OSR / z times of the symbol rate. Considering the influence of code deviation, the number of sampling points for each symbol is not necessarily OSR. The number of sampling points of the nth symbol oscillates around OSR, and the specific oscillation range is related to the stability of the crystal oscillator and the relative motion speed.

[0045] After multiple levels of z-fold downsampling, the sampling rate is reduced to 4 times the symbol rate.

[0046] Furthermore, the code deviation causes the number of sampling points of each symbol to be not necessarily OSR, and the first filtering and extraction process of each symbol is inconsistent with the subsequent process, which specifically includes the following steps:

[0047] According to the value of the number of sampling points, the first filtering result can be obtained after the current symbol is input with m sampling points. There is a corresponding relationship between m and the number of sampling points.

[0048] When the current sampling point outputs the filtering result, a convolution operation is performed, and the filter order is set to N. fir , the filter coefficients are fir_coe0, fir_coe1, fir_coe2, ..., fir_coe Nfir-1 If the current sampling point is the kth sampling point, the first-stage filter outputs the lth result.

[0049] The current symbol starts from the second segment, and a filtering result is output for every z sampling points input.

[0050] As an optional implementation, timing error estimation is performed based on the output result of matched filtering, and the sampling deviation of the next segment of data is adjusted using the timing error estimation result, including: timing error estimation is performed based on the output result of matched filtering, the timing error estimation result is converted into the number of sampling points to be adjusted and the sampling deviation of the next segment of data is adjusted.

[0051] As an optional implementation, a timing error estimation is performed based on the output result of matched filtering, and the timing error estimation result is converted into the number of sampling points to be adjusted and the sampling deviation of the next segment of data is adjusted, including: using the output result of matched filtering to perform timing error estimation and obtain a normalized symbol timing error estimation result; calculating the sampling deviation of the next segment of data based on the timing error estimation result and determining the corresponding address adjustment amount.

[0052] A normalized symbol timing error estimation is performed based on the output of the matched filter, and the obtained normalized symbol timing error estimation result is used to compensate for the timing deviation of the next segment of L symbols, that is, the address adjustment amount.

[0053] As an optional implementation, a phase-locked value is obtained based on the optimal sampling point, and the phase-locked value is used to adjust the digital frequency conversion of the next segment of data, including: inputting the optimal sampling point into a phase lock detector to obtain a phase-locked value; inputting the phase-locked value into a loop filter, and using the frequency deviation estimate value output by the loop filter to control the digital frequency conversion frequency of the next segment of data.

[0054] As an optional implementation, the optimal sampling point is input into a phase detector to obtain a phase detection value, including: inputting the optimal sampling point into an arctan inverse tangent phase detector, compensating the phase detection result, and compensating again after the loop is locked to obtain a phase detection value that eliminates sudden phase detection errors.

[0055] Phase detection is performed based on the optimal sampling point of the matched filter output, and the phase detection result is fed back to the digital frequency conversion after loop filtering, which specifically includes the following steps:

[0056] S6.1: Input the best sampling point of the matched filter output in step S4 into the phase detector to obtain the phase detection value.

[0057] S6.2: Input the phase detection result into the loop filter, and the output of the loop filter is fed back to step S2 to control the frequency of the digital frequency conversion. The phase detection result is updated once for each symbol, and the frequency deviation estimation value output by the loop filter is updated accordingly, and fed back to the digital frequency conversion module to eliminate the frequency deviation of the next symbol and eliminate the frequency deviation effect caused by high dynamics.

[0058] Considering that the frequency offset estimation value is updated once per symbol, step S2 performs frequency offset compensation for each sampling point, and the result fcw fed back to S2 is n and the frequency deviation estimation result fd n There is a corresponding relationship, and the phase deviation caused by the frequency deviation within a symbol is distributed to each sampling point.

[0059] In step S6.1, the phase detection process includes arctan calculation and two phase compensations, which specifically includes the following steps:

[0060] S6.1.1: Use the arctan inverse tangent phase detector, which is insensitive to 180° phase jumps and is suitable for BPSK signals.

[0061] S6.1.2: Compensate the obtained phase detection result to break through the limit of -π / 2 to π / 2 of arctan inverse tangent phase detection result. S6.1.3: After the loop is locked, further compensate the obtained phase to reduce the sudden phase detection error caused by noise under low signal-to-noise ratio and improve the stability of the loop. No compensation is performed when the loop is not locked. After the loop is locked, it is believed that each phase deviation will not exceed the limit of -π / 2 to π / 2, so compensation is performed to limit the phase detection result to the range of -π / 2 to π / 2.

[0062] In a specific embodiment, the communication system uses a non-spread spectrum BPSK continuous wave signal, and the symbol rate is denoted by R s , the sampling rate is recorded as f s , the Doppler frequency deviation caused by high dynamics is [-f dmax ,f dmax ] range, the frequency deviation acceleration is in [-f amax ,f amax ] range. Among them, the sampling rate f s =40000×R s , which is 40000 times the symbol rate, OSR = 40000 represents the oversampling multiple, f dmax =780×R s The maximum frequency separation is greater than the symbol rate, f amax =16×R s The maximum frequency deviation acceleration is greater than the symbol rate. The frequency deviation changes in a sinusoidal waveform. When the frequency deviation is maximum, the frequency deviation acceleration is minimum, and when the frequency deviation is minimum, the frequency deviation acceleration is maximum, which conforms to the actual satellite communication scenario.

[0063] The high dynamic stability tracking method of the non-spread spectrum continuous wave signal in this embodiment is not limited to the following. Figure 2 As shown, it includes: S1, data cache; S2, digital frequency conversion; S3, multi-stage extraction filtering; S4, matching filtering; S5, timing error estimation; S6, phase detection and loop filtering. Specifically, the steps are as follows:

[0064] S1: Cache the digital signal sampled by the ADC in the RAM, and read certain sampling points from the RAM when needed.

[0065] Further, S1.1: Buffer the digital signal sampled by the ADC in the RAM. Specifically, each time a sampling point is input, the address of the RAM storage number is increased by one.

[0066] S1.2: Read certain sampling points from RAM when needed. Specifically, the addresses of the sampling points read from RAM are as follows:

[0067] Further, S1.2.1: determine the starting address of the first sampling point of the first symbol according to the capture result, and start from this address and continuously read OSR=40000 sampling points as one symbol.

[0068] S1.2.2: Update the address information of the current sampling point according to the address of the previous symbol sampling point and the address adjustment amount obtained by the timing error estimation module, and read the corresponding sampling point from the RAM.

[0069] Further, S1.2.2.1: update the starting address of the current symbol according to the starting address of the previous symbol and the address adjustment amount obtained by the timing error estimation module. Specifically, the update formula is:

[0070]

[0071] In the above formula, AddrStart n Indicates the starting address of the nth symbol, AddrStart n-1 Indicates the starting address of the n-1th symbol, AddrAdjust n represents the address adjustment amount estimated by the timing error estimation module, Indicates the rounding down operation. According to S1.2.1, the starting address of the first symbol is given by the capture result; starting from the second symbol, the starting position is calculated by formula (1).

[0072] S1.2.2.2: Update the number of sampling points of the current symbol according to the starting position of the current symbol and the starting position of the previous symbol. Specifically, the update formula is:

[0073] ΔSampleNum n =AddrStart n -AddrStart n-1 (2)

[0074] In the above formula, ΔSampleNum n Indicates the number of sampling points of the nth symbol. According to S1.2.1, the number of sampling points of the first symbol is OSR=40000; starting from the second symbol, the sampling points are calculated by formula (2).

[0075] S1.2.2.3: According to the starting address AddrStart of the current symbol n and the number of sampling points of the current symbol ΔSampleNum n , to read all sampling points of the current symbol from RAM.

[0076] S2: According to the current estimated frequency result, all sampling points of the current symbol are digitally converted to achieve frequency offset compensation, eliminate the frequency offset effect caused by high dynamics, and obtain a digital baseband signal.

[0077] S3: After multiple levels of downsampling, the sampling rate of the digital baseband signal is gradually reduced, and finally reduced to 4 times oversampling.

[0078] Further, S3.1: After 10 times downsampling, the sampling rate is reduced to 4000 times the symbol rate. Considering the influence of code deviation, the number of sampling points of each symbol is not necessarily OSR. The number of sampling points of the nth symbol ΔSampleNum n Oscillates around OSR=40000, and the specific oscillation range is related to the stability of the crystal oscillator and the relative movement speed.

[0079] Further, S3.1.1: According to ΔSampleNum n The value of the current symbol is obtained by inputting m sampling points and then outputting the first filtering result. If the current symbol has ΔSampleNum n =osr=40000 sampling points, then the first-stage filter of the current symbol outputs a filtering result every 10 sampling points for the first time; if the current symbol has ΔSampleNum n =osr-1=39999 sampling points, then the first-stage filter of the current symbol outputs a filtering result every 9 sampling points for the first time; if the current symbol has ΔSampleNum n =osr+1=40001 sampling points, then the first-stage filter of the current symbol outputs a filtering result every 11 sampling points for the first time; ΔSampleNum n The same applies to other values. m and ΔSampleNum n The specific corresponding relationship is as follows:

[0080]

[0081] S3.1.2: When the current sampling point needs to output the filtering result, a convolution operation is performed, and the filter order is set to N fir , the filter coefficients are If the current sampling point is the kth sampling point, the first-stage filter outputs the lth result, and the convolution operation process is as follows:

[0082]

[0083] S3.1.3: The current symbol starts from the second segment, and outputs a filtering result every 10 sampling points. After adjustment in S3.1.1, there are 39990 sampling points left in the current symbol, and a total of 3999 filtering results are output. Filtering is a convolution operation process, as shown in formula (4).

[0084] S3.2: After 10 times downsampling, the sampling rate is reduced to 400 times the symbol rate. After 10 times decimation filtering in step S3.1, there are 4000 sampling points left in the current symbol. After another 10 times downsampling, the second stage decimation filtering outputs a total of 400 filtering results. Filtering is a convolution operation process, which is the same as shown in equation (4).

[0085] S3.3: After 10 times downsampling, the sampling rate is reduced to 40 times the symbol rate. After 10 times decimation filtering in step S3.2, there are 400 sampling points left in the current symbol. After another 10 times downsampling, the third-level decimation filtering outputs a total of 40 filtering results. Filtering is a convolution operation process, which is the same as shown in equation (4).

[0086] S3.4: After 10 times downsampling, the sampling rate is reduced to 4 times the symbol rate. After 10 times decimation filtering in step S3.2, there are 400 sampling points left in the current symbol. After another 10 times downsampling, the fourth level decimation filtering outputs a total of 4 filtering results. Filtering is a convolution operation process, which is the same as shown in formula (4).

[0087] S4: Perform matched filtering on the digital baseband signal under 4 times oversampling. The 0th, 4th, 8th, 12th, 16th, ... output results of the matched filtering output are the best sampling points. The best sampling points are given to the subsequent decoding modules as the output of the tracking loop. The best sampling points of the matched filtering output are not limited to the following: Figure 3 As shown, the specific display is the constellation diagram simulation result.

[0088] S5: Make a timing error estimate based on the result of matched filtering, convert the timing error estimation result into the number of sampling points to be adjusted and feed it back to step S1.2 to adjust the sampling deviation of the next segment of data, so as to achieve stable tracking of the code deviation of high dynamic BPSK continuous signal. The specific steps are not limited to the following: Figure 4 shown.

[0089] Further, S5.1: Use x i represents the output of the matched filter, and the normalized symbol timing error estimation formula is

[0090]

[0091] As shown in equation (5), the normalized symbol timing error is updated once every 1024 symbols (corresponding to 1024×4 sampling points), and the estimated result is in the range of -0.5-0.5. Due to the influence of relative motion, the normalized symbol timing error will change over time. The error can be considered unchanged in a short period of time. Here, it is assumed that the timing error within 1024 symbols will not change.

[0092] S5.2: Based on the normalized symbol timing error estimation result obtained in S5.1, compensate the timing deviation of the next 1024 symbols, that is, calculate the address adjustment required in step S1.2.2.1. The specific calculation formula is:

[0093]

[0094] The value range of n is (b+1)×1024≤n≤(b+2)×1024-1. The address adjustment value of each received signal segment adopts the normalized symbol timing error value obtained in the previous segment. The initial time is not limited to the address adjustment value of the received signal segment. n Equal to OSR, the timing error estimate is considered to be 0 at the equivalent initial moment.

[0095] After the timing deviation is adjusted, the number of sampling points for each symbol is not limited to Figure 5 shown.

[0096] S6: The residual phase deviation is estimated according to the optimal sampling point of the matched filter output in step S4 to obtain the phase discrimination value, and the phase discrimination value is fed back to the digital frequency conversion in step S2 after passing through the loop filter to build a phase-locked loop feedback loop to achieve stable tracking of the frequency deviation of the high-dynamic BPSK continuous signal. The specific steps are not limited to the following. Figure 6 Shown

[0097] Further, S6.1: input the best sampling point output by the matched filter in step S4 into the phase detector module to obtain the phase detection value.

[0098] Specifically, S6.1.1: Use an arctan inverse tangent phase detector, which is insensitive to 180° phase jumps and is suitable for BPSK signals:

[0099] θ1(n)=arctanx n (7)

[0100] In formula (7), x n Represents the optimal sampling point of the matched filter output. Each symbol has one optimal sampling point.

[0101] S6.1.2: Compensate the phase detection result obtained by equation (7) to break through the limitation of arctan inverse tangent phase detection result -π / 2 to π / 2. The compensation formula is as follows:

[0102]

[0103] S6.1.3: After the loop is locked, the phase obtained by equation (8) is further compensated to reduce the sudden phase detection error caused by noise under low signal-to-noise ratio and improve the stability of the loop. The compensation formula is as follows:

[0104]

[0105] When the loop is not locked, the compensation shown in equation (9) is not performed. After the loop is locked, it is assumed that each phase deviation will not exceed the limit of -π / 2 to π / 2, so compensation is required to limit the phase detection result to the range of -π / 2 to π / 2.

[0106] S6.2: Input the phase detection result into the loop filter, and the output of the loop filter is fed back to step S2 to control the frequency of the digital frequency conversion. The phase detection result is updated once for each symbol, and the frequency deviation estimation value output by the loop filter is updated accordingly, which is fed back to the digital frequency conversion module to eliminate the frequency deviation of the next symbol and the frequency deviation effect caused by high dynamics. Considering that the frequency deviation estimation value is updated once for each symbol, and step S2 needs to compensate for the frequency deviation of each sampling point, the result fcw fed back to S2 is n and the frequency deviation estimation result fd nThe relationship is:

[0107]

[0108] Formula (10) is used to evenly distribute the phase deviation caused by the frequency offset within a symbol to each sampling point.

[0109] The output of the phase-locked loop phase detector is not limited to Figure 7 shown.

[0110] In the embodiment of the present application, multi-stage filtering and extraction are used to gradually improve the signal-to-noise ratio of the received signal, thereby ensuring the accuracy of the timing deviation estimation; the timing deviation estimation value is re-estimated at regular intervals, and the number of sampling points of each symbol is adjusted accordingly, thereby achieving stable tracking and compensation of the code deviation in high-dynamic scenarios of non-spread spectrum continuous wave signals. The number of sampling points of each symbol is adjusted at a high oversampling multiple, which reduces the phase error caused by adjusting the number of sampling points, thereby reducing the impact on the carrier loop, improving the stability of the carrier tracking loop, and achieving stable tracking and compensation of the frequency deviation in high-dynamic scenarios; a combination of a code deviation tracking loop and a frequency deviation tracking loop is used to achieve stable tracking and compensation of the code deviation and frequency deviation in high-dynamic scenarios of non-spread spectrum continuous wave signals, which can be used for reliable tracking under low signal-to-noise ratios, and can be used for demodulation of non-spread spectrum continuous wave signals. The embodiment of the present application is simple to implement, highly operable and portable, occupies few resources, is very suitable for FPGA implementation, and has good engineering benefits.

[0111] Example 2

[0112] A high dynamic and stable tracking system for non-spread spectrum continuous wave signals, such as Figure 8 As shown, the system includes:

[0113] The digital frequency conversion module 802 is used to obtain a digital signal based on ADC sampling, and to perform digital frequency conversion on all sampling points of the current symbol to obtain a digital baseband signal according to the current estimated frequency result;

[0114] A matched filtering module 804 is used to obtain a digital baseband signal with 4 times oversampling through multiple stages of downsampling, and to perform matched filtering on the digital baseband signal with 4 times oversampling, and to determine the output result of a preset position in the matched filtering output as the optimal sampling point as the output of the tracking loop;

[0115] The timing error estimation module 806 is used to estimate the timing error according to the output result of the matched filter, and use the timing error estimation result to adjust the sampling deviation of the next segment of data;

[0116] The phase detector module 808 is used to obtain a phase detection value according to the optimal sampling point, and use the phase detection value to adjust the digital frequency conversion of the next segment of data.

[0117] Optionally, after obtaining a digital signal based on ADC sampling, the high dynamic and stable tracking system for the above-mentioned non-spread spectrum continuous wave signal further includes: caching the digital signal obtained by ADC sampling in RAM, wherein, for each sampling point input, the address of the RAM storage number is increased by one; and reading the corresponding sampling point from RAM.

[0118] Optionally, the high dynamic stability tracking system of the above-mentioned non-spread spectrum continuous wave signal reads the corresponding sampling points from RAM, including: determining the starting address of the first sampling point of the first symbol, and determining a preset number of continuous sampling points as a symbol based on the starting address; updating the address information of the current sampling point according to the address of the sampling point of the previous symbol and the timing error estimation result, and reading the corresponding sampling point from RAM.

[0119] Optionally, the matched filter module 804 undergoes multiple levels of downsampling to obtain a digital baseband signal under 4 times oversampling, including: reducing the sampling rate of the digital baseband signal by a preset multiple step by step, and after multiple levels of downsampling by preset multiples, reducing the sampling rate to 4 times the symbol rate, to obtain a digital baseband signal under 4 times oversampling.

[0120] Optionally, the above-mentioned matched filtering module 804 reduces the sampling rate of the digital baseband signal by a preset multiple step by step, including: obtaining the first filtering result after the current symbol is input with a preset number of sampling points according to the number of sampling points of the current symbol; performing a convolution operation when outputting the filtering result at the current sampling point; starting from the second segment of data of the current symbol, outputting a filtering result every time a preset number of sampling points are input.

[0121] Optionally, the timing error estimation and phase detector module 806 performs timing error estimation based on the output result of matched filtering, and uses the timing error estimation result to adjust the sampling deviation of the next segment of data, including: performing timing error estimation based on the output result of matched filtering, converting the timing error estimation result into the number of sampling points to be adjusted and adjusting the sampling deviation of the next segment of data.

[0122] Optionally, the timing error estimation module 806 performs timing error estimation based on the output result of matched filtering, converts the timing error estimation result into the number of sampling points to be adjusted and adjusts the sampling deviation of the next segment of data, including: using the output result of matched filtering to perform timing error estimation to obtain the normalized symbol timing error estimation result; calculating the sampling deviation of the next segment of data based on the timing error estimation result, and determining the corresponding address adjustment amount.

[0123] Optionally, the above-mentioned phase detector module 808 obtains a phase detection value according to the optimal sampling point, and uses the phase detection value to adjust the digital frequency conversion of the next segment of data, including: inputting the optimal sampling point into the phase detector to obtain the phase detection value; inputting the phase detection value into the loop filter, and using the frequency deviation estimation value output by the loop filter to control the digital frequency conversion frequency of the next segment of data.

[0124] Optionally, the phase detector module 808 inputs the optimal sampling point into the phase detector to obtain a phase detection value, including: inputting the optimal sampling point into the arctan inverse tangent phase detector, compensating the phase detection result, and compensating again after the loop is locked to obtain a phase detection value that eliminates sudden phase detection errors.

[0125] In the embodiment of the present application, a digital signal is obtained based on ADC sampling, and according to the current estimated frequency result, all sampling points of the current symbol are digitally converted to obtain a digital baseband signal; after multi-stage downsampling, a digital baseband signal with 4 times oversampling is obtained, and the digital baseband signal with 4 times oversampling is processed by matched filtering, and the output result of a preset position in the matched filtering output is determined as the optimal sampling point to serve as the output of the tracking loop; a timing error is estimated according to the output result of the matched filtering, and the sampling deviation of the next segment of data is adjusted using the timing error estimation result, and a phase detection value is obtained according to the optimal sampling point, and the phase detection value is used Adjust the digital frequency conversion of the next segment of data, thereby gradually improving the signal-to-noise ratio of the received signal through multi-stage filtering extraction, improving the accuracy of timing deviation estimation, adjusting the sampling deviation through the timing deviation estimation result, adjusting the number of sampling points of the symbol under high oversampling multiples, reducing the phase error caused by adjusting the number of sampling points, and combining the sampling code deviation tracking loop and the frequency deviation tracking loop to achieve stable tracking and compensation of code deviation in high dynamic scenarios of non-spread spectrum continuous wave signals, solving the technical problem that non-spread spectrum continuous wave signals cannot be continuously and stably tracked in high dynamic scenarios, and achieving the technical effect of continuously and stably tracking non-spread spectrum continuous wave signals in high dynamic scenarios.

[0126] Example 3

[0127] Another aspect of the embodiments of the present invention further provides an electronic device for implementing the above-mentioned high dynamic stability tracking method of non-spread spectrum continuous wave signals, and the electronic device is not limited to being a terminal device or a server in the system. The electronic device is not limited to including a memory and a processor, the memory stores a computer program, and the processor is configured to execute the steps in any of the above-mentioned method embodiments through the computer program.

[0128] Example 4

[0129] Another aspect of the embodiments of the present invention provides a computer program product or a computer program, which includes computer instructions stored in a computer-readable storage medium. A processor of a computer device reads the computer instructions from the computer-readable storage medium, and the processor executes the computer instructions, so that the computer device executes the method provided in various optional embodiments of the high dynamic stability tracking method for non-spread spectrum continuous wave signals. The computer program is configured to execute the steps of any of the above method embodiments when running.

Claims

1. A high dynamic stability tracking method for non-spread spectrum continuous wave signals, characterized in that: include: A digital signal is obtained based on ADC sampling, and according to the current estimated frequency result, all sampling points of the current symbol are digitally converted to obtain a digital baseband signal; After multiple levels of downsampling, a digital baseband signal with 4 times oversampling is obtained, and the digital baseband signal with 4 times oversampling is subjected to matched filtering processing, and the output result of a preset position in the matched filtering output is determined as the optimal sampling point to serve as the output of the tracking loop; The timing error is estimated based on the output result of the matched filter, and the sampling deviation of the next data segment is adjusted using the timing error estimation result. At the same time, the phase detection value is obtained based on the optimal sampling point, and the digital frequency conversion of the next data segment is adjusted using the phase detection value.

2. The high dynamic stability tracking method of non-spread spectrum continuous wave signal according to claim 1, characterized in that: After obtaining the digital signal based on ADC sampling, it also includes: The digital signal sampled by the ADC is cached in the RAM, wherein the address of the RAM storage number is increased by one each time a sampling point is input; Read the corresponding sampling points from RAM.

3. The high dynamic stability tracking method of non-spread spectrum continuous wave signal as claimed in claim 2, characterized in that: Read the corresponding sampling points from RAM, including: Determine a starting address of a first sampling point of a first symbol, and determine a preset number of consecutive sampling points as one symbol based on the starting address; According to the address of the previous symbol sampling point and the timing error estimation result, the address information of the current sampling point is updated, and the corresponding sampling point is read from the RAM.

4. The high dynamic stability tracking method of non-spread spectrum continuous wave signal according to claim 1, characterized in that: After multiple levels of downsampling, a 4-fold oversampled digital baseband signal is obtained, including: The sampling rate of the digital baseband signal is reduced by a preset multiple step by step. After multiple steps of downsampling by the preset multiples, the sampling rate is reduced to 4 times of the symbol rate, thereby obtaining a digital baseband signal under 4 times oversampling.

5. The high dynamic stability tracking method of non-spread spectrum continuous wave signal as claimed in claim 4, characterized in that: The sampling rate of the digital baseband signal is gradually reduced by a preset multiple, including: According to the number of sampling points of the current symbol, the first filtering result after the current symbol is input with a preset number of sampling points is obtained; Perform convolution operation when outputting the filtering result at the current sampling point; Starting from the second segment of data of the current symbol, a filtering result is output every time a preset number of sampling points are input.

6. The high dynamic stability tracking method of non-spread spectrum continuous wave signal according to claim 1, characterized in that: The timing error is estimated based on the output result of the matched filter, and the sampling deviation of the next segment of data is adjusted using the timing error estimation result, including: The timing error is estimated based on the output result of the matched filter, and the timing error estimation result is converted into the number of sampling points to be adjusted and the sampling deviation of the next segment of data is adjusted.

7. The high dynamic stability tracking method of non-spread spectrum continuous wave signal according to claim 6, characterized in that: The timing error is estimated according to the output result of the matched filter, and the timing error estimation result is converted into the number of sampling points to be adjusted and the sampling deviation of the next segment of data is adjusted, including: The output result of the matched filter is used to estimate the timing error and obtain the normalized symbol timing error estimation result; The sampling deviation of the next segment of data is calculated based on the timing error estimation result, and the corresponding address adjustment amount is determined.

8. The high dynamic stability tracking method of non-spread spectrum continuous wave signal according to claim 1, characterized in that: The phase-locked value is obtained according to the optimal sampling point, and the digital frequency conversion of the next segment of data is adjusted using the phase-locked value, including: Input the best sampling point into the phase detector to obtain the phase detection value; The phase detection value is input into the loop filter, and the frequency deviation estimation value output by the loop filter is used to control the digital frequency conversion frequency of the next segment of data.

9. The high dynamic stability tracking method of non-spread spectrum continuous wave signal according to claim 8, characterized in that: Input the best sampling point into the phase detector to obtain the phase detection value, including: The best sampling point is input into the arctan inverse tangent phase detector, the phase detection result is compensated, and the compensation is performed again after the loop is locked to obtain the phase detection value that eliminates the sudden phase detection error.

10. A high dynamic stability tracking system for non-spread spectrum continuous wave signals, characterized in that: include: A digital frequency conversion module is used to obtain a digital signal based on ADC sampling, and to perform digital frequency conversion on all sampling points of the current symbol to obtain a digital baseband signal according to the current estimated frequency result; A matched filter module is used to obtain a digital baseband signal with 4 times oversampling through multiple stages of downsampling, and to perform matched filtering on the digital baseband signal with 4 times oversampling, and to determine the output result of a preset position in the matched filter output as the optimal sampling point as the output of the tracking loop; A timing error estimation module is used to estimate the timing error according to the output result of the matched filter, and use the timing error estimation result to adjust the sampling deviation of the next segment of data; The phase detector module is used to obtain the phase detection value according to the optimal sampling point, and use the phase detection value to adjust the digital frequency conversion of the next segment of data.

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