Off-grid Inverter Control Method for Resisting Three-phase Unbalanced Loads

Abstract

The present invention discloses a control method for an off-grid inverter against three-phase unbalanced loads, belonging to the technical field of wind power generation. The method includes: separating the three-phase signals output by a three-level three-phase four-leg inverter to obtain single-phase signals, and respectively performing virtual coordinate transformation on the single-phase signals; performing double closed-loop control on the single-phase signals after virtual coordinate transformation; performing rotating coordinate transformation on the single-phase signals after double closed-loop control to obtain single-phase signals in the α-β coordinate system; respectively inputting the modulation waves of the three single-phase signals into an SPWM modulation module to obtain drive signals; and obtaining complementary control signals for the switches of the fourth leg through a fourth-leg control strategy. The present invention can ensure that the midpoint potential is quickly stabilized during dynamic changes of the load, effectively eliminate the inter-phase coupling effect, realize independent regulation of each phase load, significantly improve the operation stability of the system under three-phase unbalanced loads, and provide strong support for the development of off-grid wind power generation systems.

Description

Technical Field

[0001] The present invention relates to the technical field of wind power generation, and particularly to a control method for an off-grid inverter against three-phase unbalanced loads. Background Art

[0002] With the rapid development of renewable energy, wind power generation, as a clean energy technology, has been widely applied in off-grid independent power supply systems. However, the off-grid wind power generation system often faces the problem of three-phase load imbalance during actual operation, resulting in phenomena such as asymmetric output voltage of the inverter and midpoint potential shift of the three-phase system, which seriously affect the stability of the system and the power supply quality. At present, for the problem of three-phase unbalanced loads, the existing solutions mainly focus on grid-connected inverters or balanced load scenarios. However, these methods have obvious limitations in off-grid wind power generation systems. In solving the midpoint potential shift problem, the existing technologies mostly rely on complex software control algorithms. Although these methods can improve the midpoint potential shift problem to a certain extent, the implementation process is complex, the calculation amount is large, the performance requirements for the controller are high, which increases the system cost and implementation difficulty. The software control method has a slow response speed under dynamic load changes and is difficult to meet the requirements of real-time performance and stability of off-grid wind power generation systems. Summary of the Invention

[0003] The purpose of this application is to overcome the defects of the existing technology and provide a control method for an off-grid inverter against three-phase unbalanced loads, so as to effectively cope with three-phase unbalanced loads in an off-grid wind power generation system, and at the same time solve the midpoint potential shift problem through a simple and reliable hardware circuit method.

[0004] In a first aspect, this application provides a control method for an off-grid inverter against three-phase unbalanced loads. The off-grid inverter against three-phase unbalanced loads includes: a wind turbine, a diode rectifier, a DC converter, a three-level three-phase four-leg inverter, and a three-phase unbalanced load. The wind turbine is connected to the first end of the diode rectifier, the second end of the diode rectifier is connected to the first end of the DC converter, the second end of the DC converter is connected to the first end of the three-level three-phase four-leg inverter, and the second end of the three-level three-phase four-leg inverter is connected to the three-phase unbalanced load. The three-level three-phase four-leg inverter includes: a fourth leg circuit, the first end of the fourth leg circuit is connected to the DC voltage output by the DC converter; a T-type midpoint clamped three-level circuit, the first end of the T-type midpoint clamped three-level circuit is connected to the second end of the fourth leg circuit, and the T-type midpoint clamped three-level circuit includes three legs; a filtering circuit, the first end of the filtering circuit is connected to the second end of the T-type midpoint clamped three-level circuit, and the second end of the filtering circuit is connected to the three-phase unbalanced load.

[0005] The fourth bridge arm circuit includes: a first switch, the first end of the first switch is connected to the first end of the DC voltage; a second switch, the first end of the second switch is connected to the second end of the first switch, and the second end of the second switch is connected to the second end of the DC voltage; a first inductor, the first end of the first inductor is connected to the series midpoint of the first switch and the second switch, and the second end of the first inductor is connected to the first end of the T-type neutral point clamped three-level circuit;

[0006] A control method for an off-grid inverter against three-phase unbalanced loads includes the following steps:

[0007] Separate the three-phase signals output by the three-level three-phase four-bridge arm inverter to obtain single-phase signals, and perform virtual coordinate transformation on the single-phase signals respectively to obtain the single-phase signals after virtual coordinate transformation; the single-phase signals after virtual coordinate transformation include three-phase output voltage, filter capacitor voltage, load current, and filter inductor current;

[0008] Perform double closed-loop control on the single-phase signals after virtual coordinate transformation;

[0009] Perform rotational coordinate transformation on the single-phase signals after double closed-loop control to obtain single-phase signals in the α-β coordinate system, and take the α-axis component as the modulation wave of the corresponding single-phase signal;

[0010] Input the modulation waves of the three single-phase signals into the SPWM modulation module respectively to obtain switch drive signals;

[0011] Through the fourth bridge arm control strategy, obtain the complementary control signals of the fourth bridge arm switch, and realize the control of the neutral point potential offset of the three-level three-phase four-bridge arm inverter.

[0012] Optionally, separating the three-phase signals output by the three-level three-phase four-bridge arm inverter to obtain single-phase signals, and performing virtual coordinate transformation on the single-phase signals respectively to obtain the single-phase signals after virtual coordinate transformation includes:

[0013] Separate the three-phase signals output by the three-level three-phase four-bridge arm inverter to obtain single-phase signals;

[0014] Construct a virtual signal that is strictly orthogonal to the single-phase signal in phase;

[0015] Take the single-phase signal as the α-axis component in the α-β coordinate system, and take the virtual signal as the β-axis component in the α-β coordinate system;

[0016] Introduce a synchronous rotation angle, and transform the α-axis component and β-axis component in the α-β coordinate system into the d-axis component and q-axis component in the d-q coordinate system to obtain the single-phase signals after virtual coordinate transformation.

[0017] Optionally, a virtual signal that is strictly orthogonal in phase to the single-phase signal is constructed, including: delaying the single-phase signal by T / 4 through an all-pass filter, that is, performing a 90° lag, to obtain a 90° lag virtual signal.

[0018] Optionally, double closed-loop control is performed on the single-phase signal after virtual coordinate transformation, including:

[0019] Taking the filtered capacitor voltage after virtual coordinate transformation as a feedforward quantity and inputting it into the current inner-loop control circuit to obtain a current inner-loop reference value;

[0020] Taking the load current after virtual coordinate transformation as a feedforward quantity and adding it to the voltage outer-loop control circuit to obtain a voltage outer-loop reference value.

[0021] Optionally, the rotation coordinate transformation expression is:

[0022]

[0023] where, is the α-axis component, is the β-axis component, is the d-axis component, is the q-axis component, and θ is the synchronous rotation angular velocity.

[0024] Optionally, through the fourth bridge arm control strategy, complementary control signals for the fourth bridge arm switches are obtained to achieve the control of the three-level three-phase four-bridge arm inverter, including:

[0025] Detecting the first capacitor voltage and the second capacitor voltage, subtracting the first capacitor voltage from the second capacitor voltage and passing it through a PI controller to obtain the set value of the first inductor current;

[0026] Detecting the first inductor current, subtracting the detected first inductor current from the set value of the first inductor current and passing it through a PI controller to obtain the fourth bridge arm modulation wave;

[0027] Generating a triangular wave through a triangular wave generator, comparing the fourth bridge arm modulation wave with the triangular wave to obtain complementary control signals for the first switch and the second switch, and realizing the control of the midpoint potential offset of the three-level three-phase four-bridge arm inverter.

[0028] Optionally, the expression for the set value of the current of the first inductor is:

[0029]

[0030] where, is the set value of the current of the first inductor, 、 are the proportional and integral parameters of the PI controller respectively, and s is the Laplace operator. is the first capacitor voltage, is the second capacitor voltage.

[0031] Optionally, the modulation wave expression of the fourth bridge arm is:

[0032]

[0033] wherein, is the modulation wave of the fourth bridge arm, and are the proportional and integral parameters of the PI controller respectively, is the current of the first inductor, s is the Laplace operator, is the set value of the first inductor current.

[0034] This application provides a control method for an off-grid inverter against three-phase unbalanced loads. Through the design of the fourth bridge arm hardware circuit, the neutral point potential is directly controlled, which can avoid the dependence on high-performance controllers in traditional software control methods, simplify the control logic and improve the response speed; by using virtual coordinate transformation, the d-q axis components of single-phase signals are independently separated and adjusted, which can reduce the algorithm complexity and the calculation burden of the controller; through the three-phase independent control method, independent control loops are designed for each phase voltage, which can eliminate the inter-phase coupling effect, realize the independent adjustment of each phase load, and significantly improve the operation stability of the system under three-phase unbalanced loads; through the collaborative optimization of hardware and software, the complexity and manufacturing cost of the system are reduced, while the power quality and operation efficiency are improved, which can provide strong support for the development of off-grid wind power generation systems.

[0035] To make the above features and advantages of the invention more obvious and understandable, specific embodiments are given below and will be described in detail in conjunction with the accompanying drawings as follows. Brief Description of the Drawings

[0036] In order to more clearly illustrate the technical solutions in the embodiments of the present application or related technologies, the following will briefly introduce the drawings required for use in the description of the embodiments or related technologies. Obviously, the drawings in the following description are only some embodiments of the present application. For those of ordinary skill in the art, other drawings can be obtained based on these drawings without creative efforts.

[0037] Figure 1 is a schematic structural diagram of an off-grid wind power generation system.

[0038] Figure 2 is a schematic structural diagram of an off-grid inverter system against three-phase unbalanced loads provided in an embodiment of the present application.

[0039] Figure 3Flow chart of the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0040] Figure 4 Virtual coordinate transformation control block diagram in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0041] Figure 5 Flow chart of step S4 in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0042] Figure 6 Dual closed-loop control block diagram in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0043] Figure 7 Flow chart of step S5 in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0044] Figure 8 Fourth bridge arm midpoint potential balance control block diagram in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0045] Figure 9 Three-phase output voltage waveform diagram of a three-level three-phase four-bridge arm inverter under three-phase unbalanced load conditions in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0046] Figure 10 Midpoint potential offset waveform diagram of a three-level three-phase four-bridge arm inverter under three-phase unbalanced load conditions in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0047] Figure 11 Three-phase phase voltage waveform diagram of a three-level three-phase four-bridge arm inverter with a single-phase load in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application.

[0048] Figure 12 Midpoint potential offset waveform diagram of a three-level three-phase four-bridge arm inverter with a single-phase load in the off-grid inverter control method for resisting three-phase unbalanced loads provided in another embodiment of the present application. Specific implementation manner

[0049] To make the objectives and technical solutions of the embodiments of this application clearer, the following will clearly and completely describe the technical solutions of the embodiments of this application with reference to the accompanying drawings of the embodiments of this application. Obviously, the described embodiments are some, but not all, of the embodiments of this application. All other embodiments obtained by those of ordinary skill in the art based on the described embodiments of this application without creative efforts fall within the scope of protection of this application.

[0050] As an example, please refer to Figure 1 , an off-grid inverter system for resisting three-phase unbalanced loads, which may include: a wind turbine 1, a diode rectifier 2, a DC converter 3, a three-level three-phase four-leg inverter 4, and a three-phase unbalanced load 5. Among them, the wind turbine 1 is connected to the first end of the diode rectifier 2, the second end of the diode rectifier 2 is connected to the first end of the DC converter 3, the second end of the DC converter 3 is connected to the first end of the three-level three-phase four-leg inverter 4, and the second end of the three-level three-phase four-leg inverter 4 is connected to the three-phase unbalanced load 5.

[0051] In one embodiment, please refer to Figure 2 , the three-level three-phase four-leg inverter 4 may include: a T-type neutral-point clamped three-level circuit 41, a fourth-leg circuit 42, and a filtering circuit 43. Among them, the first end of the fourth-leg circuit 42 is connected to the DC voltage V dc output by the DC converter 3, the second end of the fourth-leg circuit 42 is connected to the first end of the T-type neutral-point clamped three-level circuit 41, the second end of the T-type neutral-point clamped three-level circuit 41 is connected to the first end of the filtering circuit 43, and the second end of the filtering circuit 43 is connected to the three-phase unbalanced load 5.

[0052] As an example, the fourth-leg circuit 42 includes: a switch T1, a switch T2, and an inductor L. Among them, the first end of the switch T1 is connected to the first end of the DC voltage V dc , the second end of the switch T1 is connected to the first end of the switch T2, the second end of the switch T2 is connected to the second end of the DC voltage V dc , the first end of the inductor L is connected to the series midpoint of the switch T1 and the switch T2, the second end of the inductor L is connected to the first end of the T-type neutral-point clamped three-level circuit 41, and the output current of the inductor L is .

[0053] As an example, the T-type neutral-point clamped three-level circuit 41 includes: a capacitor C1, a capacitor C2, an A-phase leg, a B-phase leg, and a C-phase leg. Each leg includes four switches, and the leg-side voltages of the three legs are respectively , , . Specifically, the first end of the capacitor C1 is connected to the first end of the switch T1, the second end of the capacitor C1 is connected to the first end of the capacitor C2, the second end of the capacitor C2 is connected to the second end of the switch T2, and the voltage across the capacitor C1 is , and the voltage across the capacitor C2 is ; The A-phase bridge arm includes switches T3, T4, T9, and T10. Among them, the first end of the switch T3 is connected to the second end of the inductor L, the second end of the switch T3 is connected to the second end of the switch T4, the first end of the switch T4 is connected to the second end of the switch T9, the first end of the switch T9 is connected to the first end of the capacitor C1, the first end of the switch T10 is connected to the second end of the switch T9, and the second end of the switch T10 is connected to the second end of the capacitor C2; The B-phase bridge arm includes switches T5, T6, T11, and T12. Among them, the first end of the switch T5 is connected to the second end of the inductor L, the second end of the switch T5 is connected to the second end of the switch T6, the first end of the switch T6 is connected to the second end of the switch T11, the first end of the switch T11 is connected to the first end of the capacitor C1, the first end of the switch T12 is connected to the second end of the switch T11, and the second end of the switch T12 is connected to the second end of the capacitor C2; The C-phase bridge arm includes switches T7, T8, T13, and T14. Among them, the first end of the switch T7 is connected to the second end of the inductor L, the second end of the switch T7 is connected to the second end of the switch T8, the first end of the switch T8 is connected to the second end of the switch T13, the first end of the switch T13 is connected to the first end of the capacitor C1, the first end of the switch T14 is connected to the second end of the switch T13, and the second end of the switch T14 is connected to the second end of the capacitor C2.

[0054] As an example, the fourth bridge arm circuit 42 realizes the neutral point potential balance by alternately turning on the switches T1 and T2. Specifically, when the switch T1 is turned on, the switch T2 is in the off state. At this time, the DC-side capacitor C1, the switch T1, and the inductor L form a current loop. During this process, the energy stored in the capacitor C1 is released through the inductor L, resulting in a gradual decrease in the voltage of the capacitor C1; at the same time, the inductor L absorbs energy, and its current gradually increases accordingly.

[0055] Furthermore, when the switch T2 is turned on, the switch T1 is turned off, and the current loop switches to be composed of the DC-side capacitor C2, the switch T2, and the inductor L. Due to the freewheeling characteristic of the inductor, the energy stored in the inductor L is transferred to the capacitor C2, causing the voltage of the capacitor C2 to gradually increase; at the same time, the inductor L releases energy, and its current gradually decreases.

[0056] As an example, from the perspective of energy transfer, the inductor L can serve as a medium for energy exchange, and its core role is to facilitate the dynamic transfer of energy between the capacitor C1 and the capacitor C2. Through a reasonable control strategy, by adjusting the conduction timing and duty cycle of the switch T1 and the switch T2, the energy exchange between the capacitor C1 and the capacitor C2 can reach a dynamic balance state, that is, the voltages of the two tend to be equal, achieving precise control of the midpoint potential, effectively suppressing the offset of the midpoint potential, and thus enhancing the stability and reliability of the system.

[0057] As an example, the filter circuit 43 includes: a phase-A sub-filter circuit, a phase-B sub-filter circuit, and a phase-C sub-filter circuit. Specifically, the phase-A sub-filter circuit includes a resistor , an inductor , and a capacitor . Among them, the first end of the resistor is connected to the midpoint of the phase-A bridge arm, that is, the series midpoint of the switch T4 and the switch T9. The second end of the resistor is connected to the first end of the inductor . The second end of the inductor is connected to the three-phase unbalanced load 5. The first end of the capacitor is connected to the second end of the inductor . The second end of the capacitor is grounded. The phase-A inductor current output by the inductor is . The phase-A capacitor current flowing into the capacitor is . The phase-A capacitor voltage output by the capacitor is . The phase-A load current output is . The phase-B sub-filter circuit includes a resistor , an inductor , and a capacitor . Among them, the first end of the resistor is connected to the midpoint of the phase-B bridge arm, that is, the series midpoint of the switch T6 and the switch T11. The second end of the resistor is connected to the first end of the inductor . The second end of the inductor is connected to the three-phase unbalanced load 5. The first end of the capacitor is connected to the second end of the inductor . The second end of the capacitor is grounded. The phase-B inductor current output by the inductor is . The phase-B capacitor current flowing into the capacitor is . The phase-B capacitor voltage output by the capacitor is . The phase-B load current output is . The C-phase sub-filter circuit includes a resistor , an inductor , and a capacitor . Among them, the first end of the resistor is connected to the midpoint of the C-phase bridge arm, that is, the series midpoint of switch T8 and switch T13. The second end of the resistor is connected to the first end of the inductor . The second end of the inductor is connected to the three-phase unbalanced load 5. The first end of the capacitor is connected to the second end of the inductor . The second end of the capacitor is grounded. The C-phase inductor current output by the inductor is . The C-phase capacitor current flowing into the capacitor is . The C-phase capacitor voltage output by the capacitor is . The C-phase load current output is .

[0058] As an example, the wind turbine 1 can be a permanent magnet synchronous generator; the diode rectifier 2 can include six high-power diodes to form a three-phase full-bridge topology.

[0059] As an example, the DC converter 3 can be a Boost DC converter; the DC converter 3 includes: an inductor L1, a diode D, a switch T15, and a capacitor C. Among them, the first end of the inductor L1 is connected to the first end of the diode rectifier 2. The second end of the inductor L1 is connected to the first end of the switch T15. The second end of the switch T15 is connected to the second end of the diode rectifier 2. The first end of the diode D is connected to the second end of the inductor L1. The second end of the diode D is connected to the first end of the capacitor C. The second end of the capacitor C is connected to the second end of the diode rectifier 2. The voltage across the capacitor C is , and a DC voltage V dc is connected in parallel across the two ends of the capacitor C.

[0060] As an example, the wind turbine 1 generates three-phase alternating current, which is converted into direct current by the diode rectifier 2 and then adjusted to a DC voltage level matching the energy storage system by the DC converter 3, so as to provide a stable DC source for the subsequent inversion link. Since the DC bus voltage is a constant value, the output power of the wind turbine 1 can be flexibly adjusted by adjusting the conduction time of the switching device in the DC converter 3, so as to maximize the wind energy utilization efficiency.

[0061] As an example, a diode rectifier 2 is used to rectify the three-phase alternating current output by the wind turbine 1 to obtain a direct current voltage, and a DC converter 3 is used to keep the direct current voltage obtained by the wind turbine 1 through the diode rectifier 2 constant. A three-level three-phase four-leg inverter 4 is used to perform virtual coordinate transformation on the output voltage and current for three-phase independent control. While ensuring the elimination of the inter-phase coupling effect, realizing the independent regulation of each phase load, and improving the operation stability and power quality of the off-grid wind power generation system under unbalanced load conditions, it also realizes the control of the neutral point potential to avoid the dependence on high-performance controllers by traditional software control methods, simplifies the control logic, and improves the response speed, ensuring that the neutral point potential is quickly stabilized when the load dynamically changes, and further improving the system reliability.

[0062] In the above off-grid inverter system for resisting three-phase unbalanced loads, by setting the fourth leg circuit 42 and the T-type neutral point clamped three-level circuit 41 to work together, the problem of three-phase unbalanced loads can be effectively addressed. The switches T1 and T2 in the fourth leg circuit 42 conduct alternately, which can achieve precise control of the neutral point potential, effectively suppress the offset of the neutral point potential, and further improve the stability and reliability of the system.

[0063] In another embodiment, please refer to Figure 3 , the present application provides a control method for an off-grid inverter for resisting three-phase unbalanced loads. The control method for an off-grid inverter for resisting three-phase unbalanced loads may include the following steps: Step S1 to Step S5.

[0064] Step S1: Separate the three-phase signals output by the three-level three-phase four-leg inverter to obtain single-phase signals, and perform virtual coordinate transformation on the single-phase signals respectively to obtain the single-phase signals after virtual coordinate transformation; the single-phase signals after virtual coordinate transformation include three-phase output voltage, filter capacitor voltage, load current, and filter inductor current.

[0065] Step S2: Perform double closed-loop control on the single-phase signals after virtual coordinate transformation.

[0066] Step S3: Perform rotation coordinate transformation on the single-phase signals after double closed-loop control to obtain single-phase signals in the α-β coordinate system, and take the α-axis component as the modulation wave of the corresponding single-phase signal.

[0067] Step S4: Input the modulation waves of the three single-phase signals into the SPWM modulation module respectively to obtain switch drive signals.

[0068] Step S5: Through the fourth leg control strategy, obtain the complementary control signals of the fourth leg switches to realize the control of the neutral point potential offset of the three-level three-phase four-leg inverter.

[0069] In the off-grid inverter control method for resisting three-phase unbalanced loads in this application, by separating the three-phase signals output by the three-level three-phase four-leg inverter into single-phase signals, performing virtual coordinate transformation, double closed-loop control, and rotating coordinate transformation, finally obtaining the modulation wave of the single-phase signal in the α-β coordinate system; inputting the modulation wave into the SPWM modulation module to obtain the switch drive signal, and combining with the control strategy of the fourth leg, it can effectively cope with three-phase unbalanced loads, improve the power quality of the inverter output, enhance the stability and reliability of the system, reduce the harmonic content, and ensure the normal and stable operation of electrical equipment.

[0070] In step S1, refer to Figure 3 Step S1 in, separate the three-phase signals output by the three-level three-phase four-leg inverter to obtain single-phase signals, and perform virtual coordinate transformation on the single-phase signals respectively to obtain the single-phase signals after virtual coordinate transformation; the single-phase signals after virtual coordinate transformation include filter capacitor voltage, load current, and filter inductor current.

[0071] As an example, the three-phase alternating current output by the wind turbine 1 is processed by the diode rectifier 2. When the three-phase alternating voltage output by the wind turbine 1 is applied to the input end of the diode rectifier 2, the diode conducts when it is forward-biased and cuts off when it is reverse-biased. Through this periodic on-off switching, the input sinusoidal alternating current is converted into direct current.

[0072] Further, the DC converter 3 performs closed-loop control on the output DC-side capacitor voltage Specifically, subtract the DC-side capacitor voltage from the given DC-side capacitor voltage reference value , and input the difference between the two into the PI controller, thereby generating a PWM modulation wave, and the expression is:

[0073]

[0074] Wherein, is the switch modulation signal of the DC converter, , are the proportional and integral parameters of the PI controller respectively.

[0075] Further, control the on and off of the switch in the DC converter through the duty cycle generated by the PWM modulation wave to keep the DC voltage constant. The three-level three-phase four-leg inverter continuously outputs three-phase symmetrical voltages with stable amplitude and frequency.

[0076] As an example, after the output voltage under the balanced load condition undergoes Clark and Park coordinate transformations, it can be converted into a DC quantity in the d-q synchronous rotating coordinate system. This characteristic enables the system to adopt a traditional PI control strategy to achieve accurate tracking of the reference signal and ensure that the steady-state error is zero. When the load is unbalanced, the output voltage and current will simultaneously contain positive-sequence, negative-sequence, and zero-sequence components, and the system needs to uniformly regulate them. The expression is as follows:

[0077]

[0078] Among them, e a 、e b 、e c represent the voltage components of phase A, phase B, and phase C respectively, E m represents the voltage amplitude, and the superscripts P, N, and 0 represent the positive-sequence, negative-sequence, and zero-sequence components respectively. This method has problems such as complex algorithms, large computational amounts, and slow dynamic responses. Especially when the load changes suddenly, it is difficult to quickly track the load changes, resulting in asymmetric output voltage and current distortion. In a three-phase power system, the voltages and currents of each phase exhibit obvious time-varying characteristics and symmetric distribution characteristics in the stationary a-b-c coordinate system. Specifically, the voltages and currents of each phase not only change periodically with time, but also, under ideal conditions, maintain a strict symmetric relationship with equal amplitudes and a phase difference of 120° between the three phases. By applying the Clark transformation and Park transformation, these time-varying AC quantities can be transformed from the a-b-c coordinate system to the synchronous rotating d-q coordinate system. In this new coordinate system, the originally sinusoidally varying AC quantities with time are transformed into DC quantities synchronized with the fundamental frequency of the power grid:

[0079]

[0080]

[0081] Among them, u d 、u q respectively represent the DC voltage components obtained through the Park transformation under the condition of constant amplitude, i d 、i q respectively represent the corresponding DC current components obtained through the Park transformation under the condition of constant amplitude, 、 、 are the three-phase voltages on the arm side of the three bridge arms, 、 、 are the corresponding three-phase currents on the arm side of the three bridge arms, is the synchronous rotation angular velocity. This transformation converts the time-varying quantities in a three-phase AC system into DC quantities in a synchronous rotating coordinate system, thus simplifying the design of the control system. For the converted DC quantities, a PI controller can be used to achieve precise tracking control, which can not only ensure the dynamic response performance of the system but also effectively eliminate the steady-state error.

[0082] Based on the above principle, the three-phase signals output by the three-level three-phase four-leg inverter are separated to obtain single-phase signals, and virtual coordinate transformation is performed on the single-phase signals respectively, and a mathematical model in an independent single-phase d-q rotating coordinate system can be established. In this application, the three-phase signals output by the three-level three-phase four-leg inverter in step S1 include: three-phase output voltage, filter capacitor voltage, load current, and filter inductor current.

[0083] Specifically, step S1 may include the following steps: separating the three-phase signals output by the three-level three-phase four-leg inverter to obtain single-phase signals; constructing a virtual signal that is strictly orthogonal to the single-phase signal in phase; using the single-phase signal as the α-axis component in the α-β coordinate system and the virtual signal as the β-axis component in the α-β coordinate system; introducing the synchronous rotation angle to transform the α-axis component and β-axis component in the α-β coordinate system into the d-axis component and q-axis component on the d-q coordinate system, and obtaining the single-phase signal after virtual coordinate transformation.

[0084] As an example, the single-phase signal includes: the output voltage of phase A u a , the filter capacitor voltage of phase A , the load current of phase A , the inductor current of phase A ; the output voltage of phase B u b , the filter capacitor voltage of phase B , the load current of phase B , the inductor current of phase B ; the output voltage of phase C u c , the filter capacitor voltage of phase C , the load current of phase C , the load current of phase C .

[0085] As an example, the single-phase signal after virtual coordinate transformation includes: the d-axis component of the output voltage of phase A , the q-axis component of the output voltage of phase A , the d-axis component of the filter capacitor voltage of phase A , the q-axis component of the filter capacitor voltage of phase A , the d-axis component of the load current of phase A , the q-axis component of the load current of phase A , d-axis component of phase-A inductor current , q-axis component of phase-A inductor current ; d-axis component of phase-B output voltage , q-axis component of phase-B output voltage , d-axis component of phase-B filter capacitor voltage , q-axis component of phase-B filter capacitor voltage , d-axis component of phase-B load current , q-axis component of phase-B load current , d-axis component of phase-B inductor current , q-axis component of phase-B inductor current ; d-axis component of phase-C output voltage , q-axis component of phase-C output voltage , d-axis component of phase-C filter capacitor voltage , q-axis component of phase-C filter capacitor voltage , d-axis component of phase-C load current , q-axis component of phase-C load current , d-axis component of phase-C inductor current , q-axis component of phase-C inductor current 。

[0086] The following takes the phase-A signal as an example to specifically introduce step S1.

[0087] As an example, please refer to Figure 4 , construct a virtual alternating quantity that is strictly orthogonal to the original phase-A signal in phase, which can not only maintain the amplitude characteristics of the original signal, but also ensure that the phase difference between the virtual signal and the original signal is constantly 90°.

[0088] Specifically, use an all-pass filter to lag the phase-A signal by 90° (i.e., delay T / 4) to obtain a 90°-lagged phase-A virtual signal. The transfer function of the all-pass filter in the s domain The expression is:

[0089]

[0090] where f is the fundamental frequency, which is 50 Hz, and s is the Laplace operator.

[0091] As an example, assume that the phase-A output voltage of a three-level three-phase four-leg inverter u a , the expression is:

[0092]

[0093] where is the voltage amplitude, is the angular frequency, is the phase angle, is the time.

[0094] As an example, the phase-A output voltage is delayed by T / 4 through an all-pass filter, that is, a 90° lag is performed to obtain a 90°-lagged virtual phase-A output voltage :

[0095]

[0096] wherein, is the voltage amplitude, is the angular frequency, is the phase angle, is the time.

[0097] Furthermore, the original phase-A output voltage is used as the α-axis component in the α-β coordinate system , and the 90°-lagged virtual phase-A output voltage is regarded as the β-axis component . By introducing θ as the synchronous rotation angular velocity, the α-axis component and the β-axis component in the α-β coordinate system are transformed to the d-q coordinate system to obtain the voltage signals in the synchronous rotation coordinate system and the stationary coordinate system:

[0098]

[0099] wherein, is the d-axis voltage component, is the q-axis voltage component, is the α-axis voltage component, is the β-axis voltage component, and θ is the synchronous rotation angular velocity.

[0100] Similarly, the current signals in the synchronous rotation coordinate system and the stationary coordinate system can be obtained as:

[0101]

[0102] wherein, is the d-axis voltage component, is the q-axis voltage component, is the α-axis voltage component, is the β-axis voltage component, and θ is the synchronous rotation angular velocity.

[0103] Similarly, the B-phase voltage signal, B-phase current signal, C-phase voltage signal, and C-phase current signal in the d-q coordinate system can be obtained. Through virtual coordinate transformation, not only the control design of the single-phase system is simplified, but also it provides convenience for the dynamic performance analysis and optimization of the system. By converting AC quantities into DC quantities, various control strategies can be more effectively applied to achieve precise regulation and stable operation of the system.

[0104] In step S2, refer to Figure 3 step S2 in

[0105] For example, refer to Figure 5 and step S2 may include the following steps: step S21 to step S22.

[0106] Step S21: Take the filtered capacitor voltage after virtual coordinate transformation as the feedforward quantity and input it into the current inner-loop control circuit to obtain the current inner-loop reference value, and perform decoupling control on the filtered inductor current 、 .

[0107] Step S22: Take the load current after virtual coordinate transformation as the feedforward quantity and add it to the voltage outer-loop control circuit to obtain the voltage outer-loop reference value, and perform decoupling control on the filtered capacitor voltage after virtual coordinate transformation.

[0108] For example, the d-axis components 、 of the filtered capacitor voltage and inductor current in the complex frequency domain are expressed as:

[0109]

[0110] where is the d-axis component of the filtered capacitor voltage, is the q-axis component of the filtered capacitor voltage, is the d-axis component of the voltage on the bridge arm side of the three-level three-phase four-leg inverter, is the d-axis component of the inductor current, is the d-axis component of the load current, is the q-axis component of the inductor current, is the value of the filtered capacitor, is the value of the filtered inductor, is the equivalent resistance of the filtered inductor, is the d-axis fundamental angular frequency, and s is the Laplace operator.

[0111] Similarly, the q-axis components 、 of the filtered capacitor voltage and inductor current in the complex frequency domain are expressed as:

[0112]

[0113] Among them, is the d-axis filter capacitor voltage, is the q-axis component of the filter capacitor voltage, is the q-axis component of the arm-side voltage of the three-level three-phase four-leg inverter, is the d-axis component of the inductor current, is the q-axis component of the load current, is the q-axis component of the inductor current, is the value of the filter capacitor, is the value of the filter inductor, is the equivalent resistance of the filter inductor, is the d-axis fundamental angular frequency, and s is the Laplace operator.

[0114] As an example, it can be seen from the expressions of the d-axis component in the complex frequency domain and the q-axis component in the complex frequency domain that after the rotation transformation, a significant cross-coupling relationship is formed between the d-axis and the q-axis of the system. Specifically, the output inductor current , is not only directly regulated by the arm-side voltage , of the three-level three-phase four-leg inverter, but also affected by the coupling voltage , introduced by the rotating coordinate system and the interaction of the filter capacitor voltage , . At the same time, the filter capacitor voltage , is controlled by the inductor current , , and is also affected by the coupling current components , and the dynamic disturbances of the load current , . This complex multivariable coupling relationship makes it impossible to completely decouple the control loops of the d-axis and the q-axis, and an appropriate decoupling compensation strategy must be introduced to eliminate the cross-axis interference, so as to ensure that each axis can achieve precise independent control.

[0115] As an example, the double closed-loop control block diagram is as shown in Figure 6 . The following takes signal A as an example to specifically introduce steps S21 - S22.

[0116] Specifically, in step S21, the d-axis component of the A-phase filter capacitor voltage and the q-axis component after the virtual coordinate transformation are introduced into the current inner-loop control loop as feedforward quantities to obtain the current inner-loop reference value, and the d-axis component of the filter inductor current is realized. Decoupling control of the q-axis component of the filter inductor current has the following expression:

[0117]

[0118] where is the d-axis component of the given reference value of the inductor current, is the q-axis component of the given reference value of the inductor current, , respectively correspond to the proportional and integral coefficients of the current inner-loop PI regulator, is the d-axis component of the filter capacitor voltage, is the q-axis component of the filter capacitor voltage, is the d-axis component of the voltage on the arm side of the three-level three-phase four-leg inverter, is the q-axis component of the voltage on the arm side of the three-level three-phase four-leg inverter, is the d-axis component of the inductor current, is the d-axis component of the load current, is the q-axis component of the inductor current, is the fundamental d-axis angular frequency, is the value of the filter inductor.

[0119] As an example, the proportional and integral coefficients , of the current inner-loop PI regulator can be calculated by the following expression:

[0120]

[0121]

[0122] where is the value of the filter inductor, is the equivalent resistance of the filter inductor, is the switching period, .

[0123] Furthermore, in step S22, the d-axis component and the q-axis component of the A-phase load current after the virtual coordinate transformation are added as feedforward quantities to the voltage outer-loop control loop to obtain the voltage outer-loop reference value, realizing decoupling control of the d-axis component and the q-axis component of the filter capacitor voltage, and the expression is:

[0124]

[0125] where is the d-axis component of the given reference value of the capacitor voltage, is the q - axis component of the given reference value of the capacitor voltage, and correspond to the proportional and integral coefficients of the voltage outer - loop PI regulator respectively, is the d - axis component of the filter capacitor voltage, is the q - axis component of the filter capacitor voltage, is the d - axis component of the inductor current, is the d - axis component of the load current, is the q - axis component of the load current, is the q - axis component of the inductor current, is the d - axis fundamental angular frequency, is the value of the filter capacitor.

[0126] As an example, the proportional and integral coefficients and of the voltage outer - loop PI regulator can be calculated by the following expressions:

[0127]

[0128]

[0129] where is the value of the filter inductor, is the value of the filter capacitor, and correspond to the proportional and integral coefficients of the current inner - loop PI regulator respectively, , .

[0130] As an example, by adopting the feed - forward decoupling control strategy and introducing the corresponding compensation terms in the control loop, the mutual interference between the d - axis and q - axis is effectively eliminated, and finally the independent and precise control of the single - phase output variable in the d - q coordinate system is realized. This double - closed - loop control method can effectively cancel the cross - coupling terms introduced by the rotating coordinate system, which can not only improve the dynamic response performance of the system, but also enhance the anti - interference ability of the system, providing a reliable guarantee for achieving high - quality inverter output.

[0131] In step S3, refer to Figure 3 step S3 therein, perform a rotating coordinate transformation on the single - phase signal after double - closed - loop control to obtain the single - phase signal in the α - β coordinate system, and take the α - axis component as the modulation wave of the corresponding single - phase signal.

[0132] As an example, the modulation waves of the single - phase signal include: the modulation wave of the A - phase signal, the modulation wave of the B - phase signal, and the modulation wave of the C - phase signal.

[0133] Specifically, taking the A-phase signal as an example, the d-axis component of the A-phase voltage signal on the arm side of the three-level three-phase four-leg inverter after double closed-loop control , the q-axis component of the A-phase voltage undergoes a rotational coordinate transformation to obtain the A-phase voltage components in the α-β coordinate system , , and the expressions are as follows:

[0134]

[0135] where, is the α-axis voltage component, is the β-axis voltage component, is the matrix of the d-axis component of the A-phase voltage , is the matrix of the q-axis component of the A-phase voltage , and θ is the synchronous rotational angular velocity.

[0136] Further, take the α-axis voltage component as the modulation wave of the A-phase signal .

[0137] As an example, similarly, according to steps S1 to S3, the modulation wave of the B-phase signal , and the modulation wave of the C-phase signal can be obtained.

[0138] In step S4, refer to Figure 3 for step S4 therein, and input the modulation waves of the three single-phase signals into the SPWM modulation module respectively to obtain the switch drive signals.

[0139] Specifically, input the modulation waves of the three-phase signals , , into the SPWM modulation module (not shown) to obtain the switch drive signals. The SPWM modulation module generates a high-frequency triangular carrier signal with a fixed frequency and amplitude. The modulation waves of the three-phase signals , , are respectively compared with the high-frequency triangular carrier signal. When the amplitude of the modulation wave is greater than the amplitude of the high-frequency triangular carrier signal, a high-level drive signal is output; when the amplitude of the modulation wave is less than the amplitude of the high-frequency triangular carrier signal, a low-level drive signal is output. The switch conducts and turns off according to the high and low level states of the drive signal. In this way, the inverter can convert the DC power supply into an AC output, and the amplitude, frequency, and phase of the output AC voltage can be controlled by the modulation wave.

[0140] In step S5, refer to Figure 3In step S5, through the control strategy of the fourth bridge arm, complementary control signals for the switches of the fourth bridge arm are obtained to achieve the control of the three-level three-phase four-bridge-arm inverter.

[0141] For example, please refer to Figure 7 , step S5 may include the following steps: step S51 to step S53.

[0142] Step S51: Detect the capacitor voltage and , and after the voltage values of the two are subtracted and passed through a PI controller, the set value of the inductor current of the fourth bridge arm is obtained.

[0143] Step S52: Detect the inductor current of the fourth bridge arm , and subtract the detected inductor current of the fourth bridge arm from the set value of the inductor current of the fourth bridge arm , and then pass the result through a PI controller to obtain the modulation wave of the fourth bridge arm.

[0144] Step S53: Generate a triangular wave through a triangular wave generator, compare the modulation wave of the fourth bridge arm with the triangular wave, and obtain the complementary control signals for the switches of the fourth bridge arm.

[0145] For example, Figure 8 is the control block diagram for the midpoint potential balance of the fourth bridge arm. In step S51, detect the voltage of capacitor C1 and the voltage of capacitor C2, subtract the voltage from the voltage , input the difference between the voltage and the voltage into the PI controller, and calculate the set value of the inductor current of the fourth bridge arm to reduce the difference between the voltages of the two capacitors. The expression is as follows:

[0146]

[0147] where , are the proportional and integral parameters of the PI controller respectively, and s is the Laplace operator.

[0148] Furthermore, in step S52, detect the inductor current of the fourth bridge arm , compare the detected current of the inductor L of the fourth bridge arm with the set value of the inductor current of the fourth bridge arm, input the difference between the two into the PI controller to obtain the modulation wave of the fourth bridge arm to further refine the modulation signal and ensure the inductor current Track its set value more precisely, and the expression is as follows:

[0149]

[0150] Wherein, 、 are the proportional and integral parameters of the PI controller respectively, and s is the Laplace operator.

[0151] Further, in step S53, a triangular wave with a set frequency is generated by a triangular wave generator, and the refined modulation wave of the fourth bridge arm is compared with the triangular wave to obtain complementary control signals for switches T1 and T2, precisely controlling the on and off states of the switching tubes, and achieving precise control of the inductor current of the fourth bridge arm . Through the control strategy of the fourth bridge arm, the energy flow between the capacitors of the fourth bridge arm can be effectively managed to ensure that the voltage and the voltage are maintained in the desired balanced state, which can improve the stability of the system and optimize the utilization efficiency of electric energy, and is crucial for the performance of power electronic devices.

[0152] In one example, the present application also builds a simulation model of an off-grid wind power generation system inverter for resisting three-phase unbalanced loads in Matlab / Simulink simulation software. Figure 9 is the waveform diagram of the three-phase output voltage of the three-level three-phase four-bridge-arm inverter under the condition of three-phase unbalanced loads, Figure 10 is the waveform diagram of the midpoint potential offset of the three-level three-phase four-bridge-arm inverter with three-phase unbalanced loads. Among them, three-phase unbalanced loads and single-phase unbalanced loads are connected at 0.5 s respectively, Figure 9 showing the three-phase output voltage waveforms of the inverter under the condition of three-phase unbalanced loads, Figure 10 shows the dynamic offset characteristics of the midpoint potential under load imbalance. It can be clearly observed that after adopting the control method proposed in the present application, the dynamic response performance of the off-grid wind power generation system under load mutation conditions has been significantly improved. The three-level three-phase four-bridge-arm inverter can quickly suppress the fluctuation of the midpoint potential and quickly restore to a stable state during the dynamic change of the load, thereby effectively improving the operation reliability of the off-grid wind power generation system. Even under severe three-phase load imbalance conditions, the three-level three-phase four-bridge-arm inverter can still continuously output three-phase symmetrical voltages with stable amplitude and frequency.

[0153] In another example, the present application also simulates the operating conditions of the most severe three-phase unbalanced loads in Matlab / Simulink simulation software, that is, the operating conditions of single-phase loads (the other two-phase loads are both open). Figure 11The waveform diagram of the neutral point potential offset of the three-level three-phase four-leg inverter under the condition of three-phase unbalanced load, Figure 12 is the waveform diagram of the neutral point potential offset of the three-level three-phase four-leg inverter with a single-phase load. It can be seen that even in the face of the most severe three-phase unbalanced load condition, after the inverter of the wind power generation system adopts the control method proposed in this application, it can still ensure that the neutral point potential is quickly stabilized during the dynamic change of the load and can continuously output three-phase symmetrical voltages with stable amplitude and frequency.

[0154] In the off-grid inverter control method for resisting three-phase unbalanced loads of this application, through virtual coordinate transformation, the d-q axis components of single-phase signals are independently separated and adjusted, which can avoid the redundant operations brought by coordinate transformation in traditional three-phase unbalanced load problems, reduce the algorithm complexity and the calculation burden of the controller; through the voltage-current double closed-loop control method with three-phase independent control, an independent control loop is designed for each phase to eliminate the inter-phase coupling effect and realize the independent adjustment of each phase load, which can significantly improve the dynamic response ability of the system under load mutation conditions; through the fourth leg control strategy, the neutral point potential is directly controlled to ensure that the neutral point potential is quickly stabilized during the dynamic change of the load, further improving the system reliability; through the collaborative optimization of hardware and software, the complexity and manufacturing cost of the system are reduced, while the power quality and operation efficiency are improved, filling the gaps in the prior art in off-grid wind power generation systems; it can make full use of wind energy resources, reduce the dependence on traditional energy, and thus reduce the energy cost.

[0155] It should be understood that although each step in the flowchart of the accompanying drawings is displayed in sequence according to the indication of the arrow, these steps do not necessarily need to be executed in the order indicated by the arrow. Unless there is a clear indication in this article, the execution of these steps has no strict order restriction, and these steps can be executed in other orders. Moreover, at least a part of the steps in the accompanying drawings may include multiple sub-steps or multiple stages. These sub-steps or stages do not necessarily need to be completed at the same moment, but can be executed at different moments. The execution order of these sub-steps or stages does not necessarily need to be sequential, but can be executed alternately or alternately with at least a part of other steps or sub-steps or stages of other steps.

[0156] The technical features of the above embodiments can be combined arbitrarily. For the sake of concise description, not all possible combinations of the technical features in the above embodiments are described. However, as long as the combination of these technical features does not conflict, it should be considered as the scope recorded in this specification.

[0157] Although the present application has been disclosed above by way of examples, it is not intended to limit the present application. Any person having ordinary knowledge in the relevant technical field may make some modifications and refinements without departing from the spirit and scope of the present application. Therefore, the scope of protection of the present application shall be subject to that defined by the appended claims for patent application.

Claims

1. A control method for an off-grid inverter against three-phase unbalanced loads, the off-grid inverter against three-phase unbalanced loads comprising: Wind turbine, diode rectifier, DC converter, three-level three-phase four-leg inverter, three-phase unbalanced load. The wind turbine is connected to the first end of the diode rectifier. The second end of the diode rectifier is connected to the first end of the DC converter. The second end of the DC converter is connected to the first end of the three-level three-phase four-leg inverter. The second end of the three-level three-phase four-leg inverter is connected to the three-phase unbalanced load. The three-level three-phase four-leg inverter includes: a fourth leg circuit, the first end of the fourth leg circuit is connected to the DC voltage output by the DC converter; a T-type neutral point clamped three-level circuit, the first end of the T-type neutral point clamped three-level circuit is connected to the second end of the fourth leg circuit, and the T-type neutral point clamped three-level circuit includes three legs; a filter circuit, the first end of the filter circuit is connected to the second end of the T-type neutral point clamped three-level circuit, and the second end of the filter circuit is connected to the three-phase unbalanced load; The fourth leg circuit includes: a first switch, the first end of the first switch is connected to the first end of the DC voltage; a second switch, the first end of the second switch is connected to the second end of the first switch, and the second end of the second switch is connected to the second end of the DC voltage; a first inductor, the first end of the first inductor is connected to the series midpoint of the first switch and the second switch, and the second end of the first inductor is connected to the first end of the T-type neutral point clamped three-level circuit. It is characterized by including the following steps: Separate the three-phase signals output by the three-level three-phase four-leg inverter to obtain single-phase signals, and perform virtual coordinate transformation on the single-phase signals respectively to obtain the single-phase signals after virtual coordinate transformation; the single-phase signals after virtual coordinate transformation include three-phase output voltage, filter capacitor voltage, load current, and filter inductor current; Perform double closed-loop control on the single-phase signals after virtual coordinate transformation; Perform rotating coordinate transformation on the single-phase signals after double closed-loop control to obtain single-phase signals in the α-β coordinate system, and take the α-axis component as the modulation wave of the corresponding single-phase signal; Input the modulation waves of the three single-phase signals into the SPWM modulation module respectively to obtain switch drive signals; Obtain the complementary control signals of the fourth leg switches through the fourth leg control strategy to achieve the control of the neutral point potential offset of the three-level three-phase four-leg inverter.

2. The off-grid inverter control method for resisting three-phase unbalanced loads according to claim 1, wherein, Separate the three-phase signals output by the three-level three-phase four-leg inverter to obtain single-phase signals, and perform virtual coordinate transformation on the single-phase signals respectively to obtain the single-phase signals after virtual coordinate transformation, including: Separate the three-phase signals output by the three-level three-phase four-leg inverter to obtain single-phase signals; Construct a virtual signal that is strictly orthogonal to the single-phase signal in phase; Take the single-phase signal as the α-axis component in the α-β coordinate system, and take the virtual signal as the β-axis component in the α-β coordinate system; Introduce a synchronous rotation angle to transform the α-axis component and β-axis component in the α-β coordinate system into the d-axis component and q-axis component in the d-q coordinate system to obtain the single-phase signals after virtual coordinate transformation.

3. The off-grid inverter control method for resisting three-phase unbalanced loads according to claim 2, wherein, Construct a virtual signal that is strictly orthogonal in phase to the single-phase signal, including: delaying the single-phase signal by T / 4 through an all-pass filter, that is, performing a 90° lag, to obtain a 90°-lag virtual signal.

4. The off-grid inverter control method for resisting three-phase unbalanced loads according to claim 1, wherein, Perform double closed-loop control on the single-phase signal after virtual coordinate transformation, including: Input the filtered capacitor voltage after virtual coordinate transformation as a feedforward quantity into the current inner-loop control circuit to obtain the current inner-loop reference value; Add the load current after virtual coordinate transformation as a feedforward quantity to the voltage outer-loop control circuit to obtain the voltage outer-loop reference value.

5. The off-grid inverter control method for resisting three-phase unbalanced loads according to claim 1, characterized in that, The expression of the rotational coordinate transformation is: Among them, is the α-axis component, is the β-axis component, is the d-axis component, is the q-axis component, and θ is the synchronous rotational angular velocity.

6. The off-grid inverter control method for resisting three-phase unbalanced loads according to claim 1, characterized in that Through the fourth-bridge-arm control strategy, obtain the complementary control signal of the fourth-bridge-arm switch to achieve the control of the neutral-point potential offset of the three-level three-phase four-bridge-arm inverter, including: Detect the first capacitor voltage and the second capacitor voltage, subtract the first capacitor voltage from the second capacitor voltage and then pass it through a PI controller to obtain the set value of the first inductor current; Detect the first inductor current, subtract the detected first inductor current from the set value of the first inductor current and then pass it through a PI controller to obtain the fourth-bridge-arm modulation wave; Generate a triangular wave through a triangular wave generator, compare the fourth-bridge-arm modulation wave with the triangular wave to obtain the complementary control signal of the first switch and the second switch, and achieve the control of the three-level three-phase four-bridge-arm inverter.

7. The off-grid inverter control method for resisting three-phase unbalanced loads according to claim 6, wherein, The expression of the set value of the first inductor current is: Among them, is the set value of the first inductor current, , are the proportional and integral parameters of the PI controller respectively, s is the Laplace operator, is the first capacitor voltage, is the second capacitor voltage.

8. The off-grid inverter control method for resisting three-phase unbalanced loads according to claim 6, characterized in that The expression of the fourth-bridge-arm modulation wave is: Among them, is the modulation wave of the fourth bridge arm, , are the proportional and integral parameters of the PI controller respectively, is the current of the first inductor, s is the Laplace operator, is the set value of the current of the first inductor.

Images