Dual Power Superposition Synchronization and Feedforward Passive Remodeling Control Method and System for Grid-Forming Inverters

Abstract

The present application discloses a dual-power superposition synchronization and feed-forward passive reshaping control method and system for a network-forming inverter, belonging to the field of power electronics technology. The method includes measuring the capacitor voltage, grid current, and inductor current, and calculating the actual active power and actual reactive power; calculating an error signal based on the reactive power reference and the actual reactive power, and feeding it forward to the active power controller through a dual-power superposition branch to calculate the reference angle; calculating the voltage amplitude based on the active power reference and the actual active power; performing coordinate transformation on the grid current to obtain the transformed grid current, and obtaining the inductor current reference value through a feed-forward controller; and providing a converter control signal through sinusoidal pulse width modulation according to the inductor current reference value and the coordinate-transformed inductor current. The present application can improve the synchronization stability of the network-forming inverter, reduce the risk of system wide-frequency oscillation, and ensure stability under various operating conditions.

Description

Technical Field

[0001] The present application relates to the field of power electronics technology, and particularly to a dual-power superposition synchronization and feed-forward passive reshaping control method and system for a grid-forming inverter. Background Art

[0002] With the large-scale grid connection of distributed energy and renewable energy in the power system, the importance of grid-forming inverters in improving power quality, enhancing grid stability, and realizing multi-energy complementarity has become increasingly prominent. The grid-connected inverters in the prior art usually adopt multi-time-scale control systems such as power control loops, AC control loops, and synchronization loops to control active and reactive power through sequential regulation. However, power electronic grid-connected devices interact with the AC grid within a relatively wide frequency band (from a few Hz to several kHz), resulting in the wide-band characteristic of the power system oscillation frequency. With the increase in the proportion of renewable energy and distributed power generation, the equivalent grid impedance at the point of common coupling gradually becomes significant, leading to a closer dynamic coupling between the inverter and the grid, thus increasing the risk of wide-band oscillation in the system, directly affecting the synchronization stability of the inverter and the safety of power feeding.

[0003] In the prior art, PI control or cascade feedback control is mostly used to regulate active and reactive power respectively, and there is insufficient coordination between the control loops. Under the conditions of grid disturbances or load mutations, the inherent time-delay effect of feedback control makes the response to sudden disturbances untimely, which may trigger wide-band oscillation and then reduce the dynamic stability of the system. At the same time, traditional control strategies lack sufficient damping in suppressing high-frequency dynamic disturbances and are difficult to eliminate high-frequency oscillations in a timely manner. In the case of a large change in grid impedance or the existence of wide-band disturbances, traditional control strategies are difficult to effectively suppress power coupling oscillations, which may lead to problems such as synchronization instability or wide-band resonance, resulting in a decline in the quality of the inverter output voltage and current and possibly threatening the safe operation of the entire power system. In summary, the grid-forming inverters in the prior art have significant technical problems in aspects such as wide-band oscillation risk, insufficient response speed, and lack of high-frequency dynamic suppression ability. Therefore, there is an urgent need for a new control method to solve the above problems and improve the overall performance of the inverter. Summary of the Invention

[0004] The present application aims to provide a dual-power superposition synchronization and feed-forward passive reshaping control method and system for a grid-forming inverter to solve the problems of wide-band oscillation risk, insufficient response speed, and lack of high-frequency dynamic suppression ability of the grid-forming inverter.

[0005] To achieve the above object, the technical solution of the present application is:

[0006] A dual-power superposition synchronization and feed-forward passive reshaping control method for a grid-forming inverter, comprising

[0007] Step S1: Measure capacitor voltage v o , grid current i g , after abc / αβ transformation, the actual active power is calculated using the instantaneous power calculation theory p and actual reactive power q ;

[0008] Step S2: Based on reactive power reference Q set The actual reactive power q , calculate the error signal e Q , and through the proportional link K Q After processing, pass through a high-pass filter G HPF Feed forward to active power controller G APC Output signal;

[0009] Step S3: Based on the active power reference P set The actual active power p and error signal e Q , calculate the reference angle required for Clarke transform or inverse Clarke transform θ ;

[0010] Step S4: Based on reactive power reference Q set and actual reactive power q , through the reactive power controller G PRC Calculate voltage amplitude V , combined with the reference angle required for the Clarke transform or inverse Clarke transform θ , the α-axis component of the voltage reference value is obtained through voltage reference calculation v orefα , the β-axis component of the voltage reference value v orefβ ;

[0011] Step S5: Based on the reference angle required for Clarke transform or Clarke inverse transform θ , the grid current i g , capacitor voltage v o and the inductor current i L After abc / αβ transformation, the α-axis component of the transformed grid current is obtained i gα , the β-axis component of the grid currenti gβ , the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ and the α-axis component of the inductor current i Lα , the β-axis component of the inductor current i Lβ , the α-axis component of the grid current i gα , the β-axis component of the grid current i gβ are processed by a grid current feedforward controller G f to obtain a current feedforward signal; based on the α-axis component of the voltage reference value v orefα , the β-axis component of the voltage reference value v orefβ , the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ , a voltage controller G V and the current feedforward signal, calculate the α-axis component of the current reference value i Lrefα 、 the β-axis component of the current reference value i Lrefβ ;

[0012] Step S6: Based on the α-axis component of the inductor current i Lα , the β-axis component of the inductor current i Lβ and the α-axis component of the current reference value i Lrefα 、 the β-axis component of the current reference value i Lrefβ , through a current controller G i , obtain the α-axis component of the reference voltage required for sinusoidal pulse width modulation v modα 、 the β-axis component of the reference voltage v modβ , after αβ / abc transformation, obtain the transformed reference voltage required for sinusoidal pulse width modulation v mod abc , and generate a three-phase inverter control signal through sinusoidal pulse width modulation.

[0013] Optionally, in step S2, the error signal is calculated e Q using the formula:

[0014]

[0015] where Q set is the reactive power reference, q is the actual reactive power;

[0016] The proportional link K Q has the expression:

[0017]

[0018] where Δθ max is the maximum allowable angle deviation of the network-forming inverter, Q max is the maximum allowable reactive power deviation of the network-forming inverter, is the proportional coefficient;

[0019] The high-pass filter G HPF has the transfer function:

[0020]

[0021] where ω c is the cut-off angular frequency of the high-pass filter G HPF ζ is the damping ratio, s s is the Laplace transform variable;

[0022] The low-pass filtering link of the active power controller G APC has the expression:

[0023]

[0024] where ω p is the cut-off angular frequency of the low-pass filter in the active power error path, m p is the active power droop coefficient, s is the Laplace transform variable;

[0025] The active power droop coefficient m p has the following expression:

[0026]

[0027] Among them, Δω max is the maximum allowable frequency deviation of the network-forming inverter, P max is the maximum allowable active power deviation of the network-forming inverter under specific working conditions, k 1 is the proportionality coefficient.

[0028] Optionally, in the step S3, the reference angle θ for Clarke transformation or Clarke inverse transformation is calculated by the formula:

[0029]

[0030]

[0031] Among them, Δω is the angular frequency adjustment amount of the network-forming inverter, ω n is the reference angular frequency of the network-forming inverter, ω p is the cut-off angular frequency of the low-pass filter in the active power error path, m p is the active power droop coefficient, P set is the active power reference, p is the actual active power, s is the Laplace transform variable.

[0032] Optionally, in the step S4, the low-pass filtering link of the reactive power controller G RPC is expressed as:

[0033]

[0034] Among them, ω q is the cut-off frequency of the low-pass filter in the reactive power error path, n q is the reactive power droop coefficient, s is the Laplace transform variable;

[0035] The reactive power droop coefficient n q is expressed as follows:

[0036]

[0037] Among them, ΔV maxis the maximum voltage amplitude regulation amount allowed for the network-forming inverter, Q max is the maximum reactive power deviation allowed for the network-forming inverter, k 2 is the proportionality coefficient;

[0038] Calculate the voltage amplitude V The formula is:

[0039]

[0040]

[0041] Wherein, n q is the reactive power droop coefficient, V n is the reference voltage of the network-forming inverter, ω q is the cut-off frequency of the low-pass filter in the reactive power error path, ΔV is the voltage amplitude V regulation amount;

[0042] Calculate the α-axis component of the voltage reference value v orefα and the β-axis component of the voltage reference value v orefβ The formula is:

[0043]

[0044]

[0045] Wherein, v orefα and v orefβ are the voltage reference values, θ is the reference angle for Clarke transformation or Clarke inverse transformation.

[0046] Optionally, in step S5, the grid current feedforward controller G f The formula is:

[0047]

[0048] Wherein, K pf is the proportional gain of the grid current feedforward controller, K df is the differential gain of the grid current feedforward controller, is the time constant of the differential filter, the time constant of the differential filter, sis the Laplace transform variable;

[0049] Voltage controller G v The formula for is:

[0050]

[0051] Wherein, K pv is the proportionality coefficient of the voltage controller, K r is the resonance gain coefficient, ω r is the bandwidth coefficient, ω 0 is the resonance center frequency, s is the Laplace transform variable;

[0052] Calculate the α-axis component of the current reference value i Lrefα 、 β-axis component of the current reference value i Lrefβ The formula for is:

[0053]

[0054]

[0055] Wherein, i Lrefα is the α-axis component of the inductor current reference value, i Lrefβ is the β-axis component of the inductor current reference value, v orefα is the α-axis component of the voltage reference value, v orefβ is the β-axis component of the voltage reference value, v oα is the α-axis component of the capacitor voltage, v oβ is the β-axis component of the capacitor voltage.

[0056] Optionally, in step S6, the current controller G i The formula for is:

[0057]

[0058] Wherein, K pi is the proportionality coefficient of the current controller;

[0059] Calculate the α-axis component of the reference voltage required for sinusoidal pulse width modulation vmodα 、 β-axis component of the reference voltage v modβ The formula for is:

[0060]

[0061]

[0062] Wherein, i Lα is the α-axis component of the inductor current, i Lβ is the β-axis component of the inductor current, i Lrefα is the α-axis component of the reference value of the inductor current, i Lrefβ is the β-axis component of the reference value of the inductor current;

[0063] Obtain the reference voltage required for the transformed sinusoidal pulse width modulation v mod abc The formula for is:

[0064]

[0065] Wherein, v moda is the a-axis component of the reference voltage, v modb is the b-axis component of the reference voltage, v modc is the c-axis component of the reference voltage, v modα is the α-axis component of the reference voltage, v modβ is the β-axis component of the reference voltage.

[0066] Optionally, in the step S1, calculate the actual active power p and the actual reactive power q The formula for is:

[0067]

[0068]

[0069] Where ,voα is the α-axis component of the capacitor voltage, voβ is the β-axis component of the capacitor voltage, igα is the α-axis component of the grid current, igβ is the β-axis component of the grid current.

[0070] A dual - power superposition synchronization and feed - forward passive reshaping control system for a network - forming inverter, which executes a dual - power superposition synchronization and feed - forward passive reshaping control method for a network - forming inverter as described in any one of the above, includes: a power synchronization loop and an output voltage loop, and the power synchronization loop and the output voltage loop are connected in sequence.

[0071] Optionally, the power synchronization loop includes: an instantaneous power calculation module, an active - power control branch, a dual - power superposition branch, a reactive - power control branch, and a voltage reference calculation module;

[0072] The first end of the instantaneous power calculation module is respectively connected to the first ends of the active - power control branch and the reactive - power control branch. The second ends of the active - power control branch and the reactive - power control branch are respectively connected to the first end of the voltage reference calculation module. The second end of the voltage reference calculation module is connected to the first end of the output voltage loop. The third end of the reactive - power control branch is connected to the first end of the dual - power superposition branch, and the second end of the dual - power superposition branch is connected to the third end of the active - power control branch.

[0073] Optionally, the output voltage loop includes: a current feed - forward branch;

[0074] The second end of the voltage reference calculation module passes through a voltage controller G v and then is connected to the first end of the current feed - forward branch. The second end of the current feed - forward branch is respectively connected to a current controller G i .

[0075] The dual - power superposition synchronization and feed - forward passive reshaping control method and system for a network - forming inverter provided by this application can achieve precise and independent control of the actual active power and the actual reactive power under various working conditions through the dual - power superposition of the actual active power and the actual reactive power, avoid the synchronous frequency resonance phenomenon of the network - forming inverter, and improve the synchronous stability and anti - disturbance ability of the inverter. By adopting a control method of coordinated regulation of feed - forward and feedback, it can effectively reduce the risk of broadband oscillation and ensure that the network - forming inverter maintains high - efficiency and stable grid - connection performance under dynamic load changes and disturbances.

[0076] To make the above - mentioned features and advantages of the application more obvious and understandable, specific embodiments are given below and are described in detail in conjunction with the accompanying drawings as follows. Brief Description of the Drawings

[0077] Figure 1 It is a flowchart of the dual - power superposition synchronization and feed - forward passive reshaping control method for a network - forming inverter provided by this application.

[0078] Figure 2This is the control block diagram of the dual-power superposition synchronization and feed-forward passive reshaping control system for the network-forming inverter proposed in this application.

[0079] Figure 3 This is the schematic diagram of the dual-power superposition of the power synchronization loop under the method provided in this application.

[0080] Figure 4 This is the simulation waveform diagram of the network-forming inverter using the traditional method.

[0081] Figure 5 This is the simulation waveform diagram of the network-forming inverter using the method provided in this application.

[0082] Explanation of reference numerals:

[0083] DC power supply 21, inverter bridge 22, LC filter 23, common power grid 24, dual-power superposition synchronization and feed-forward passive reshaping control system 25 of the network-forming inverter;

[0084] Power synchronization loop 251, output voltage loop 252;

[0085] Instantaneous power calculation module 2511, active power control branch 2512, dual-power superposition branch 2513, reactive power control branch 2514, voltage reference calculation module 2515, current feed-forward branch 2521. Detailed implementation manners

[0086] To make the objectives and technical solutions of the embodiments of this application clearer, the technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings of the embodiments of this application. Obviously, the described embodiments are some but not all of the embodiments of this application. All other embodiments obtained by those of ordinary skill in the art based on the described embodiments of this application without creative efforts shall fall within the scope of protection of this application.

[0087] In one embodiment of this application, please refer to Figure 1 , Figure 1 This is the flowchart of the dual-power superposition synchronization and feed-forward passive reshaping control method for the network-forming inverter provided in this application. The dual-power superposition synchronization and feed-forward passive reshaping control method for the network-forming inverter provided in this application includes the following steps:

[0088] Step S1: Measure the capacitor voltage v o , grid current i g , and calculate the actual active power p and actual reactive power q using the instantaneous power calculation theory after abc / αβ transformation;

[0089] Step S2: Based on the reactive power reference Q set and the actual reactive power q , calculate the error signal e Q , and after being processed by the proportional link K Q , feed it forward to the signal output by the active power controller G HPF through a high-pass filter G APC ;

[0090] Step S3: Based on the active power reference P set and the actual active power p and the error signal e Q , calculate the reference angle θ required for Clarke transformation or inverse Clarke transformation;

[0091] Step S4: Based on the reactive power reference Q set and the actual reactive power q , calculate the voltage amplitude G PRC through the reactive power controller V , combine it with the reference angle θ required for Clarke transformation or inverse Clarke transformation, and obtain the α-axis component v orefα of the voltage reference value, v orefβ the β-axis component

[0092] of the voltage reference value through voltage reference calculation; θ Step S5: According to the reference angle i g required for Clarke transformation or inverse Clarke transformation, transform the grid current v o , the capacitor voltage i L and the inductor current i gα through abc / αβ transformation to obtain the α-axis component i gβ of the transformed grid current, v oα the β-axis component v oβ of the grid current, i Lα the α-axis componenti Lβ , the α-axis component of the grid current i gα , and the β-axis component of the grid current i gβ are processed by a grid current feedforward controller G f to obtain a current feedforward signal; based on the α-axis component of the voltage reference value v orefα , the β-axis component of the voltage reference value v orefβ , the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ , a voltage controller G V , and the current feedforward signal, calculate the α-axis component of the current reference value i Lrefα 、 and the β-axis component of the current reference value i Lrefβ ;

[0093] Step S6: Based on the α-axis component of the inductor current i Lα , the β-axis component of the inductor current i Lβ and the α-axis component of the current reference value i Lrefα 、 and the β-axis component of the current reference value i Lrefβ , through a current controller G i , obtain the α-axis component of the reference voltage required for sinusoidal pulse width modulation v modα 、 and the β-axis component of the reference voltage v modβ , after αβ / abc transformation, obtain the reference voltage required for the transformed sinusoidal pulse width modulation v mod abc , and generate a three-phase inverter control signal through sinusoidal pulse width modulation.

[0094] Specifically, in the step S1, the formulas for calculating the actual active power p and the actual reactive power q are:

[0095]

[0096]

[0097] wherein ,voα is the α-axis component of the capacitor voltage, voβ is the β-axis component of the capacitor voltage, igα is the α-axis component of the grid current, igβ is the β-axis component of the grid current.

[0098] Specifically, in the step S2, the calculation formula of the error signal e Q is:

[0099]

[0100] wherein, Q set is the reactive power reference, q is the actual reactive power.

[0101] The expression of the proportional link K Q is:

[0102]

[0103] wherein, Δθ max is the maximum angle deviation allowed by the network-forming inverter, Q max is the maximum reactive power deviation allowed by the network-forming inverter, is the proportional coefficient.

[0104] The transfer function of the high-pass filter G HPF is:

[0105]

[0106] wherein, ω c is the cut-off angular frequency of the high-pass filter G HPF , ζ is the damping ratio, s is the Laplace transform variable.

[0107] The expression of the low-pass filtering link of the active power controller G APC is:

[0108]

[0109] wherein, ω p is the cut-off angular frequency of the low-pass filter in the active power error path, mp is the active power droop coefficient, s is the Laplace transform variable.

[0110] Active power droop coefficient m p has the following expression:

[0111]

[0112] wherein, Δω max is the maximum allowable frequency deviation of the network-forming inverter, P max is the maximum allowable active power deviation of the network-forming inverter under specific working conditions, k 1 is the proportionality coefficient.

[0113] Specifically, in the step S3, the reference angle θ for Clark transformation or inverse Clark transformation is calculated by the formula:

[0114]

[0115]

[0116] wherein, Δω is the angular frequency adjustment amount of the network-forming inverter, ω n is the reference angular frequency of the network-forming inverter, ω p is the cut-off angular frequency of the low-pass filter in the active power error path, m p is the active power droop coefficient, P set is the active power reference, p is the actual active power, s is the Laplace transform variable.

[0117] Specifically, in the step S4, the low-pass filtering link of the reactive power controller G RPC has the following expression:

[0118]

[0119] wherein, ω q is the cut-off frequency of the low-pass filter in the reactive power error path, n q is the reactive power droop coefficient, s is the Laplace transform variable.

[0120] Reactive power droop coefficient n q The expression is as follows:

[0121]

[0122] Wherein, ΔV max is the maximum voltage amplitude adjustment amount allowed for the grid-forming inverter, Q max is the maximum reactive power deviation allowed for the grid-forming inverter, k 2 is the proportionality coefficient.

[0123] Calculate the voltage amplitude V The formula is:

[0124]

[0125]

[0126] Wherein, n q is the reactive power droop coefficient, V n is the reference voltage of the grid-forming inverter, ω q is the cut-off frequency of the low-pass filter in the reactive power error path, ΔV is the voltage amplitude V The adjustment amount.

[0127] Calculate the α-axis component of the voltage reference value v orefα , the β-axis component of the voltage reference value v orefβ The formula is:

[0128]

[0129]

[0130] Wherein, v orefα and v orefβ are the voltage reference values, θ is the reference angle for Clarke transformation or Clarke inverse transformation.

[0131] Specifically, in the step S5, the grid current feed-forward controller G f The formula is:

[0132] Wherein, K pfis the proportional gain of the grid current feedforward controller, K df is the derivative gain of the grid current feedforward controller, is the time constant of the derivative filter, s is the Laplace transform variable.

[0133] Voltage controller G v The formula is:

[0134]

[0135] Wherein, K pv is the proportional coefficient of the voltage controller, K r is the resonance gain coefficient, ω r is the bandwidth coefficient, ω 0 is the resonance center frequency, s is the Laplace transform variable.

[0136] Calculate the α-axis component of the current reference value i Lrefα 、 β-axis component of the current reference value i Lrefβ The formula is:

[0137]

[0138]

[0139] Wherein, i Lrefα is the α-axis component of the inductor current reference value, i Lrefβ is the β-axis component of the inductor current reference value, v orefα is the α-axis component of the voltage reference value, v orefβ is the β-axis component of the voltage reference value, v oα is the α-axis component of the capacitor voltage, v oβ is the β-axis component of the capacitor voltage.

[0140] Specifically, in the step S6, the current controller G i The formula is:

[0141]

[0142] Among them, K pi is the proportional coefficient of the current controller.

[0143] Calculate the α-axis component of the reference voltage required for sinusoidal pulse width modulation v modα 、 The β-axis component of the reference voltage v modβ The formula is:

[0144]

[0145]

[0146] Among them, i Lα is the α-axis component of the inductor current, i Lβ is the β-axis component of the inductor current, i Lrefα is the α-axis component of the reference value of the inductor current, i Lrefβ is the β-axis component of the reference value of the inductor current.

[0147] Obtain the reference voltage required for the transformed sinusoidal pulse width modulation v mod abc The formula is:

[0148]

[0149] Among them, v moda is the a-axis component of the reference voltage, v modb is the b-axis component of the reference voltage, v modc is the c-axis component of the reference voltage, v modα is the α-axis component of the reference voltage, v modβ is the β-axis component of the reference voltage.

[0150] In another embodiment of the present application, please refer to Figure 2 , Figure 2 is the control block diagram of the dual-power superposition synchronization and feed-forward passive reshaping control system of the network-forming inverter proposed in the present application. The dual-power superposition synchronization and feed-forward passive reshaping control system 25 of the network-forming inverter includes: a power synchronization loop 251 and an output voltage loop 252, and the power synchronization loop 251 and the output voltage loop 252 are connected in sequence.

[0151] As an example, the power synchronization loop 251 includes: an instantaneous power calculation module 2511, an active power control branch 2512, a dual power superposition branch 2513, a reactive power control branch 2514, and a voltage reference calculation module 2515. The first end of the instantaneous power calculation module 2511 is respectively connected to the first end of the active power control branch 2512 and the first end of the reactive power control branch 2514. The second ends of the active power control branch 2512 and the reactive power control branch 2514 are respectively connected to the first end of the voltage reference calculation module 2515. The second end of the voltage reference calculation module 2515 is connected to the first end of the output voltage loop 252. The third end of the reactive power control branch 2514 is connected to the first end of the dual power superposition branch 2513. The second end of the dual power superposition branch 2513 is connected to the third end of the active power control branch 2512.

[0152] The output voltage loop 252 includes: a current feedforward branch 2521. The second end of the voltage reference calculation module 2515 is respectively connected to the first end of the current feedforward branch 2521 after passing through a voltage controller G v and then. The second end of the current feedforward branch 2521 is respectively connected to a current controller G i .

[0153] Please continue to refer to Figure 2 , the dual power superposition synchronization and feedforward passive reshaping control system of the grid-forming inverter proposed in this application is used for the grid-forming inverter. The grid-forming inverter includes: a DC power supply 21, an inverter bridge 22, an LC filter 23, and a common power grid 24. The DC power supply 21, the inverter bridge 22, the LC filter 23, and the common power grid 24 are connected in sequence. Among them, the inverter bridge 22 includes switching tubes and their corresponding freewheeling diodes; the LC filter 23 includes a filter inductor L and a filter capacitor C. The first end of the filter inductor L is connected to the first end of the inverter bridge 22. The second end of the filter inductor L is connected to the first end of the filter capacitor C. The second end of the filter capacitor C is grounded; the common power grid 24 includes a three-phase voltage source v g , an equivalent grid impedance z g and a transformer K T . The first end of the transformer K T is connected to the first end of the filter capacitor C. The second end of the transformer K T is connected to the first end of the equivalent grid impedance z g . The second end of the equivalent grid impedance z g is connected to the three-phase voltage source vg is connected to the first end of the three-phase voltage source v g The second end is grounded. The voltage across the DC power supply 21 is V dc .

[0154] As an example, the power synchronization loop 251 is used to feed the reactive power reference Q set , through the proportional link K Q and the high-pass filter G HPF , and feed it forward to the active power controller G APC .

[0155] Specifically, the instantaneous power calculation module 2511 is used to utilize the capacitor voltage v o and the grid current i g to calculate the actual active power of the grid-forming inverter p and the actual reactive power q .

[0156] The reactive power control branch 2514 is used to utilize the reactive power reference Q set and the actual reactive power q to obtain the error signal e Q . And based on the reactive power reference Q set and the actual reactive power q , through the reactive power controller G PRC calculate the voltage amplitude V .

[0157] The double power superposition branch 2513 is used to process the error signal e Q through the proportional link K Q , and then through the high-pass filter G HPF process it, and feed it forward to the active power controller G APC of the active power control branch 2512 to achieve fast dynamic compensation

[0158] The active power control branch 2512 is used to calculate the reference angle for the Clarke transform or the inverse Clarke transform based on the active power reference P set , the actual active power p and the error signal e Q ​θ 。

[0159] The voltage reference calculation module 2515 is used to calculate the voltage reference value based on the reference angle for Clarke transformation or inverse Clarke transformation θ and the voltage amplitude V , and send it to the output voltage loop 252 v orefα 、v orefβ .

[0160] As an example, the output voltage loop 252 is used to perform abc / αβ transformation on the grid current i g , capacitor voltage v o and inductor current i L to obtain the α-axis component of the transformed grid current i gα , the β-axis component of the grid current i gβ , the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ and the α-axis component of the inductor current i Lα , the β-axis component of the inductor current i Lβ ; detect the α-axis component of the voltage reference value v orefα , the β-axis component of the voltage reference value v orefβ , the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ , subtract the α-axis component of the voltage reference value v orefα , the β-axis component of the voltage reference value v orefβ from the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ , and after passing through the voltage controller G V , perform current feedforward in the current feedforward branch 2521 to obtain the α-axis component of the current reference value i Lrefα 、 the β-axis component of the current reference value i Lrefβ . The α-axis component of the current reference valuei Lrefα 、 β-axis component of the current reference value i Lrefβ and the α-axis component of the inductor current i Lα and the β-axis component of the inductor current i Lβ are subtracted, and through the current controller G i to obtain the α-axis component of the reference voltage required for sinusoidal pulse width modulation v modα 、 β-axis component of the reference voltage v modβ After αβ / abc transformation, the reference voltage required for the transformed sinusoidal pulse width modulation is obtained v mod abc Through sinusoidal pulse width modulation, the sinusoidal pulse width control signals of the switching tubes of the inverter bridge 22 are generated. Finally, the control signals after sinusoidal pulse width modulation are input into the inverter bridge 22 to control the switching tubes of the inverter bridge 22.

[0161] Specifically, the current feedforward branch 2521 is used for the α-axis component of the grid current i gα and the β-axis component of the grid current i gβ are processed through the grid current feedforward controller G f to obtain the current feedforward signal; the voltage reference value v oref is subtracted from the capacitor voltage v o and then subtracted from the current feedforward signal through the voltage controller G V to obtain the α-axis component of the current reference value i Lrefα 、 β-axis component of the current reference value i Lrefβ .

[0162] As an example, please refer to Figure 3 , Figure 3 which is a schematic diagram of double power superposition of the power synchronization loop under the method provided by this application. Among them, in the active power control branch 2512, the active power controller G APC includes a low-pass filtering link m p ω p / s+ ωp , in the reactive power control branch 2514, the reactive power controller G RPC includes a low-pass filtering section n q ω q / s + ω q ; in the conventional droop control with a low-pass filter (LPF), in the active power control branch 2512 and the reactive power control branch 2514, by feeding the reactive power error signal e Q forward to the dynamic decoupling mechanism of the active power control branch 2512 through the double power superposition branch 2513, the coordinated regulation of active and reactive power is achieved.

[0163] As an example, please refer to Figure 4 and Figure 5 , Figure 4 is the simulation waveform diagram of the grid-forming inverter using the traditional method, Figure 5 is the simulation waveform diagram of the grid-forming inverter using the method provided in this application. Through the MATLAB / Simulink software, the operation of the grid-forming inverter is simulated and verified by the double power superposition synchronization and feedforward passive reshaping control method of the grid-forming inverter. Under specific grid conditions, such as when the grid impedance is purely inductive and the equivalent short-circuit capacity ratio is 30, the traditional control method and the control method proposed in this application are respectively used to verify the effectiveness of the control method proposed in this application. The simulation results are as shown in Figure 4 and Figure 5 . It can be seen from the figure that under this grid condition, there are obvious oscillations in the grid-forming inverter using the traditional control method, while for the grid-forming inverter using the control method proposed in this application, both the output voltage and the output current of the grid-forming inverter present sinusoidal stable waveforms. The experimental results show that the double power superposition synchronization and feedforward passive reshaping control method and system of the grid-forming inverter provided in this application can, through the double power superposition of the actual active power and the actual reactive power, achieve precise and independent control of the actual active power and the actual reactive power under various conditions, avoid the synchronous frequency resonance phenomenon of the grid-forming inverter, and improve the synchronous stability and anti-disturbance ability of the inverter. By adopting the control method of coordinated regulation of feedforward and feedback, the risk of broadband oscillation can be effectively reduced, ensuring that under dynamic load changes and disturbances, the grid-forming inverter maintains high-efficiency and stable grid-connected performance, enabling the grid-forming inverter to operate reliably under complex grid conditions and realizing the plug-and-play function of the grid-forming inverter.

[0164] Although the present application has been disclosed above by way of examples, it is not intended to limit the present application. Any person with ordinary knowledge in the relevant technical field may make some modifications and refinements without departing from the spirit and scope of the present application. Therefore, the protection scope of the present application shall be subject to that defined by the appended claims for patent application.

Claims

1. A dual-power superposition synchronization and feedforward passive reshaping control method for a network-forming inverter, characterized in that including, Step S1: Measure the capacitor voltage v o , the grid current i g , after abc / αβ transformation, use the instantaneous power calculation theory to calculate the actual active power p and the actual reactive power q ; Step S2: Based on the reactive power reference Q set and the actual reactive power q , calculate the error signal e Q , and after processing through the proportional link K Q , feed it forward to the signal output by the active power controller G HPF via a high-pass filter G APC ; Step S3: Based on the active power reference P set , the actual active power p and the error signal after proportional-integral and high-pass filtering processing e Q , calculate the reference angle required for Clarke transformation or inverse Clarke transformation θ ; Step S4: Based on the reactive power reference Q set and the actual reactive power q , calculate the voltage amplitude G PRC through the reactive power controller V , combine with the reference angle required for Clarke transformation or inverse Clarke transformation θ , and obtain the α-axis component v orefα of the voltage reference value and the β-axis component v orefβ ; Step S5: According to the reference angle required for Clarke transformation or inverse Clarke transformation θ , transform the grid current i g , capacitor voltage v o and inductor current i L through abc / αβ transformation to obtain the α-axis component of the transformed grid current i gα , the β-axis component of the grid current i gβ , the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ , the α-axis component of the inductor current i Lα and the β-axis component of the inductor current i Lβ . The α-axis component of the grid current i gα and the β-axis component of the grid current i gβ are processed by the grid current feedforward controller G f to obtain the current feedforward signal; Based on the α-axis component of the voltage reference value v orefα , the β-axis component of the voltage reference value v orefβ , the α-axis component of the capacitor voltage v oα , the β-axis component of the capacitor voltage v oβ , the voltage controller G V and the current feedforward signal, calculate to obtain the α-axis component of the current reference value i Lrefα 、 and the β-axis component of the current reference value i Lrefβ ; Step S6: Based on the α-axis component of the inductor current i Lα , the β-axis component of the inductor current i Lβ and the α-axis component of the current reference value i Lrefα 、 the β-axis component of the current reference value i Lrefβ , through the current controller G i , obtain the α-axis component of the reference voltage required for sinusoidal pulse width modulation v modα 、 the β-axis component of the reference voltage v modβ , after αβ / abc transformation, obtain the transformed reference voltage v required for sinusoidal pulse width modulation mod abc , and generate the three-phase inverter control signal through sinusoidal pulse width modulation.

2. The dual-power superposition synchronization and feedforward passive reshaping control method for the network-forming inverter according to claim 1, wherein In the step S2, the error signal is calculated e Q using the formula: Among them, Q set is the reactive power reference, q is the actual reactive power; Proportional link K Q The expression is: Among them, Δθ max is the maximum angle deviation allowed for the network-forming inverter, Q max is the maximum reactive power deviation allowed for the network-forming inverter, k q is the proportionality coefficient; High-pass filter G HPF The transfer function of is as follows: Among them, ω c is a high-pass filter G HPF is the cut-off angular frequency, ζ is the damping ratio, s is the Laplace transform variable; Active power controller G APC The expression of the low-pass filtering section is as follows: Among them, ω p is the cut-off angular frequency of the low-pass filter in the active power error path, m p is the active power droop coefficient, s is the Laplace transform variable; Active power droop coefficient m p The expression is as follows: Among them, Δω max is the maximum frequency deviation allowed for the grid-forming inverter, P max is the maximum active power deviation allowed for the grid-forming inverter under specific operating conditions, k 1 is the proportionality coefficient.

3. The dual-power superposition synchronization and feedforward passive reshaping control method for the network-forming inverter according to claim 2, characterized in that In the step S3, the reference angle for Clarke transformation or inverse Clarke transformation is calculated θ The formula is as follows: Among them, Δω is the angular frequency adjustment amount of the network-forming inverter, ω n is the reference angular frequency of the network-forming inverter, ω p is the cut-off angular frequency of the low-pass filter in the active power error path, m p is the active power droop coefficient, P set is the active power reference, p is the actual active power, s is the Laplace transform variable.

4. The dual-power superposition synchronization and feedforward passive reshaping control method for the network-forming inverter according to claim 1, characterized in that In the step S4, the reactive power controller G RPC The expression of the low-pass filtering link is: Among them, ω q is the cut-off frequency of the low-pass filter in the reactive power error path, n q is the reactive power droop coefficient, s is the Laplace transform variable; Reactive power droop coefficient n q The expression is as follows: Among them, ΔV max is the maximum voltage amplitude adjustment amount allowed for the grid-forming inverter, Q max is the maximum reactive power deviation allowed for the grid-forming inverter, k 2 is the proportionality coefficient; Calculate the voltage amplitude V The formula is as follows: Among them, n q is the reactive power droop coefficient, V n is the reference voltage of the network-forming inverter, ω q is the cut-off frequency of the low-pass filter in the reactive power error path, ΔV is the voltage amplitude V adjustment amount; Calculate the α-axis component of the voltage reference value v orefα , the β-axis component of the voltage reference value v orefβ The formula is as follows: Wherein, v orefα and v orefβ are voltage reference values, θ is the reference angle for Clarke transformation or inverse Clarke transformation.

5. The double-power superposition synchronization and feed-forward passive reshaping control method of the network-forming inverter according to claim 4, characterized in that, In the step S5, the grid current feedforward controller G f has the formula as follows: Among them, K pf is the proportional gain of the grid current feedforward controller, K df is the derivative gain of the grid current feedforward controller, is the time constant of the derivative filter, s is the Laplace transform variable; Voltage controller G v The formula is: Among them, K pv is the proportional coefficient of the voltage controller, K r is the resonant gain coefficient, ω r is the bandwidth coefficient, ω 0 is the resonant center frequency, s is the Laplace transform variable; α-axis component of the calculated current reference value i Lrefα 、 β-axis component of the current reference value i Lrefβ The formula for: wherein, i Lrefα is the α-axis component of the inductor current reference value, i Lrefβ is the β-axis component of the inductor current reference value, v orefα is the α-axis component of the voltage reference value, v orefβ is the β-axis component of the voltage reference value, v oα is the α-axis component of the capacitor voltage, v oβ is the β-axis component of the capacitor voltage.

6. The dual-power superposition synchronization and feedforward passive reshaping control method for the network-forming inverter according to claim 1, characterized in that, In the step S6, the current controller G i has the formula as follows: Among them, K pi is the proportional coefficient of the current controller; Calculate the α-axis component of the reference voltage required for sinusoidal pulse width modulation v modα 、 β-axis component of the reference voltage v modβ The formula for is: Among them, i Lα is the α-axis component of the inductor current, i Lβ is the β-axis component of the inductor current, i Lrefα is the α-axis component of the reference value of the inductor current, i Lrefβ is the β-axis component of the reference value of the inductor current; Obtain the reference voltage required for the transformed sinusoidal pulse width modulation v mod abc The formula is as follows: wherein, v moda is the a-axis component of the reference voltage, v modb is the b-axis component of the reference voltage, v modc is the c-axis component of the reference voltage, v modα is the α-axis component of the reference voltage, v modβ is the β-axis component of the reference voltage.

7. The dual-power superposition synchronization and feedforward passive reshaping control method for the network-forming inverter according to claim 1, characterized in that In the step S1, the actual active power is calculated p and the actual reactive power q are calculated according to the following formula: wherein , voα is the α-axis component of the capacitor voltage, voβ is the β-axis component of the capacitor voltage, igα is the α-axis component of the grid current, igβ is the β-axis component of the grid current.

8. A dual-power superposition synchronization and feedforward passive reshaping control system for a grid-forming inverter, which executes a dual-power superposition synchronization and feedforward passive reshaping control method for a grid-forming inverter as described in any one of claims 1-7, characterized in that, including: a power synchronization loop and an output voltage loop, the power synchronization loop and the output voltage loop are connected in sequence.

9. The dual-power superposition synchronization and feedforward passive reshaping control system of the network-forming inverter according to claim 8, characterized in that the power synchronization loop includes: an instantaneous power calculation module, an active power control branch, a dual-power superposition branch, a reactive power control branch, and a voltage reference calculation module; the first end of the instantaneous power calculation module is respectively connected to the first end of the active power control branch and the first end of the reactive power control branch, the second ends of the active power control branch and the reactive power control branch are respectively connected to the first end of the voltage reference calculation module, the second end of the voltage reference calculation module is connected to the first end of the output voltage loop, the third end of the reactive power control branch is connected to the first end of the dual-power superposition branch, and the second end of the dual-power superposition branch is connected to the third end of the active power control branch.

10. The dual-power superposition synchronization and feedforward passive reshaping control system of the network-forming inverter according to claim 9, characterized in that the output voltage loop includes: a current feedforward branch; The second end of the voltage reference calculation module passes through a voltage controller respectively G v and then is connected to the first end of the current feedforward branch, and the second end of the current feedforward branch is connected to a current controller respectively G i .

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