Multi-section constant-current constant-voltage wireless power transmission system based on quasi-Z-source inverter

Abstract

The invention discloses a quasi-Z-source inverter-based multi-section constant-current constant-voltage wireless power transmission system, which comprises a direct-current power supply, a quasi-Z-source inverter, a PWM (Pulse Width Modulation) modulation control circuit, a transmitting coil, a receiving coil, a resonant network, a synchronous rectifier and a load, and is characterized in that the direct-current power supply, the quasi-Z-source inverter, the primary side of the resonant network and the transmitting coil are connected in series; the receiving coil, the secondary side of the resonance network, the synchronous rectifier and the load are connected in series; and the PWM modulation control circuit is used for changing the boosting capability of the quasi-Z-source inverter and controlling the synchronous rectifier to realize multi-section constant-current and constant-voltage output of a system load side. Multi-section constant-current constant-voltage output is achieved, boosting of primary side power input, boosting of secondary side current and multi-section constant-current constant-voltage output of a load side are achieved, meanwhile, the system charging speed is increased, the service life of a lithium battery is prolonged, and the multi-section constant-current constant-voltage power supply is suitable for application occasions such as electric automobiles.

Description

Technical Field

[0001] The present invention relates to the field of wireless power transmission and the field of power converters, and particularly relates to a multi-segment constant current and constant voltage wireless power transmission system based on a quasi-Z-source inverter. Background Art

[0002] Wireless Power Transfer (WPT) technology uses a charging pile and a wireless transmitting device buried underground. Non-contact charging avoids manual operation of wired charging and eliminates the need for wires and interfaces, thus eliminating the risk of electric shock and facilitating automated charging, which conforms to the future development trend.

[0003] Currently, a large number of studies have been conducted on traditional WPT systems at home and abroad. However, with the wide application of WPT technology, many high-power application scenarios such as electric vehicles have emerged. Conditions such as charging speed have become factors restricting its further development. Concerns about the service life of lithium batteries are also a major reason affecting people's choice between electric vehicles and traditional fuel vehicles. At the same time, in order to meet the needs of different power levels, a DC chopper circuit link often needs to be added to the entire system to achieve functions such as boosting, which also makes the circuit more complex and increases the system cost. Summary of the Invention

[0004] Object of the Invention: To overcome the defects of the above-mentioned traditional WPT system, the present invention proposes a multi-segment constant current and constant voltage wireless power transmission system based on a quasi-Z-source inverter. This system can boost the DC input voltage of the system without adding a primary-side DC boost chopper circuit. At the same time, no dead time needs to be set, making the system operation safer. The synchronous rectification method adopted on the secondary side can be used to control the multi-segment constant current and constant voltage output of the system load side.

[0005] Technical Solution: To achieve the above object, the technical solution adopted by the present invention is as follows:

[0006] A multi-segment constant current and constant voltage wireless power transmission system based on a quasi-Z-source inverter includes a DC power supply, a quasi-Z-source inverter, a PWM modulation control circuit, a transmitting coil, a receiving coil, a resonant network, a synchronous rectifier, and a load. The DC power supply, the quasi-Z-source inverter, the primary side of the resonant network, and the transmitting coil are connected in series. The receiving coil, the secondary side of the resonant network, the synchronous rectifier, and the load are connected in series. The PWM modulation control circuit is used to change the boosting ability of the quasi-Z-source inverter and to control the synchronous rectifier to achieve multi-segment constant current and constant voltage output on the system load side.

[0007] Preferably: The quasi-Z-source inverter includes diode D1, inductor L z1 and inductor L z2, Capacitor 1 C z1 , Capacitor 2 C z2 , Switching transistor 1 S 1 , Switching transistor 2 S 2 , Switching transistor 3 S 3 , Switching transistor 4 S 4 , The DC power supply, Inductor 1 L z1 , Diode 1 D1, Inductor 2 L z2 , Switching transistor 1 S 1 , Switching transistor 2 S 2 are connected in series, and the Capacitor 1 C z1 is connected in parallel across both ends of Diode 1 D1 and Inductor 2 L z2 , and one end of the Capacitor 2 C z2 is connected to the connection line between Diode 1 D1 and Inductor 2 L z2 , and the other end is connected to the negative pole of the DC power supply; the Switching transistor 3 S 3 and the Switching transistor 4 S 4 are connected in series and then connected in parallel across both ends of the Switching transistor 1 S 1 and the Switching transistor 2 S 2 .

[0008] Preferably: The primary side of the resonant network includes a first compensation inductor L 1 , a first compensation capacitor 1 C 1 , a first compensation capacitor 2 C p , one end of the first compensation inductor L 1 , the first compensation capacitor 2 C p , and one end of the transmitting coil are connected together, the other end of the first compensation inductor L 1 is connected to the connection line between the Switching transistor 1 S 1 and the Switching transistor 2 S 2 , the other end of the transmitting coil is connected to the connection line between the Switching transistor 3 S 3 and the Switching transistor 4 S 4 .

[0009] Preferably: The synchronous rectifier includes a third inductor 1 L t1 , a third inductor 2 L t2 , a switching transistor 5 S 5 , a switching transistor 6 S 6 , a third capacitor C 0 , the third inductor 1 L t1 , the load, and the switching transistor 5 S 5 are connected in series, the third inductor 2 L t2 , and the switching transistor 6 S 6 are connected in series and then connected in parallel across both ends of the load, and the third capacitor C 0 is connected in parallel across both ends of the load.

[0010] Preferably, the primary side of the resonant network includes a second compensation inductor L 2 , a first second compensation capacitor C 2 , a second second compensation capacitor C s . One end of the second compensation inductor L 2 , one end of the second second compensation capacitor C s , and one end of the receiving coil are connected together. The other end of the second compensation inductor L 2 is connected to the connection line between the first third inductor L t1 and the fifth switching transistor S 5 . The other end of the receiving coil is connected to the connection line between the second third inductor L t2 and the sixth switching transistor S 6 .

[0011] Preferably, the system input power is:

[0012]

[0013] Wherein, is the system input power, U inv is the high-frequency AC voltage output by the system through the inverter. M represents the mutual inductance between the primary coil and the secondary coil. R eq is the equivalent AC load, ω 0 represents the corresponding resonance frequency, L 1 is the first compensation inductor, L 2 is the second compensation inductor.

[0014] Preferably, the system output power is:

[0015]

[0016] Wherein, is the system output power, k represents the coupling coefficient, L p is the self-inductance of the transmitting coil, L s is the self-inductance of the receiving coil, U inv is the high-frequency AC voltage output by the system through the inverter, ω 0 represents the corresponding resonance frequency, L 1 is the first compensation inductor, L 2 is the second compensation inductor, R eq is the equivalent AC load.

[0017] Preferably, the primary input impedance Z in and the secondary equivalent impedance Z s are expressed as follows:

[0018]

[0019]

[0020] Among them, Z in represents the primary-side input impedance, and Z s represents the secondary-side equivalent impedance. / / represents impedance parallel connection, represents the phase difference in an AC circuit, represents the angular frequency, and L 1 is the first compensation inductor, and L p is the self-inductance of the transmitting coil, and C p represents the second first compensation capacitor, represents the secondary-side reflected impedance, and C 1 represents the first first compensation capacitor, and L s is the self-inductance of the receiving coil, and C s represents the second second compensation capacitor, and C 2 represents the second first compensation capacitor, and L 2 represents the second compensation inductor, and R eq is the equivalent AC load.

[0021] Preferably, when both the primary and secondary resonant networks of the system are in resonance, the system satisfies the following relationship:

[0022]

[0023] Among them, ω 0 represents the corresponding resonant frequency, and L 1 is the first compensation inductor, and C 1 represents the first first compensation capacitor, and L 2 represents the second compensation inductor, and C 2 represents the second first compensation capacitor, and L p is the self-inductance of the transmitting coil, and C p represents the second first compensation capacitor, and L s is the self-inductance of the receiving coil, and C s represents the second second compensation capacitor.

[0024] Preferably, the multi-stage constant current and constant voltage charging strategy is optimized by improving the particle swarm optimization algorithm. The method for optimizing the multi-stage constant current and constant voltage charging strategy by improving the particle swarm optimization algorithm is as follows:

[0025] Step 1: Introduce an adaptive PID controller, and comprehensively consider the overshoot, settling time, and steady-state error of the system for optimization design. The global optimal solution Pg is obtained with the maximum fitness function value as the optimization goal. The expression of the objective fitness function is as follows:

[0026]

[0027] Among them, represents the fitness function value, It represents the steady-state error of the system, T represents the system stabilization time, ess represents the system steady-state error, a is the weight coefficient of the system steady-state error, b is the weight coefficient of the system stabilization time, and c is the weight coefficient of the system steady-state error;

[0028] Step 2, the limiting expression for the particle velocity is as follows:

[0029]

[0030]

[0031]

[0032] Among them, V n represents the search velocity of the particle at the nth charging switching point, V n，max represents the upper limit of the search velocity, V n，min represents the lower limit of the search velocity, represents the variance of the fitness function, is the introduced initial threshold, represents the mean value of the system fitness function, represents the number of particles in the particle population, represents the value of the fitness function, represents the search velocity of the particle at the 1st charging switching point, represents the search velocity of the particle at the 2nd charging switching point, represents the search velocity of the particle at the 3rd charging switching point;

[0033] Step 3, the dynamic inertia weight is as follows:

[0034]

[0035] Among them, represents the dynamic inertia weight, ω max represents the maximum limit of the inertia factor, ω min represents the minimum limit of the inertia factor, represents the current maximum value of the fitness function, represents the current minimum value of the fitness function;

[0036] Step 4, the individual average optimal solution is as follows:

[0037]

[0038] Among them, represents the individual average optimal solution, N represents the number of particles in the particle population, represents the individual optimal solution;

[0039] Step 5, Gaussian perturbation is added to both the individual average optimal solution and the global optimal solution as follows:

[0040]

[0041]

[0042] Among them, represents the individual average optimal solution, represents the individual average optimal solution after updating by introducing Gaussian perturbation, is the perturbation parameter, is a proportionality coefficient set according to specific circumstances, and are the upper and lower limits of the perturbation parameter set according to actual requirements, represents the number of iterations, represents setting the upper limit of the number of iterations, represents the global optimal solution, represents the global optimal solution after updating by introducing Gaussian perturbation.

[0043] Compared with the prior art, the present invention has the following beneficial effects:

[0044] 1. The quasi-Z-source inverter used in the present invention has a lower requirement for the DC power supply voltage level, and can achieve several times the increase of the input DC voltage. Therefore, when applied to larger power occasions, there is no need to worry about the insufficient DC voltage level, and there is no need to add a boost chopper circuit and other links when boosting, making the circuit more simplified.

[0045] 2. The synchronous rectifier used in the present invention, compared with the traditional diode rectification, this technology is not affected by the Schottky barrier voltage and does not generate dead zone voltage, thereby reducing energy loss. This makes it more widely used in low-voltage and high-current applications. The added switching tubes also make it possible to achieve closed-loop control.

[0046] 3. The multi-stage constant current and constant voltage charging strategy adopted by the present invention has a higher charging rate and a longer service life for lithium batteries.

[0047] 4. The present invention optimizes the multi-stage constant current and constant voltage charging strategy by improving the particle swarm algorithm, making the charging strategy more reliable and practical. BRIEF DESCRIPTION OF THE DRAWINGS

[0048] Figure 1 is the overall structure of the multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z-source inverter involved in the embodiment;

[0049] Figure 2 is the equivalent circuit diagram of the primary side of the resonant network in the embodiment;

[0050] Figure 3 is the equivalent circuit diagram of the secondary side of the resonant network in the embodiment;

[0051] Figure 4 is the working schematic diagram of the synchronous rectifier circuit; among which Figure 4 in Figure (a) is the working schematic diagram of the positive half cycle, Figure 4 in Figure (b) is the working schematic diagram of the negative half cycle;

[0052] Figure 5 is the flowchart of calculating the switching condition of the constant current and constant voltage charging stage by the proposed improved particle swarm optimization algorithm;

[0053] Figure 6 is the comparison diagram of the DC-link output voltage and the DC power input voltage of the quasi-Z-source inverter in the simulation;

[0054] Figure 7 is the waveform diagram of the output voltage and current of the synchronous rectifier circuit. Specific embodiments

[0055] The following further clarifies the present invention in conjunction with the accompanying drawings and specific embodiments. It should be understood that these embodiments are only used to illustrate the present invention and not to limit the scope of the present invention. After reading the present invention, various equivalent forms of modification by those skilled in the art fall within the scope defined by the appended claims of this application.

[0056] A multi-stage constant current and constant voltage wireless power transmission system based on a quasi-Z-source inverter, as Figure 1 shown, includes a DC power supply, a quasi-Z-source inverter, a PWM modulation control circuit, a transmitting coil, a receiving coil, a resonant network, a synchronous rectifier, and a load. The DC power supply, the quasi-Z-source inverter, the primary side of the resonant network, and the transmitting coil are connected in series. The receiving coil, the secondary side of the resonant network, the synchronous rectifier, and the load are connected in series. The PWM modulation control circuit is used to change the boosting ability of the quasi-Z-source inverter and to control the synchronous rectifier to achieve multi-stage constant current and constant voltage output on the load side of the system.

[0057] The PWM modulation control circuit includes two sets. One set is used to control the direct-conduction duty ratio of the quasi-Z-source inverter on the primary side to adjust the boosting ability, and the other set realizes different outputs by detecting the voltage on the load side and according to the constant current and constant voltage control strategy optimized and designed by the particle swarm optimization algorithm.

[0058] The quasi-Z-source inverter is obtained by adding a series of passive devices to the ordinary full-bridge inverter. It can eliminate the dead time, which reduces the risk of abnormal operation of the system. According to the size of the direct-conduction duty ratio, it can also realize the boosting function without introducing an additional switching tube like the boost chopper circuit, which simplifies the system complexity and saves costs.

[0059] The quasi-Z-source inverter includes diode D1, inductor L z1 、inductor Lz2 、 Capacitor 1 C z1 、 Capacitor 2 C z2 、 Switching transistor 1 S 1 、 Switching transistor 2 S 2 、 Switching transistor 3 S 3 、 Switching transistor 4 S 4 , the DC power supply, Inductor 1 L z1 、 Diode 1 D1, Inductor 2 L z2 、 Switching transistor 1 S 1 、 Switching transistor 2 S 2 are connected in series, and the Capacitor 1 C z1 is connected in parallel across both ends of Diode 1 D1 and Inductor 2 L z2 , and one end of the Capacitor 2 C z2 is connected to the connection line between Diode 1 D1 and Inductor 2 L z2 , and the other end is connected to the negative pole of the DC power supply; the Switching transistor 3 S 3 and the Switching transistor 4 S 4 are connected in series and then connected in parallel across both ends of the Switching transistor 1 S 1 and the Switching transistor 2 S 2 .

[0060] The primary side of the resonant network includes a first compensation inductor L 1 、 First compensation capacitor 1 C 1 、 First compensation capacitor 2 C p , one end of the first compensation inductor L 1 , the first compensation capacitor 2 C p , and one end of the transmitting coil are connected together. The other end of the first compensation inductor L 1 is connected to the connection line between the Switching transistor 1 S 1 and the Switching transistor 2 S 2 , and the other end of the transmitting coil is connected to the connection line between the Switching transistor 3 S 3 and the Switching transistor 4 S 4 .

[0061] The synchronous rectifier is different from the traditional diode uncontrollable rectification. It does not have the dead zone voltage caused by the Schottky barrier voltage, reduces losses, and is especially more widely used in the field of low voltage and large current. However, adding active devices will make the circuit design more complex, but the need for closed-loop control can also be achieved by adding switching transistors.

[0062] The synchronous rectifier includes 2 inductors and 2 switching transistors, namely the third inductor 1 L t1 、 Third inductor 2 L t2 、 Switching transistor 5 S 5 、 Switching transistor 6 S 6 、 Third capacitor C 0 , the third inductor 1 Lt1 and the load, switch tube five S 5 are connected in series, and the third inductor two L t2 and switch tube six S 6 are connected in series and then connected in parallel across the load, and the third capacitor C 0 is connected in parallel across the load.

[0063] The primary side of the resonant network includes a second compensation inductor L 2 , a first second compensation capacitor C 2 , a second second compensation capacitor C s . One end of the second compensation inductor L 2 , one end of the second second compensation capacitor C s , and one end of the receiving coil are connected together. The other end of the second compensation inductor L 2 is connected to the connection line between the third inductor one L t1 and switch tube five S 5 . The other end of the receiving coil is connected to the connection line between the third inductor two L t2 and switch tube six S 6 .

[0064] The LCC-LCC high-order compensation topology structure selected for the resonant network part shows better robustness when facing system detuning, which means it can better adapt to non-ideal conditions such as coil offset and frequency change. Compared with the SS compensation network, the LCC-LCC compensation network has better constant current characteristics, can cope with the fluctuation of the coupling coefficient between coils, and has good anti-interference ability. For the primary side, the PWM modulation control circuit gives control signals to the 4 switch tubes of the quasi-Z-source inverter to change the boost capacity of the inverter. For the secondary side, the PWM modulation control circuit gives signals to 2 switch tubes to achieve multi-stage constant current and constant voltage output on the system load side.

[0065] The PWM modulation control circuit is used to realize the closed-loop control of the system. The voltage on the load side is collected through the voltage acquisition circuit, and constant current and constant voltage control in different stages is realized according to the voltage level on the load side.

[0066] The primary side of the overall wireless power transfer system uses a quasi-Z-source inverter, which realizes the boost of the input power supply voltage through an impedance network composed of passive devices; allows the devices on the same phase bridge arm to conduct simultaneously, and the drive signal does not need to insert a dead time, avoiding the voltage and current harmonics caused by the dead time; compared with the Z-source inverter, the input power supply and the inverter of the quasi-Z-source share the same ground, and the electromagnetic interference problem is smaller. At the same input voltage and boost capacity, the capacitor in the quasi-Z-source topology has a smaller voltage stress. Its boost factor B is shown in the following formula, where D represents the direct-on duty ratio:

[0067]

[0068] The peak value of the inverter output voltage is:

[0069]

[0070] Among them, m is the inverter modulation ratio, and E is the input voltage of the DC power supply.

[0071] According to the relevant content of Fourier decomposition, we have:

[0072]

[0073] Among them, U invrms represents the effective value of the output voltage of the system inverter. The output voltage not only includes the fundamental component but also a certain amount of odd harmonics. However, these harmonics will be greatly attenuated after being filtered by the resonant network. Therefore, the output of the inverter bridge can be equivalent to a sinusoidal voltage source with the same frequency as the fundamental wave. The equivalent expression is as follows, where Vp represents the effective value of the fundamental wave voltage output by the inverter.

[0074]

[0075] To simplify the analysis, all passive components are ideal devices, and the parasitic resistances of the inductors and capacitors in the circuit are ignored. Among them, U inv is the high-frequency AC voltage output by the system through the inverter, I L1 represents the fundamental wave phase current of the inverter output, L p is the self-inductance of the transmitting coil, L s is the self-inductance of the receiving coil, I p is the current on the transmitting coil, I s is the current on the receiving coil, M represents the mutual inductance between the primary coil and the secondary coil, , k represents the coupling coefficient, the compensation inductors L 1 、L 2 , the compensation capacitors C 1 、C P 、C 2 、C S constitute an LCC-LCC type resonant compensation network, Z in is the primary input impedance U rec is the voltage before rectification output through the resonant network. R L is the DC load of the system, R eq is the equivalent AC load, U o is the voltage on the load side.

[0076] Its equivalent circuit on the primary side is as Figure 2 shown,

[0077] Among them, Z r is the reflected impedance of the secondary side referred to the primary side, and the expression is as follows:

[0078]

[0079]

[0080] f represents the inverter switching frequency, and Z s represents the secondary-side impedance.

[0081] Its secondary-side equivalent circuit is as shown in Figure 3 where U oc represents the open-circuit voltage of the secondary-side equivalent circuit, and R eq represents the equivalent resistance of the subsequent circuit of the secondary-side resonant network

[0082]

[0083] where Z r is the reflected impedance of the secondary side referred to the primary side, represents the angular frequency, M represents the mutual inductance between the primary coil and the secondary coil, and Z s represents the secondary-side impedance, represents the phase difference in the AC circuit, represents the current in the primary transmitting coil.

[0084] According to Kirchhoff's theorem, we can obtain:

[0085]

[0086] When the primary and secondary resonant networks of the system are both in the resonant state, the system satisfies the following relationship:

[0087]

[0088] where ω 0 represents the corresponding resonant frequency, L 1 is the first compensation inductor, C 1 represents the first compensation capacitor, L 2 represents the second compensation inductor, C 2 represents the second compensation capacitor, L p is the self-inductance of the transmitting coil, C p represents the second compensation capacitor, L s is the self-inductance of the receiving coil, C s represents the second compensation capacitor.

[0089] The primary input impedance Z in and the secondary equivalent impedance Z s are expressed as follows:

[0090]

[0091]

[0092] Among them, Z in represents the primary-side input impedance, Z s represents the secondary-side equivalent impedance, and / / represents impedance parallel connection, represents the phase difference in an AC circuit, represents the angular frequency, L 1 is the first compensation inductor, L p is the self-inductance of the transmitting coil, C p represents the second compensation capacitor two, represents the secondary-side reflected impedance, C 1 represents the first compensation capacitor one, L s is the self-inductance of the receiving coil, C s represents the second compensation capacitor two, C 2 represents the second compensation capacitor one, L 2 represents the second compensation inductor, R eq is the equivalent AC load.

[0093] According to the above formulas for calculation and simplification, finally, when reaching the resonance state, the expressions of the system input power and output power are as follows:

[0094]

[0095]

[0096] Among them, is the system input power, U inv is the high-frequency AC voltage output by the system through the inverter, M represents the mutual inductance between the primary-side coil and the secondary-side coil, R eq is the equivalent AC load, ω 0 represents the corresponding resonance frequency, L 1 is the first compensation inductor, L 2 is the second compensation inductor. is the system output power, k represents the coupling coefficient, L p is the self-inductance of the transmitting coil, L s is the self-inductance of the receiving coil, U inv is the high-frequency AC voltage output by the system through the inverter.

[0097] Figure 4 The working principle of the synchronous rectifier is shown. Positive half-cycle: Q 5 conducts, Q 6 turns off, and the inductor stores energy; Q 5 turns off, Q 6 conducts for freewheeling, and the inductor releases energy to R L ; Negative half-cycle: Q 5 turns off, Q6 Conduct, and the inductor stores energy; Q 5 Conduct for freewheeling, Q 6 Turn off, and the inductor releases energy to R L 。

[0098] Optimize the multi-stage constant current and constant voltage charging strategy through an improved particle swarm algorithm. An adaptive PID controller is introduced into the system, and optimization design is carried out by comprehensively considering three aspects of the system: overshoot, settling time, and steady-state error. The expression of the objective fitness function is as follows:

[0099]

[0100] Where represents the steady-state error of the system, T represents the settling time of the system, ess represents the steady-state error of the system, and a, b, and c are their weight coefficients respectively. The global optimal solution Pg is obtained with the maximum fitness function value as the optimization goal.

[0101] The maximum particle velocity V max Determines the resolution of the region between the current position and the best position. If the maximum velocity value V max is too large, it is easy to cross the optimal region. If V max is too small, the particles cannot conduct sufficient exploration, resulting in being trapped in the local optimal value. To avoid the blind search of the particles, it is necessary to limit the particle velocity to achieve a better particle search effect. The limiting expression is as follows:

[0102]

[0103]

[0104] Where, V n represents the search velocity of the particle at the nth charging switching point, V nmax and V nmax represent its upper and lower limits of the search velocity, represents the variance of its fitness function, and its expression is as follows, where represents the mean value of the system fitness function. is the introduced initial threshold, which is set according to specific requirements. At the same time, the search velocities of each charging switching point show an overall downward trend.

[0105]

[0106] The inertia weight ω reflects the ability of particles to inherit the previous velocity. A larger inertia weight is beneficial for global search, while a smaller inertia weight is more conducive to local search. Although the particle swarm optimization algorithm with a constant inertia weight ω has a fast convergence speed, it is prone to falling into local optimum in the later stage and has low solution accuracy. To better balance the global search and local search capabilities of the algorithm, that is, because the local optimum credibility of the inertia weight is not high in the initial stage, in order to prevent falling into the local optimum solution. It is necessary to make the inertia weight ω larger in the initial stage of iteration and smaller in the later stage of iteration. Set the dynamic inertia weight as follows:

[0107]

[0108] Among them, ω max represents the maximum value limit of the inertia factor, ω min represents the minimum value limit of the inertia factor, represents the current maximum value of the fitness function, represents the current minimum value of the fitness function. Through this kind of dynamic inertia weight, it can be adaptively adjusted according to the problem characteristics to balance the system exploration ability.

[0109] Then calculate the individual average optimal solution as follows:

[0110]

[0111] Then add Gaussian perturbations to both the individual average optimal solution and the global optimal solution as follows:

[0112]

[0113]

[0114] Among them, represents the individual average optimal solution, represents the updated individual average optimal solution after introducing Gaussian perturbation, is the perturbation parameter, is the proportionality coefficient set according to the specific situation, and are the upper and lower limits of the perturbation parameter set according to the actual needs, represents the number of iterations, represents setting the upper limit of the number of iterations, represents the global optimal solution, represents the updated global optimal solution after introducing Gaussian perturbation. By perturbing the average individual optimal and global optimal at all times, the activity of particles is maintained, effectively avoiding the decline of diversity and the occurrence of premature convergence, enabling the algorithm to obtain better performance, and the final result can better escape from the local optimum, thus obtaining a more reasonable charging stage switching point.

[0115] It adopts stepped multi-stage constant current charging and ends with constant voltage charging in the last stage. The selection of the charging stage switching point is optimized by an improved particle swarm optimization algorithm: by establishing training set, test set, and validation set data, establishing an algorithm model and testing it, and then updating the velocity and position of the particles based on the dynamic inertia weight to finally update the global optimum, so as to optimize the particle swarm optimization algorithm. Through the improved particle swarm optimization algorithm, the multi-stage constant current and constant voltage charging strategy is optimized and designed, such as Figure 5 shown, first initialize the particles (initialize the current group) to make them as evenly distributed in the whole space as possible. Then substitute the initialized current into the model for calculation to obtain parameters such as voltage, temperature, SOC, and SOH, and judge whether the voltage reaches the threshold voltage and switch to the next stage of constant current charging. Judge whether the conditions simultaneously satisfy SOC = 90%, V ch = 4.2V, and switch to constant voltage charging.

[0116] According to the inertia weight coefficient set by the user's requirements, calculate the objective function value. Compare the objective function values of 100 particles to obtain the individual and global optimum values P i 、P g . Compare the P g of the current particle swarm with the previous particle swarm. If the difference is less than the set threshold. Then the program terminates, otherwise go to the next step. Use the update formula to update the positions r and velocities of all particles, so that all particles approach the vicinity of the optimal solution.

[0117] At the same time, specific parameters are brought in, and a certain technical analysis of the technical effect of the present invention is carried out through simulation. Figure 6 shown is the DC-link voltage output of the quasi-Z-source inverter when the boost factor is B = 1.5 under a 100V DC input, Figure 7 shown are the voltage and current waveforms of the system output after synchronous rectification.

[0118] In the case of a small-voltage DC power supply or DC power obtained directly by rectifying industrial-frequency alternating current, it can still be used in larger-power occasions. The LCC-LCC resonant compensation network adopted by the resonant network part has better constant current characteristics and shows better robustness in the face of system detuning. At the same time, the multi-stage constant current and constant voltage charging method adopted on the secondary side can improve the charging speed and extend the service life.

[0119] The present invention realizes the boosting of DC input by controlling the shoot-through ratio of the quasi-Z-source inverter, enabling the rectified civil alternating current to be directly used in high-voltage input scenarios. The electrical energy is converted into magnetic energy by the primary transmitting coil, and the magnetic energy is converted back into electrical energy by the secondary receiving coil. The reactive power is compensated and the electrical energy of other frequencies is filtered out with the help of a resonant network. Then, the switching devices of the secondary active rectifier are used to control the load side to achieve multi-stage constant current and constant voltage output, so as to improve the charging speed of the system and increase the service life of the lithium battery. The present invention realizes the multi-stage constant current and constant voltage output of the wireless charging system based on the quasi-Z-source inverter, realizes the boost of the power input on the primary side, the increase of the current on the secondary side, and the multi-stage constant current and constant voltage output on the load side, improves the charging speed of the system, extends the service life of the lithium battery, and is applicable to application scenarios such as electric vehicles.

[0120] The above are only the preferred embodiments of the present invention. It should be pointed out that for those of ordinary skill in the art in this technical field, several improvements and refinements can be made without departing from the principle of the present invention, and these improvements and refinements should also be regarded as the protection scope of the present invention.

Claims

1. A multi-stage constant current and constant voltage wireless power transmission system based on a quasi-Z source inverter, characterized in that: The invention comprises a direct current power supply, a quasi-Z source inverter, a PWM modulation control circuit, a transmitting coil, a receiving coil, a resonant network, a synchronous rectifier and a load. The direct current power supply, the quasi-Z source inverter, the primary side of the resonant network and the transmitting coil are connected in series, and the receiving coil, the secondary side of the resonant network, the synchronous rectifier and the load are connected in series. The PWM modulation control circuit is used to change the boosting capability of the quasi-Z source inverter and to control the synchronous rectifier to realize multi-stage constant current and constant voltage output on the load side of the system.

2. According to claim 1, the multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter is characterized in that: The quasi-Z source inverter includes a diode D1, an inductor L z1 、Inductor 2 L z2 , Capacitor - C z1 , Capacitor C z2 , switch tube 1 S1, switch tube 2 S2, switch tube 3 S3, switch tube 4 S4, the DC power supply, inductor 1 L z1 , diode 1 D1, inductor 2 L z2 , switch tube 1 S1, switch tube 2 S2 are connected in series, the capacitor C z1 Connect the diode D1 and the inductor L in parallel. z2 The two ends of the capacitor C z2 One end is connected between diode D1 and inductor L z2 The switch tube three S3 and the switch tube four S4 are connected in series and then connected in parallel at both ends of the switch tube one S1 and the switch tube two S2.

3. According to claim 2, the multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter is characterized in that: The primary side of the resonant network includes a first compensation inductor L1, a first compensation capacitor C1, a first compensation capacitor C2, and a first compensation capacitor C p , one end of the first compensation inductor L1 and the first compensation capacitor C p , one end of the transmitting coil is connected together, the other end of the first compensation inductor L1 is connected to the connecting line between the switch tube 1 S1 and the switch tube 2 S2, and the other end of the transmitting coil is connected to the connecting line between the switch tube 3 S3 and the switch tube 4 S4.

4. According to claim 3, the multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter is characterized in that: The synchronous rectifier includes a third inductor L t1 、The third inductor L t2 , switch tube five S5, switch tube six S6, third capacitor C0, and the third inductor L t1 , load, and switch tube S5 are connected in series, and the third inductor L t2 , the switch tube S6 is connected in series and then connected in parallel at both ends of the load, and the third capacitor C0 is connected in parallel at both ends of the load.

5. According to claim 4, the multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter is characterized in that: The primary side of the resonant network includes a second compensation inductor L2, a second compensation capacitor C2, and a second compensation capacitor C s , one end of the second compensation inductor L2, the second compensation capacitor C s , one end of the receiving coil is connected together, the other end of the second compensation inductor L2 is connected to the third inductor L t1 The other end of the receiving coil is connected to the connecting line between the switch tube S5 and the third inductor L. t2 Connected to the connecting line between switch tube six S6.

6. The multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter according to claim 5, characterized in that: The system input power is: in, is the system input power, U inv is the high-frequency AC voltage output by the system through the inverter, M represents the mutual inductance between the primary coil and the secondary coil, R eq is an equivalent AC load, ω0 represents the corresponding resonant frequency, L1 is the first compensation inductor, and L2 is the second compensation inductor.

7. The multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter according to claim 6, characterized in that: The system output power is: in, is the system output power, k represents the coupling coefficient, L p is the self-inductance of the transmitting coil, L s is the receiving coil self-inductance, U inv is the high-frequency AC voltage output by the system through the inverter, ω0 represents the corresponding resonant frequency, L1 is the first compensation inductor, L2 is the second compensation inductor, R eq is the equivalent AC load.

8. The multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter according to claim 7, characterized in that: Primary input impedance Z in And the secondary equivalent impedance Z s The expression is as follows: Among them, Z in Represents the primary input impedance, Z s Indicates the equivalent impedance of the secondary side, / / indicates impedance in parallel, Represents the phase difference in an AC circuit, represents the angular frequency, L1 is the first compensation inductor, L p is the self-inductance of the transmitting coil, C p represents the first compensation capacitor two, represents the secondary side reflection impedance, C1 represents the first compensation capacitor, L s is the receiving coil self-inductance, C s represents the second compensation capacitor 2, C2 represents the second compensation capacitor 1, L2 represents the second compensation inductor, R eq is the equivalent AC load.

9. The multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter according to claim 8, characterized in that , when the primary and secondary resonant networks of the system are both in a resonant state, the system satisfies the following relationship: Wherein, ω0 represents the corresponding resonant frequency, L1 represents the first compensation inductor, C1 represents the first compensation capacitor, L2 represents the second compensation inductor, C2 represents the second compensation capacitor, L p is the self-inductance of the transmitting coil, C p Represents the first compensation capacitor, L s is the receiving coil self-inductance, C s Represents the second compensation capacitor two.

10. The multi-stage constant current and constant voltage wireless power transmission system based on the quasi-Z source inverter according to claim 9, characterized in that , The method for optimizing the design of multi-stage constant current and constant voltage charging strategy by improving the particle swarm algorithm is as follows: Step 1: Introduce an adaptive PID controller, comprehensively consider the system's overshoot, stabilization time, and steady-state error to optimize the design, and take the maximum fitness function value as the optimization goal to obtain the global optimal solution Pg; the target fitness function expression is as follows: in, represents the fitness function value, represents the system steady-state error, T represents the system stabilization time, ess represents the system stable error, a represents the system steady-state error weight coefficient, b represents the system stable time weight coefficient, and c represents the system stable error weight coefficient; Step 2: The limiting expression for particle velocity is as follows: Among them, V n represents the search speed of the nth charging switching point particle, V n，max Represents the upper limit of search speed, V n，min Represents the lower limit of search speed, represents the variance of the fitness function, is the initial threshold introduced, represents the mean value of the system fitness function, represents the number of particle populations, represents the fitness function value, represents the search speed of the first charging switching point particle, represents the search speed of the second charging switching point particle, Indicates the search speed of the third charging switching point particle; Step 3, the dynamic inertia weight is as follows: in, represents the dynamic inertia weight, ω max Represents the maximum value of the inertia factor, ω min Represents the minimum value of the inertia factor, Represents the current maximum value of the fitness function, Represents the current minimum value of the fitness function; Step 4, the individual average optimal solution is as follows: in, represents the average optimal solution of individuals, N represents the number of particle populations, represents the individual optimal solution; step 5, add Gaussian perturbation to the individual average optimal solution and the global optimal solution at the same time as follows: in, represents the average optimal solution of individuals, represents the individual average optimal solution after the introduction of Gaussian perturbation update, is the disturbance parameter, It is a proportional coefficient set according to specific circumstances. and The upper and lower limits of disturbance parameters are set according to actual needs. represents the number of iterations, Indicates setting the upper limit of the number of iterations. represents the global optimal solution, Represents the global optimal solution after introducing Gaussian perturbation update.

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