Asymmetric dual-active bridge direct current converter and modulation method thereof

Abstract

The invention relates to an asymmetric dual-active bridge direct current converter and a modulation method thereof. A secondary side of the converter comprises a second active full bridge and a semi-active bridge which are connected in parallel; the second active full bridge comprises active switches S21 to S24, the active switches S21 and S22 are connected in series between the output end of the converter and the reference ground, the active switches S23 and S24 are connected in series between the output end of the converter and the reference ground, and the common end of the S21 and the common end of the S22 are coupled to the common end of the S23 and the common end of the S24 through two groups of inductors and the secondary side of the transformer; the semi-active bridge comprises diodes D31 and D33 and active switches S32 and S34; the D31 and the S32 are connected in series between the output end of the converter and the reference ground, the D33 and the S34 are connected in series between the output end of the converter and the reference ground, and the common end of the diode D31 and the active switch S32 is coupled to the common end of the diode D33 and the active switch S34 through the group inductor and the secondary side of the transformer. Compared with the prior art, the method has the advantages of realizing two-way power differential transmission, reducing loss and the like.

Description

Technical Field

[0001] The present invention relates to the technical field of dual-active-bridge DC converters, and in particular to an asymmetric dual-active-bridge DC converter and a modulation method thereof. Background Art

[0002] Solid-state transformers (SSTs), based on series-parallel connections of DC converter modules, not only perform the basic functions of traditional power-frequency transformers, such as voltage conversion and electrical isolation, but also incorporate new features such as active / reactive power decoupling control, bidirectional power flow regulation, fault ride-through, and fault-tolerant operation. They are widely used in new power systems. For example, in new photovoltaic power plants, SSTs enable direct integration of photovoltaic arrays and battery energy storage systems on the low-voltage DC side. New energy vehicle charging stations use SSTs to aggregate DC energy from power batteries and photovoltaic storage systems onto the medium-voltage busbar.

[0003] In these application scenarios, SST exhibits significant bidirectional power flow asymmetry. For example, an electric vehicle charging station's primary function is to rapidly charge electric vehicles at rated power, while its secondary function is to provide power services. When the grid is overloaded or emergency power is required, energy stored in the station's energy storage and power batteries is fed back into the grid through a vehicle-grid interaction mechanism. The feed-in power is typically set at 30%-60% of the charging power. Conventional bidirectional DC converter circuit topologies, including dual active bridge converters, three-level half-bridge converters, current-fed converters, and CLLC bidirectional resonant converters, share a common characteristic: symmetrical circuits on both sides of the transformer, with component selection and circuit parameter design based on rated capacity. In asymmetric power transmission applications, the mismatch between the forward and reverse rated capacities of the converter can lead to underutilization of some components, resulting in redundant and wasted reverse capacity. Two currently feasible approaches to addressing asymmetric power transmission applications are the use of hybrid single and bidirectional DC converter modules and differentiated designs for the active bridge and uncontrolled rectifier bridge within a single DC converter module. The former method has problems such as low device reuse rate and poor flexibility and adaptability; the latter method has high peak current and the circulating power will cause additional loss.

[0004] In summary, asymmetric dual-active-bridge DC converters are an important class of converters for asymmetric power transmission applications. However, existing asymmetric converter topologies, modeling, and modulation methods require further research and optimization. Currently, research on the topology and modulation methods of asymmetric dual-active-bridge DC converters is limited. To rationally control asymmetric dual-active-bridge DC converters, a modulation method and power transmission model must be developed.

[0005] After searching, Chinese invention patent application publication number CN115498893A discloses a dual-active bridge converter, which includes a primary full bridge, a secondary full bridge, an inductor, and a transformer. The primary full bridge includes four switches S1 to S4, and the secondary full bridge includes four switches Q1 to Q4. Switches S1 and S2 are connected in series between the converter input and a reference ground, while switches S3 and S4 are connected in series between the converter input and a reference ground. The common terminal of switches S1 and S2 is coupled to the common terminal of switches S3 and S4 via the inductor and the primary side of the transformer. Switches Q1 and Q2 are connected in series between the converter output and a reference ground, while switches Q3 and Q4 are connected in series between the converter output and a reference ground. The common terminal of switches Q1 and Q2 is coupled to the common terminal of switches Q3 and Q4 via the secondary side of the transformer. This existing patent application suffers from the problem that the converter has a symmetrical structure and cannot achieve differentiated power conversion.

[0006] How to realize a flexible and low-loss asymmetric dual active bridge DC converter has become a technical problem that needs to be solved. Summary of the Invention

[0007] The purpose of the present invention is to overcome the defects of the above-mentioned prior art and provide an asymmetric dual active bridge DC converter topology and a modulation method thereof.

[0008] The purpose of the present invention can be achieved by the following technical solutions:

[0009] According to one aspect of the present invention, an asymmetric dual-active bridge DC converter is provided, wherein the primary side of the converter is a first active full bridge, the primary side and the secondary side are connected via a transformer, and the secondary side of the converter includes a second active full bridge and a half active bridge connected in parallel;

[0010] The second active full bridge includes an active switch S 21 To S 24 , active switch S 21 and S 22 Connected in series between the output of the converter and the reference ground, the active switch S 23 and S 24 Connected in series between the output of the converter and the reference ground, the active switch S 21 and S 22 The common end of the two inductors and the secondary side of the transformer are coupled to the active switch S 23 and S 24 The public end;

[0011] The semi-active bridge includes a diode D 31 、D 33 With active switch S 32 、S34 ; diode D 31 and active switch S 32 Connected in series between the output of the converter and the reference ground, the diode D 33 and active switch S 34 Connected in series between the output of the converter and the reference ground, the diode D 31 and active switch S 32 The common end of the diode is coupled to the secondary side of the diode D through the inductor and transformer. 33 and active switch S 34 The public end of the

[0012] Preferably, the secondary winding of the transformer is connected to the second active full bridge via a first inductor and a second inductor; the secondary winding of the transformer is connected to the midpoint of the bridge arm of the half-active bridge via a third inductor and a fourth inductor;

[0013] The sum of the first inductor and the third inductor is denoted as L2, which is called the inductor group L2; the sum of the second inductor and the fourth inductor is denoted as L3, which is called the inductor group L3.

[0014] Preferably, the converter includes a forward working mode and a reverse working mode;

[0015] In the forward operating mode, power flows through the first active full bridge, the second active full bridge and the half active bridge;

[0016] In the reverse operating mode, power flows through the second active full bridge and the first active full bridge.

[0017] Preferably, the first active full bridge comprises an active switch S 11 To S 14 , active switch S 11 and S 12 Connected in series between the input of the converter and the reference ground, the active switch S 13 and S 14 Connected in series between the input of the converter and the reference ground, the active switch S 11 and S 12 The common end of the active switch S is coupled to the active switch S through the inductor L1 and the primary side of the transformer. 13 and S 14 The public end of the

[0018] According to another aspect of the present invention, a modulation method for an asymmetric dual active bridge DC converter is provided, wherein the modulation method includes a forward mode modulation sub-method, specifically:

[0019] The second active full bridge and half active bridge adopt synchronous phase shift modulation strategy, and the active switch S 21 and S 23Working in synchronous rectification state, the switching period is recorded as T S , where the active switch S 11 、S 14 、S 24 、S 34 Driven by a square wave with a duty cycle of 50%, the active switch S 12 、S 13 、S 22 、S 32 The driving signal is complementary to the active switch S 21 and S 23 Driven by a square wave with a duty cycle of δ;

[0020] Assume that the converter has reached steady state, define S 11 and S 14 The PWM drive signal is turned on at t0 as the starting moment. Within a switching cycle, seven time points, t0 to t6, are defined in sequence according to the inductor current slope change points. These time points divide the switching cycle into six intervals, the first three of which are positive half-cycles and the last three are negative half-cycles.

[0021] Definition U i Indicates input voltage, U o Indicates the output voltage. The voltage conversion ratio of the converter is k, which is U o / (nU i ), the equivalent inductance L on the secondary side 11 =n 2 L1, n is the transformer ratio, L1 represents the primary inductance of the transformer, L2 and L3 represent the inductance group L2 and inductance group L3 of the secondary side of the transformer respectively;

[0022] Define the current flowing through the leakage inductor L1 as i L1 , the current flowing through the second active full bridge is i L2 , the current flowing through the semi-active bridge is i L3 ;

[0023] In the interval [t0, t1), the voltage U applied to the inductor group L2 at time t L2 (t) and i L2 The expression is:

[0024]

[0025] In the interval [t1, t2), the voltage U applied to the inductor group L2 at time t L2 (t) and i L2 The expression is:

[0026]

[0027] In the interval [t2, t3), the voltage U applied to the inductor group L2 at time tL2 (t) and i L2 The expression is:

[0028]

[0029] The current i in the negative half cycle L2 and voltage U L2 (t) Symmetric to the first half cycle.

[0030] Preferably, the current i flowing into the semi-active bridge L3 The current i flowing into the second active full bridge L2 Similar patterns of change.

[0031] Preferably, the power distribution between the second active full bridge and the half active bridge is related to the ratio between the inductors L2 and L3, that is,

[0032]

[0033] The power values ​​shared by the second active full bridge and the half active bridge are distributed by adjusting the parameters of L2 and L3.

[0034] Preferably, the modulation method includes a reverse mode modulation sub-method, including: in the reverse mode, the current i L3 The uncontrollable circulating current power P caused by C ;

[0035] Using the unidirectional conduction characteristics of the uncontrolled rectifier bridge, the midpoint voltage U H3 Clamped to output voltage U o , the circulating power P can be eliminated C , the following formula must be satisfied:

[0036]

[0037] Further simplification yields

[0038] k≥1

[0039] Where k is the voltage conversion ratio.

[0040] More preferably, in the reverse mode, the reverse equivalent leakage inductance L is defined as eq =(L 11 +L2), active switch S 11 、S 14 、S 21 、S 24 Driven by a square wave with a duty cycle of 50%, S 12 、S 13 、S 22 、S 23 The driving signal complements it and blocks the S in the semi-active bridge at the same time.32 、S 34 The driving signal;

[0041] Define θ as S 11 Hysteresis S 21 The phase shift duty cycle, that is, θ = (t1-t0) / Ts;

[0042] Assuming the circuit has reached steady state, define S 21 and S 24 The PWM drive signal is turned on at t0 as the starting moment. Within a switching cycle, five time points, t0 to t4, are defined in sequence based on the inductor current slope change points. These time points divide the switching cycle into four intervals, the first two of which are positive half-cycles and the last two are negative half-cycles:

[0043] In the interval (t0, t1), the reverse equivalent leakage inductance L is applied eq Voltage U on Leq and i L2 The expression is:

[0044] U Leq (t)=-(nU i +U o )

[0045]

[0046] In the interval (t1, t2), the reverse equivalent leakage inductance L is applied eq Voltage U on Leq and i L2 The expression is:

[0047] U Leq (t)=(nU i -U o )

[0048]

[0049] The negative half cycle is applied to the reverse equivalent leakage inductance L eq Voltage U on Leq and i L2 Symmetrical to the first half of the cycle.

[0050] Preferably, the working mode selection and power regulation of the converter are completed by unifying the phase shift duty cycle η, which is defined as For S 11 Advanced S 24 / S 34 The phase shift duty cycle is

[0051] When η>0, the converter works in the forward mode. and through The phase shift duty cycle δ = f(η) is calculated, where k is the voltage conversion ratio;

[0052] When η<0, the converter operates in reverse mode, blocking the active switch S 32 and S 34 The driving signal of is given, and -θ = η, the semi-active bridge on the secondary side of the converter stops running;

[0053] When the output voltage U o When the forward transmission power is reduced, the total forward transmission power P is increased by increasing η in the forward mode. FB To stabilize the output voltage U o In the reverse mode, the reverse transmission power P3 is reduced by increasing η, that is, reducing θ to stabilize the output voltage U o .

[0054] Compared with the prior art, the present invention has the following beneficial effects:

[0055] 1) The secondary side of the DC converter of the present invention includes a second active full bridge and a half-active bridge connected in parallel. The two active full bridges are multiplexed modules in two working modes. When transmitting power in the forward direction, electric energy flows through full bridge one, full bridge two and half bridge one; when transmitting power in the reverse direction, electric energy flows through full bridge two and full bridge one, and half bridge one does not participate, thereby realizing bidirectional power differentiated transmission.

[0056] 2) The semi-active bridge includes diodes and active switches. Compared with similar DC-DC converter circuits composed of three full-bridge circuits or conventional bidirectional DC converters, the use of diodes instead of two fully controlled semiconductor switching devices can save device materials and reduce system costs.

[0057] 3) In the forward mode, the DC converter of the present invention can distribute the power shared by its second active full bridge and half active bridge by adjusting the ratio of the two inductor groups; in the reverse mode, the reference power is adjusted to eliminate the commutation power, thereby realizing power-controllable, differentiated forward and reverse transmission power DC-DC power conversion.

[0058] 4) The topology of the converter of the present invention, combined with the modulation algorithm, has the ability to suppress circulating current power, thereby reducing transmission losses and improving the overall efficiency of the converter. BRIEF DESCRIPTION OF THE DRAWINGS

[0059] Figure 1 Schematic diagram of the topology of the asymmetric dual active bridge DC converter in the present invention;

[0060] Figure 2 Schematic diagram of the power transmission operation mode of the converter in the present invention;

[0061] Figure 3Schematic diagram of the switch tube driving signal and typical waveform in the forward mode of the present invention;

[0062] Figure 4 Schematic diagram of the switch tube driving signal and typical waveform in the reverse mode of the present invention;

[0063] Figure 5 Schematic diagram of the control algorithm of the converter modulation method of the present invention;

[0064] Figure 6 Schematic diagram of the forward and reverse mode switching simulation waveforms of the converter modulation method of the present invention;

[0065] Figure 7(a) shows the S 11 Schematic diagram of drain-source / gate-source voltage and inductor current;

[0066] Figure 7(b) shows the S 24 Schematic diagram of drain-source / gate-source voltage and inductor current;

[0067] Figure 8(a) shows the S 11 Schematic diagram of drain-source / gate-source voltage and inductor current;

[0068] Figure 8(b) shows the S 24 Schematic diagram of drain-source / gate-source voltage and inductor current;

[0069] Figure 9(a) is a schematic diagram of the transmission steady-state experimental waveform in the forward mode;

[0070] Figure 9(b) is a schematic diagram of the transmission steady-state experimental waveform in reverse mode. DETAILED DESCRIPTION

[0071] The following will clearly and completely describe the technical solutions in the embodiments of the present invention in conjunction with the accompanying drawings. Obviously, the described embodiments are part of the embodiments of the present invention, not all of them. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without making creative efforts should fall within the scope of protection of the present invention.

[0072] This application designs a novel asymmetric dual-active-bridge DC converter. It establishes mathematical models for both forward and reverse power flow, and derives power transfer expressions. Furthermore, a dual-mode modulation method based on a uniform phase-shift duty cycle is designed for the modeling of the asymmetric dual-active-bridge DC converter to achieve differentiated control of forward and reverse power transfer. When transferring power in the forward direction, the converter operates in dual-half-active-bridge mode, and switches to dual-active-bridge mode when transferring power in the reverse direction.

[0073] This embodiment relates to an asymmetric dual active bridge DC converter, such as Figure 1 ,The DC converter consists of three parts: a transformer, a first active full bridge, a second active full bridge and a half active bridge.

[0074] The primary side is the first active full bridge (abbreviated as: first full bridge), and the secondary side is composed of the second active full bridge (abbreviated as: full bridge 2) and the half active bridge (abbreviated as: half bridge 1) circuit in parallel. Among them, the first active full bridge includes active switches S 11 To S 14 , the second active full bridge includes active switches S 21 To S 24 The semi-active bridge includes diode D 31 、D 33 With active switch S 32 、S 34 .

[0075] Active switch S 11 and S 12 Connected in series between the input of the converter and the reference ground, the active switch S 13 and S 14 Connected in series between the input of the converter and the reference ground, the active switch S 11 and S 12 The common end of the active switch S is coupled to the active switch S through the inductor L1 and the primary side of the transformer. 13 and S 14 The public end of the

[0076] Active switch S 21 and S 22 Connected in series between the output of the converter and the reference ground, the active switch S 23 and S 24 Connected in series between the output of the converter and the reference ground, the active switch S 21 and S 22 The common end of the transformer is coupled to the active switch S through the first inductor, the secondary side of the transformer and the second inductor. 23 and S 24 The public end of the

[0077] In a semi-active bridge, diode D 31 and active switch S 32 Connected in series between the output of the converter and the reference ground, the diode D 33 The active switch S34 is connected in series between the output terminal of the converter and the reference ground, and the diode D 31 and active switch S 32 The common end of the transformer is coupled to the diode D through the third inductor, the secondary side of the transformer and the fourth inductor. 33 and active switch S 34 The public end of the

[0078] Figure 1 Middle,U i Indicates input voltage, U o Indicates the output voltage, the input side capacitance and the output side capacitance are respectively denoted as C i and C o The two circuits are connected via a high-frequency transformer with a turns ratio of N1:N2 (definition transformer ratio n = N2 / N1). The transformer leakage inductance is denoted as L1, and the secondary winding is connected to the midpoint of the bridge arms of full bridge 2 and half bridge 1 via two sets of inductors (the inductance of each set is the same, the sum of the first and third inductors is denoted as L2, called inductance group L2; the sum of the second and fourth inductors is denoted as L3, called inductance group L3). The current flowing through the leakage inductance L1 is defined as i L1 , the current flowing through the inductor group L2 is i L2 , the current flowing through the inductor group L3 is i L3 The voltage conversion ratio of the converter is defined as k, which is U o / (nU i ).

[0079] This DC-DC converter enables bidirectional, differentiated power transmission. Forward operating mode is defined as power flowing from the input to the output, while reverse operating mode is defined as power flowing from the input to the output. Therefore, full-bridge 1 and full-bridge 2 are multiplexed modules for both modes. During forward power transmission, energy flows through full-bridge 1, full-bridge 2, and half-bridge 1; during reverse power transmission, energy flows through full-bridge 2 and full-bridge 1, with half-bridge 1 not participating. Therefore, compared to a similar DC-DC converter circuit consisting of three full-bridge circuits, the use of diodes in half-bridge 1 instead of two fully controlled semiconductor switching devices saves device materials.

[0080] This topology itself has the advantages of bidirectional power transmission and wide-range zero-voltage soft switching, which can save devices and reduce system costs.

[0081] This embodiment also relates to a dual-mode modulation method for an asymmetric dual active bridge DC converter. To facilitate analysis, the parameters are converted to the secondary side, and the transformer's magnetizing inductance and winding parasitic resistance are ignored. The star equivalent circuit model of the converter in the two modes can be obtained, as shown in FIG. Figure 2 As shown. P1 and P2 are the power flowing through the full bridge 2 and half bridge 1 modules in the forward mode, respectively. P3 is the power flowing into the full bridge 1 in the reverse mode. C Indicates the circulating current power that needs to be suppressed in reverse mode. H1 、U H2 、U H3 They represent the high-frequency quasi-square wave voltage at the midpoint of the full bridge arm, full bridge arm 2, and half bridge arm 1, respectively. The equivalent inductance converted to the secondary side is L 11 =n 2 L1, where n is the transformer ratio.

[0082] Applying the superposition theorem, we can deduce that Figure 2 The reference point voltage U r The expression is

[0083]

[0084] The following analyzes the operating principles of the converter's two working modes: Figure 3 As shown, at this moment, the power is transmitted in the forward direction and the converter operates in dual half-active bridge mode. The specific modulation method is: full bridge 2 and half bridge 1 adopt synchronous phase shift modulation strategy, active switch S 21 and S 23 Works in synchronous rectification state. The switching period is recorded as T S , where the active switch S 11 、S 14 、S 24 、S 34 Driven by a square wave with a duty cycle of 50%, the active switch S 12 、S 13 、S 22 、S 32 The driving signal is complementary to it (complementary means the opposite level state appears); S 21 and S 23 It is driven by a square wave with a duty cycle of δ. For S 11 Advanced S 24 / S 34 The phase shift duty cycle is At the same time S 21 With S 24 Assuming the circuit has reached steady state, define S 11 and S 14 The PWM drive signal is turned on at t0 as the starting moment. Within a switching cycle, 7 time points, t0 to t6, are defined in sequence according to the inductor current slope change points. These time points divide the switching cycle into 6 intervals:

[0085] Among them, in the period t0~t1, the active switch S 11 、S 14 、S 22 、S 32 、S 23 and diode D 33 conduction, S 12 、S 13 、S 24 、S 34 、S 21 Shutdown, D 11 、D 14 、D 24 、D 34 Freewheeling;

[0086] During the period t1-t2, the active switch S 11 、S 14 、S 22 、S 32 conduction, S 12 、S 13 、S 24 、S 34 、S 21 、S 23 Turn off, diode D 24 、D 34 Freewheeling;

[0087] During the period t2-t3, the active switch S 11 、S 14 、S 24 、S 34 、S 21 Conducting, diode D 31 conduction, S 12 、S 13 、S 23 、S 22 、S 32 Shutdown, D 24 、D 34 Freewheeling;

[0088] During the period t3-t4, the active switch S 24 、S 34 、S 21 、S 12 、S 13 Conducting, diode D 31 conduction, S 22 、S 32 、S 23 、S 11 、S 14 Shutdown, D 12 、D 13 、D 24 、D 34 Freewheeling;

[0089] During the period t4-t5, the active switch S 12 、S 13 、S 24 、S 34 conduction, S 21 、S 11 、S 14 、S 22 、S 32 、S 23 Turn off, diode D 22 、D 32 Freewheeling;

[0090] During the period t5-t6, the active switch S12 、S 13 、S 22 、S 32 、S 23 Conducting, diode D 33 conduction, S 24 、S 34 、S 11 、S 14 、S 21 Shutdown, D 22 、D 32 Continued flow.

[0091] The typical operating waveform of the converter in forward mode is as follows: Figure 3 shown.

[0092] Taking t0 as the starting time, a detailed analysis of the circuit within a switching cycle is carried out.

[0093] Interval (t0, t1): At time t0, S 12 and S 13 Shutdown, S 11 and S 14 Open, S 23 In the on state, i L1 By D 11 and D 14 Freewheeling, i L2 Flowing through S 23 and D 22 , U H1 By -U i Change to U i , the positive direction of the inductor current refers to Figure 1 Annotation.

[0094] U L2 (t) is the voltage applied to L2 at time t, then the current i applied to L2 at time t is L2 (i.e. the current flowing into full bridge 2) is expressed as:

[0095]

[0096] Interval (t1, t2): At time t1, i L1 From positive to negative, flows through S 11 and S 14 ;i L2 Flowing through S 22 and D 24 At this time U H2 By -U o becomes zero, the output current i o is zero, S 23 Achieve zero current shutdown. Because before current commutation, D 11 、D 14、D 22 Early conduction, S 11 、S 14 、S 22 It provides the conditions for zero voltage switching. L2 The expression is:

[0097]

[0098] Interval (t2, t3): At time t2, S 22 Shutdown, i L1 、i L2 The forward current reaches its peak value, at which point D 21 、D 24 conduction, due to S 21 Using synchronous rectification modulation, i L2 With S 21 and D 24 Form a freewheeling circuit. H2 From zero to U o .i L2 The expression is:

[0099]

[0100] The circuit operating principle during the negative half cycle, i.e., at time t3-t6, is similar to the above analysis.

[0101] Depend on Figure 2 It can be seen that the voltage U applied to L2 L2 It can be expressed as U r -U H2 According to the symmetry of the inductor current in a switching cycle, the inductor current satisfies i L2 (t0)=-i L2 (t3), and at the same time, t0 = 0 and Substituting into equations (2)-(4), we can obtain the expression of t1:

[0102]

[0103] Combining equations (2)-(5), in the forward mode, the voltage U L2 and current i L2 The specific expressions in the positive half cycle are listed in Table 1.

[0104] Table 1

[0105]

[0106] Then we get S 21 Driving signal duty cycle δ and phase shift duty cycle The relationship:

[0107]

[0108] The average current value I flowing through L2 is calculated from this 2_avg for:

[0109]

[0110] From this, the power P1 flowing into full bridge 2 can be further calculated as:

[0111] P1=I 2_avg U o (8)

[0112] Under synchronous phase shift modulation, U H2 and U H3 The phase is kept consistent, and the inductor current i flowing into half bridge one L3 Showing the same L2 Therefore, by exchanging the parameters of L2 and L3 in equations (7) and (8), the average current value I flowing into half-bridge 1 can be derived. SB1_avg And the transmission power P2. The total forward transmission power P FB It is equal to the algebraic sum of the power of full bridge 2 and half bridge 1.

[0113]

[0114] At the same time, from the above analysis, it can be seen that the power distribution between full bridge 2 and half bridge 1 is related to the ratio between inductors L2 and L3, that is,

[0115]

[0116] When inductor L2 = L3, the output power of the two modules is the same. By adjusting the parameters of L2 and L3, the power shared by full-bridge 2 and half-bridge 1 can be properly distributed.

[0117] In reverse mode, the power is transmitted in reverse, and the current i L3 The uncontrollable circulating current power P caused by C Utilizing the unidirectional conduction characteristic of the uncontrolled rectifier bridge, the bridge arm midpoint voltage U H3 Clamped to output voltage U o , the circulating power P can be eliminated C . The conditions given in formula (11) must be met

[0118]

[0119] Further simplification yields

[0120] k≥1 (12)

[0121] Where k is the voltage conversion ratio.

[0122] At this time, the first half-bridge module stops running and the converter switches to the dual active bridge mode. Inductors L1 and L2 are in series, and the reverse equivalent leakage inductance L is defined as eq =(L 11 +L2). Figure 4 As shown, the specific modulation method is: the active switch S 11 、S 14 、S 21 、S 24 Driven by a square wave with a duty cycle of 50%, S 12 、S 13 、S 22 、S 23 The driving signal complements it and blocks the S 32 、S 34 The driving signal of θ is defined as S 11 Hysteresis S 21 The phase shift duty cycle (θ is always positive), that is, θ = (t1-t0) / Ts. Assuming that the circuit has reached a steady state, define S 21 and S 24 The PWM drive signal is turned on at t0 as the starting time. Within a switching cycle, five time points, t0 to t4, are defined in sequence based on the inductor current slope change points. These time points divide the switching cycle into four intervals:

[0123] Among them, in the time interval t0~t1, t0′ is defined as the current i L2 When the value is 0, the active switch S 12 、S 13 、S 21 、S 24 conduction, S 11 、S 14 、S 22 、S 23 Turn off, diode D 12 、D 13 Continue current in the interval t0~t0′;

[0124] During the period t1 to t2, the active switch S 11 、S 14 、S 21 、S 24 conduction, S 12 、S 13 、S 22 、S 23 Turn off, diode D 11 、D 14 Freewheeling;

[0125] In the time interval t2 to t3, define t2′ as i in this time interval. L2 When the value is 0, the active switch S11 、S 14 、S 22 、S 23 conduction, S 12 、S 13 、S 21 、S 24 Turn off, diode D 11 、D 14 Continue current in the interval t2-t2′;

[0126] During the period t3-t4, the active switch S 12 、S 13 、S 22 、S 23 conduction, S 11 、S 14 、S 21 、S 24 Turn off, diode D 12 、D 13 Continued flow.

[0127] The typical operating waveform of the converter in reverse mode is as follows: Figure 4 shown.

[0128] Continuing the forward mode analysis method, with t0 as the starting time, the steady-state analysis of the positive half cycle of the converter circuit is carried out as follows:

[0129] Interval (t0, t1): At time t0, S 22 and S 23 Shutdown, i L2 By D 21 and D 24 Freewheeling, causing U H2 By -U o Change to U o . Because I L3 is zero, the reference point voltage U r with U H3 Same, so U H2 The change will also lead to U H3 Change, i L1 By D 12 and D 13 Continue current. At t0', the current crosses zero. During this period, the reverse equivalent leakage inductance L eq Voltage U on Leq The expression of: U Leq (t)=-(nU i +U o ), i L2 The expression is:

[0130]

[0131] Interval (t1, t2): At time t1, S 12 and S 13 Shutdown, i L1 By D 11 and D 14 Freewheeling, U H1 By -U i Change to U i , similarly, U H1 Changes will cause U H3 The full-bridge second current path remains unchanged, and the reverse equivalent leakage inductance L eq Voltage U on Leq The expression of: U Leq (t)=(nU i -U o ), i L2 The expression is:

[0132]

[0133] In the reverse mode, the voltage and current waveforms also have symmetry between the positive and negative half cycles, so the circuit analysis principle in the negative half cycle, that is, the time interval [t2~t4], is similar. In summary, the bridge arm voltage U in the reverse mode is H3 and the inductor current i L2 The expression of is shown in Table 2.

[0134] The average current I3 flowing through L2 in reverse mode avg for:

[0135]

[0136] Then the reverse transmission power P3 is calculated as:

[0137]

[0138] Table 2

[0139]

[0140] The forward / reverse working mode selection and power regulation of the converter can be achieved by unifying the phase shift duty cycle η.

[0141] When η>0, the converter works in the forward mode. And the phase shift duty ratio δ=f(η) is calculated by formula (6);

[0142] When η<0, the converter works in reverse mode, blocking S 32 and S 34 The driving signal is set to -θ=η, and the half-bridge module on the secondary side of the converter stops running.

[0143] The rated output voltage of the converter is U o * , the actual output voltage is U o The voltage stabilization mechanism is: when the output voltage U o When the forward transmission power is reduced, the total forward transmission power P is increased by increasing η in the forward mode. FB To stabilize the voltage U o In the reverse mode, the reverse transmission power P3 is reduced by increasing η or decreasing θ, thereby stabilizing the voltage U o .

[0144] The specific value of η can be obtained by calculating the proportional-integral (PI) regulator of the output voltage control loop. Figure 5 The figure shows the overall block diagram of the voltage feedback control and modulation algorithm based on η and PI regulators.

[0145] This embodiment also relates to a simulation experiment of an asymmetric dual active bridge DC converter. Figure 1 The DC converter topology is used to build a simulation model. Input voltage U i =1200V, output voltage U o =750V, switching frequency f s =10kHz, dead time t d =1μs, transformer ratio 1:n=1:0.5, DC side capacitance C i =C o = 330μF. Based on the differentiated power transmission requirements, the converter's forward power rating is set at 160kW and its reverse power rating is set at 96kW. Therefore, the transformer leakage inductances L1 = 20μH, L2 = 45μH, and L3 = 30μH are set.

[0146] Forward mode / reverse mode switching simulation: A dual-mode modulation strategy based on a unified phase shift duty cycle η is used, and the resulting simulation waveform is as follows: Figure 6 As shown. Before time t, the converter operates in forward mode, transmitting at rated power, and the uniform phase shift duty cycle η is 0.2. When the circuit is in steady state operation, the average DC input current i1 = 133.3A. When t = 0.25s, the converter switches to reverse mode with a rated power of 96kW. At this time, the uniform phase shift duty cycle η changes from 0.2 to -0.1, and the inductor current i L3 When the input current drops to zero, the half-bridge module 1 in the converter stops operating. The average input current i1 = -80A. The above simulation results are basically the same as the theoretical values.

[0147] Forward mode / reverse mode soft switching simulation: Figure 7(a) shows S 11 Drain-source / gate-source voltage and inductor current waveforms; Figure 7(b) shows the S 24Drain-source / gate-source voltage and inductor current waveforms. Figure 8(a) shows the S 11 Drain-source / gate-source voltage and inductor current waveforms; Figure 8(b) shows the S 24 Drain-source / gate-source voltage and inductor current waveforms. It can be seen that the switches in the input full-bridge, output full-bridge, and output half-bridge can all achieve zero-voltage turn-on in both modes.

[0148] Based on the above working principle analysis, a low-power experimental prototype with a bidirectional rated power of 140W / 60W (forward / reverse mode) was built to implement and experimentally verify the proposed topology and modulation method. The experimental results show that the bridge arm midpoint voltage U in the forward / reverse mode is H1 、U H2 And the inductor current i L2 、i L3 The oscilloscope waveforms are shown in FIG9( a ) and FIG9 ( b ), which can verify the feasibility of the PWM modulation algorithm proposed in this application.

[0149] The secondary side of this DC converter adopts a parallel architecture of two full-bridges and one half-bridge, achieving a differentiated power level design. By adjusting the parameters of inductors L2 and L3 to optimize the power distribution ratio, full-bridge two and half-bridge one handle most of the forward transmission power, and full-bridge two handles part of the reverse transmission power. While this architecture improves the overall capacity, it can distribute the total power transmission capacity to a single switch tube, thereby reducing the overall cost of the converter. In addition, combined with an optimized modulation strategy to achieve a balanced distribution of device current stress, the negative impact of high current stress on device failure rate can be reduced. In differentiated application scenarios for bidirectional power transmission, this DC converter can achieve asymmetric power transmission to reduce system cost. In addition, the device current stress of this converter is low, the system failure rate is low, and it has high reliability.

[0150] The above description is merely a specific embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any person skilled in the art can easily conceive of various equivalent modifications or substitutions within the technical scope disclosed in the present invention, and such modifications or substitutions are intended to be within the scope of protection of the present invention. Therefore, the scope of protection of the present invention shall be subject to the scope of protection of the claims.

Claims

1. An asymmetric dual active bridge DC converter, wherein the primary side of the converter is a first active full bridge, and the primary side and the secondary side are connected through a transformer, characterized in that: The secondary side of the converter includes a second active full bridge and a half active bridge connected in parallel; The second active full bridge includes an active switch S 21 To S 24 , active switch S 21 and S 22 Connected in series between the output of the converter and the reference ground, the active switch S 23 and S 24 Connected in series between the output of the converter and the reference ground, the active switch S 21 and S 22 The common end of the two inductors and the secondary side of the transformer are coupled to the active switch S 23 and S 24 The public end; The semi-active bridge includes a diode D 31 、D 33 With active switch S 32 、S 34 ; diode D 31 and active switch S 32 Connected in series between the output of the converter and the reference ground, the diode D 33 and active switch S 34 Connected in series between the output of the converter and the reference ground, the diode D 31 and active switch S 32 The common end of the diode is coupled to the secondary side of the diode D through the inductor and transformer. 33 and active switch S 34 The public end.

2. The asymmetric dual active bridge DC converter according to claim 1, characterized in that: The secondary winding of the transformer is connected to the second active full bridge via the first inductor and the second inductor; the secondary winding of the transformer is connected to the midpoint of the bridge arm of the half-active bridge via the third inductor and the fourth inductor; The sum of the first inductor and the third inductor is denoted as L2, which is called the inductor group L2; the sum of the second inductor and the fourth inductor is denoted as L3, which is called the inductor group L3.

3. The asymmetric dual active bridge DC converter according to claim 1, characterized in that: The converter includes a forward working mode and a reverse working mode; In the forward operating mode, power flows through the first active full bridge, the second active full bridge and the half active bridge; In the reverse operating mode, power flows through the second active full bridge and the first active full bridge.

4. The asymmetric dual active bridge DC converter according to claim 1, characterized in that: The first active full bridge comprises an active switch S 11 To S 14 , active switch S 11 and S 12 Connected in series between the input of the converter and the reference ground, the active switch S 13 and S 14 Connected in series between the input of the converter and the reference ground, the active switch S 11 and S 12 The common end of the active switch S is coupled to the active switch S through the inductor L1 and the primary side of the transformer. 13 and S 14 The public end.

5. A modulation method using the asymmetric dual active bridge DC converter according to any one of claims 1 to 4, characterized in that: The modulation method includes a forward mode modulation sub-method, specifically: The second active full bridge and half active bridge adopt synchronous phase shift modulation strategy, and the active switch S 21 and S 23 Working in synchronous rectification state, the switching period is recorded as T S , where the active switch S 11 、S 14 、S 24 、S 34 Driven by a square wave with a duty cycle of 50%, the active switch S 12 、S 13 、S 22 、S 32 The driving signal is complementary to the active switch S 21 and S 23 Driven by a square wave with a duty cycle of δ; Assume that the converter has reached steady state, define S 11 and S 14 The PWM drive signal is turned on at t0 as the starting moment. Within a switching cycle, seven time points, t0 to t6, are defined in sequence according to the inductor current slope change points. These time points divide the switching cycle into six intervals, the first three of which are positive half-cycles and the last three are negative half-cycles. Definition U i Indicates input voltage, U o Indicates the output voltage. The voltage conversion ratio of the converter is k, which is U o / (nU i ), the equivalent inductance L on the secondary side 11 =n 2 L1,n is the transformer ratio, L1 represents the primary inductance of the transformer, L2 and L3 represent the inductance group L2 and inductance group L3 of the secondary side of the transformer respectively; Define the current flowing through the leakage inductor L1 as i L1 , the current flowing through the second active full bridge is i L2 , the current flowing through the semi-active bridge is i L3 ; In the interval [t0, t1), the voltage U applied to the inductor group L2 at time t L2 (t) and i L2 The expression is: In the interval [t1, t2), the voltage U applied to the inductor group L2 at time t L2 (t) and i L2 The expression is: In the interval [t2, t3), the voltage U applied to the inductor group L2 at time t L2 (t) and i L2 The expression is: The current i in the negative half cycle L2 and voltage U L2 (t) Symmetric to the first half cycle.

6. The modulation method according to claim 5, characterized in that: The current i flowing into the semi-active bridge L3 The current i flowing into the second active full bridge L2 Similar patterns of change.

7. The modulation method according to claim 5, characterized in that: The power distribution between the second active full bridge and the half active bridge is related to the ratio between the inductors L2 and L3, that is, The power values ​​shared by the second active full bridge and the half active bridge are distributed by adjusting the parameters of L2 and L3.

8. The modulation method according to claim 5, characterized in that: The modulation method includes a reverse mode modulation sub-method, including: in the reverse mode, the current i L3 The uncontrollable circulating current power P caused by C ; Using the unidirectional conduction characteristics of the uncontrolled rectifier bridge, the midpoint voltage U H3 Clamped to output voltage U o , the circulating power P can be eliminated C , the following formula must be satisfied: Further simplification yields k≥1 Where k is the voltage conversion ratio.

9. The modulation method according to claim 8, characterized in that: In reverse mode, define the reverse equivalent leakage inductance L eq =(L 11 +L2), active switch S 11 、S 14 、S 21 、S 24 Driven by a square wave with a duty cycle of 50%, S 12 、S 13 、S 22 、S 23 The driving signal complements it and blocks the S in the semi-active bridge at the same time. 32 、S 34 The driving signal; Define θ as S 11 Hysteresis S 21 The phase shift duty cycle, that is, θ = (t1-t0) / Ts; Assuming the circuit has reached steady state, define S 21 and S 24 The PWM drive signal is turned on at t0 as the starting moment. Within a switching cycle, five time points, t0 to t4, are defined in sequence based on the inductor current slope change points. These time points divide the switching cycle into four intervals, the first two of which are positive half-cycles and the last two are negative half-cycles: In the interval (t0, t1), the reverse equivalent leakage inductance L is applied eq Voltage U on Leq and i L2 The expression is: YOU Leq (t)=-(nU i +U o ) In the interval (t1, t2), the reverse equivalent leakage inductance L is applied eq Voltage U on Leq and i L2 The expression is: YOU Leq (t)=(nU i -YOU o ) The negative half cycle is applied to the reverse equivalent leakage inductance L eq Voltage U on Leq and i L2 Symmetrical to the first half of the cycle.

10. The modulation method according to claim 5, characterized in that: The working mode selection and power regulation of the converter are completed by unifying the phase shift duty cycle η, which is defined as For S 11 Advanced S 24 / S 34 The phase shift duty cycle is When η>0, the converter works in the forward mode. and through The phase shift duty cycle δ = f(η) is calculated, where k is the voltage conversion ratio; When η<0, the converter operates in reverse mode, blocking the active switch S 32 and S 34 The driving signal of is given, and -θ = η, the semi-active bridge on the secondary side of the converter stops running; When the output voltage U o When the forward transmission power is reduced, the total forward transmission power P is increased by increasing η in the forward mode. FB To stabilize the output voltage U o In the reverse mode, the reverse transmission power P3 is reduced by increasing η, that is, reducing θ to stabilize the output voltage U o .

Images