Semiconductor laser systems, control methods, and atomic magnetometers

Abstract

This application provides a semiconductor laser system, control method, and atomic magnetometer, applicable to the field of atomic magnetometer technology. The system includes a control module, a constant current source circuit module, and a laser module. The constant current source circuit module includes a coarse adjustment circuit module and a fine adjustment circuit module. The fine adjustment circuit module includes a main circuit and a compensation circuit. The first terminal of the main circuit is connected to the second output terminal of the control module, and the second terminal is connected to the input terminal of the laser module. The input terminal of the compensation circuit is connected to the input terminal of the laser module, and the output terminal is connected to the first terminal of the main circuit. The compensation circuit is configured such that, when the second signal is at a zero level, the main current in the main circuit and the compensation current in the compensation circuit cancel each other out at the input terminal of the laser module. This application avoids the influence of voltage changes in the coarse adjustment circuit module on the fine adjustment circuit module, ensuring current accuracy and improving the stability of the semiconductor laser.

Description

Technical Field

[0001] This application relates to the field of atomic magnetometer technology, specifically to a semiconductor laser system, control method, and atomic magnetometer. Background Technology

[0002] Spin Exchange Relaxation Free (SERF) atomic magnetometers are highly sensitive magnetic field detection devices that rely on circularly polarized lasers to pump atoms to achieve polarization and measure the magnetic field through changes in atomic energy levels. The control performance of the laser has a crucial impact on the measurement accuracy and stability of the SERF atomic magnetometer. Distributed Bragg Reflector (DBR) lasers are currently commonly used semiconductor lasers in SERF atomic magnetometers; they achieve population inversion and laser emission through current injection.

[0003] Improving the frequency stability of DBR lasers is crucial for ensuring the performance of atomic magnetometers. The output frequency of DBR lasers is highly sensitive to driving current and operating temperature; fluctuations in current or changes in temperature can cause laser frequency drift, affecting the resonance matching between the laser and atoms, and consequently reducing the polarization efficiency and magnetic sensitivity of the magnetometer. However, current adjustment range and precision of DBR lasers in existing technologies are limited, making it difficult to meet the long-term frequency stability requirements of magnetometers. Summary of the Invention

[0004] In view of the above problems, this application provides a semiconductor laser system, control method and atomic magnetometer for improving laser stability.

[0005] According to a first aspect of this application, a semiconductor laser system is provided, comprising: a control module, a constant current source circuit module, and a laser module; the constant current source circuit module includes a coarse adjustment circuit module and a fine adjustment circuit module; the input terminal of the coarse adjustment circuit module is connected to a first output terminal of the control module, and the output terminal is connected to the input terminal of the laser module, for outputting a first current to the laser module according to a first signal output from the first output terminal; the input terminal of the fine adjustment circuit module is connected to a second output terminal of the control module, and the output terminal is connected to the input terminal of the laser module, for outputting a second current to the laser module according to a second signal output from the second output terminal; wherein the second current is less than the first current; the laser module is used to emit laser light under the injection of the first current and the second current; the fine adjustment circuit module includes a main circuit and a compensation circuit; a first terminal of the main circuit is connected to the second output terminal of the control module, and a second terminal is connected to the input terminal of the laser module; the input terminal of the compensation circuit is connected to the input terminal of the laser module, and the output terminal is connected to the first terminal of the main circuit; the compensation circuit is configured such that, when the second signal is at a zero level, the main current in the main circuit and the compensation current in the compensation circuit cancel each other out at the input terminal of the laser module.

[0006] According to an embodiment of this application, the compensation circuit includes an inverting amplifier circuit and a first resistor; the first end of the first resistor is connected to the output end of the inverting amplifier circuit, and the second end is connected to the first end of the main circuit; the inverting input end of the inverting amplifier circuit is connected to the input end of the laser module; the amplification factor of the inverting amplifier circuit is -k; where k is a positive number; the resistance value of the first resistor is k times the resistance value of the main circuit.

[0007] According to an embodiment of this application, the compensation circuit includes a non-inverting amplifier circuit, an inverting amplifier circuit, and a first resistor; the non-inverting input terminal of the non-inverting amplifier circuit is connected to the input terminal of the laser module, the output terminal of the non-inverting amplifier circuit is connected to the inverting input terminal of the inverting amplifier circuit, the output terminal of the inverting amplifier circuit is connected to the first terminal of the first resistor, and the second terminal of the first resistor is connected to the first terminal of the main circuit; the amplification factor of the non-inverting amplifier circuit is k, and the amplification factor of the inverting amplifier circuit is -1; where k is a positive number; the resistance value of the first resistor is k times the resistance value of the main circuit.

[0008] According to an embodiment of this application, the coarse adjustment circuit module includes a voltage module, a subtractor circuit module, and a current output circuit module. The non-inverting input terminal of the subtractor circuit module is connected to the output terminal of the voltage module, the inverting input terminal is connected to the first output terminal of the control module, and the output terminal is connected to the current output circuit module to output a differential signal between a first signal and a reference voltage signal to the current output circuit module. The reference voltage signal is provided by the voltage module. The current output module includes a comparison feedback circuit, a first sampling resistor, and a power transistor. The first terminal of the first sampling resistor is connected to the output terminal of the voltage module, and the second terminal is connected to the source of the power transistor. The comparison feedback circuit includes an operational amplifier. The non-inverting input terminal of the operational amplifier is connected to the output terminal of the subtractor circuit module, the inverting input terminal is connected to the second terminal of the first sampling resistor, and the output terminal is connected to the gate of the power transistor. The drain of the power transistor is connected to the input terminal of the laser module.

[0009] According to an embodiment of this application, the first sampling resistor includes a plurality of shunt resistors, which are connected in parallel or in a combination of series and parallel connections.

[0010] According to an embodiment of this application, the system further includes a monitoring circuit module. The input terminal of the monitoring circuit module is connected to the laser module, and the output terminal is connected to the first input terminal of the control module. The monitoring circuit module is used to obtain the current value of the laser module so that the control module can monitor the working status of the laser module according to the relationship between the current value and the target current value, wherein the target current value is the sum of the first current and the second current.

[0011] According to an embodiment of this application, the system further includes a temperature control module; the temperature control module includes a thermoelectric cooler, a first temperature sampling module, and a temperature control circuit module; the thermoelectric cooler is disposed within the laser module; the input terminal of the first temperature sampling module is disposed within the laser module, and its output terminal is connected to the control module; the input terminal of the temperature control circuit module is connected to the third output terminal of the control module, and its current output terminal is connected to the thermoelectric cooler, for changing the magnitude and / or direction of the current input to the thermoelectric cooler according to the third signal output from the third output terminal; the control module is used to output a third signal to the temperature control circuit module through the third output terminal according to the temperature sampling signal output by the first temperature sampling module, so as to change the operating state of the thermoelectric cooler by changing the magnitude and / or direction of the current input to the thermoelectric cooler.

[0012] According to an embodiment of this application, the temperature control circuit module further includes a second sampling resistor and a current sampling circuit module; the second sampling resistor is connected in series with the thermoelectric cooler to obtain the magnitude of the current passing through the thermoelectric cooler; the input terminal of the current sampling circuit module is connected to the second sampling resistor, and the output terminal is connected to the control module to feed back the obtained magnitude of the current of the thermoelectric cooler to the control module.

[0013] According to an embodiment of this application, the temperature control module further includes a second temperature sampling module, which is used to acquire the ambient temperature at the constant current source circuit module and feed the ambient temperature back to the control module; the control module has a pre-stored current temperature compensation lookup table, which includes current compensation values ​​corresponding to different temperatures; the control module is configured to look up the corresponding current compensation value in the current temperature compensation lookup table based on the ambient temperature obtained by the second temperature sampling module, and correct the output current of the constant current source circuit module by changing the second signal.

[0014] A second aspect of this application provides a semiconductor laser control method for the aforementioned semiconductor laser system, comprising: acquiring a required current of a laser module; splitting the required current into a first current and a second current; outputting a first signal through a first output terminal of a control module to output the first current based on a coarse adjustment circuit module; and outputting a second signal through a second output terminal of the control module to output the second current based on a fine adjustment circuit module.

[0015] According to an embodiment of this application, the method further includes: acquiring a temperature signal sampled from the ambient temperature; acquiring a current compensation value corresponding to the temperature signal based on the temperature signal and a pre-stored current-temperature compensation lookup table; and generating a second signal based on the current compensation value.

[0016] A third aspect of this application provides an atomic magnetometer, including the semiconductor laser system described above.

[0017] The above-described one or more embodiments have the following beneficial effects: A constant current source circuit module including a coarse adjustment circuit module and a fine adjustment circuit module is designed. The coarse adjustment circuit module can output a wide range of currents, while the fine adjustment circuit module can output a small range of high-precision currents, thereby achieving wide-range and high-resolution current adjustment and meeting the laser's requirements for current adjustment range and accuracy. In the fine adjustment circuit module, a compensation circuit is designed so that the voltage at the circuit output terminal cancels out the current generated in the main circuit and the compensation circuit, adjusting the potential at the first terminal of the main circuit to 0, thereby forming an equivalent virtual ground at the input terminal of the fine adjustment circuit module. This avoids the influence of voltage changes in the coarse adjustment circuit module on the fine adjustment circuit module, ensuring current accuracy. Attached Figure Description

[0018] The above-mentioned contents, other objects, features and advantages of this application will become clearer from the following description of embodiments with reference to the accompanying drawings, in which:

[0019] Figure 1 This schematic diagram illustrates the overall structure of a semiconductor laser system according to an embodiment of this application.

[0020] Figure 2A schematic diagram of the current control section of a semiconductor laser system according to an embodiment of this application is shown.

[0021] Figure 3 The diagram schematically illustrates a circuit diagram of a constant current source circuit module of a semiconductor laser system according to an embodiment of this application;

[0022] Figure 4 A schematic diagram of the temperature control section of a semiconductor laser system according to an embodiment of this application is shown.

[0023] Figure 5 A circuit diagram of a temperature control circuit module of a semiconductor laser system according to an embodiment of this application is shown schematically.

[0024] Figure 6 A schematic diagram illustrating the temperature control flowchart of a semiconductor laser system according to an embodiment of this application is shown. Detailed Implementation

[0025] The embodiments of this application will now be described with reference to the accompanying drawings. However, it should be understood that these descriptions are exemplary only and are not intended to limit the scope of this application. In the following detailed description, numerous specific details are set forth to provide a thorough understanding of the embodiments of this application for ease of explanation. However, it will be apparent that one or more embodiments may be implemented without these specific details. Furthermore, descriptions of well-known structures and technologies are omitted in the following description to avoid unnecessarily obscuring the concepts of this application.

[0026] The terminology used herein is for the purpose of describing particular embodiments only and is not intended to limit the scope of this application. The terms “comprising,” “including,” etc., as used herein indicate the presence of the stated features, steps, operations, and / or components, but do not exclude the presence or addition of one or more other features, steps, operations, or components.

[0027] All terms used herein (including technical and scientific terms) have the meanings commonly understood by those skilled in the art, unless otherwise defined. It should be noted that the terms used herein are to be interpreted in a manner consistent with the context of this specification, and not in an idealized or overly rigid way.

[0028] When using expressions such as "at least one of A, B and C", they should generally be interpreted in accordance with the meaning that is commonly understood by those skilled in the art (e.g., "a system having at least one of A, B and C" should include, but is not limited to, a system having A alone, a system having B alone, a system having C alone, a system having A and B, a system having A and C, a system having B and C, and / or a system having A, B and C, etc.).

[0029] To address the problem that the current adjustment range and accuracy of existing DBR lasers are limited, making it difficult to meet the long-term stability requirements of magnetometers for laser frequency, this application provides a semiconductor laser system, including: a control module, a constant current source circuit module, and a laser module. The constant current source circuit module includes a coarse adjustment circuit module and a fine adjustment circuit module. The input terminal of the coarse adjustment circuit module is connected to the first output terminal of the control module, and the output terminal is connected to the input terminal of the laser module. It is used to output a first current to the laser module according to the first signal output from the first output terminal. The input terminal of the fine adjustment circuit module is connected to the second output terminal of the control module, and the output terminal is connected to the input terminal of the laser module. It is used to output a second current to the laser module according to the second signal output from the second output terminal. The second current is less than the first current. The laser module is used to emit laser light under the injection of the first current and the second current. The fine adjustment circuit module includes a main circuit and a compensation circuit. The first terminal of the main circuit is connected to the second output terminal of the control module, and the second terminal is connected to the input terminal of the laser module. The input terminal of the compensation circuit is connected to the input terminal of the laser module, and the output terminal is connected to the first terminal of the main circuit. The compensation circuit is configured such that when the second signal is at a zero level, the main current in the main circuit and the compensation current in the compensation circuit cancel each other out at the input terminal of the laser module.

[0030] In the embodiments of this application, a constant current source circuit module is designed, including a coarse adjustment circuit module and a fine adjustment circuit module. The coarse adjustment circuit module can output a wide range of currents, while the fine adjustment circuit module can output a small range of high-precision currents, thereby achieving wide-range and high-resolution current regulation to meet the laser's requirements for current regulation range and accuracy. In the fine adjustment circuit module, a compensation circuit is designed so that the main current in the main circuit and the compensation current in the compensation circuit cancel each other out at the input terminal of the laser module, adjusting the potential of the first terminal of the main circuit to 0. This forms an equivalent virtual ground at the input terminal of the fine adjustment circuit module, avoiding the influence of voltage changes in the coarse adjustment circuit module on the fine adjustment circuit module and ensuring current accuracy.

[0031] Figure 1 A schematic diagram of the overall structure of a semiconductor laser system according to an embodiment of this application is shown. Figure 1 As shown, the semiconductor laser system includes a host computer, a control module, a constant current source circuit module, and a laser module. In some embodiments, a temperature control module is also included to further achieve temperature control and improve laser accuracy.

[0032] In the embodiments of this application, the control module is implemented based on FPGA (Field Programmable Gate Array), integrates specific control algorithms, is responsible for the real-time scheduling and management of current and / or temperature control signals, and can communicate with the host computer in real time to meet the real-time monitoring requirements.

[0033] Figure 2 A schematic diagram of the current control section of a semiconductor laser system according to an embodiment of this application is shown. Figure 2 As shown in the embodiments of this application, the current control includes two parts: coarse adjustment control and fine adjustment compensation. The first output terminal of the control module uses a setting DAC (Digital-to-Analog Converter) control circuit to output a first signal for coarse current adjustment. The setting DAC control circuit includes a DAC conversion circuit, whose input terminal is connected to the output terminal of the control module, and whose output terminal outputs the first signal, which is connected to the input terminal of the coarse adjustment circuit module, to convert the digital signal output by the control module into an analog voltage signal. The second output terminal of the control module uses a fine adjustment DAC control circuit to output a second signal for fine current adjustment. The fine adjustment DAC control circuit includes a DAC conversion circuit, whose input terminal is connected to the output terminal of the control module, and whose output terminal outputs the second signal, which is connected to the input terminal of the fine adjustment circuit module, to convert the digital signal output by the control module into an analog voltage signal. In some embodiments, the setting DAC control circuit and the fine adjustment DAC control circuit can adopt the same circuit structure.

[0034] The first signal is a voltage signal. The coarse adjustment circuit module receives the first signal, generates a corresponding first current based on the voltage magnitude of the first signal, and outputs it to the laser module. The second signal is also a voltage signal. The fine adjustment circuit module receives the second signal, generates a corresponding second current based on the voltage magnitude of the second signal, and outputs it to the laser module. The magnitude of the second current is smaller than the first current, but its accuracy is higher than the first current, to further improve the current accuracy input to the laser module. In the embodiments of this application, the output range of the first current (IB) is 0~200mA. Based on the implementation of a 16-bit DAC (reference voltage 3.3V) and a 10Ω sampling resistor (RS), its theoretical adjustment accuracy is approximately 5.04μA, used for a wide range of power settings for the laser. The output range of the second current (IL) is -3.3mA~+3.3mA. Based on the implementation of a 16-bit DAC (reference voltage 3.3V) and a 1kΩ sampling resistor (R19), its theoretical adjustment accuracy is as high as 50.35nA. With an accuracy approximately two orders of magnitude higher than that of the coarse adjustment channel, the fine adjustment channel can accurately compensate for and correct the quantization step (5.04μA) of the coarse adjustment current, thereby achieving high-resolution current control across the entire range.

[0035] Considering that the outputs of both the coarse and fine adjustment circuit modules are connected to the laser module to supply current, the coarse adjustment circuit module will generate a voltage at the input of the laser module, which is the operating voltage of the laser module. To prevent this voltage from interfering with the output of the fine adjustment circuit module, a compensation circuit is designed in the fine adjustment circuit module. This ensures that the current generated by the compensation circuit cancels out the current generated in the main circuit. When the second signal voltage is zero, the fine adjustment circuit module does not draw current from the laser module (i.e., the current flowing from the laser module to the fine adjustment circuit module is zero). This allows the second current generated by the second signal through the main circuit to flow directly into the laser without interference, isolating the fine and coarse adjustment loops. This achieves high-precision, low-interference laser current fine-tuning control, ensuring the independence and stability of the coarse and fine adjustment circuit modules during operation.

[0036] Figure 3 A schematic diagram of a constant current source circuit module for a semiconductor laser system according to an embodiment of this application is shown. Figure 3 As shown in the embodiment of this application, the compensation circuit includes a non-inverting amplifier circuit, an inverting amplifier circuit, and a first resistor. The non-inverting input terminal of the non-inverting amplifier circuit is connected to the input terminal of the laser module, the output terminal of the amplifier circuit is connected to the inverting input terminal of the inverting amplifier circuit, the output terminal of the inverting amplifier circuit is connected to the first terminal of the first resistor, and the second terminal of the first resistor is connected to the first terminal of the main circuit. The amplification factor of the non-inverting amplifier circuit is k, and the amplification factor of the inverting amplifier circuit is -1. At this time, the resistance value of the first resistor is k times the resistance value of the main circuit.

[0037] The non-inverting amplifier circuit consists of operational amplifier U3, resistors R14 and R15. To provide a current path, resistor R16 is also connected to the output of operational amplifier U3. Specifically, the non-inverting input of operational amplifier U3 is connected to the input of the laser module, the inverting input is grounded through resistor R14, and the output is connected to the inverting input through resistor R15 to form a feedback loop. The first end of resistor R16 is connected to the output of operational amplifier U3, and the second end is connected to the input of the laser module. The amplification factor k of the non-inverting amplifier circuit is determined by the ratio of the resistance values ​​of resistors R15 and R14, k = 1 + R15 / R14.

[0038] The inverting amplifier circuit consists of operational amplifier U4, resistors R17, R18, and R20. Specifically, the non-inverting input of operational amplifier U4 is grounded through resistor R18, the inverting input is connected to the output of the non-inverting amplifier circuit through resistor R17, and the output is connected to the inverting input through resistor R20 to form a feedback loop. The amplification factor of the inverting amplifier circuit is determined by the ratio of the resistance values ​​of resistors R20 and R17. In the embodiment of this application, R17 = R20, so the amplification factor of the inverting amplifier circuit is -1, forming a unity-gain inverter.

[0039] Let the input voltage of the laser module be VLD. After passing through the non-inverting and inverting amplifier circuits, the voltage output to the first terminal of the first resistor R21 is -kVLD. Therefore, the current from the compensation circuit through the first resistor R21 to the first terminal of the main circuit (i.e., the input terminal of the fine-tuning circuit module) is -kVLD / R21. To cancel the current in the main circuit, the potential at the first terminal of the main circuit needs to be 0. In the embodiment of this application, the main circuit includes a resistor R19, so the resistance of the main circuit is equal to the resistance of R19. The current generated by the main circuit through VLD is VLD / R19. Therefore, as long as VLD / R19 = kVLD / R21, the potential at the first terminal of the main circuit can be made 0. Rearranging, we get R21 = kR19, meaning the resistance of the first resistor is k times the resistance of the main circuit. Furthermore, the current through resistor R16 is VLD / R16, which is equal to the current flowing through resistor R19, so R16 = R19. At this time, an equivalent virtual ground is formed at the input of the fine-tuning circuit module, avoiding the influence of the main circuit voltage change on the fine-tuning circuit; the constant current source fine-tuning voltage, that is, the voltage value VL of the second signal output by the control module through the second output terminal, can be directly injected into the laser module by the second current IL=VL / R19 generated by the resistor R19, acting only on the laser, without affecting the path of the first current, and is not affected by the first current.

[0040] This design effectively isolates the fine-tuning circuit module from the coarse-tuning circuit module, achieving high-precision, low-interference laser current fine-tuning control.

[0041] In some other embodiments, an inverting amplifier circuit and a first resistor can be used to build a compensation circuit instead of a non-inverting amplifier circuit. In this case, the first terminal of the first resistor is connected to the output terminal of the inverting amplifier circuit, and the second terminal is connected to the first terminal of the main circuit. The inverting input terminal of the inverting amplifier circuit is connected to the input terminal of the laser module. The amplification factor of the inverting amplifier circuit is -k, where k is a positive number. The resistance value of the first resistor is k times the resistance value of the main circuit. The difference from the previous embodiment is that the resistance values ​​of resistor R20 and resistor R17 are different, and the amplification factor of the inverting amplifier circuit is not -1, but -k, thus eliminating the need to build an additional non-inverting amplifier circuit.

[0042] In the embodiments of this application, the coarse adjustment circuit module includes a voltage module, a subtractor circuit module, and a current output circuit module. The non-inverting input of the subtractor circuit module is connected to the output of the voltage module, the inverting input is connected to the first output of the control module, and the output is connected to the current output circuit module to output a differential signal between the first signal and the reference voltage signal to the current output circuit module. The reference voltage signal is provided by the voltage module and input to the subtractor circuit module via its output. The current output module includes a comparison feedback circuit, a first sampling resistor, and a power transistor. The first end of the first sampling resistor is connected to the output of the voltage module, and the second end is connected to the source of the power transistor. The comparison feedback circuit includes an operational amplifier. The non-inverting input of the operational amplifier is connected to the output of the subtractor circuit module, the inverting input is connected to the second end of the first sampling resistor, and the output is connected to the gate of the power transistor. This generates a control signal based on the difference between the differential signal output by the subtractor circuit module and the voltage at the second end of the sampling resistor. The magnitude of the control signal output controls the current flowing through the source and drain of the power transistor. The drain of the power transistor serves as the output of the current output circuit module and is connected to the input of the laser module.

[0043] In the embodiments of this application, the voltage module employs a low dropout regulator (LDO) to provide a stable reference voltage VCC and limit the input voltage to protect the laser and prevent overcurrent. In some embodiments, the voltage module further includes a soft-start circuit and an inductor L1. The soft-start circuit is connected in series with the LDO between the first sampling resistor and the LDO, and is used to gradually change the output voltage during circuit power-on by utilizing the charging and discharging characteristics of the capacitor, so as to suppress the surge current at the moment of power-on and ensure smooth circuit startup. The inductor L1 is connected in series with the soft-start circuit to further smooth the current and reduce current surges.

[0044] In the embodiments of this application, the voltage output of the subtractor circuit module composed of operational amplifier U1, resistors R1, R2, and R3 is the difference between the voltage value VCC of the voltage module and the voltage value VB of the first signal, VCC-VB. This voltage value is connected to the second terminal of the first sampling resistor Rs. The first terminal of the first sampling resistor Rs is connected to VCC. Therefore, the magnitude of the first current IB flowing through the first sampling resistor Rs can be expressed as: IB=(VCC-(VCC-VB)) / Rs=VB / Rs. This allows control of the first current by the magnitude of the input first signal, thereby achieving coarse current adjustment.

[0045] To suppress current drift caused by temperature changes and improve the thermal stability of the output current, this application employs temperature compensation technology. The core of this technology lies in ensuring that the voltage value VB of the first signal and the first sampling resistor Rs have matched, unidirectional temperature coefficient change characteristics. That is, when the temperature changes, the values ​​of VB and Rs change in the same direction (either increasing or decreasing simultaneously), thereby producing a mutually canceling effect in the current calculation formula IB = VB / Rs, maintaining current stability.

[0046] In the embodiments of this application, an RC low-pass filter circuit composed of resistor R4, capacitor C1, resistor R5, and capacitor C2 is further provided between the subtractor circuit module and the current output circuit module to perform low-pass filtering on the output signal of the subtractor circuit, thereby suppressing high-frequency noise and improving system stability.

[0047] To reduce heat dissipation and the risk of thermal runaway, the first sampling resistor Rs is connected in parallel to shunt a large current. The first sampling resistor includes multiple shunt resistors, which are connected in parallel or in a combination of series and parallel connections. Specifically, in the embodiments of this application, the first sampling resistor uses a combination of series and parallel dual resistors to shunt a large current. The shunt resistors include resistors R8, R9, R10, and R11, where resistors R8 and R9 are connected in series, and resistors R10 and R11 are connected in parallel. The resistance value of the first sampling resistor Rs can be expressed as Rs = (R8 + R9) / / (R10 + R11), where / / indicates that the resistors are connected in parallel.

[0048] In some implementations, current can be shunted using multiple resistors connected in parallel. However, compared to shunting current using only parallel resistors, using a combination of series and parallel resistors offers advantages such as distributing power consumption, preventing overheating, and enhancing reliability. Specifically, regarding power distribution, resistors in a circuit consume power and generate heat. Using a combination of multiple resistors distributes the total power consumption across multiple resistors, preventing a single resistor from bearing excessive power, thereby improving the circuit's reliability and stability. Regarding preventing overheating, the resistance value changes with temperature. A single high-power resistor generates concentrated heat, resulting in a higher temperature rise and more severe resistance drift. Multiple resistors distribute heat, with each resistor experiencing a smaller temperature rise, helping to maintain the overall resistance value's stability and preventing overheating from affecting accuracy. Regarding enhanced reliability, if one resistor fails (open circuit or short circuit), in a series and parallel combination structure, the entire circuit may not completely fail, or its performance change will not be as drastic, improving the circuit's robustness and avoiding single-point failures.

[0049] In the embodiments of this application, the comparison feedback circuit includes an operational amplifier U2, a resistor R6, and a capacitor C3. The power transistor is a PMOS (P-channel Metal-Oxide-Semiconductor Field-Effect Transistor) Q1. The output of the subtractor circuit module is connected to the non-inverting input of the operational amplifier U2, while the inverting input of the operational amplifier U2 is connected to the second terminal of the first sampling resistor Rs, i.e., the source of the PMOS transistor Q1. The output of the operational amplifier U2 is connected to the gate of the PMOS transistor Q1 through the resistor R6. One end of the capacitor C3 is connected between the resistor R6 and the PMOS transistor Q1, and the other end is connected to the inverting input of the operational amplifier U2. The drain of the PMOS transistor Q1 is connected to the input of the laser module.

[0050] PMOS transistor Q1 is used to enhance current output capability, enabling rapid turn-on and turn-off of the circuit. Resistor R6 is used to suppress oscillation and improve circuit stability. Due to the parasitic capacitance at the base of the transistor, oscillation (high-frequency self-oscillation) can easily occur when the op-amp output directly drives a capacitive load. The series resistor R6 isolates the capacitive load and forms a damping effect with the parasitic capacitance, effectively suppressing oscillation and making the system operate more stably. The value of resistor R6 determines the magnitude of the gate current flowing into PMOS transistor Q1, achieving current limiting protection.

[0051] In the embodiments of this application, a resistor R7 is also connected in parallel across the first sampling resistor Rs. The resistor R7, together with the capacitor C3, can further filter out high-frequency ripple and noise in the output signal of the operational amplifier U2.

[0052] The voltage VCC-VB output by the subtractor circuit module can control the conduction state of PMOS transistor Q1. Under the action of operational amplifier U2, the voltage across the first sampling resistor Rs is equal to VB, thereby stabilizing the output current.

[0053] Specifically, the comparison feedback circuit and PMOS transistor Q1 form a constant current source circuit. The drain output current of PMOS transistor Q1 is the first current IB flowing through the first sampling resistor Rs. The voltage at the second terminal of the first sampling resistor Rs can be expressed as VCC - IB * Rs. This voltage is fed back to the inverting input terminal of operational amplifier U2 and compared with the voltage value VCC - VB at the non-inverting input terminal of the subtractor circuit module. As a differential amplifier, operational amplifier U2 continuously adjusts its output voltage (i.e., the gate voltage of PMOS transistor Q1), thereby changing the drain current of PMOS transistor Q1. The adjustment goal of operational amplifier U2 is to make the voltages at its two input terminals equal, i.e., VCC - VB = VCC - IB * Rs.

[0054] When an increase in output current is required, i.e., when the voltage value VB of the first signal increases, the circuit adjusts through the following negative feedback process: First, the increase in VB causes the output voltage of the subtractor circuit to decrease, and this voltage serves as the voltage V at the non-inverting input terminal of operational amplifier U2. + (V) + =VCC-VB), when the voltage decreases, V + Instantaneously lower than the voltage V at the inverting input terminal of operational amplifier U2 - (V) - =VCC-IB×Rs), thus generating a negative error voltage at the input of operational amplifier U2. Subsequently, operational amplifier U2, as a high-gain error amplifier, will immediately respond to this negative error. Its output voltage will decrease significantly, and this voltage is applied to the gate of PMOS transistor Q1. For PMOS transistors, the increase in gate voltage leads to an increase in the gate-source voltage |V_GS| of Q1, thereby deepening its conduction channel and causing an increase in drain current IB. As IB increases, the voltage drop across sampling resistor Rs, IB×Rs, increases accordingly, thereby increasing the voltage V at the inverting input of operational amplifier U2. - Decrease. This causes V to... - The voltage automatically adjusts and follows the reduced V. + Voltage change. The above adjustment process continues, forming a negative feedback loop, until V... - With V + Once the voltages are approximately equal again, the error voltage approaches zero, and the circuit reaches a stable state once more. At this point, the magnitude of the first current IB output from the coarse adjustment circuit module to the laser module can be expressed as: IB = (VCC - (VCC - VB)) / Rs = VB / Rs.

[0055] Similarly, when the voltage value VB of the first signal decreases, VCC-VB increases, that is, the voltage at the non-inverting input terminal of operational amplifier U2 increases. At this time, the output voltage of operational amplifier U2 decreases, the gate voltage of PMOS transistor Q1 decreases, and the drain current of PMOS transistor Q1 decreases accordingly. This leads to an increase in VCC-IB*Rs, that is, an increase in the voltage at the inverting input terminal of operational amplifier U2, until the voltages at the non-inverting and inverting input terminals of operational amplifier U2 are approximately equal and reach a stable state. At this time, the magnitude of the first current IB output from the coarse adjustment circuit module to the laser module can also be expressed as: IB=(VCC-(VCC-VB)) / Rs=VB / Rs.

[0056] The constant current source circuit module utilizes a high-precision DAC, low-temperature drift precision resistors, and a low-noise operational amplifier, significantly improving the resolution and stability of current control, effectively suppressing system noise, and providing a reliable guarantee for the long-term stable operation of the SERF atomic magnetometer. The circuit provides a high-precision, low-noise, adjustable constant current source output to the laser. Its output current is set via an FPGA-controlled DAC, achieving a two-stage current control mechanism of coarse and fine adjustment to meet the current regulation requirements under different operating conditions.

[0057] The laser module includes a laser diode D2, which serves as the laser light source for laser emission. A resistor R13 is connected in series with the laser diode D2 for current limiting protection. Capacitors C4 and C5 are connected in parallel with the laser diode D2 for filtering and noise reduction. A diode D1 is also connected in reverse parallel across the laser diode D2 to discharge reverse current and protect the circuit from damage caused by reverse induced voltage. A resistor R12 and an inductor L2 are also connected in parallel at the input of the laser module for filtering and current limiting protection.

[0058] In the embodiments of this application, the constant current source circuit module further includes a monitoring circuit module. The input terminal of the monitoring circuit module is connected to the laser module, specifically to the cathode of the laser diode D2, with the location of the cathode of the laser diode D2 serving as the current sampling point. The output terminal of the monitoring circuit module is connected to the first input terminal of the control module. The monitoring circuit module is used to acquire the current value of the laser module, i.e., the current value flowing through the laser diode D2, which is the sum of the first current output by the coarse adjustment circuit module and the second current output by the fine adjustment circuit module. This current is then fed back to the control module so that the control module can monitor the current value flowing through the laser diode D2 in real time. Based on the relationship between this current value and the target current value, the control module monitors the operating status of the laser module. The target current value is the sum of the first current corresponding to the first signal and the second current corresponding to the second signal. This allows for the immediate identification of abnormal states such as overcurrent, open circuit, or short circuit, triggering corresponding alarms and executing corresponding hardware protection mechanisms, thereby effectively preventing permanent damage to the laser due to unexpected operating conditions.

[0059] In the embodiments of this application, the monitoring circuit includes a buffer module and an amplifier module. The buffer module is connected to the cathode of the laser diode D2 and is used to isolate the impedance before and after the sampling point to avoid measurement errors caused by impedance mismatch; the amplifier module is connected to the buffer module and is used to amplify the sampling voltage at the sampling point to facilitate subsequent signal conversion operations.

[0060] Specifically, the buffer module includes operational amplifier U5. The non-inverting input of operational amplifier U5 is connected to the cathode of laser diode D2, and the inverting input is directly connected to the output. The buffer module utilizes the high input impedance of the operational amplifier to effectively isolate the impedance influence of subsequent circuits at the sampling point, fundamentally avoiding measurement errors caused by impedance mismatch. The amplifier module includes operational amplifier U6, resistors R22 and R23. The non-inverting input of operational amplifier U6 is connected to the output of operational amplifier U5, and the inverting input is grounded through resistor R22. The inverting input is also connected to the output through resistor R23. The output is connected to the first input of the control module through an ADC (Analog-to-Digital Converter) circuit, outputting the current monitoring signal ILD. The amplification gain of the amplifier module is determined by the resistance values ​​of resistors R22 and R23. The amplification factor is equal to the ratio of the resistance values ​​of resistors R23 and R22, thereby amplifying the small-range sampled voltage signal to an ideal range suitable for ADC acquisition. This allows the control module to accurately obtain the current magnitude at the sampling point for monitoring.

[0061] Besides current fluctuations, temperature changes also affect the accuracy of DBR lasers. The output frequency of DBR lasers is highly sensitive to operating temperature; any temperature change will cause laser frequency drift, affecting the laser's resonance matching with atoms, and thus reducing the polarization efficiency and magnetic sensitivity of the magnetometer. Furthermore, the internal temperature of the laser gradually increases during operation, and even small temperature fluctuations can cause significant changes in the laser frequency; at the same time, external magnetic fields, circuit noise, and environmental interference will further exacerbate frequency instability.

[0062] Based on this, in the embodiments of this application, a temperature control module is further designed to realize a DBR laser control system design with high-precision temperature control, wide-range low-noise current drive capability, and excellent anti-interference performance and protection mechanism, thereby improving the measurement accuracy and long-term stability of the atomic magnetometer.

[0063] In embodiments of this application, the semiconductor laser system further includes a temperature control module; the temperature control module includes a thermoelectric cooler, a first temperature sampling module, and a temperature control circuit module; the thermoelectric cooler is disposed within the laser module; the input terminal of the first temperature sampling module is disposed within the laser module, and its output terminal is connected to the control module; the input terminal of the temperature control circuit module is connected to the third output terminal of the control module, and its current output terminal is connected to the thermoelectric cooler, used to change the magnitude and direction of the current input to the thermoelectric cooler according to the third signal output from the third output terminal; the control module is used to output a third signal to the temperature control circuit module through the third output terminal according to the temperature sampling signal output by the first temperature sampling module, so as to change the operating state of the thermoelectric cooler by changing the magnitude and direction of the current input to the thermoelectric cooler.

[0064] Figure 4 A schematic diagram of the temperature control section of a semiconductor laser system according to an embodiment of this application is shown. Figure 1 , Figure 4 As shown, the laser module contains a thermoelectric cooler (TEC) and an NTC thermistor (Negative Temperature Coefficient Thermistor). The thermoelectric cooler is used to heat or cool the laser module, ensuring temperature stability within the module. The first temperature sampling module includes a first NTC thermistor, a temperature measurement circuit, and an ADC sampling module. The first NTC thermistor is located inside the laser module and is used to acquire the internal temperature of the laser module in real time, i.e., the junction temperature of the laser module, and generate a temperature signal. The temperature measurement circuit filters and amplifies the temperature signal generated by the first NTC thermistor and inputs it to the ADC sampling module. The ADC sampling module performs ADC conversion on the processed temperature signal and sends the converted signal to the control module, thereby controlling the thermoelectric cooler to achieve temperature feedback control. The control module can generate a PID (Proportion Integration Differentiation) output based on the converted signal, and output a third signal through the third output terminal. After being converted by a DAC and processed by the temperature control circuit, the output signal sets the current required by the TEC. By limiting the DAC output, current limiting and voltage limiting functions can be achieved, which can protect the TEC of the DFB laser from damage due to excessive current.

[0065] In the embodiments of this application, the control module compares the temperature signal with the reference voltage corresponding to the set temperature to obtain a temperature error signal. The temperature error signal is then processed by a PID based on Kalman filtering to generate a drive signal (i.e., the third signal) to adjust the current direction and magnitude of the TEC to achieve cooling or heating. The role of the TEC is to change the operating temperature of the laser.

[0066] Figure 5 A schematic diagram of a temperature control circuit module for a semiconductor laser system according to an embodiment of this application is shown. Figure 5 As shown, the temperature control circuit module includes a main control amplifier circuit, a gate drive circuit, and an H-bridge circuit. The H-bridge circuit includes switching elements located on the four bridge arms and a thermoelectric cooler (TEC) located at the center. In embodiments of this application, the switching elements are metal-oxide-semiconductor field-effect transistors (MOSFETs). The on / off states of the diagonally located switching elements are configured to be the same to achieve a change in the current direction for the TEC at the center. Figure 5 Taking the positions shown as an example, the switching elements in the upper left and upper right bridge arms are PMOS transistors, while the switching elements in the lower left and lower right bridge arms are NMOS transistors (N-channel Metal-Oxide-Semiconductor Field-Effect Transistors). The PMOS transistor Q2 in the upper left bridge arm and the NMOS transistor Q5 in the lower right bridge arm have the same on / off state, as do the PMOS transistor Q4 in the upper right bridge arm and the NMOS transistor Q3 in the lower left bridge arm. Furthermore, when Q2 and Q5 are on, the current direction within the TEC is opposite to when Q3 and Q4 are on, thus controlling the heating or cooling of the TEC by controlling the on / off states of Q2, Q3, Q4, and Q5. The center position of the H-bridge circuit serves as the current output terminal of the temperature control circuit module, connected to the thermoelectric cooler to achieve drive control of the thermoelectric cooler.

[0067] In the embodiments of this application, the main control amplifier circuit includes an operational amplifier U12, a resistor R38, and a capacitor C11. The output terminal of the main control amplifier circuit is connected to the gates of PMOS transistors Q4 and Q5, and is used to control the on / off state of PMOS transistors Q4 and Q5, driving the right half-bridge of the H-bridge circuit. Specifically, the non-inverting input terminal of operational amplifier U12 serves as the input terminal of the temperature control circuit module, connected to the third output terminal of the control module, and is used to receive the third signal VIN; the inverting input terminal is connected to the H-bridge circuit, and is used to obtain the output state of the H-bridge circuit; the output terminal is connected to the inverting input terminal through capacitor C11, and is also connected to the gates of PMOS transistors Q4 and Q5 to control the on / off state of PMOS transistors Q4 and Q5. Capacitor C11 is the frequency compensation capacitor in the main control amplifier circuit, and its core function is to ensure system stability and prevent self-oscillation. In the feedback path of operational amplifier U12, C11 introduces a low-frequency zero in the loop gain frequency response. This zero-point, by providing phase lead at critical frequencies, can effectively compensate for the inherent phase lag of the amplification loop itself and other parasitic components, thereby expanding the phase margin of the system and preventing the total phase shift from reaching 180° at frequencies with a gain greater than 1, fundamentally eliminating oscillation conditions.

[0068] The output of the main control amplifier circuit is also connected to the gate drive circuit, so as to control the switching of PMOS transistor Q2 and NMOS transistor Q3 through the gate drive circuit, and drive the left half bridge of the H-bridge circuit.

[0069] The gate drive circuit includes operational amplifier U7, resistor R24, and resistor R25. The non-inverting input of operational amplifier U7 can be connected to a reference voltage VREF, and the inverting input is connected to the output of the main control amplifier circuit.

[0070] In the embodiments of this application, the operational amplifier U12 receives feedback signals from the H-bridge circuit and works together with the external control input (the third signal VIN output from the third output terminal) to set the drive level of the temperature control circuit, thereby precisely controlling the magnitude and direction of the current flowing through the TEC.

[0071] The main control amplifier circuit drives the right half-bridge, and the gate drive circuit drives the left half-bridge. The output of the gate drive circuit is adjusted by a DC bias provided by a reference voltage. In the embodiments of this application, the resistance values ​​of resistors R25 and R24 are equal, and the input and output of operational amplifier U7 satisfy the following relationship:

[0072] U7_VOUT=2VREF-U12_VOUT;

[0073] Where U7_VOUT represents the output voltage of operational amplifier U7, VREF represents the reference voltage, and U12_VOUT represents the output voltage of operational amplifier U12. This relationship ensures that the outputs of operational amplifier U7 and operational amplifier U12 change in opposite phase and linearly, that is, when the output of operational amplifier U12 increases, the output of operational amplifier U7 decreases.

[0074] In the embodiments of this application, the main control amplifier circuit is connected to PMOS transistors Q4 and Q5 via resistor R39, and the gate drive circuit is connected to PMOS transistors Q2 and Q3 via resistor R27. Resistors R39 and R27 are gate drive resistors, and their values ​​can be 10Ω to 100Ω depending on the actual needs. They are used to suppress LC oscillations formed by gate-source parasitic capacitance and PCB trace parasitic inductance, effectively attenuating gate voltage ringing and improving circuit stability. The gates of PMOS transistors Q4 and Q5 are grounded via resistor R40, and the gates of PMOS transistors Q2 and Q3 are grounded via resistor R28. Resistors R40 and R28 provide bias voltages for the switching elements to prevent false turn-on, and their values ​​can be 5kΩ to 10kΩ depending on the actual needs.

[0075] To achieve safe commutation and prevent shoot-through of switching elements on the same side, a fast recovery diode D3 and a bleeder resistor R26 are connected in reverse parallel across resistor R27, and a fast recovery diode D4 and a bleeder resistor R41 are connected in reverse parallel across resistor R39, constructing a fast discharge path and forming an asymmetrical charge-discharge path. The function of the bleeder resistor is to prevent the front-end operational amplifier from burning out due to excessive current when it is turned off; the resistance value can be 0.1Ω to 10Ω depending on the actual needs.

[0076] When the switching elements are turned on, the switching elements of the left half-bridge are slowly charged through the gate drive resistor R27, delaying the turn-on of the switching elements. The switching elements of the right half-bridge are slowly charged through the gate drive resistor R39, delaying the turn-on of the switching elements, thereby effectively suppressing current surges and ringing. When turned off, the switching elements of the left half-bridge quickly discharge the gate charge through the fast recovery diode D3 and the discharge resistor R26, and the switching elements of the right half-bridge quickly discharge the gate charge through the fast recovery diode D4 and the discharge resistor R41, thus realizing the logic of fast turn-off followed by delayed turn-on. This "slow turn-on, fast turn-off" characteristic naturally creates an inherent dead time during the switching process, ensuring that the other pair of switching elements only begins to conduct after one pair of switching elements is completely turned off, fundamentally avoiding shoot-through and ensuring the safe commutation of the H-bridge.

[0077] In the embodiments of this application, the output level of the main control amplifier circuit directly determines the conduction state of the switching elements in the right half-bridge: when the output of the main control amplifier circuit is high, NMOS transistor Q5 is turned on; when the output of the main control amplifier circuit is low, PMOS transistor Q4 is turned on. The output level of the gate drive circuit directly determines the conduction state of the switching elements in the left half-bridge: when the output level of the gate drive circuit is high, NMOS transistor Q3 is turned on; when the output level of the gate drive circuit is low, PMOS transistor Q2 is turned on.

[0078] Based on the above inverted design, the gate-source voltage of the switching element in the left half-bridge can be expressed as:

[0079] V_GS_Q2=VCC–(2VREF–U12_VOUT);

[0080] V_GS_Q3=2VREF–U12_VOUT;

[0081] Where V_GS_Q2 represents the gate-source voltage of PMOS transistor Q2, and V_GS_Q3 represents the gate-source voltage of NMOS transistor Q3.

[0082] In the embodiments of this application, the power supply voltage VCC of the circuit is 3.3V, and the minimum threshold voltage |Vth| of the selected switching element is ≥2V. This voltage relationship can effectively prevent two switching elements in the same half-bridge from conducting simultaneously, thereby fundamentally preventing the occurrence of shoot-through current and avoiding damage to the circuit.

[0083] Ideally, the H-bridge circuit conducts diagonally, meaning only Q2 and Q5 or Q3 and Q4 are simultaneously conducting. This inverting drive mechanism, combined with the aforementioned voltage relationship, ensures that only one pair of diagonal switching elements is effectively conducting during circuit operation, guaranteeing the proper formation of the current path and preventing shoot-through.

[0084] In embodiments of this application, the temperature control circuit module further includes a second sampling resistor R31 and a current sampling circuit module. The second sampling resistor is connected in series with the thermoelectric cooler to obtain the current flowing through the thermoelectric cooler. The input terminal of the current sampling circuit module is connected to the second sampling resistor, and the output terminal is connected to the control module, outputting a current sampling signal IOUT. This signal is used to feed back the obtained current of the thermoelectric cooler to the control module, thereby achieving precise control of the thermoelectric cooler. Based on this, the system uses a first NTC thermistor to feed back the temperature change of the laser module after the thermoelectric cooler is adjusted to the control module via the temperature sampling module. The control module then outputs a third signal to the main control circuit module, forming a closed-loop control.

[0085] The current sampling circuit module includes a subtractor circuit. The non-inverting and inverting inputs of the subtractor circuit are respectively connected to the two ends of the second sampling resistor R31 to obtain the current signal of the second sampling resistor R31 and generate a feedback voltage. The current signal is then buffered and sent to the ADC conversion circuit for analog-to-digital conversion. The result of the analog-to-digital conversion is input to the control module for real-time acquisition of the current signal. In the embodiments of this application, to avoid errors caused by impedance mismatch across the second sampling resistor R31, the non-inverting and inverting inputs of the subtractor circuit are each connected to the two ends of the sampling resistor through a buffer circuit. The non-inverting input of the buffer circuit is connected to the second sampling resistor R31, and the inverting input is connected to the output. The output is also connected to the subtractor circuit, effectively isolating impedance, improving signal accuracy and stability, thereby achieving high-precision current detection.

[0086] Specifically, the buffer circuit can be a voltage follower circuit. The non-inverting input of the subtractor circuit is connected to one end of the second sampling resistor R31 through a voltage follower circuit composed of operational amplifier U8, and the inverting input of the subtractor circuit is connected to the other end of the second sampling resistor R31 through a voltage follower circuit composed of operational amplifier U9.

[0087] The subtractor circuit includes operational amplifier U11, resistors R34, R35, R36, and R37. The non-inverting input of operational amplifier U11 is connected to the output of a voltage follower circuit composed of operational amplifier U8 through resistor R35. The inverting input of operational amplifier U11 is connected to the output of a voltage follower circuit composed of operational amplifier U9 through resistor R34. The output of operational amplifier U11 is connected to the inverting input of operational amplifier U11 through resistor R36. The non-inverting input of operational amplifier U11 is also grounded through resistor R37.

[0088] The inverting input of operational amplifier U12 is connected to the output of the subtractor circuit to obtain the current of the thermoelectric cooler. The direction and magnitude of the current in the H-bridge circuit are controlled by comparing the difference between the current of the thermoelectric cooler and the input third signal.

[0089] In the embodiments of this application, the H-bridge circuit further includes a filter network comprising inductors L3 and L4, capacitors C8, C9, and C10. Inductors L3 and L4 are connected in series with the thermoelectric cooler (TEC), and capacitor C9 is connected in parallel with the TEC and the second sampling resistor R31. This filter network has multiple functions: firstly, it effectively suppresses high-frequency noise and voltage ripple caused by H-bridge switching, reducing electromagnetic interference in the system; secondly, by limiting the rate of change of the TEC load current, it acts as a current buffer and device protection, extending the TEC's lifespan; and thirdly, it improves the dynamic response performance of the current control loop, contributing to a more stable and accurate bidirectional constant current output.

[0090] In the embodiments of this application, in the H-bridge circuit composed of PMOS transistor Q2, NMOS transistor Q3, PMOS transistor Q4 and NMOS transistor Q5, resistors and capacitors are connected in parallel between the source and gate of each MOS transistor, which helps to buffer the switching speed, limit the surge current, suppress the spike oscillation, and improve the system reliability.

[0091] In the embodiments of this application, a rail-to-rail input / output bipolar operational amplifier is used to achieve high-precision linear detection and control of the bidirectional current of the TEC. Compared with traditional single-supply op-amps, which are limited by input common-mode range and output swing and require a bias voltage to measure bidirectional current, the bipolar op-amp can operate near zero potential without an additional reference voltage, achieving linear amplification within the operating current range. To avoid sampling errors caused by impedance mismatch across the sampling resistor, an op-amp buffer circuit is added across the sampling resistor to effectively isolate the impedance, ensuring the accuracy and stability of the sampling signal. Finally, the buffered signal is input to the ADC to achieve high-precision current acquisition. At the same time, the system operates stably at a lower H-bridge supply voltage, significantly reducing power consumption and system complexity, improving energy efficiency and reliability, reducing MOSFET conduction losses and heat dissipation, alleviating heat dissipation pressure, and optimizing overall performance.

[0092] Figure 6 A schematic flowchart illustrating the temperature control process of a semiconductor laser system according to an embodiment of this application is shown. Figure 6As shown in the embodiments of this application, in order to improve the measurement accuracy and stability of the laser temperature by the temperature control system, a Kalman filter is introduced into the temperature acquisition channel of the control module. The temperature signal generated by the first NTC thermistor is converted by the ADC sampling module and then input to the control module. The control module first performs Kalman filtering on the signal to filter out interference signals such as power supply noise, environmental disturbances and sampling jitter during the ADC sampling process in real time. Then, the temperature data is extracted from the filtered signal and compared with the set temperature. The comparison result is processed by the PID module to generate a PID control signal, which is then converted by the DAC to generate a third signal and input to the temperature control circuit to realize the control of the laser temperature.

[0093] Assuming the true temperature of the laser module is The measured voltage value is the observation value. The Kalman filter model can be established as the following state-space equation:

[0094] State update equation: ;

[0095] Observation equation: ;

[0096] in, This represents the system state at the current time k, i.e., the estimated temperature of the laser module; This represents the system state at the previous time step k-1; B represents the control input at the current time k, which can be ignored in the case of pure filtering; B represents the control input matrix, which will contain the control quantity. The effect of this is mapped onto the state variables. In a pure filtering model without control input, B=0. This represents the control input at the previous time step k-1; This represents the process noise (also known as system noise) at the previous time k-1. It represents the uncertainty of the state prediction model itself, such as factors that the model fails to describe, such as internal heating of the laser or fluctuations in ambient temperature. This represents the voltage value obtained by sampling via a thermistor, which is the observed temperature of the laser module; This represents the state transition matrix (it can be set to 1, indicating a steady temperature change). This represents the observation matrix (usually 1, indicating directly observed temperature). Let N(0,Q) represent process noise, which follows a normal distribution, where Q represents the process noise covariance matrix, which defines the process noise. The strength of the Q value reflects the level of confidence in the state prediction model. The larger the Q value, the greater the uncertainty of the model prediction, and the more the filter will depend on the observations. Let represent the measurement noise, which follows a normal distribution N(0,R), where R represents the observation noise covariance matrix, which defines the observation noise. The strength of R reflects the noise level of the ADC measurement system (including thermistors, amplifier circuits, and the ADC itself). The larger the R value, the less reliable the observation is, and the more the filter will rely on its own state prediction.

[0097] In the embodiments of this application, the recursive formula for the standard Kalman filter is as follows.

[0098] In the forecasting phase, the formula for temperature prediction can be expressed as: The formula for covariance prediction can be expressed as: .

[0099] Update Phase: Kalman Gain The state update formula can be expressed as: The covariance update formula can be expressed as: .

[0100] The physical meanings of each parameter are as follows:

[0101] The prior state estimate (or predicted value) of the laser temperature at current time k. The superscript "-" indicates that this is incorporating the actual observation value z at current time k. k Previously, predictions were based solely on system models.

[0102] : The posterior state estimate of the laser temperature at the previous time k-1 (i.e., the optimal result of the previous round of filtering).

[0103] The prior estimate error covariance of the current time k. It measures the error of the predicted value before incorporating new observations. The degree of uncertainty.

[0104] : The posterior estimation error covariance of the previous time step k-1.

[0105] A: State transition matrix. It describes how the system state evolves naturally over time. For a slowly changing physical quantity like temperature, A=1 is usually set, indicating that the model assumes that the laser temperature remains essentially constant within adjacent sampling periods without any drastic external disturbances.

[0106] Q: Process noise covariance. It reflects the uncertainty of the state prediction model itself, such as unmodeled thermal dynamics or environmental disturbances. A larger Q value indicates less reliable model predictions, and the filter will rely more heavily on subsequent observations. Considering the slow temperature change of the laser, Q is typically chosen to be a small value, such as 10.-4 This is to avoid the filter overreacting to small fluctuations in the model's predictions.

[0107] : The Kalman gain at the current time k. It is a weighting coefficient that determines whether the current observation is given more weight during state updates. Or do we place more trust in the model's predictions? .

[0108] : The observed value at the current time k, that is, the temperature-related voltage measurement obtained by sampling through a thermistor and converting it through an ADC.

[0109] H: Observation matrix. It establishes a linear relationship between the system state (temperature) and the observed value (voltage). In this embodiment, since the ADC measurement voltage and temperature have an approximately linear relationship, H can usually be set to 1 (note the calibration coefficient in actual applications).

[0110] R: Observation noise covariance. It reflects the noise level of the temperature measurement system (including thermistors, amplifier circuits, and ADC). The larger the R value, the less reliable the observation. Setting example: R needs to be determined based on the noise of the actual measurement system. For example, by sampling a stable temperature multiple times with the ADC and calculating its variance, its magnitude can be used as the initial value of R, such as 0.01.

[0111] The posterior state estimate of the laser temperature at time k (i.e., the optimal result of this round of filtering) indicates that it incorporates the observations at the current time. The optimal estimate was then obtained.

[0112] The posterior estimation error covariance at time k. It reflects the optimal estimate. The uncertainty is used to predict the next moment.

[0113] I: Identity matrix.

[0114] In the embodiments of this application, the control module is implemented using an FPGA. The Kalman filter is implemented in the FPGA by using fixed-point / floating-point computing resources (such as a DSP core) to periodically execute the prediction and update process. The parameter values ​​of the Kalman filter can be set according to actual needs. In some embodiments, A=1 indicates that the laser temperature changes little within the sampling period; H=1 indicates direct observation; Q can take a smaller value (e.g., ...). To avoid over-response; R can be set according to the ADC sampling noise level; initial state settings. (Initial Temperature Estimate): This can be set to a rough temperature measurement taken at system startup, or an expected ambient temperature (e.g., 25°C). Initial estimate error covariance. Because the uncertainty of the actual temperature is greatest during system startup, It should be set to a large value for fast convergence. For example, It can be set to 10 or 100, which is much larger than the process noise Q, thus indicating that the initial confidence level is extremely low and the filter will quickly rely on the first few observations to correct the estimate.

[0115] In the embodiments of this application, the temperature control system is built based on discrete components, which offers superior heat dissipation performance compared to integrated chips, making it particularly suitable for laser applications with extremely high requirements for temperature control linearity, response speed, and stability. Furthermore, the Kalman filter built into the FPGA significantly improves the quality of the temperature sampling signal. By deploying the Kalman filter within the FPGA, the system effectively suppresses high-frequency measurement jitter and power supply noise interference, significantly improving the stability and reliability of the temperature signal. Simultaneously, the filter enhances the robustness of the temperature control system to dynamic disturbances, ensuring that the control module receives a smooth and reliable temperature error signal, thereby achieving fully hardware closed-loop control. This solution is particularly suitable for applications with extremely high requirements for temperature control accuracy, stability, and response speed, meeting the needs for low-latency and high-reliability temperature control.

[0116] In the embodiments of this application, a PID temperature control method based on Kalman filtering is designed to achieve high-response and high-precision temperature control.

[0117] The laser module integrates a first NTC thermistor, which is connected to a high-precision 18-bit ADC acquisition module via a voltage divider with a 10k resistor to acquire the divided voltage V across the thermistor in real time. NTC The current temperature can be calculated using the following formula:

[0118] ;

[0119] ;

[0120] in, , , , This indicates the resistance value of the first NTC thermistor. This represents the calculated temperature (in K). This indicates the reference voltage of the first NTC thermistor, which can be 3.3V.

[0121] Considering the quantization noise, power supply noise, and environmental interference present in ADC measurements, directly using the raw temperature data may lead to control loop oscillation or erroneous adjustment. Therefore, a first-order Kalman filter was implemented inside the FPGA to filter the temperature sampling data in real time.

[0122] The Kalman filter, based on a prediction-update architecture, achieves dynamic estimation and noise suppression of temperature values ​​by introducing the system state covariance matrix. The filter effectively reduces random fluctuations in temperature data, improving the smoothness and robustness of the control system response.

[0123] This mechanism ensures that the temperature control loop can still stably control the TEC cooling module even in the presence of measurement noise and system disturbances, maintaining the laser's operating point within the ideal temperature range, thereby further improving the stability of the laser output frequency and intensity.

[0124] In some embodiments, in addition to setting a first NTC thermistor inside the laser module, a second temperature sampling module is further provided to collect the ambient temperature in order to compensate for the temperature drift of the device caused by changes in ambient temperature, and further improve the stability of the laser frequency output by the laser module.

[0125] Specifically, the temperature control module also includes a second temperature sampling module. This second temperature sampling module comprises a second NTC thermistor and a temperature feedback module. The second NTC thermistor is mounted on the circuit board of the constant current source circuit module to acquire the ambient temperature of the circuit board. The temperature feedback module is connected to the second NTC thermistor and the control module, and transmits the temperature signal output by the second NTC thermistor to the control module. The second temperature sampling module acquires the ambient temperature at the constant current source circuit module and feeds it back to the control module to compensate for circuit parameter drift caused by changes in ambient temperature (such as changes in reference voltage, operational amplifier bias, and resistor values), further improving the long-term stability and accuracy of the output current.

[0126] The control module has a pre-stored current-temperature compensation lookup table, which includes current compensation values ​​corresponding to different temperatures. The control module is configured to look up the corresponding current compensation value in the current compensation lookup table based on the ambient temperature (i.e., the output temperature signal) obtained by the second temperature sampling module, so as to correct the output current of the constant current source circuit module, thereby offsetting the characteristic drift of resistors and other devices caused by temperature changes and ensuring the stability of the output current.

[0127] Specifically, the relationship between current and temperature was first established through temperature experiments. Under experimental conditions, the current output of the constant current source circuit module was tested at different temperature points. The set current of the first signal of the constant current source circuit module was kept constant, and the actual sampled current value at each temperature point was recorded. The deviation ΔI (unit: mA) between the actual current and the set value was calculated. This yielded the curve showing the relationship between the current deviation ΔI and temperature, which serves as the data basis for subsequent temperature drift compensation.

[0128] Subsequently, based on the relationship between the voltage value VL of the second signal of the fine-tuning circuit module and the current converted through resistor R19: The required voltage compensation value VL1 (in V) is calculated at each temperature point. In the embodiments of this application, the second signal is output by a 16-bit bipolar DAC with an output voltage range of ±3.3V, corresponding to a digital range of -32768 to +32767. Therefore, the voltage compensation value VL1 can be converted into a DAC digital correction value according to the following formula. :

[0129] ;

[0130] in, This is the reference voltage for the DAC, which can be 3.3V. To fine-tune the resistance value of the main circuit in the circuit module, specifically the resistance value of resistor R19. Each temperature point ΔI corresponds to a unique DAC digital correction value. All DAC digital correction values ​​are used as an index to construct a current-temperature compensation lookup table, which is then stored in the ROM or register array inside the FPGA.

[0131] During actual operation, the ambient temperature is acquired in real time through the second NTC thermistor, and this temperature is used as an index to look up the required DAC digital correction value in a table. This DAC digital correction value is superimposed on the original second signal to obtain a new second signal, which is then output to the fine-tuning circuit module to output a second current, thereby dynamically correcting the current error caused by temperature drift.

[0132] This method can effectively improve the output stability and current control accuracy of the constant current source over a wide temperature range, providing a more reliable driving current for the laser and meeting the stringent requirements of the SERF atomic magnetometer for long-term stability of current control.

[0133] This application also provides a semiconductor laser control method for the semiconductor laser system provided above, comprising: acquiring the required current of the laser module; splitting the required current into a first current and a second current; outputting a first signal through a first output terminal of the control module to output the first current based on a coarse adjustment circuit module; and outputting a second signal through a second output terminal of the control module to output the second current based on a fine adjustment circuit module.

[0134] In the embodiments of this application, the coarse adjustment circuit module can output a wide range of current, while the fine adjustment circuit module can output a small range of high-precision current, thereby achieving wide-range and high-resolution current adjustment to meet the laser's requirements for current adjustment range and accuracy. The required current can be split into different magnitudes, with the coarse adjustment circuit module outputting a large current and the fine adjustment circuit module outputting a fine-tuning current, thus simultaneously meeting the requirements for current adjustment range and accuracy.

[0135] In the embodiments of this application, current compensation can be further achieved through temperature acquisition and feedback. In this case, the control method further includes: acquiring a temperature signal sampled from the ambient temperature; acquiring a current compensation value corresponding to the temperature signal based on the temperature signal and a pre-stored current-temperature compensation lookup table; and generating a second signal output through the second output terminal of the control module based on the current compensation value.

[0136] Embodiments of this application also provide an atomic magnetometer, including the semiconductor laser system described above.

[0137] The semiconductor laser system provided in this application achieves wide-range and high-resolution current regulation through the design of coarse-tuning and fine-tuning circuit modules. By designing a compensation circuit in the fine-tuning circuit module, the potential at the first terminal of the main circuit is adjusted to 0, thereby forming an equivalent virtual ground at the input terminal of the fine-tuning circuit module. This avoids the influence of voltage changes in the coarse-tuning circuit module on the fine-tuning circuit module, ensuring current accuracy. By introducing a temperature drift compensation mechanism, the influence of ambient temperature changes on the laser current is effectively suppressed, ensuring the long-term stability and accuracy of the drive current. The system is constructed using discrete components, possessing good heat dissipation capabilities. Combined with NTC thermistor feedback, closed-loop PID control, and FPGA internal Kalman filtering, temperature measurement noise can be effectively filtered out, improving temperature control accuracy and system response speed. The overall solution meets the high requirements of lasers for both current and temperature stability and is suitable for critical applications such as high-precision atomic magnetometers.

[0138] Those skilled in the art will understand that the features described in the various embodiments of this application can be combined and / or combined in various ways, even if such combinations or combinations are not explicitly described in this application. In particular, the features described in the various embodiments of this application can be combined and / or combined in various ways without departing from the spirit and teachings of this application. All such combinations and / or combinations fall within the scope of this application.

Claims

1. A semiconductor laser system, characterized by, The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module.

2. The system of claim 1, wherein, The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module.

3. The system of claim 1, wherein, The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module. The application relates to a laser current source circuit module.

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The application The current output module comprises a comparison feedback circuit, a first sampling resistor and a power tube, a first end of the first sampling resistor is connected with an output end of the voltage module, a second end is connected with a source electrode of the power tube, the comparison feedback circuit comprises an operational amplifier, a non-inverting input end of the operational amplifier is connected with an output end of the subtraction circuit module, an inverting input end is connected with the second end of the first sampling resistor, an output end is connected with a gate electrode of the power tube, a drain electrode of the power tube is connected with an input end of the laser module.

5. The system of claim 4, wherein, The first sampling resistor comprises a plurality of shunt resistors, and the plurality of shunt resistors are connected in parallel or in series and in parallel combination.

6. The system of claim 1, wherein, Further comprising a monitoring circuit module, an input end of the monitoring circuit module is connected with the laser module, an output end is connected with a first input end of the control module, and the current value of the laser module is acquired, so that the control module monitors the working state of the laser module according to the size relationship between the current value and a target current value, wherein the target current value is the sum of the first current and the second current.

7. The system of claim 1, wherein, Further comprising a temperature control module; The temperature control module comprises a thermoelectric cooler, a first temperature sampling module and a temperature control circuit module; The thermoelectric cooler is arranged in the laser module; An input end of the first temperature sampling module is arranged in the laser module, and an output end is connected with the control module; An input end of the temperature control circuit module is connected with a third output end of the control module, a current output end is connected with the thermoelectric cooler, and the current size and / or direction input into the thermoelectric cooler are changed according to a third signal output by the third output end; The control module is used for outputting the third signal to the temperature control circuit module through the third output end according to a temperature sampling signal output by the first temperature sampling module, so as to change the working state of the thermoelectric cooler by changing the current size and / or direction input into the thermoelectric cooler.

8. The system of claim 7, wherein, The temperature control circuit module further comprises a second sampling resistor and a current sampling circuit module; The second sampling resistor is connected with the thermoelectric cooler in series, and is used for acquiring the current size passing through the thermoelectric cooler; An input end of the current sampling circuit module is connected with the second sampling resistor, and an output end is connected with the control module, and the current size of the thermoelectric cooler acquired is fed back to the control module.

9. The system of claim 7, wherein, The temperature control module further comprises a second temperature sampling module, and the second temperature sampling module is used for acquiring the ambient temperature at the constant current source circuit module and feeding back the ambient temperature to the control module; The control module pre-stores a current temperature compensation lookup table, and the current temperature compensation lookup table comprises current compensation values corresponding to different temperatures; The control module is configured to find the corresponding current compensation value in the current temperature compensation lookup table according to the ambient temperature obtained by the second temperature sampling module, and correct the output current of the constant current source circuit module by changing the second signal.

10. A method of controlling a semiconductor laser for use in a semiconductor laser system as claimed in any one of claims 1 to 9, characterized in that It comprises: Acquiring the required current of the laser module; splitting the required current into a first current and a second current; outputting a first signal through a first output terminal of the control module to output the first current based on the coarse adjustment circuit module; outputting a second signal through a second output terminal of the control module to output the second current based on the fine adjustment circuit module.

11. The method of claim 10, wherein, The method further comprises: acquiring a temperature signal sampled from an ambient temperature; acquiring a current compensation value corresponding to the temperature signal according to the temperature signal and a pre-stored current-temperature compensation lookup table; generating the second signal according to the current compensation value.

12. An atomic magnetometer, comprising: A semiconductor laser system comprising any one of claims 1-9.

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