Full-bridge soft switching inverter circuit based on silicon carbide power device and control method thereof

Abstract

The invention discloses a silicon carbide power device-based full-bridge soft switching inverter circuit and a control method thereof, and belongs to the technical field of power electronics, the silicon carbide power device-based full-bridge soft switching inverter circuit comprises a soft switching inverter module, the module comprises a full-bridge inverter circuit, a resonant inductor, a resonant capacitor and a main transformer, the full-bridge inverter circuit is composed of four silicon carbide power switch tubes Q1-Q4, q1 and Q2 are connected in series to form a leading arm, Q3 and Q4 are connected in series to form a lagging arm, two ends of the leading arm Q1 and Q2 are respectively connected in parallel with capacitors C2 and C3, a midpoint of the leading arm, a midpoint of the lagging arm and a resonant inductor, and the resonant capacitors are connected in series and then connected with a primary side loop of a main transformer; and the control module generates a driving signal, and drives stable high-frequency on-off of the silicon carbide power switch tube after being isolated and optimized by the driving circuit, so that high-frequency switching action is realized. By optimizing the design of the soft switch and the drive circuit and introducing the common mode inductor, the problems of drive crosstalk and low reliability of the silicon carbide power device in the application of the welding machine power supply are solved.

Description

TECHNICAL FIELD

[0001] The present application belongs to the technical field of power electronics, and particularly relates to a full-bridge soft-switching inverter circuit based on a silicon carbide power device and a control method thereof. BACKGROUND

[0002] The statements in this section merely provide background information related to the present disclosure and do not necessarily constitute the prior art.

[0003] Electric welding machines are indispensable equipment in the field of industrial production and processing. Among them, the inverter welding machine has become the mainstream development direction of welding machine products due to its small size, light weight, good control performance, fast dynamic response, and easy realization of real-time control of the welding process. At present, the IGBT full-bridge soft-switching inverter welding machine has become the main application trend in the current inverter welding machine field due to its small power device voltage rating and small filter inductance.

[0004] With the development of power switch tube devices, silicon carbide power transistors (SiC MOSFET) have become the ideal choice for the miniaturization and high efficiency of welding machine power supplies due to their low loss, high frequency, and high temperature characteristics. However, due to the characteristics of silicon carbide power switch tubes themselves, it is difficult to directly apply them to existing full-bridge soft-switching inverter power supplies, and high-frequency reliable switching of silicon carbide power switch tubes cannot be achieved. The main reasons are as follows: (1) Compared with traditional silicon-based transistors, silicon carbide wafer materials have more impurities and lower yield. Due to the physical characteristics of silicon carbide power switch tubes, the gate oxide layer is weaker. The weak gate oxide layer is very sensitive to negative gate voltage, resulting in low negative voltage resistance of silicon carbide power switch tubes and making them more susceptible to negative voltage spikes. Negative voltage spikes (caused by Miller capacitance coupling) will generate high electric fields on the gate oxide layer during the application of negative voltage or in the soft-switching process, increasing the risk of gate oxide breakdown and threshold voltage drift, thereby affecting the long-term reliability of the device.

[0005] (2) High voltage rate of change of silicon carbide power switch tubes ( ) is easily coupled to adjacent switch tubes or drive circuits through parasitic capacitance, forming high-frequency crosstalk, which can easily cause misfires and also cause device gate oxide damage or gate breakdown, which will affect the reliability of the device.

[0006] (3) At the moment of switching of the silicon carbide power switch tube, the output capacitance forms an oscillation circuit with the loop stray inductance and the parallel capacitance of the leading arm, which produces high-frequency oscillation at the switching moment, thereby causing significant high-frequency drive crosstalk. This will further increase the power consumption of the parallel capacitance of the leading arm, affecting the reliability of the device.

[0007] (4)Although the full-bridge soft switching scheme using silicon carbide power devices is also proposed at present, such as full-bridge LLC resonant soft switching or phase-shifted full-bridge soft switching, the adaptability of the existing soft switching topologies is poor: the input voltage range of the full-bridge LLC resonant soft switching is narrow, the frequency modulation needs to work in the inductive region (the soft switching fails when working in the capacitive region), the load mainly works above half load, and the load changes relatively little, which is not suitable for high-frequency welding machine power supply; the existing phase-shifted full-bridge soft switching scheme does not consider the problems of the lead leg capacitor and stray inductance, high-frequency oscillation of the switching tube junction capacitor and derived drive crosstalk, the reliability is poor, and the load range is narrow, it is difficult to realize soft switching control under light load and no load, which also cannot meet the application of high-frequency welding machine power supply. SUMMARY

[0008] In view of the deficiencies of the prior art, the present application provides a full-bridge soft switching inverter circuit based on silicon carbide power devices and a control method thereof, optimizes the design of the drive circuit, improves the anti-interference ability and reliability of the pulse transformer drive, and ensures the stability of the negative voltage during switching; the design of the lead leg circuit is optimized, a common mode inductor is added at the parallel capacitor of the lead leg to suppress the high-frequency oscillation in the parallel capacitor of the lead leg; by optimizing the design of the soft switching technology and control method suitable for the characteristics of the silicon carbide power switching tube, the high-frequency crosstalk is suppressed, the switching stress is reduced, the problems of serious high-frequency crosstalk of the silicon carbide inverter, insufficient switching reliability and high-frequency oscillation of the parallel capacitor of the lead leg are solved, and the silicon carbide inverter is suitable for high-frequency welding machine power supply, industrial pulse power supply and other scenes.

[0009] To achieve the above object, one or more embodiments of the present application provide the following technical solutions: In a first aspect, the present application provides a full-bridge soft switching inverter circuit based on silicon carbide power devices.

[0010] A full-bridge soft switching inverter circuit based on silicon carbide power devices, comprising a soft switching inverter module, the soft switching inverter module comprising a full-bridge inverter circuit, a resonant inductor, a resonant capacitor and a main transformer, the full-bridge inverter circuit being composed of four silicon carbide power switching tubes Q1-Q4, Q1 and Q2 being connected in series as a lead leg, Q3 and Q4 being connected in series as a lag leg, capacitors C2 and C3 being connected in parallel across the lead leg Q1 and Q2 respectively, and the midpoint of the lead leg, the midpoint of the lag leg and the resonant inductor and the resonant capacitor being connected in series and then connected to the primary side loop of the main transformer.

[0011] Further technical solutions also include a drive module; the drive module optimizes the drive signal through a drive circuit or a drive chip to drive the on-off of the silicon carbide power switching tube.

[0012] Further, the driving module is realized by a driving circuit, the driving circuit comprises an isolated pulse transformer, a primary side of the isolated pulse transformer is connected with a driving signal, and a secondary side of the isolated pulse transformer is connected with a Miller clamp circuit, a negative voltage circuit and a pre-charging circuit; the Miller clamp circuit comprises a driving MOSFET and a diode D1 between a source and a gate of the driving MOSFET, and a capacitor C11 between a drain and a gate of the driving MOSFET; a drain of the driving MOSFET is connected with a gate of a silicon carbide power switch tube; the driving MOSFET and the diode D1 clamp a stable negative voltage generated by the pre-charging circuit and the negative voltage circuit in a negative voltage state; and the capacitor C11 absorbs transient overvoltage of the gate of the driving MOSFET.

[0013] Further, a same-name end of the secondary side of the isolated pulse transformer is connected with a driving resistor, and the driving resistor and the capacitor C11 form an RC absorption circuit. The negative voltage circuit is arranged between the Miller clamp circuit and the pre-charging circuit, and comprises a zener ZD1 and a capacitor C12 connected in parallel; an anode of the zener ZD1 is connected with an anode of the diode D1 and one end of the capacitor C12; and a cathode of the zener ZD1 is connected with the other end of the capacitor C12. The pre-charging circuit comprises a diode D2 and a resistor R2 connected in series; an anode of the diode D2 is connected with the driving resistor; a cathode of the diode D2 is connected with the resistor R2; and the other end of the resistor R2 is connected with the cathode of the zener ZD1.

[0014] Further, the driving circuit further comprises a gate overvoltage protection circuit and a gate capacitor C13 connected with the secondary side of the isolated pulse transformer. The gate overvoltage protection circuit comprises two transient suppression diodes ZD2 and ZD3, and the transient suppression diodes ZD2 and ZD3 are combined into a bidirectional transient suppression diode; a cathode of the transient suppression diode ZD2 is connected with the driving resistor R1; an anode of the transient suppression diode ZD2 is connected with an anode of the transient suppression diode ZD3; and a cathode of the transient suppression diode ZD3 is connected with the cathode of the zener ZD1.

[0015] Further, the control module comprises a sampling circuit and a control chip. The sampling circuit comprises a current sampling circuit and a voltage sampling circuit, and is used for collecting output current of the soft-switching inverter module, output voltage and output current of the welding power supply, and feeding back the collected voltage and current signals to the control chip. The control chip is used for preliminarily generating a PWM driving pulse based on a preset driving strategy, judging a current working condition and dynamically adjusting dead time and PWM pulse width of the driving pulse according to the collected voltage and current signals, and generating a driving signal.

[0016] Further technical solutions, the soft switch inverter module further comprises a three-phase rectifier bridge, a bus capacitor, a commutation inductor, and a secondary rectifier circuit. The three-phase rectifier bridge, the bus capacitor, and the full-bridge inverter circuit are connected in parallel, and an external low-frequency alternating current is connected to the three-phase rectifier bridge. The secondary side of the main transformer is connected to the commutation inductor and the secondary rectifier circuit, respectively; the commutation inductor is used to introduce reactive current to perform commutation of the parallel capacitor of the leading arm, and to provide a commutation path for the no-load working condition; the secondary rectifier circuit includes two parallel diodes, and the two parallel diodes are connected in series with the output reactance, and are connected to the external load as the output end of the silicon carbide inverter to supply power to the external load.

[0017] Further technical solutions, the resonant inductor is at least one magnetic body structure sleeved on a conductor in the AC loop of the switching power supply; or, in the AC loop of the switching power supply, a conductor is connected in series, and at least one magnetic body structure is sleeved on the conductor. The magnetic body structure is a closed structure, and the shape of the magnetic body structure is annular, spherical, cylindrical, square, or other special-shaped structures.

[0018] In a second aspect, the present application provides a full-bridge soft switch inverter circuit based on silicon carbide power devices.

[0019] A full-bridge soft switch inverter circuit based on silicon carbide power devices, in the first aspect, the full-bridge soft switch inverter circuit structure is connected in series with the inductor between the midpoint of the leading arm and the parallel capacitor C2, C3 of the leading arm.

[0020] Further technical solutions, the impedance of the inductor is positively correlated with the frequency, and the inductor has low impedance at low frequency and high impedance at high frequency.

[0021] Further technical solutions, the inductor is any one of a ferrite core inductor, a nickel-zinc ferrite magnetic bead, or a manganese-zinc ferrite magnetic bead.

[0022] Further technical solutions, the inductor is a common mode inductor, and the common mode inductor is connected in series at the same end of the two parallel capacitors of the leading arm, and then connected to the midpoint of the leading arm through the common mode inductor.

[0023] Further technical solutions, the inductor is two independent and uncoupled differential mode inductors, and the two parallel capacitors of the leading arm are connected in series with the corresponding differential mode inductors at the same end, and then connected to the midpoint of the leading arm through the differential mode inductor.

[0024] In a third aspect, the present application provides a control method for a full-bridge soft switch inverter circuit based on silicon carbide power devices.

[0025] A control method of a full-bridge soft switching inverter circuit based on silicon carbide power devices, for controlling the operation of the full-bridge soft switching inverter circuit proposed in the first aspect or the second aspect, the control method comprising: Collecting the output current of the soft switching inverter module, the output voltage and output current of the welding power source; Based on the preset driving strategy, the PWM driving pulse is initially generated, and then the current working condition is judged and the dead time and PWM pulse width of the driving pulse are dynamically adjusted according to the collected voltage and current signals to generate a driving signal; Based on the optimized driving signal, the stable high-frequency on-off of the silicon carbide power switch tube is driven to realize high-frequency switching action.

[0026] Further technical solutions, the driving strategy is: The first silicon carbide power switch tube Q1 and the fourth silicon carbide power switch tube Q4 are turned on at the same time, the first silicon carbide power switch tube Q1 is turned off in advance, and the fourth silicon carbide power switch tube Q4 is turned off with a delay; The second silicon carbide power switch tube Q2 and the third silicon carbide power switch tube Q3 are turned on at the same time, the second silicon carbide power switch tube Q2 is turned off in advance, and the third silicon carbide power switch tube Q3 is turned off with a delay; A dead zone is provided between the third silicon carbide power switch tube Q3 and the fourth silicon carbide power switch tube Q4, and all power switch tubes in the dead zone are turned off.

[0027] Further technical solutions, according to the collected voltage and current signals, the current working condition is judged and the driving pulse is dynamically adjusted according to the current working condition, comprising: When the output voltage is the no-load voltage and the output current is 0, it is judged as a no-load state; when the current working condition is the no-load state, the leading arm pulse width is controlled to be a set small pulse width, the leading arm and the lagging arm are fixed to a set dead zone, and the leading arm parallel capacitor completes the commutation through the commutation circuit; When the output voltage is lower than the set value and the output current is greater than the rated current, it is judged as a short circuit state; when the current working condition is the short circuit state, the leading arm pulse width is controlled to be a set small pulse width, and the lagging arm zero current is turned off; When the output power reaches the preset maximum value, it is judged as a maximum power point state; when the current working condition is the maximum power point state, the leading arm pulse width is controlled to be maximum, and a fixed dead zone is set between the leading arm and the lagging arm, so that the lagging arm is turned off with a small current at the maximum power point.

[0028] Further technical solutions, in the process of controlling the operation of the full-bridge soft switching inverter circuit with common mode inductance, the common mode inductance alternates between normal working mode and high-frequency oscillation working mode, comprising: At the moment when the leading arm power switch tube Q1, Q2 is turned on and off, high-frequency oscillation is generated in the loop formed by the junction capacitance of the leading arm power switch tube Q1, Q2, the corresponding parallel capacitance and loop inductance, and the common-mode inductance enters the high-frequency oscillation working mode, the common-mode inductance presents high impedance and the impedance increases with the increase of the common-mode current in the loop; At other times, the common-mode inductance enters the normal working mode, the common-mode inductance presents low impedance, and has no additional effect on the operation of the full-bridge soft switching inverter circuit.

[0029] The above one or more technical solutions have the following beneficial effects: 1. The full-bridge soft switching inverter circuit based on silicon carbide power devices and the control method thereof are provided, the soft switching technology suitable for the characteristics of the silicon carbide power switch tube is designed, the soft switching of the silicon carbide power switch tube is realized, the high-frequency crosstalk problem is greatly reduced, the switching stress is reduced, and the switching reliability of the silicon carbide power switch tube is ensured; meanwhile, the design of the driving circuit is optimized, the anti-interference ability and reliability of the pulse transformer driving are improved, and the stability of the negative voltage in the switching process is ensured; in addition, an improved driving control method is designed, the influence of driving crosstalk is reduced, and the long-term reliability of the silicon carbide power device is improved. In the above manner, the silicon carbide full-bridge inverter can be applied to high-frequency welding machine power supplies, industrial pulse power supplies and the like, and the problems of serious high-frequency crosstalk of the silicon carbide inverter, insufficient switching reliability and high-frequency oscillation of the parallel capacitance of the leading arm are solved in the field of welding machines.

[0030] 2. The silicon carbide full-bridge inverter provided by the application can make the voltage across the power switch tube or the current flowing through the power switch tube change regularly during the switching process of the silicon carbide power switch tube by introducing a resonant circuit, so that the opening and closing of the power switch tube at zero current or zero voltage can be realized, the crosstalk problem can be greatly reduced, and the switching reliability of the silicon carbide power switch tube can be ensured. The voltage gain characteristic of the structure is more gentle compared with LLC, and the sensitivity to input voltage change is lower. In addition, the application also sets a commutation inductance and an adaptive control strategy, introduces a reactive current through the commutation inductance, completes the commutation of the parallel capacitance of the leading arm, realizes the soft switching of the silicon carbide inverter under no-load and light load, avoids the direct discharge of the power switch tube by the divided capacitance of the leading arm under no-load, and makes it adapt to all working conditions of the welding machine from no-load to short circuit.

[0031] 3、The silicon carbide full-bridge inverter provided by the application, the designed driving circuit can effectively solve the high-frequency reliable switching problem of the silicon carbide power switch tube. The negative voltage pre-charging circuit is designed, the diode and the resistance series circuit are adopted to quickly complete the pre-charging of the negative voltage circuit, the rapid establishment of the negative voltage is ensured, the negative voltage establishment time can be below 10 µS, the negative voltage power-off drop time is above 500 mS, the stability of the negative voltage in the switching process of the silicon carbide power switch tube start-stop moment can be effectively ensured; the negative voltage off and the Miller clamp circuit are designed, the driving MOSFET and the negative voltage generation circuit are adopted to form the off discharge circuit, the gate voltage is pulled down to the negative voltage in the off stage, the off is accelerated and the oscillation is inhibited, the circuit is stable and reliable, the driving MOSFET is connected in parallel between the gate-source of the silicon carbide power switch tube (i.e. the main power switch tube), the driving MOSFET is turned on in the main power switch tube off process and during the off period, the gate-source voltage can be effectively clamped in the negative voltage state, and the problem of the mis-conduction caused by the Miller effect and the stray inductance oscillation in the existing driving circuit is solved; in addition, the driving resistance and the capacitor in the Miller clamp circuit form the RC absorption protection circuit, the gate of the driving MOSFET is protected, the overvoltage breakdown is prevented, and the stability and reliability of the driving circuit are ensured.

[0032] 4、The silicon carbide full-bridge inverter provided by the application, the high-frequency common-mode inductance is connected in series between the two leading arms parallel capacitors of the full-bridge soft switching variable current circuit and the bridge arm midpoint, based on the low-frequency low-impedance, high-frequency high-impedance and common-mode inductance characteristics of the high-frequency common-mode inductance, the high-frequency common-mode oscillation can be filtered out without affecting the normal soft switching work, the high-frequency crosstalk is further reduced, the driving crosstalk problem is improved, the leading arm parallel capacitor loss caused by the high-frequency oscillation is reduced, the temperature rise of the leading arm parallel capacitor is reduced, and the reliability of the soft switching is further improved.

[0033] 5、The silicon carbide full-bridge inverter driving control method provided by the application can realize the effective soft switching control of the silicon carbide power switch tube in the full-bridge inverter circuit, ensure the zero current of the leading arm and the lagging arm opening of the inverter, the near zero voltage of the leading arm off, and the zero current of the lagging arm off in most working conditions, there is no sharp voltage and current change in the switching process, and the crosstalk problem can be effectively reduced. BRIEF DESCRIPTION OF DRAWINGS

[0034] The drawings accompanying the specification of the present application form a part thereof, serve to provide further understanding of the present application, and together with the description of the exemplary embodiments of the present application, explain the present application, and do not constitute any improper limitation to the present application.

[0035] Figure 1 It is the typical topology structure diagram of the full-bridge soft switching inverter circuit based on the silicon carbide power device provided by the embodiment of the present application; Figure 2is a topological structure diagram of a driving circuit in an embodiment of the present application; Figure 3 is an equivalent circuit and a current loop diagram of a working mode 1 stage under a typical topological structure in an embodiment of the present application; Figure 4 is an equivalent circuit and a current loop diagram of a working mode 2 stage under a typical topological structure in an embodiment of the present application; Figure 5 is an equivalent circuit and a current loop diagram of a working mode 3 stage under a typical topological structure in an embodiment of the present application; Figure 6 is an equivalent circuit and a current loop diagram of a working mode 4 stage under a typical topological structure in an embodiment of the present application; Figure 7 is an equivalent circuit and a current loop diagram of a working mode 5 stage under a typical topological structure in an embodiment of the present application; Figure 8 is an equivalent circuit and a current loop diagram of a working mode 6 stage under a typical topological structure in an embodiment of the present application; Figure 9 is a driving voltage waveform diagram of a full-bridge soft-switching inverter circuit in an embodiment of the present application; Figure 10 is an adjusted driving voltage waveform diagram of a full-bridge soft-switching inverter circuit in an embodiment of the present application; Figure 11 is a typical topological structure diagram of a full-bridge soft-switching inverter circuit with high-frequency common-mode inductance compensation in an embodiment of the present application; Figure 12 is an equivalent circuit and a current loop diagram of a normal working mode 1 stage under a typical topological structure with high-frequency common-mode inductance compensation in an embodiment of the present application; Figure 13 is an equivalent circuit and a current loop diagram of a normal working mode 2 stage under a typical topological structure with high-frequency common-mode inductance compensation in an embodiment of the present application; Figure 14 is a high-frequency oscillation equivalent circuit and a current loop diagram under a typical topological structure with high-frequency common-mode inductance compensation in an embodiment of the present application. DETAILED DESCRIPTION

[0036] It should be noted that the following detailed description is exemplary in nature and is intended to provide further description of the present application. Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this application belongs.

[0037] It is to be understood that the terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of example embodiments in accordance with the present application. As used herein, the singular forms "a", "an" and "the" are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms "comprises" and / or "comprising," when used in this specification, specify the presence of stated features, steps, operations, devices, components and / or combinations thereof, but do not preclude the presence or addition of one or more other features, steps, operations, devices, components and / or combinations thereof.

[0038] Embodiment One The embodiment proposes a full-bridge soft switching inverter circuit based on silicon carbide power devices, which mainly includes a soft switching inverter module, a driving module and a control module. The soft switching inverter module includes a full-bridge inverter circuit composed of four silicon carbide power switching tubes Q1-Q4. The driving signal generated by the control module is isolated and optimized by the driving module, and then is used to drive the stable high-frequency on-off of the silicon carbide power switching tube in the soft switching inverter module, so as to realize the high-frequency switching action. Through the above-mentioned circuit and soft switching inverter technology proposed by the embodiment, the soft switching of the silicon carbide power switching tube is realized, the crosstalk problem is greatly reduced, the switching reliability of the silicon carbide power switching tube is ensured, the driving circuit and the control method are optimized, the reliability problem of the pulse transformer driving is solved, the successful application of the silicon carbide inverter pulse transformer driving scheme is realized, and then the effective application of the silicon carbide inverter in the high-frequency welding machine power supply is realized.

[0039] As shown in Figure 1 The soft switching inverter module includes a three-phase rectifier bridge TB1, a bus capacitor C1, a full-bridge inverter circuit composed of silicon carbide power switching tubes Q1-Q4, a resonance capacitor Cx, a resonance inductor Lx, a main transformer T, a secondary rectifier circuit connected with a load, an output reactance Lo, and a commutation inductor L1.

[0040] Specifically, in the soft switching inverter, the full-bridge inverter circuit is composed of two silicon carbide half-bridge modules, mainly including a leading arm composed of a first silicon carbide power switching tube Q1 and a second silicon carbide power switching tube Q2 in series, and a lagging arm composed of a third silicon carbide power switching tube Q3 and a fourth silicon carbide power switching tube Q4 in series. The leading arm Q1 is connected in parallel with a capacitor C2, and the leading arm Q2 is connected in parallel with a capacitor C3. The three-phase rectifier bridge TB1 and the bus capacitor C1 are connected in parallel with the full-bridge inverter circuit. The external low-frequency alternating current is converted into direct current by the three-phase rectifier bridge TB1, and then is output to the full-bridge inverter circuit through the bus capacitor C1. The output end of the full-bridge inverter circuit is connected with a resonance circuit, which includes a resonance inductor Lx and a resonance capacitor Cx. The leading arm midpoint, the lagging arm midpoint, the resonance inductor Lx and the resonance capacitor Cx are connected in series, and then are connected with the primary side loop of the main transformer.

[0041] Through the above circuit design, soft switching of the silicon carbide power switch tube can be realized, and the principle is as follows: crosstalk refers to electromagnetic interference coupling to the driving circuit caused by the rapid change of voltage or current in the switching process of the power switch tube (or switch tube); when the switch tube bridge arm is turned on, due to the influence of the left and right saturation inductances of the resonant inductor, the loop current rises slowly, the switch tube is turned on with zero current, and there is no rapid current interference in the conversion process from the cutoff state to the amplification state and the saturation state, so the crosstalk can be effectively reduced; when the leading arm is turned off, due to the existence of the parallel capacitor of the leading arm, the voltage change rate of the switch tube is low, and the crosstalk is reduced; when the lagging arm is turned off, due to the action of the resonant capacitor, the freewheeling current is reduced to zero or a very small value, and there is no rapid voltage and current change when the switch is turned off, so the crosstalk is effectively reduced. Therefore, by designing the above resonant circuit, the voltage across the power switch tube or the current flowing through the power switch tube can change according to the above rules during the switching process of the silicon carbide power switch tube, so that the switch can be turned on and off at zero current or zero voltage, which can greatly reduce the crosstalk problem and ensure the switching reliability of the silicon carbide power switch tube.

[0042] Further, the working frequency of the main transformer T is 20 kHz-100 kHz, the secondary side of the transformer is connected with a secondary rectification circuit, the secondary rectification circuit includes two parallel diodes D5, D6, and the two diodes D5, D6 are connected in series with the output reactance Lo and connected to the external load as the output end of the silicon carbide inverter to supply power to the external load, which includes a high-frequency welding machine and the like. In addition, the secondary side of the main transformer T is also connected with a commutation inductor L1 to provide a commutation path for the no-load working condition. Under the no-load condition of the silicon carbide inverter, the no-load working condition is introduced through the commutation inductor L1, the commutation of the leading arm parallel capacitor C2, C3 is completed, the no-load soft switching of the silicon carbide inverter is realized, and the direct discharge of the leading arm capacitor to the power switch tube under no-load is avoided.

[0043] As shown in Figure 2 The driving module optimizes the driving signal through the driving circuit or the driving chip to drive the on-off of the silicon carbide power switch tube. In the embodiment, the driving module is realized by using a driving circuit, which can also be called a pulse transformer driving circuit, for optimizing the driving signal and driving the on-off of the silicon carbide power switch tubes Q1-Q4 to realize high-frequency switching action.

[0044] Specifically, the driving circuit comprises an isolation pulse transformer, a driving resistor, a negative voltage circuit, a pre-charge circuit, a Miller clamp circuit, a gate overvoltage protection circuit and a Miller capacitor C13. The primary side of the isolation pulse transformer is connected with the control module, and the primary side is connected with the driving signal output by the control module. The secondary side of the isolation pulse transformer is connected with the Miller clamp circuit, the pre-charge circuit, the gate overvoltage protection circuit and the Miller capacitor in parallel. The Miller clamp circuit and the pre-charge circuit are connected in series with the negative voltage circuit. The same end of the secondary side of the isolation pulse transformer is connected to one end of the driving resistor R1. The Miller clamp circuit comprises a driving MOSFET Q5, a capacitor C11 and a diode D1. The other end of the driving resistor is connected with the drain of the driving MOSFET Q5 and one end of the capacitor C11 respectively. The other end of the capacitor C11 is connected with the gate of the driving MOSFET Q5. The capacitor C11 and the driving resistor R1 form an RC absorption circuit for protecting the gate of the driving MOSFET Q5, preventing overvoltage breakdown and ensuring the stability and reliability of the driving circuit.

[0045] In addition, the gate of the driving MOSFET Q5 is also connected with the cathode of the diode D1, and the source of the driving MOSFET Q5 is connected with the anode of the diode D1. The drain-source of the driving MOSFET Q5 is connected in parallel with the gate-source of the driven silicon carbide power switch (i.e. Q1-Q4). That is, the gate of the silicon carbide power switch is connected with the drain of the driving MOSFET Q5, and the source of the driving MOSFET Q5 is connected with the source of the silicon carbide power switch through a voltage stabilizing tube. Preferably, the driving MOSFET in the embodiment is an N-type MOS tube.

[0046] Through the design of the above-mentioned Miller clamp circuit, the driving MOSFET is connected in parallel between the gate-source of the main power silicon carbide power switch. During the off process and period of the main power switch, the driving MOSFET is turned on, the gate-source voltage of the silicon carbide power switch is effectively clamped in a negative voltage state, and the problem of gate voltage oscillation caused by parasitic inductance and Miller effect is effectively inhibited in cooperation with the gate-source capacitor. At the same time, the resistor and the capacitor form an RC absorption circuit, which can effectively protect the gate of the driving MOSFET.

[0047] The above-mentioned negative voltage circuit comprises a voltage stabilizing tube ZD1 and a capacitor C12 connected in parallel. The anode of the voltage stabilizing tube ZD1 is connected with the anode of the diode D1 and one end of the capacitor C12. The cathode of the voltage stabilizing tube ZD1 is connected with the other end of the capacitor C12. Through this design, the driving MOSFET and the negative voltage generation circuit form an off discharging circuit, which can pull down the gate voltage to a negative voltage (such as -5V) in the off stage, accelerate the off process and inhibit oscillation, and the circuit is stable and reliable.

[0048] The pre-charge circuit includes a diode D2 and a resistor R2 connected in series, the anode of the diode D2 is connected to the driving resistor R1, the cathode of the diode D2 is connected to the resistor R2, and the other end of the resistor R2 is connected to the cathode of the ZD1 voltage stabilizing tube. Through the design of the negative voltage and the pre-charge circuit, the pre-charge of the instantaneous negative voltage circuit can be quickly completed by using the diode and resistor series circuit, and the rapid establishment of the negative voltage is ensured.

[0049] The gate overvoltage protection circuit includes two transient suppression diodes ZD2 and ZD3, which are combined into a bidirectional transient suppression diode, the cathode of the transient suppression diode ZD2 is connected to the driving resistor R1, the anode of the transient suppression diode ZD2 is connected to the anode of the transient suppression diode ZD3, and the cathode of the transient suppression diode ZD3 is connected to the cathode of the ZD1 voltage stabilizing tube. Through the design of the above-mentioned gate overvoltage protection circuit, the gate of the silicon carbide power switch tube can be protected, so that the silicon carbide power switch tube is not affected by voltage spikes, overvoltage breakdown is prevented, and the normal and reliable operation of the silicon carbide power switch tube is protected.

[0050] The primary side of the isolation pulse transformer is connected to the output driving signal of the control module, and signal and energy transmission are performed through the pulse transformer. When the secondary side of the pulse transformer outputs a high level, the silicon carbide power switch tube is turned on, and vice versa. The working mode of the driving circuit is as follows: Working mode 1: The same name end of the isolation pulse transformer T1 outputs a voltage, and the current path is isolation pulse transformer T1→driving resistor R1→capacitor C11→isolation pulse transformer T1, and isolation pulse transformer T1→driving resistor R1→capacitor C13→capacitor C12→diode D1→isolation pulse transformer T1. In this process, the voltages of capacitors C11, C12 and C13 linearly rise.

[0051] Working mode 2: After the voltages of capacitors C11 and C13 reach a certain value, the diode D2 and the resistor R2 in the pre-charge circuit and the two transient suppression diodes ZD2 and ZD3 in the gate overvoltage protection circuit work to complete the charging of the negative voltage circuit C12, thereby rapidly establishing a negative voltage. The current path is: isolation pulse transformer T1→driving resistor R1→diode D2→resistor R2→capacitor C12→diode D1→isolation pulse transformer T1, isolation pulse transformer T1→driving resistor R1→transient suppression diode ZD2→transient suppression diode ZD3→capacitor C12→diode D1→transient suppression diode T1.

[0052] Working mode 3: the primary side inverter of the isolation pulse transformer T1 is turned off, the secondary side of the transformer T1 is short-circuited, the voltage of the capacitor C13 charges the gate of Q5 through the driving resistor R1 and the isolation pulse transformer T1, and Q5 is turned on quickly. At this time, the current path is: capacitor C13→driving resistor R1→isolation pulse transformer T1→driving MOSFET Q5→voltage stabilizing tube ZD1→capacitor C13. Wherein, after Q5 is turned on, the voltage of the capacitor C11 is positive at the top and negative at the bottom, the voltage of the gate of Q5 is positive at the top and negative at the bottom, the voltage at the top of C11 is slightly higher than the voltage at the gate of Q5, and C11 cannot charge the gate capacitor of Q5. After Q5 is turned on, the capacitor C13 is discharged quickly through Q5, the current path is capacitor C13→MOS tube Q5→voltage stabilizing tube ZD1→capacitor C13, Q5 is turned on at this time, and the capacitor C13 is discharged quickly and the charge in the capacitor C12 below is transferred to the capacitor C13, realizing negative voltage clamping. Through the above process, the gate voltage can be pulled down to negative voltage (such as-5V), so that the voltage between the gate and the source is lower than the opening voltage, thereby quickly turning off the main power switch tube.

[0053] Working mode 4: the isolation pulse transformer T1 outputs voltage at the opposite end, Q5 is turned on, the capacitor C12 in the negative voltage circuit is discharged to the capacitor C13, the negative voltage is added to the gate of the main power switch tube, and the Miller clamp is performed to prevent mis-conduction. The current path is capacitor C12→capacitor C13→driving MOSFET Q5.

[0054] Based on the above switching process, the pre-charge circuit can realize the rapid establishment of the negative voltage circuit voltage (within 10µS), and at the same time, the Miller clamp circuit ensures the reliability of the high-speed switching process of the main power switch tube.

[0055] Preferably, by reasonably selecting the magnetic core material, the number of turns ratio, the winding wire diameter and other parameters of the pulse transformer, optimizing the winding method and shielding structure, the driving ability and anti-interference ability of the pulse transformer can be improved, and it can be ensured that the driving signal can be reliably transmitted to the silicon carbide power switch tube. For example, by reducing the pulse transformer turn ratio, and cooperating with the voltage stabilizing tube, the negative voltage of the power switch tube can be realized, and sufficient negative voltage can ensure that the switch tube will not be mis-triggered; at the same time, the on-resistance is increased and the off-resistance is reduced, so that the turn-on speed of the power switch tube is reduced, the voltage change rate of the power switch tube is reduced, and the crosstalk is reduced.

[0056] As another embodiment, the resonant inductor for the inverter power supply is at least one magnetic body structure sleeved on the wire of the switching power supply AC loop; or in the switching power supply AC loop, a wire is connected in series, and at least one magnetic body structure is sleeved on the wire.

[0057] In the embodiment, the magnetic body structure is sleeved on the wire of the high-frequency AC loop of the inverter power supply, and specifically, it is sleeved on the wire of the primary side loop of the main transformer as the resonant inductor Lx.

[0058] Further, each magnet structure is independently fixed, and a gap can be reserved between two adjacent magnet structures, which is beneficial to heat dissipation of each magnet structure; in general, the interval between two adjacent magnet structures is greater than 5 mm. The shape of the magnet structure can be ring-shaped, cylindrical, square, spherical or other special-shaped structures, and a through hole for passing a wire is left in the middle. For the ring-shaped magnet structure, the shape can also be a circular ring, a square ring, a triangular ring or other polygonal ring structures, which can be selected according to actual needs.

[0059] The magnet structure of the embodiment can be a completely closed ring, cylinder, square, sphere or other special-shaped structure, which is equivalent to a single-turn saturated inductance; the completely closed structure makes the leakage inductance of the magnet structure very small, and the adjacent magnetic rings do not affect each other. At the same time, since the magnetic circuit is closed and has no air gap, the magnetic resistance is very small, the magnetic flux is almost completely constrained inside the magnetic ring, the magnetic field leakage can be suppressed, the circuit noise radiation can be reduced, and the electromagnetic compatibility can be optimized.

[0060] In addition, in the embodiment, the control module includes a sampling circuit and a control chip, the sampling circuit includes a current sampling circuit and a voltage sampling circuit, the current sampling circuit uses a current Hall sensor and an operational amplifier shaping circuit, and the voltage sampling circuit uses a differential circuit and an operational amplifier shaping filter circuit. The sampling circuit is used to collect the output current of the soft-switching inverter module, the output voltage and the output current of the welding power supply, and feed back the collected voltage and current signals to the control chip. The control chip is used to generate a PWM driving pulse based on a preset driving strategy, and to dynamically adjust the dead time and the PWM pulse width of the driving pulse based on the collected voltage and current signals to generate a driving signal. The driving signal generated by the above is used to drive the on-off of the silicon carbide power switch tube in the soft-switching inverter module through the driving circuit isolation, so as to realize the high-frequency soft-switching action of the soft-switching inverter module under multiple working conditions.

[0061] Embodiment two Considering that the output capacitance of the full-bridge leading arm power switch tube and the loop stray inductance form high-frequency common-mode oscillation at the moment when the switch tube is turned on or turned off, the voltage and current change sharply to cause high-frequency drive crosstalk, which affects the reliability of the silicon carbide power device, and the high-frequency oscillation of the parallel capacitor in the leading arm will cause the abnormal rise of the capacitor temperature rise, which affects the reliability and service life of the capacitor. Therefore, in order to further improve the reliability of the full-bridge soft-switching inverter circuit and avoid the additional capacitor loss caused by high-frequency oscillation, the embodiment adds an inductor in series between the midpoint of the leading arm and the parallel capacitor C2 and C3 in the leading arm based on the full-bridge soft-switching inverter circuit structure in the above embodiment one. The inductor can filter the high-frequency common-mode oscillation without affecting the normal soft-switching operation, realize low-loss and high-precision high-frequency oscillation suppression, and reduce the drive crosstalk and the capacitor temperature rise.

[0062] The series-connected inductor has frequency-selective impedance characteristics, which can be characterized by an impedance-frequency curve. The impedance of the inductor is positively correlated with the frequency. The inductor has low impedance at low frequencies and high impedance at high frequencies. Preferably, the high-frequency range is above 1 MHz. In this frequency range, the impedance of the inductor is much greater than the equivalent impedance of the high-frequency oscillation loop and the crosstalk path, so that the amplitude of the high-frequency current (such as MHz-level oscillation current) can be suppressed to a range that does not affect the reliability of the circuit. The low-frequency range is below 1 MHz, usually tens of kHz to several hundred kHz, such as 20 kHz to 100 kHz. In this frequency range, the impedance of the inductor is much smaller than the equivalent impedance of the main loop (such as the on-state impedance of a silicon carbide switch in a full-bridge inverter circuit and the impedance of a resonant circuit), so that the main power transmission efficiency is not reduced.

[0063] As an embodiment, the inductor can use a magnetic core with the above-mentioned frequency-selective impedance characteristics. For example, the inductor can be wound around a ferrite magnetic core or any of a nickel-zinc ferrite magnetic bead or a manganese-zinc ferrite magnetic bead. In this embodiment, the inductor uses a nickel-zinc ferrite magnetic bead, which has an impedance of 20 Ω or more at a frequency of 25 MHz and an impedance of 40 Ω or more at a frequency of 100 MHz. In addition, the initial permeability of the nickel-zinc ferrite magnetic bead is 850 H / m, and the saturation magnetic flux density is 0.35 T to 0.46 T.

[0064] In this embodiment, the introduced inductor can be a common-mode inductor or a differential-mode inductor. When the common-mode inductor is used, one end of each of the two parallel capacitors of the leading arms that are close to each other is connected in series as a common terminal of the common-mode inductor, and then connected to the midpoint of the leading arm through the common-mode inductor. By limiting the common terminal connected in series, it is ensured that the inductor can filter out high-frequency common-mode oscillation signals. At the same time, by limiting the common terminal, as the high-frequency common-mode signal current increases, the inductance impedance also increases significantly, further optimizing the filtering effect.

[0065] When two independent and uncoupled differential-mode inductors are used, one end of each of the two parallel capacitors of the leading arms that are close to each other is connected in series to the corresponding differential-mode inductor, and then connected to the midpoint of the leading arm through the differential-mode inductor.

[0066] Further, the common-mode inductor is used as an example for illustration. Specifically, Figure 11As shown, in the full-bridge inverter circuit, capacitors C2 and C3 are connected in parallel to the silicon carbide power switches Q1 and Q2 that form the leading arm, respectively. No capacitors are connected in parallel to the silicon carbide power switches Q3 and Q4 that form the lagging arm. One end of the parallel capacitor C2 in the leading arm is connected to the first silicon carbide power switch Q1 in the leading arm, and one end of the parallel capacitor C3 in the leading arm is connected to the second silicon carbide power switch Q2 in the leading arm. The ends of the parallel capacitors C2 and C3 that are close to each other in the leading arm serve as the same-name terminals of the high-frequency common-mode inductor, and are connected to the midpoint of the leading arm through a series common-mode inductor.

[0067] The introduced common-mode inductor has frequency-selective impedance and common-mode inductance characteristics. It exhibits low impedance at low frequencies, that is, low impedance within the normal operating frequency range of the circuit, without affecting the transmission of main power. At the same time, it exhibits high impedance at high frequencies, and the impedance increases with the increase of common-mode current, that is, it exhibits high impedance in the high-frequency oscillation range of the circuit, so as to effectively block high-frequency oscillation current.

[0068] High-frequency oscillations can be suppressed by introducing common-mode inductor compensation. The principle or mechanism is as follows: the common-mode inductor has advantages such as low permeability, high resistivity, high cutoff frequency, low high-frequency loss, and adjustable composition. As a result, it presents high impedance at high frequencies. For high-frequency common-mode signals, the magnetic fields reinforce each other, and the high impedance can effectively suppress them. For differential-mode signals that are working normally in the circuit, it presents low impedance and does not affect the normal soft-switching operation.

[0069] Similarly, the introduced differential-mode inductor also has frequency-selective impedance, exhibiting low impedance at low frequencies and high impedance at high frequencies, which will not be elaborated further here.

[0070] by Figure 12 , Figure 13 Taking the silicon carbide full-bridge soft-switching converter circuit with inductor compensation as an example, the specific process by which this circuit achieves high-frequency oscillation suppression is as follows: Normal working mode 1 like Figure 12 As shown, the lead arm Q2 is turned off early, capacitor C2 discharges, and capacitor C3 is charged. The current path is: bus capacitor C1 → power switch Q3 → resonant inductor Lx → main transformer T → oscillation capacitor Cx → common mode inductor → capacitor C3 → bus capacitor C1, capacitor C1 → power switch Q3 → resonant inductor Lx → main transformer T → oscillation capacitor Cx → common mode inductor → capacitor C2.

[0071] At this time, the common mode inductor current direction is from A to B, from A to C, for low frequency common mode current, the common mode inductor low impedance, for high frequency common mode current, the common mode inductor high impedance. The high frequency oscillation between the parallel capacitor C2, C3 of the leading arm and the junction capacitor of the power switch tube is greatly suppressed after the high frequency common mode inductor (high frequency impedance is large), the power switch tube drain-source voltage oscillation is obviously reduced, the drive crosstalk problem caused by voltage oscillation is obviously improved, and the heating of the capacitor of the leading arm is reduced.

[0072] Normal working mode 2 As shown in Figure 13 , the leading arm Q1 is turned off in advance, the capacitor C2 is charged, the capacitor C3 is discharged, and the current path is: bus capacitor C1→capacitor C2→magnetic bead common mode inductor→oscillation capacitor Cx→main transformer T→resonant inductor Lx→power switch tube Q4→bus capacitor C1, capacitor C3→common mode inductor→oscillation capacitor Cx→main transformer T→resonant inductor Lx→power switch tube Q4→capacitor C3.

[0073] At this time, the common mode inductor current direction is from B to A, from C to A, for low frequency common mode current, the common mode inductor low impedance, for high frequency common mode current, the common mode inductor high impedance. The high frequency oscillation between the parallel capacitor C2, C3 of the leading arm and the junction capacitor of the power switch tube is greatly suppressed after the high frequency common mode inductor (high frequency impedance is large), the power switch tube drain-source voltage oscillation is obviously reduced, the drive crosstalk problem caused by voltage oscillation is obviously improved, and the heating of the capacitor of the leading arm is reduced.

[0074] Further, in the normal operation process of the full-bridge soft switching inverter circuit, the common mode inductor performs the above normal working mode 1, 2, in the normal working mode, the common mode inductor almost does not work, which does not affect the normal soft switching process of the welding machine; and at the turn-on and turn-off moment of the leading arm power switch tube Q1, Q2, high frequency oscillation is generated in the loop at this time, the common mode inductor enters the high frequency oscillation working mode, the common mode inductor presents large impedance in this mode, at this time, the common mode inductor works in the inverter circuit, and precisely suppresses the high frequency common mode oscillation in the switching process, specifically: High frequency oscillation working mode As shown in Figure 14 , the junction capacitor (internal capacitor of the leading arm) of the leading arm Q1 is CQ1, the junction capacitor of the leading arm Q2 is CQ2, CQ1 and C2 and the loop inductance occur high frequency oscillation in the switching process of Q1, Q2, CQ2 and C3 and the loop inductance occur high frequency oscillation in the switching process of Q1, Q2.

[0075] In the oscillation process, the current is as shown in Figure 14As shown, the sum of CQ1 and CQ2 voltages is the bus voltage on C3, CQ1 discharges, CQ2 charges, the common-mode inductor current direction is from the super-ear parallel capacitor end to the silicon carbide power switch tube end, that is, from B to A, from C to A, the two current sizes are equal, the direction is the same, the frequency reaches MHz, at this time the common-mode inductor presents high impedance under high frequency, at the same time the common-mode current strengthens through the common-mode inductor magnetic field, the impedance is larger, and the high-frequency common-mode current can be obviously suppressed.

[0076] Correspondingly, in the oscillation process, CQ1 charges, CQ2 discharges, and the process is similar to the above, which will not be repeated here.

[0077] By connecting the common-mode inductor in series between the two super-ear parallel capacitors and the bridge arm midpoint, using the low-frequency low-impedance, high-frequency high-impedance characteristics and the common-mode inductor characteristics, the high-frequency oscillation formed by the power switch tube junction capacitor, loop inductor and super-ear parallel capacitor can be accurately suppressed without affecting the normal operation frequency of the full-bridge converter circuit, the high-frequency crosstalk problem of the super-ear parallel capacitor is solved, the influence of crosstalk on high-frequency switching is further reduced, and reliable high-frequency switching action of the full-bridge soft switching inverter circuit based on silicon carbide power devices is realized.

[0078] Embodiment three The embodiment proposes a control method of a full-bridge soft switching inverter circuit based on silicon carbide power devices, which is used to control the operation of the full-bridge soft switching inverter circuit proposed in embodiment one or embodiment two, and the control method comprises: Collecting the output current of the soft switching inverter module, the output voltage and output current of the welding power supply; Based on the preset driving strategy, the PWM driving pulse is initially generated, and then the current working condition is judged and the dead time and PWM pulse width of the driving pulse are dynamically adjusted according to the collected voltage and current signals to generate a driving signal; Based on the optimized driving signal, the stable high-frequency on-off of the silicon carbide power switch tube is driven to realize high-frequency switching action.

[0079] The preset driving strategy is that the first silicon carbide power switch tube Q1 and the fourth silicon carbide power switch tube Q4 are turned on at the same time, the first silicon carbide power switch tube Q1 is turned off in advance, and the fourth silicon carbide power switch tube Q4 is turned off with a delay; the second silicon carbide power switch tube Q2 and the third silicon carbide power switch tube Q3 are turned on at the same time, the second silicon carbide power switch tube Q2 is turned off in advance, and the third silicon carbide power switch tube Q3 is turned off with a delay; a dead zone is set between the third silicon carbide power switch tube Q3 and the fourth silicon carbide power switch tube Q4, and all power switch tubes in the dead zone are turned off.

[0080] Further, when the output voltage and current are within the rated range, it is determined that the rated working condition is reached. When the current working condition is the rated working condition, the voltage and current signals collected are used to adjust the PWM driving pulse through the double-loop control of the voltage loop and the current loop, to generate a driving signal for: The voltage loop is set as the outer control loop, and the current loop is set as the inner control loop. The feedback output voltage is compared with the set voltage, and the difference is output to the current loop given value through the PI regulator. Then, the given current is compared with the feedback output current, and the difference is output to the PWM generator through the PI regulator, to output the corresponding PWM driving pulse.

[0081] In the above manner, the PWM pulse width can be adjusted in real time according to different arc states, to adjust the working condition of the main circuit. As shown in Figure 9 and Figure 10 , the pulse width of the leading arm power switch Q1 and Q2 is adjusted. The smaller the output current is, the narrower the pulse width of the leading arm power switch Q1 and Q2 is, and the smaller the off current of the lagging arm power switch Q3 and Q4 is, and the smaller the off loss is.

[0082] As another embodiment, the driving signal of the pulse transformer driving circuit is optimized. The amplitude, width, and rise / fall time of the driving signal are adjusted, i.e., the amplitude of the driving signal is adjusted by adjusting the turns ratio of the pulse transformer, the proportion of positive and negative voltage distribution is determined by the selection of the voltage stabilizing tube, and the driving width, rise / fall time, etc. are adjusted by adjusting the driving parameters (such as driving resistance, gate capacitance, etc.), so as to further improve the reliability of the driving circuit.

[0083] Taking the full-bridge soft-switching inverter circuit proposed in Embodiment One as an example, the working process of the full-bridge soft-switching inverter circuit is as follows: Working Mode 1 As shown in Figure 3 , the silicon carbide power switch Q2 and Q3 are turned on, and the current path is: capacitor C1→ silicon carbide power switch Q3→ resonant inductor Lx→ main transformer T→ resonant capacitor Cx→ silicon carbide power switch Q2→ bus capacitor C1. When turned on, the Q2 and Q3 current rises with a delay due to the saturation inductance effect of the resonant inductor Lx, to realize soft turn-on.

[0084] Working Mode 2 As shown in Figure 4As shown, the silicon carbide power switch Q2 is turned off early. Due to the lead-arm capacitor, the voltage across Q2 cannot change abruptly, achieving zero-voltage turn-off (ZVS) for Q2. The current path is: bus capacitor C1 → silicon carbide power switch Q3 → resonant inductor Lx → main transformer T → resonant capacitor Cx → capacitor C3 → bus capacitor C1, capacitor C2 → silicon carbide power switch Q3 → resonant inductor Lx → main transformer T → resonant capacitor Cx → capacitor C2.

[0085] Working Mode 3 like Figure 5 As shown, the voltage of capacitor C3 rises and the voltage of capacitor C2 falls. When the voltage of C2 drops to 0, the anti-parallel diode of inverter power switch Q1 is turned on. The current freewheeling path is: diode D1 → silicon carbide power switch Q3 → resonant inductor Lx → main transformer T → resonant capacitor Cx → silicon carbide power switch Q2 → capacitor C1.

[0086] Working Mode 4 like Figure 6 As shown, the silicon carbide power switch Q3 is turned off with approximately zero current. The current path is: capacitor C1 → diode D4 → resonant inductor Lx → main transformer T → resonant capacitor Cx → diode D1 → capacitor C1.

[0087] Working Mode 5 like Figure 7 As shown, silicon carbide power switches Q1 and Q4 are turned on at zero voltage. The current path is: capacitor C1 → silicon carbide power switch Q1 → resonant capacitor Cx → main transformer T → resonant inductor Lx → silicon carbide power switch Q4 → capacitor C1.

[0088] Furthermore, after completing the aforementioned current commutation, the above steps are repeated, and the driving process is as follows: Figure 9 and Figure 10 As shown, it is: At time t0, the leading arm Q1 and the lagging arm Q4 are turned on. Due to the effect of the saturated inductance, the current I rises very slowly, and the leading arm Q1 and the lagging arm Q4 are turned on with zero current. At time t1, the lead arm Q1 is turned off and enters the freewheeling stage. Due to the reverse voltage effect of the resonant capacitor, the primary current drops rapidly. At time t2, the lagging arm Q4 turns off with approximately zero current and enters the dead zone freewheeling phase td; At time t3, the current reverses direction again.

[0089] By implementing soft-switching control through the aforementioned driving strategy, zero current is ensured during the conduction of the inverter's leading and lagging arms (due to the effect of saturated inductance), near-zero voltage is achieved during the leading arm's turn-off, and zero current is achieved during the lagging arm's turn-off under most operating conditions. Throughout the switching process, there are no drastic voltage or current changes. While the changing current generates a magnetic field and the changing voltage generates an electric field, soft switching effectively reduces crosstalk. Furthermore, during the soft-switching process, considering the small capacitance of the leading arm during turn-off and the reduced rate of voltage change, some crosstalk still exists, and the Miller effect can still generate drive spikes. Simultaneously, under some operating conditions, the lagging arm experiences a certain current during turn-off, with a rapid rate of change, which also generates crosstalk. To address these issues, this embodiment further incorporates negative voltage turn-off and negative voltage clamping into the driving circuit to effectively control drive spikes generated by crosstalk and ensure the reliability of the switching process.

[0090] As one implementation method, this embodiment can also dynamically adjust the dead time of the drive pulse and the PWM pulse width based on the collected voltage and current signals to generate a drive signal. Specifically, the load is the electric arc during welding, and the operating condition refers to the circuit state. The entire circuit can operate in states such as no-load, short circuit, and maximum power. The control process is as follows: (1) When the output voltage is the no-load voltage and the output current is 0, it is judged to be in no-load state. When the working state is no-load state, the current loop is given a fixed value, and the PWM generator outputs a small pulse width for the leading arm, thereby controlling the dead zone of the leading arm and the lagging arm, ensuring that the dead zone of the leading arm and the lagging arm is large enough under no-load conditions. This design of fixed pulse width for the leading arm under no-load conditions is an optimized design based on the commutation problem of the soft-switching no-load leading arm capacitor. Compared with the existing method, this design can ensure that the leading arm capacitor completes commutation through the commutation circuit, preventing discharge to the switching transistor.

[0091] Specifically, under no-load conditions, the operation process of the full-bridge soft-switching inverter circuit is as follows: Working Mode 6 like Figure 8 As shown, when the power supply output is unloaded, in order to ensure that the inverter operates in a soft-switching state and complete the commutation of the lead-arm capacitors C2 and C3, the commutation inductor L1 acts as a dummy load to provide a commutation current loop for C2 and C3. The current path is as follows: capacitor C1 → silicon carbide power switch Q3 → resonant inductor Lx → main transformer T → resonant capacitor Cx → capacitor C3 → capacitor C1, capacitor C2 → silicon carbide power switch Q3 → resonant inductor Lx → main transformer T → resonant capacitor Cx → capacitor C2.

[0092] (2) When the output voltage is lower than the set value (10V in this embodiment) and the output current is greater than the rated current, it is judged to be in a short circuit state. When the working state is in a short circuit state, the pulse width of the leading arm is controlled to be very small (such as the set small pulse width state), and the lagging arm is a zero current switch.

[0093] (3) When the output power reaches the preset maximum value, it is judged to be in the maximum power point state. When the working state is the maximum power point state, the pulse width of the lead arm is the maximum. A fixed dead zone is set between the lead arm and the lagging arm. This dead zone can ensure that the lagging arm is turned off with a small current when it is turned off at the maximum power point, reduce the drive crosstalk spike when the lagging arm is turned off, keep the drive crosstalk within a controllable range, ensure that the spike is not too high, avoid misleading turn-on causing bridge arm shoot-through, or affect the long-term life of silicon carbide power devices.

[0094] The above control strategy is mainly achieved by setting limit values. For example, under no-load conditions, there is a limit pulse width to ensure commutation; under short-circuit conditions, the short-circuit pulse width is determined according to the set short-circuit current, with no minimum limit; under maximum power point conditions, the leading arm pulse width has a maximum adjustment limit to ensure soft turn-off of the lagging arm and ensure reliability.

[0095] By combining the above control methods, complete control of the welding process of the welding machine can be achieved, specifically including control strategies under no-load, short-circuit, maximum power point, and rated output conditions.

[0096] As one implementation method, during the operation of a full-bridge soft-switching inverter circuit with a common-mode inductor controlled by the above-described control method, the common-mode inductor alternates between a normal operating mode and a high-frequency oscillation operating mode, including: During the operation of the full-bridge soft-switching inverter circuit, at the instants when the power switches Q1 and Q2 are turned on and off, high-frequency oscillations are generated in the loop formed by the junction capacitance of the power switches Q1 and Q2, their corresponding parallel capacitance, and the loop inductance. The common-mode inductor enters the high-frequency oscillation working mode. In this mode, the common-mode inductor has high impedance and the impedance increases with the increase of the high-frequency common-mode current, which is used to suppress high-frequency oscillations and high-frequency common-mode current. At other times, the common-mode inductor enters normal operating mode, in which it exhibits low impedance and has no additional effect on the operation of the full-bridge soft-switching inverter circuit.

[0097] By using the above methods, high-frequency common-mode oscillations can be filtered out without affecting normal soft-switching operation, achieving low-loss, high-precision high-frequency oscillation suppression, and reducing drive crosstalk and capacitor temperature rise.

[0098] In summary, the full-bridge soft-switching inverter circuit based on silicon carbide power devices proposed in this embodiment can be applied to scenarios such as welding machine power supplies. Through the design of the soft-switching inverter, the optimization of the drive circuit, and the improved control method, a reliable and effective solution can be provided for the high-frequency application of silicon carbide power devices in welding machine power supplies.

[0099] The above description is merely a preferred embodiment of the present invention and is not intended to limit the invention. Various modifications and variations can be made to the present invention by those skilled in the art. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

[0100] While the specific embodiments of the present invention have been described above in conjunction with the accompanying drawings, this is not intended to limit the scope of protection of the present invention. Those skilled in the art should understand that various modifications or variations that can be made by those skilled in the art without creative effort based on the technical solutions of the present invention are still within the scope of protection of the present invention.

Claims

1. A full-bridge soft-switching inverter circuit based on silicon carbide power devices, characterized in that, The system includes a soft-switching inverter module, which comprises a full-bridge inverter circuit, a resonant inductor, a resonant capacitor, and a main transformer. The full-bridge inverter circuit consists of four silicon carbide power switching transistors Q1 to Q4. Q1 and Q2 are connected in series to form the leading arm, and Q3 and Q4 are connected in series to form the lagging arm. Capacitors C2 and C3 are connected in parallel across the two ends of the leading arm Q1 and Q2, respectively. The midpoint of the leading arm, the midpoint of the lagging arm, the resonant inductor, and the resonant capacitor are connected in series to the primary circuit of the main transformer.

2. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 1, characterized in that, It also includes a drive module; the drive module isolates and optimizes the drive signal through a drive circuit or drive chip to drive the silicon carbide power switch to switch on and off.

3. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 2, characterized in that, The driving module is implemented using a driving circuit, which includes an isolation pulse transformer. The primary side of the isolation pulse transformer is connected to the driving signal, and the secondary side is connected to a Miller clamping circuit, a negative voltage circuit, and a pre-charge circuit. The Miller clamping circuit includes a driving MOSFET and a diode D1 between its source and gate, and a capacitor C11 between its drain and gate. The drain of the driving MOSFET is connected to the gate of the silicon carbide power switch. The driving MOSFET, together with the diode D1, clamps the stable negative voltage generated by the pre-charge circuit and the negative voltage circuit into a negative voltage state. The capacitor C11 absorbs the instantaneous overvoltage of the gate of the driving MOSFET.

4. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 3, characterized in that, The secondary side of the isolation pulse transformer is connected to a driving resistor, and the driving resistor is connected to capacitor C11 to form an RC absorption circuit. The negative pressure circuit is located between the Miller clamping circuit and the pre-charge circuit, and includes a Zener diode ZD1 and a capacitor C12 connected in parallel; the anode of the Zener diode ZD1 is connected to the anode of the diode D1 and one end of the capacitor C12, and the cathode of the Zener diode ZD1 is connected to the other end of the capacitor C12. The pre-charge circuit includes a diode D2 and a resistor R2 connected in series. The anode of the diode D2 is connected to the driving resistor, the cathode of the diode D2 is connected to the resistor R2, and the other end of the resistor R2 is connected to the cathode of the Zener diode ZD1.

5. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 3, characterized in that, The driving circuit also includes a gate overvoltage protection circuit and a gate capacitor connected to the secondary side of the isolation pulse transformer. The gate overvoltage protection circuit includes two transient suppression diodes ZD2 and ZD3, which are combined into a bidirectional transient suppression diode. The cathode of transient suppression diode ZD2 is connected to the driving resistor R1, the anode of transient suppression diode ZD2 is connected to the anode of transient suppression diode ZD3, and the cathode of transient suppression diode ZD3 is connected to the cathode of Zener diode ZD1.

6. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 1, characterized in that, It also includes a control module, which comprises a sampling circuit and a control chip; The sampling circuit includes a current sampling circuit and a voltage sampling circuit, which are used to collect the output current of the soft-switching inverter module, the output voltage and output current of the welding power supply, and feed the collected voltage and current signals back to the control chip. The control chip is used to initially generate PWM drive pulses based on a preset drive strategy, and then determine the current operating condition and dynamically adjust the dead time of the drive pulse and the PWM pulse width according to the collected voltage and current signals to generate drive signals.

7. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 1, characterized in that, The soft-switching inverter module also includes a three-phase rectifier bridge, bus capacitors, commutation inductors, and secondary rectifier circuits. The three-phase rectifier bridge, bus capacitor, and full-bridge inverter circuit are connected in parallel. External low-frequency AC power is connected to the three-phase rectifier bridge, and after being converted into DC power by the three-phase rectifier bridge, it is output to the full-bridge inverter circuit through the bus capacitor. The secondary side of the main transformer is connected to the commutation inductor and the secondary rectifier circuit respectively. The commutation inductor is used to introduce reactive current to commutate the parallel capacitor of the lead arm, providing a commutation path for no-load operation. The secondary rectifier circuit includes two parallel diodes. The two parallel diodes are connected in series with the output reactance and are connected to the external load as the output terminal of the silicon carbide inverter to supply power to the external load.

8. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 1, characterized in that, The resonant inductor is: at least one magnetic structure sleeved on a conductor in the AC circuit of the switching power supply; or, in the AC circuit of the switching power supply, a conductor is connected in series, and at least one magnetic structure is sleeved on the conductor. The magnet structure is a closed structure, and the shape of the magnet structure is ring-shaped, spherical, cylindrical, square, or other irregular shape.

9. A full-bridge soft-switching inverter circuit based on silicon carbide power devices, characterized in that, In the full-bridge soft-switching inverter circuit according to any one of claims 1-8, an inductor is connected in series between the midpoint of the lead arm and the parallel capacitors C2 and C3 of the lead arm.

10. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 9, characterized in that, The impedance of the inductor is positively correlated with the frequency; the inductor exhibits low impedance at low frequencies and high impedance at high frequencies.

11. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 10, characterized in that, The inductor is an inductor wound with a ferrite core, or any one of nickel-zinc ferrite beads or manganese-zinc ferrite beads.

12. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 10, characterized in that, The inductor is a common-mode inductor. The ends of the two parallel capacitors in the forearms that are close to each other are connected in series with the common-mode inductor as the same-name terminals of the common-mode inductor. After passing through the common-mode inductor, the capacitors are then connected to the midpoint of the forearms.

13. The full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 10, characterized in that, The inductors are two independent and uncoupled differential-mode inductors; The two parallel capacitors of the forearms are connected in series with their corresponding differential-mode inductors at their closest ends, and then connected to the midpoint of the forearms through the differential-mode inductors.

14. A control method for a full-bridge soft-switching inverter circuit based on silicon carbide power devices, characterized in that, The control method for controlling the operation of the full-bridge soft-switching inverter circuit according to any one of claims 1-13 includes: Collect the output current of the soft-switching inverter module and the output voltage and current of the welding machine power supply; Based on the preset driving strategy, a PWM driving pulse is initially generated. Then, based on the collected voltage and current signals, the current operating condition is determined and the dead time of the driving pulse and the PWM pulse width are dynamically adjusted to generate a driving signal. Based on the optimized isolation driving signal, the silicon carbide power switch is driven to achieve stable high-frequency switching, thus realizing high-frequency switching action.

15. The control method for a full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 14, characterized in that, The driving strategy is as follows: The first silicon carbide power switch Q1 and the fourth silicon carbide power switch Q4 are turned on at the same time. The first silicon carbide power switch Q1 is turned off in advance, and the fourth silicon carbide power switch Q4 is turned off with a lag. The second silicon carbide power switch Q2 and the third silicon carbide power switch Q3 are turned on at the same time. The second silicon carbide power switch Q2 is turned off in advance, and the third silicon carbide power switch Q3 is turned off with a lag. A dead zone is set between the third silicon carbide power switch Q3 and the fourth silicon carbide power switch Q4, during which all power switches are turned off.

16. The control method for a full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 14, characterized in that, Based on the collected voltage and current signals, the current operating condition is determined and the drive pulse is dynamically adjusted accordingly, including: When the output voltage is the no-load voltage and the output current is 0, it is judged to be in no-load state; when the current working condition is no-load state, the pulse width of the leading arm is controlled to the set small pulse width, and the leading arm and lagging arm are fixed to the set dead zone, so that the parallel capacitor of the leading arm completes the commutation through the commutation circuit. When the output voltage is lower than the set value and the output current is greater than the rated current, it is judged to be a short circuit state; when the current working condition is a short circuit state, the pulse width of the leading arm is controlled to be a set small pulse width, so that the lagging arm is turned off with zero current. When the output power reaches the preset maximum value, it is judged to be the maximum power point state. When the current operating condition is the maximum power point state, the pulse width of the leading arm is controlled to be the maximum, and a fixed dead zone is set between the leading arm and the lagging arm so that the lagging arm is turned off with a small current when the maximum power point is turned off.

17. The control method for a full-bridge soft-switching inverter circuit based on silicon carbide power devices as described in claim 14, characterized in that, During the operation of a full-bridge soft-switching inverter circuit that controls the introduction of a common-mode inductor, the common-mode inductor alternates between normal operating mode and high-frequency oscillation operating mode, including: At the instant when the power switches Q1 and Q2 are switched on and off, high-frequency oscillations are generated in the circuit formed by the junction capacitance of the power switches Q1 and Q2, their corresponding parallel capacitance, and the loop inductance. The common-mode inductor enters the high-frequency oscillation working mode, and the common-mode inductor exhibits high impedance, which increases with the increase of the common-mode current in the circuit. At other times, the common-mode inductor enters normal operating mode, exhibiting low impedance and having no additional effect on the operation of the full-bridge soft-switching inverter circuit.

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