Modulation method for series resonance dual-active bridge DC-AC micro inverter

Abstract

The invention relates to a modulation method for a series resonance dual-active bridge DC-AC micro-inverter, and the method comprises the following steps: obtaining the operation data of the DC-AC micro-inverter, updating the angle of output voltage and current, and obtaining the set values of the output voltage and the output current in a stable state; determining an operation quadrant and calculating a switching frequency; the switching frequency is compared with a variable frequency modulation range, if the switching frequency is within the modulation range, a variable frequency modulation method is adopted for modulation, if the switching frequency is larger than the maximum value of the modulation range, a pulse width modulation method is adopted for modulation, and if the switching frequency is smaller than the minimum value of the modulation range, a double-phase-shift control modulation method is adopted for modulation; controlling the voltage at the two ends of the resonant cavity according to the modulation process so as to realize resonant current adjustment, and modulating the output current to a set value based on the resonant current so as to complete the modulation process of the micro-inverter. Compared with the prior art, the method has the advantages that four-quadrant operation can be achieved, the output power range is expanded, the control method is simple, and calculation is convenient.

Description

Technical Field

[0001] This invention relates to the field of electrical engineering, and in particular to a modulation method for a series resonant dual active bridge DC-AC microinverter. Background Technology

[0002] Against the backdrop of the global green energy revolution, solar energy is gradually increasing its share among new energy sources. The development and utilization of solar energy has both significant economic benefits and important strategic implications. There are two main directions for the utilization of solar energy: firstly, large-scale solar power plants can promote carbon neutrality and improve energy security; secondly, small-scale photovoltaic systems can adapt to flexible residential scenarios, bringing smarter and safer green energy to households.

[0003] Grid-connected photovoltaic (PV) systems are currently the primary way to utilize solar energy. These systems can operate in island mode, supplying power to local loads independently, or they can be connected to the main power grid. There are currently three main types of grid-connected PV systems, such as... Figure 2 The diagram shows (a) centralized inverters, (b) string inverters, and (c) micro-inverter inverters. Centralized and string inverters are mainly used in large and medium-sized photovoltaic power plants and are not suitable for flexible residential applications. The main advantages of micro-inverters are: (1) they can still ensure the system operates at its maximum power point even under complex lighting conditions; (2) they are modular, easy to install, and easy to use; (3) they use high-frequency transformers, resulting in high power density and conversion efficiency, achieving electrical isolation while also solving the common-mode leakage current problem. Therefore, micro-inverters are more suitable for flexible residential applications.

[0004] Currently, microinverters face two main technical challenges. First, how to further improve conversion efficiency. Because the energy conversion rate of current photovoltaic cells is not high, the efficiency of microinverters, as grid-connected photovoltaic devices, is crucial, and improved efficiency can also bring significant economic value. Second, how to increase power density. The characteristic and advantage of microinverters is their "micro" size; by minimizing the size and thickness of the inverter while maintaining a fixed input and output power, power density can be increased.

[0005] Microinverters can be classified into single-stage and multi-stage topologies based on the number of power conversion stages. Multi-stage topologies often use a boost DC-DC microinverter in the front end and a DC-AC microinverter in the rear end, which means more losses and reduced conversion efficiency. Single-stage structures, due to their simpler structure, can improve conversion efficiency and power density through hardware design, making them more suitable for microinverter applications. Dual active bridge microinverters, as a highly efficient single-stage topology, are widely used in DC-DC and DC-AC scenarios due to their electrical isolation, high power density, and bidirectional energy flow.

[0006] The Series Resonant Dual Active Bridge (SRDAB) microinverter is an improved topology compared to the traditional Linear Dual Active Bridge (LDAB) microinverter. When the microinverter's operating frequency is much higher than the resonant frequency, the SRDAB can be considered as an LDAB. However, when the operating frequency approaches the resonant frequency, the output current waveform is no longer linear but changes in a sinusoidal shape. With the same average output current, compared to the LDAB, the SRDAB has a smaller effective current value and a smaller peak current, which reduces conduction and switching losses, thereby further improving efficiency.

[0007] The mathematical model of SRDAB is quite complex. Since the inductor current and capacitor voltage have initial values ​​at each stage, the calculations are tedious. Directly applying DC-DC modulation to DC-AC applications would result in slower response, increased costs, and ultimately reduced overall efficiency.

[0008] Existing technologies employ a phase-shift control hybrid frequency modulation method, controlling three phase shift angles and frequencies separately to achieve a large soft-switching range, zero return current power, and a minimum effective resonant cavity current. While this method can achieve high conversion efficiency in DC-DC applications, the large number of control variables and complex expressions prevent its application in DC-AC. Existing technologies also use DPS combined with fixed-frequency modulation. This results in a smaller soft-switching range and a larger effective current value.

[0009] In summary, the shortcomings of existing technologies include: 1) Low conversion efficiency. They cannot simultaneously reduce conduction and switching losses, or achieve optimal efficiency across the entire load range. 2) Inability to achieve four-quadrant operation. Most solutions can achieve bidirectional energy flow, but cannot switch between boost and buck modes. 3) Overly complex control strategies. Because series resonance involves many transient processes, the mathematical models of other solutions are quite complex, often relying on table lookups in practical applications, which leads to a decrease in output accuracy. Summary of the Invention

[0010] The purpose of this invention is to provide a modulation method for a series resonant dual active bridge DC-AC microinverter that improves conversion efficiency.

[0011] The objective of this invention can be achieved through the following technical solutions:

[0012] A modulation method for a series resonant dual active bridge DC-AC microinverter includes the following steps:

[0013] Acquire the basic operating data of the DC-AC micro-inverter, and update the angle of the output voltage and current based on the basic operating data to obtain the set values ​​of the output voltage and output current under steady state.

[0014] Based on the basic operating data and the set values ​​of output voltage and output current under steady-state conditions, the operating quadrant is determined and the switching frequency is calculated.

[0015] The switching frequency is compared with the variable frequency modulation range. If the switching frequency is within the variable frequency modulation range, the variable frequency modulation method is used for modulation. If the switching frequency is greater than the maximum value of the variable frequency modulation range, the pulse width modulation method is used for modulation. If the switching frequency is less than the minimum value of the variable frequency modulation range, the dual phase shift control modulation method is used for modulation.

[0016] The voltage across the resonant cavity in the operating quadrant is controlled according to the modulation process to achieve resonant current regulation, and the output current is modulated to the set value based on the resonant current, thus completing the modulation process of the micro-inverter.

[0017] Furthermore, the operating parameters of the DC-AC microinverter include the DC-side voltage, AC-side voltage, transformer cross ratio, and the values ​​of resonant inductance and capacitance during operation.

[0018] Furthermore, the calculation expression for the switching frequency is as follows:

[0019]

[0020] In the formula, f sw U is the switching frequency. A U B These are the voltages for segment A and segment B, respectively. ref For the required output current, L r It is a resonant inductor.

[0021] Furthermore, the specific steps for determining the operating quadrant include:

[0022] If the DC side voltage is greater than the AC side voltage and the set value of the output current is greater than 0, then the operating quadrant is the buck-positive quadrant.

[0023] If the DC side voltage is greater than the AC side voltage and the set value of the output current is less than 0, then the operating quadrant is the buck-reverse quadrant.

[0024] If the DC side voltage is less than the AC side voltage and the set value of the output current is greater than 0, then the operating quadrant is the boost-positive quadrant.

[0025] If the DC side voltage is less than the AC side voltage and the set value of the output current is less than 0, then the operating quadrant is the boost-reverse quadrant.

[0026] Furthermore, the process of achieving resonant current regulation in the step-down-positive quadrant using a variable frequency modulation method is as follows:

[0027] Phase 1: During the resonant cavity charging phase t∈(0,t1) in the positive half-cycle, according to the buck-positive quadrant operation mode, the voltage across the resonant cavity is obtained as U1-U2 through the DC side voltage U1 and the AC side voltage U2. The voltage across the resonant cavity U1-U2 acts on the resonant cavity, and the resonant current starts to increase from 0. At time t1, the resonant current increases to the positive peak value.

[0028] Phase 2: During the resonant cavity discharge phase t∈(t1,T / 2) of the positive half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U2, and the resonant current starts to decay from the positive peak value. At time T / 2, the resonant current decays to 0.

[0029] Phase 3: During the resonant cavity charging phase t∈(T / 2,T / 2+t1) in the negative half-cycle, the DC side voltage is -U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1+U2. The resonant current starts to decrease from 0 and decreases to the negative peak value at time T / 2+t1.

[0030] Phase 4: During the resonant cavity discharge phase t∈(T / 2+t1,T) in the negative half-cycle, the DC side voltage U1 is 0 and the AC side voltage is U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U2, and the resonant current starts to increase from the negative peak value. At time T, the resonant current increases to 0.

[0031] Then proceed to the next cycle and repeat the modulation process of the above six stages.

[0032] Furthermore, when the variable frequency modulation method achieves resonant current regulation in the buck-positive quadrant, the expression for the resonant current is:

[0033]

[0034] In the formula, I L For the resonant current, C r For resonant capacitor, L r For a resonant inductor, U1 and U2 are the input DC voltage and output DC voltage, respectively. A_Buck_Forward U B_Buck_Forward U represents the voltage across the resonant cavity, specifically the voltage in segment A during the forward energy transmission of the stepped-down energy and the voltage in segment B during the forward energy transmission of the stepped-down energy. cmaxFor the peak value of the resonant capacitance, ω r t1 is the resonant angular frequency, t2 is the resonant cavity charging time, t2 is the resonant cavity discharging time, and T is the period.

[0035] Furthermore, the process of achieving resonant current regulation in the buck-positive quadrant using pulse width modulation is as follows:

[0036] Phase 1: During the resonant cavity charging phase t∈(0,t1) in the positive half-cycle, according to the buck-positive quadrant operation mode, the voltage across the resonant cavity is obtained as U1-U2 through the DC side voltage U1 and the AC side voltage U2. The voltage across the resonant cavity U1-U2 acts on the resonant cavity, and the resonant current starts to increase from 0. At time t1, the resonant current increases to the positive peak value.

[0037] Phase 2: During the resonant cavity discharge phase t∈(t1,t2) of the positive half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U2, and the resonant current starts to decay from the positive peak value. At time t2, the resonant current decays to 0.

[0038] Phase 3: During the positive half-cycle phase t∈(t2,T / 2), the DC side voltage U1 is 0 and the AC side voltage is 0. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is 0 and the resonant current remains 0 until the negative half-cycle begins.

[0039] Phase 4: During the resonant cavity charging phase t∈(T / 2,T / 2+t1) in the negative half-cycle, the DC side voltage is -U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1+U2. The resonant current starts to decrease from 0 and decreases to the negative peak value at time T / 2+t1.

[0040] Phase 5: During the resonant cavity discharge phase t∈(T / 2+t1,T / 2+t2) in the negative half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U2, and the resonant current starts to increase from the negative peak value. At time T / 2+t2, the resonant current increases to 0.

[0041] Phase 6: During the resonant cavity discharge phase t∈(T / 2+t2,T) of the negative half-cycle, the DC side voltage U1 is 0 and the AC side voltage is 0. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is 0 and the resonant current remains 0 until the end of this cycle.

[0042] Then proceed to the next cycle and repeat the modulation process of the above six stages.

[0043] Furthermore, when the pulse width modulation method achieves resonant current regulation in the buck-positive quadrant, the expression for the relationship between the resonant current, charging time, and discharging time is as follows:

[0044]

[0045]

[0046]

[0047] U A_Buck_Forward =U1-U2

[0048] U B_Buck_Forward =U2

[0049] In the formula, I L For the resonant current, C r For resonant capacitor, L r For a resonant inductor, U1 and U2 are the input DC voltage and output DC voltage, respectively. A_Buck_Forward U B_Buck_Forward U represents the voltage across the resonant cavity, specifically the voltage in segment A during the forward energy transmission of the stepped-down energy and the voltage in segment B during the forward energy transmission of the stepped-down energy. cmax For the peak value of the resonant capacitance, ω r t1 is the resonant angular frequency, t2 is the resonant cavity charging time, t2 is the resonant cavity discharging time, T is the period, f is the switching frequency of the switching transistor, and I is the resonant angular frequency. out This is the output current.

[0050] Furthermore, the process of achieving resonant current regulation in the buck-positive quadrant using the dual-phase-shift control modulation method is as follows:

[0051] Phase 1: During the resonant cavity charging phase t∈(0,t1) of the positive half-cycle, according to the buck-positive quadrant operation mode, the voltage across the resonant cavity is obtained as U1-U2 through the DC side voltage U1 and the AC side voltage U2. The voltage across the resonant cavity U1-U2 acts on the resonant cavity, and the resonant current begins to increase. At time t1, the resonant current increases to the positive peak value.

[0052] Phase 2: During the resonant cavity discharge phase t∈(t1,t2) of the positive half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U2, and the resonant current begins to decay from the positive peak value.

[0053] Phase 3: During the positive half-cycle phase t∈(t2,T / 2), the DC side voltage is -U1 and the AC side voltage is U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1-U2, and the resonant current continues to decay to the negative value of the resonant current at the beginning of the phase.

[0054] Phase 4: During the resonant cavity charging phase t∈(T / 2,T / 2+t1) in the negative half-cycle, the DC side voltage is -U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1+U2, and the resonant current begins to decrease. At time T / 2+t1, the resonant current decreases to the negative peak value.

[0055] Phase 5: During the resonant cavity discharge phase t∈(T / 2+t1,T / 2+t2) in the negative half-cycle, the DC side voltage is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U2, and the resonant current starts to increase from the negative peak value.

[0056] Phase Six: During the resonant cavity discharge phase t∈(T / 2+t2,T) of the negative half-cycle, the DC side voltage is U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U1+U2. The resonant current continues to increase to the initial value of the resonant current, and this cycle ends.

[0057] Then proceed to the next cycle and repeat the modulation process of the above six stages.

[0058] Furthermore, during the process of adjusting the resonant current in the voltage-positive quadrant phase-shift control modulation method, the expression for the voltage across the resonant cavity is:

[0059]

[0060]

[0061] In the formula, U LC U is the voltage across the resonant cavity. A_Buck_Forward U B_Buck_Forward and U C_Buck_Forward U1 and U2 represent the voltages in the forward energy transmission phases A, B, and C of the resonant cavity, respectively. U1 and U2 are the input DC voltage and output DC voltage, respectively. t1 is the charging time of the resonant cavity, t2 is the discharging time of the resonant cavity, and T is the period.

[0062] Compared with the prior art, the present invention has the following beneficial effects:

[0063] (1) The hybrid modulation strategy of PWM, DPS and VFM of the present invention fully balances the characteristics of conduction loss and switching loss, and achieves high-efficiency DC-AC power conversion in the full load domain. The conversion efficiency is significantly better than other strategies.

[0064] (2) In view of the working characteristics of each quadrant, the present invention proposes a hybrid modulation strategy of PWM, DPS and VFM respectively. According to the circuit load requirements, the corresponding modulation strategy is adopted in each quadrant to realize four-quadrant operation.

[0065] (3) The hybrid modulation strategy of the present invention flexibly adopts the corresponding modulation strategy according to the circuit load requirements, which can be applied to various working occasions, realize seamless switching between working modes, and meet the working requirements of off-grid and on-grid. Attached Figure Description

[0066] Figure 1 This is a schematic diagram of the method flow of the present invention;

[0067] Figure 2 The present invention provides a conventional photovoltaic grid-connected system, wherein (a) is a centralized grid-connected system, (b) is a string grid-connected system, and (c) is a micro-inverter grid-connected system;

[0068] Figure 3 This is a topology diagram of a series resonant dual active bridge microinverter according to an embodiment of the present invention;

[0069] Figure 4 The following are schematic diagrams of the waveforms of each mode in each quadrant according to an embodiment of the present invention: (a) is a schematic diagram of the waveform of PWM used in the buck-positive quadrant, (b) is a schematic diagram of the waveform of PWM used in the buck-reverse quadrant, (c) is a schematic diagram of the waveform of VFM used in the buck-positive quadrant, (d) is a schematic diagram of the waveform of VFM used in the buck-reverse quadrant, (e) is a schematic diagram of the waveform of DPS used in the buck-positive quadrant, (f) is a schematic diagram of the waveform of DPS used in the buck-reverse quadrant, (g) is a schematic diagram of the waveform of PWM used in the boost-positive quadrant, (h) is a schematic diagram of the waveform of PWM used in the boost-reverse quadrant, (i) is a schematic diagram of the waveform of VFM used in the boost-positive quadrant, (j) is a schematic diagram of the waveform of VFM used in the boost-reverse quadrant, (k) is a schematic diagram of the waveform of DPS used in the boost-positive quadrant, and (l) is a schematic diagram of the waveform of DPS used in the boost-reverse quadrant.

[0070] Figure 5 The waveforms of current and voltage in the buck-forward converter of this invention are shown in the figure.

[0071] Figure 6 The waveforms of current and voltage in the buck-forward converter of this invention are shown in the figure.

[0072] Figure 7 The waveforms of current and voltage in the step-forward phase of this invention are shown in the figure.

[0073] Figure 8This is a schematic diagram illustrating the relationship between control variables in an embodiment of the present invention;

[0074] Figure 9 This is a schematic diagram showing the proportion of each mode under different output powers in an embodiment of the present invention;

[0075] Figure 10 This is a comparison between the average and effective values ​​of the output current in this embodiment of the invention.

[0076] Figure 11 A comparison of conduction losses in embodiments of the present invention;

[0077] Figure 12 The switch turn-off current comparison in this embodiment of the invention includes (a) a comparison of switches S1 and S2 within the ZVS range, (b) a comparison of switches S3 and S4 within the ZCS range, and (c) a comparison of switches S5 and S6 / S7 and S8 within the ZCS and ZVS ranges, respectively.

[0078] Figure 13 This is a comparison of switching losses in an embodiment of the present invention. Detailed Implementation

[0079] The present invention will now be described in detail with reference to the accompanying drawings and specific embodiments. These embodiments are based on the technical solution of the present invention and provide detailed implementation methods and specific operating procedures. However, the scope of protection of the present invention is not limited to the following embodiments.

[0080] This embodiment provides a modulation method for a series resonant dual active bridge DC-AC microinverter, such as... Figure 1 As shown, the method includes the following steps:

[0081] S1. Obtain the basic operating data of the DC-AC micro-inverter, and update the angle of the output voltage and current based on the basic operating data to obtain the set values ​​of the output voltage and output current under stable conditions.

[0082] This embodiment uses a series resonant dual active bridge converter as the topology of the DC-AC microinverter, such as... Figure 3 As shown, this topology consists of a DC-side full-bridge, an AC-side full-bridge, a folded bridge, a series resonant cavity, and a high-frequency transformer. In the forward flow of energy, the folded bridge expands the AC voltage (twice the power frequency) into a 50Hz AC mains voltage. In the reverse flow, the folded bridge folds the 50Hz AC voltage into a 100Hz multiple of the AC frequency. The DC and AC voltages are modulated into high-frequency voltages of the same frequency by the switches S1-S4 on the primary side and S5 and S8 on the secondary side of the transformer, respectively. The subtraction of the high-frequency voltages on the primary and secondary sides constitutes the input voltage of the resonant cavity Lr,Cr. r For resonant inductance, Cr This is a resonant capacitor. By controlling the phase, duty cycle, and frequency of the voltage across the resonant cavity, bidirectional energy flow and voltage boosting / debossing can be achieved.

[0083] First, obtain the basic data for circuit operation: DC side voltage, transformer turns ratio, values ​​of resonant inductance and capacitance, amplitude of output current, and power factor.

[0084] Based on this, update the angles of the output AC side voltage and current to obtain U2 and the required current I in the current steady state. ref .

[0085] For ease of description, the symbols involved in this embodiment are first explained as shown in Table 1.

[0086] Table 1 Explanation of Main Symbols

[0087] symbol name <![CDATA[S1-S8]]> MOSFET name <![CDATA[D1-D8]]> Name of a diode in parallel with a MOSFET <![CDATA[C IN ]]> Input capacitor <![CDATA[C out ]]> Output capacitor <![CDATA[L r ]]> Resonant inductor <![CDATA[C r ]]> Resonant capacitor <![CDATA[U1]]> Input DC voltage <![CDATA[U2]]> Converted to primary side output voltage <![CDATA[U LC ]]> Voltage across the resonant cavity <![CDATA[U cmax ]]> Peak resonant capacitance T Switching cycle of switching transistor f Switching frequency of switching transistor <![CDATA[t1]]> Resonant cavity charging time <![CDATA[t2]]> Resonant cavity discharge time <![CDATA[ω r ]]> Resonant angular frequency <![CDATA[I L ]]> Resonant inductor current <![CDATA[I out ]]> Output current <![CDATA[U A_Buck_Forward ]]> The resonant cavity transmits voltage segment A in the forward direction of the stepped-down energy. <![CDATA[U B_Buck_Forward ]]> The resonant cavity transmits voltage in the B segment in the forward direction of the stepped-down energy. <![CDATA[U C_Buck_Forward ]]> The resonant cavity transmits voltage in the C segment in the forward direction of the stepped-down energy. <![CDATA[U A_Boost_Forward ]]> The resonant cavity transmits voltage segment A in the forward direction of the boosted energy. <![CDATA[U B_Boost_Forward ]]> The resonant cavity transmits voltage segment B in the forward direction of the boosted energy. <![CDATA[U C_Boost_Forward ]]> The resonant cavity transmits voltage in the C segment in the forward direction of the boosted energy. <![CDATA[U A_Buck_Reverse ]]> The resonant cavity reverses the energy transfer of segment A voltage. <![CDATA[U B_Buck_Reverse ]]> The resonant cavity transmits voltage in the reverse direction of the step-down energy in segment B. <![CDATA[U C_Buck_Reverse ]]> The resonant cavity transmits voltage in the reverse direction of the step-down energy in segment C. <![CDATA[U A_Boost_Reverse ]]> The resonant cavity transmits the voltage of segment A in the reverse direction of the boosted energy. <![CDATA[U B_Boost_Reverse ]]> The resonant cavity transmits voltage in the B segment in the reverse direction of the boosted energy. <![CDATA[U C_Boost_Reverse ]]> The resonant cavity transmits voltage in the B segment in the reverse direction of the boosted energy.

[0088] S2. Based on the basic operating data and the set values ​​of output voltage and output current under stable conditions, determine the operating quadrant and calculate the switching frequency.

[0089] This step uses the data obtained in the previous step to determine the direction of energy flow and the boost / deboost mode, thereby determining which quadrant it is operating in.

[0090] The four-quadrant operating modes are obtained from Table 2.

[0091] Table 2 Four-Quadrant Operation Mode

[0092]

[0093] After determining the operating quadrant, calculate the switching frequency using the following formula:

[0094]

[0095] S3. Compare the switching frequency with the variable frequency modulation range. If the switching frequency is within the variable frequency modulation range, then use the variable frequency modulation method for modulation. If the switching frequency is greater than the maximum value of the variable frequency modulation range, then use the pulse width modulation method for modulation. If the switching frequency is less than the minimum value of the variable frequency modulation range, then use the dual phase shift control modulation method for modulation.

[0096] Based on the above circuit topology, four-quadrant operation modes are proposed, and each mode is listed in... Figure 4The system comprises 12 modes, each including Pulse Width Modulation (PWM), Variable Frequency Modulation (VFM), and Dual Phase Shift (DPS). Based on these 12 modes, detailed control strategies will be designed and combined to form a DC-AC control strategy.

[0097] The voltage across the resonant cavity varies depending on whether it is a step-down or step-up operation and whether the energy flows in the forward or reverse direction. The expressions for the voltage across the resonant cavity are shown in Table 3.

[0098] Table 3 Expressions for voltages across the resonant cavity in different modes

[0099]

[0100]

[0101] This embodiment takes positive energy flow as an example to analyze the strategy design of DC-AC.

[0102] 1. Pulse Width Modulation (PWM)

[0103] For PWM modulation, the frequency is fixed, and the resonant current is in a discontinuous state under steady-state conditions, with six different states within one cycle.

[0104] Phase 1: t∈(0,t1). A new cycle begins at t=0. The current at the end of the previous cycle was 0, therefore the switch is a zero-current switch (ZCS). For the primary side of the transformer, switches S1 and S4 are open, and S2 and S3 are closed, with the input voltage on the left side of the resonant cavity being U1. For the secondary side of the transformer, switches S5 and S8 are open, and S6 and S7 are closed, with the output voltage on the right side of the resonant cavity being U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is U1-U2, and the resonant current starts to increase from 0.

[0105] When t∈(0,t1), U1-U2=U A_Buck_Forward When applied to the resonant cavity, the current starts from 0 and increases. In the forward power transfer mode, the current increases in the forward direction; in the reverse power transfer mode, the current increases in the reverse direction.

[0106] Phase Two: t∈(t1,t2). A new phase begins at t=t1. Switch S1 is turned off and switch S2 is turned on. Due to the capacitance within the switching transistors, the switches are zero-voltage switches (ZVS). At this time, on the primary side of the transformer, S2 and S4 are on, while S1 and S3 are off, resulting in an output voltage of 0. On the secondary side of the transformer, S5 and S8 are on, while S6 and S7 are off, resulting in an output voltage of U2. According to the calculation expressions in Table 3, the voltage across the resonant cavity is -U2, and the resonant cavity current begins to decrease from its maximum value.

[0107] When t∈(t1,t2), the current will decay to zero. At the moment t=t1, the current reaches its peak value and then decays to zero.

[0108] Phase 3: t∈(t2,0.5T). Phase 3 begins when the current drops to 0, i.e., t=t2. Since the current is 0, it is a zero-current switch (ZCS). On the primary side, S2 and S4 are open, and S1 and S3 are closed, resulting in a zero output voltage. On the secondary side, S5 and S7 are closed, and S6 and S8 are open, also resulting in a zero output voltage. According to the calculation expressions in Table 3, the voltage across the resonant cavity is 0, thus clamping the resonant current to 0 until the negative half-cycle begins.

[0109] During the time interval t∈(t2,T / 2), both the primary and secondary switches will be turned off, and the resonant current will remain at 0 until the start of the second half-cycle.

[0110] The current change in the second half-cycle is the same as that in the first half-cycle, but the state is opposite.

[0111] Phase 4: t∈(0.5T, 0.5T+t1). The negative half-cycle begins at t=T / 2. The current is 0 at the end of the first half-cycle, therefore the switch is a zero-current switch (ZCS). For the primary side of the transformer, switches S2 and S3 are open, and S1 and S4 are closed, with the input voltage on the left side of the resonant cavity being -U1. For the secondary side of the transformer, switches S6 and S7 are open, and S5 and S8 are closed, with the output voltage on the right side of the resonant cavity being -U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is -U1+U2, and the resonant current decreases from 0.

[0112] Phase 5: t∈(0.5T+t1,0.5T+t2). A new phase begins at t=0.5T+t1. At this time, switch S2 is turned off and switch S1 is turned on. Due to the capacitance within the switching transistors, the switches are zero-voltage switches (ZVS). On the primary side of the transformer, S1 and S3 are on, while S2 and S4 are off, resulting in an output voltage of 0. On the secondary side, S6 and S7 are on, while S5 and S8 are off, resulting in an output voltage of -U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is U2, and the resonant cavity begins to increase from its maximum negative current value.

[0113] Phase Six: t∈(0.5T+t2,T). Phase Six begins when the current drops to 0, i.e., t=0.5T+t2. Since the current is 0, it is a zero-current switch (ZCS). On the primary side, S1 and S3 are open, and S2 and S4 are closed, resulting in a zero output voltage. On the secondary side, S5 and S7 are open, and S6 and S8 are closed, also resulting in a zero output voltage. According to the calculation expressions in Table 3, since the voltage across the resonant cavity is 0, the resonant current is clamped to 0 until the end of the cycle.

[0114] Expressions (2)-(4) give the resonant current I. L The detailed expression and the relationship between charging time t1 and discharging time t2 are shown. It can be seen that when the charging time t1 is small, the maximum value of the capacitor is also small, and the resonant current is relatively linear. In this case, the circuit can be approximated as a linear dual active bridge micro-inverter. However, when t1 increases, the capacitor voltage cannot be ignored, so the resonant circuit must be expressed using trigonometric functions.

[0115]

[0116]

[0117]

[0118] Based on PWM modulation, the waveforms of current and voltage when the circuit operates in the energy-forward buck mode were obtained, as follows: Figure 5 As shown. From Figure 5 It can be observed that in pulse width modulation (PWM), the lagging arm signals on the primary and secondary sides are consistent; the only controllable variable is the phase difference between the DC-side leading and lagging arms. Once the phase difference between the DC-side arms is known, the phase difference between the AC-side arms can also be calculated. Therefore, pulse width modulation can be considered a special case of single-phase shift control. In this mode, since the inductor's initial state is 0, zero return current power and ZCS (zero-current switching) and ZVS (zero-voltage switching) within the operating range can be achieved. However, the disadvantage is weak power output capability, making it suitable only for light load conditions.

[0119] 2. Phase-Shift Controlled Modulation (DPS)

[0120] When the circuit is under heavy load or requires a large output current, it will operate in dual-phase-shift control mode. Taking the forward transfer of buck power as an example, the frequency is fixed. Due to the large required output current, the resonant current cannot operate in discontinuous conduction mode, and the current will enter continuous conduction mode.

[0121] There are six different states within one cycle.

[0122] Phase 1: t∈(0,t1). A new cycle begins at t=0. At the end of the previous cycle, the current is small and the switching transistor has internal capacitance, therefore the switch is a zero-voltage switch (ZVS). For the primary side of the transformer, switches S1 and S4 are open, and S2 and S3 are closed, with the input voltage on the left side of the resonant cavity being U1; for the secondary side of the transformer, switches S5 and S8 are open, and S6 and S7 are closed, with the output voltage on the right side of the resonant cavity being U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is U1-U2, and the resonant current begins to increase.

[0123] As shown in equation (5), when t∈(0,t1), the voltage across the resonant cavity is U. A_Buck_Forward The direction depends on the direction of power flow, and the current will continue to increase at this time.

[0124]

[0125] By balancing the volt-seconds of the inductor and the ampere-seconds of the capacitor, we can obtain expression (5).

[0126] The expression for the average output current can be obtained through (6).

[0127] In continuous conduction mode, it is impossible to control the initial state of the inductor to be zero. In order to further reduce the peak voltage, the two phase shift angles can be optimized by equation (7) to obtain the minimum switching loss. At the same time, the expressions of the two control variables t1 and t2 can be obtained by equation (7).

[0128]

[0129]

[0130] Phase Two: t∈(t1,t2). A new phase begins at t=t1. At this time, switch S1 is turned off and switch S2 is turned on. Due to the capacitance within the switching transistors, the switches are zero-voltage switches (ZVS). On the primary side of the transformer, S2 and S4 are on, while S1 and S3 are off, resulting in an output voltage of 0. On the secondary side, S5 and S8 are on, while S6 and S7 are off, resulting in an output voltage of U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is -U2, and the resonant cavity begins to decrease from its maximum current value.

[0131] At t∈(t1,t2), the voltage across the resonant cavity becomes -U2=U B_Buck_Forward The current began to decrease.

[0132] Phase 3: t∈(t2,0.5T). When t=t2, Phase 3 begins. Because the switching transistors have internal capacitance, the switching is a zero-voltage switch (ZVS). On the primary side, S2 and S3 are on, and S1 and S4 are off, with an output voltage of -U1; on the secondary side, S6 and S7 are off, and S5 and S8 are on, with an output voltage of U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is -U1-U2, therefore the resonant current will decrease rapidly until the negative half-cycle begins.

[0133] At t∈(t2,T / 2), the voltage across the resonant cavity becomes -U1-U2=U c_Buck_Forward It quickly drops to the initial value of the inductor current.

[0134] Figure 6 These are the voltage and current waveforms of DPS in the energy positive flow buck mode; the waveforms in the other quadrants are shown in... Figure 4 As shown in Table 3, the expressions for the voltages of segments A, B, and C in different operating quadrants are given. For the pulse width modulation modes in other quadrants, simply replace the voltages of segments A, B, and C with the corresponding voltages.

[0135] Phase 4: t∈(0.5T, 0.5T+t1). The negative half-cycle begins at t=T / 2. At the end of the first half-cycle, the current is small and the switching transistor has internal capacitance, therefore the switch is a zero-voltage switch (ZVS). For the primary side of the transformer, switches S2 and S3 are open, and S1 and S4 are closed, with the input voltage on the left side of the resonant cavity being -U1. For the secondary side of the transformer, switches S6 and S7 are open, and S5 and S8 are closed. According to the calculation expression in Table 3, the output voltage on the right side of the resonant cavity is -U2. The voltage across the resonant cavity is -U1+U2, and the resonant current begins to increase in the reverse direction.

[0136] Phase 5: t∈(0.5T+t1,0.5T+t2). A new phase begins at t=0.5T+t1. At this time, switch S2 is turned off and switch S1 is turned on. Due to the capacitance within the switching transistors, the switches are zero-voltage switches (ZVS). On the primary side of the transformer, S1 and S3 are on, while S2 and S4 are off, resulting in an output voltage of 0. On the secondary side, S6 and S7 are on, while S5 and S8 are off, resulting in an output voltage of -U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is U2, and the resonant cavity begins to increase from the maximum reverse current value.

[0137] Phase Six: t∈(0.5T+t2,T). When t=0.5T+t2, Phase Six begins. Because the current is small and the switching transistors have internal capacitance, the switches are zero-voltage switches (ZVS). On the primary side, S1 and S4 are on, and S2 and S3 are off, with an output voltage of U1; on the secondary side, S5 and S8 are off, and S6 and S7 are on, with an output voltage of -U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is U1+U2, therefore the resonant current will rise rapidly until the start of the next cycle.

[0138] 3. Variable Frequency Modulation (VFM)

[0139] By removing the time periods when the current is zero from the PWM mode, a variable frequency modulation mode can be obtained. In this mode, the frequency is determined by the charging time t1. The resonant current then has four states.

[0140] Phase 1: t∈(0,t1). A new cycle begins at t=0. The current at the end of the previous cycle was 0, therefore the switch is a zero-current switch (ZCS). For the primary side of the transformer, switches S1 and S4 are open, and S2 and S3 are closed, with the input voltage on the left side of the resonant cavity being U1. For the secondary side of the transformer, switches S5 and S8 are open, and S6 and S7 are closed, with the output voltage on the right side of the resonant cavity being U2. According to the calculation expression in Table 3, the voltage across the resonant cavity is U1-U2, and the resonant current starts to increase from 0.

[0141] The resonant cavity is charged at t∈(0,t1).

[0142] Phase Two: t∈(t1,0.5T). A new phase begins at t=t1. Switch S1 is turned off and switch S2 is turned on. Due to the capacitance within the switching transistors, the switches are zero-voltage switches (ZVS). At this time, on the primary side of the transformer, S2 and S4 are on, while S1 and S3 are off, resulting in an output voltage of 0. On the secondary side of the transformer, S5 and S8 are on, while S6 and S7 are off, resulting in an output voltage of U2. According to the expression in Table 3, the voltage across the resonant cavity is -U2, and the resonant cavity current begins to decrease from its maximum value.

[0143] Discharge the resonant cavity at t∈(t1,0.5T).

[0144] Phase 3: t∈(0.5T, 0.5T+t1). When the resonant current is 0, the negative half-cycle begins immediately. The current at the end of the first half-cycle is 0, therefore the switch is a zero-current switch (ZCS). For the primary side of the transformer, switches S2 and S3 are open, and S1 and S4 are closed, with the input voltage on the left side of the resonant cavity being -U1. For the secondary side of the transformer, switches S6 and S7 are open, and S5 and S8 are closed. According to the expression in Table 3, the output voltage on the right side of the resonant cavity is -U2. The voltage across the resonant cavity is -U1+U2, and the resonant current begins to decrease from 0.

[0145] Phase 4: t∈(0.5T+t1,T). A new phase begins when t=0.5T+t1. At this time, switch S2 is turned off and switch S1 is turned on. Due to the capacitance within the switching transistors, the switches are zero-voltage switches (ZVS). On the primary side of the transformer, S1 and S3 are on, while S2 and S4 are off, resulting in an output voltage of 0. On the secondary side, S6 and S7 are on, while S5 and S8 are off, resulting in an output voltage of -U2. According to the expression in Table 3, the voltage across the resonant cavity is U2. The resonant cavity begins to increase from its maximum negative current value until the current reaches 0, at which point it enters the next cycle.

[0146] Compared to pulse width modulation, it eliminates the time period when the current is 0, which can further reduce the effective value of the output current and reduce conduction losses.

[0147] The expressions for the inductor current and frequency are shown in equations (8) and (9).

[0148]

[0149]

[0150] In variable frequency mode, the controlled variables are the charging time t1 and the switching frequency. However, the switching frequency is also determined by t1. Therefore, variable frequency modulation is still a special single-phase shift modulation method.

[0151] Based on the modulation process within one cycle described above, the voltage and current waveforms of the buck-down VFM are obtained, as shown below. Figure 7 As shown.

[0152] S4. Control the voltage across the resonant cavity in the operating quadrant according to the modulation process to achieve resonant current regulation, and based on the resonant current, modulate the output current to the set value to complete the modulation process of the micro-inverter.

[0153] The above modulation method controls the resonant current and then adjusts the output current based on the resonant current to meet the load requirements.

[0154] Based on the aforementioned DC-AC modulation strategy of the micro-inverter, this embodiment underwent simulation experiments for verification. For example... Figure 8 The diagram shows the relationship between modulation strategies and phase. In the diagram, PWM mode only controls the inner phase shift angle, VFM mode controls the frequency and inner phase shift angle, and DPS mode controls both the inner and outer phase shift angles. Figure 9 This diagram illustrates the proportion of each mode under different output power levels. Because the DPS mode has the strongest output capability, its proportion increases with increasing output power, while the proportions of PWM and VFM modes decrease.

[0155] During steady-state operation, the energy loss of a circuit is varied. In steady-state operation, according to ampere-second and volt-second balance, the changes in magnetic energy and potential energy are also zero, thus not affecting operating efficiency. The main losses are: firstly, conduction losses caused by current flowing through components with internal resistance; and secondly, switching losses caused by the short-term overlap of current and voltage waveforms during the switching process of the switching transistor. Figure 10 The effective current values ​​of the closest scheme to this invention, namely the variable frequency dual-phase-shift control scheme and the proposed modulation strategy, were compared under the same output (500W). It can be found that the waveforms of the effective values ​​of the two are not much different. Under light load conditions, DPS_VFM is slightly better, but under heavy load conditions, the effective current value of the proposed modulation scheme is smaller. Figure 11 The conduction loss curves of the two modulation strategies were compared under the same switching transistor internal resistance but different output power. It can be found that the conduction losses of the two strategies are almost the same.

[0156] Regarding switching losses Figure 12 A comparison of the turn-off currents for the two modulation methods under the condition of forward energy flow in a step-down configuration is presented. According to existing technology, in the DPS_VFM modulation method, all switches operate within the ZVS range. However, for the proposed modulation method… Figure 12 As shown in (a), switches S1 and S2 always operate within the ZVS range, while (b) and (c) show that switches S3 and S4, S5 and S6, and S7 and S8 operate within the ZCS range in PWM and VFM modes, and within the ZVS range in DPS mode. Overall, PWM and VFM modes have 6 ZCS switches and 2 ZVS switches, while DPS has 8 ZVS switches. It is well known that the switching losses of ZCS switches are much lower than those of ZVS switches; therefore, the proposed scheme has lower switching losses than the DPS_VFM scheme. Figure 13 The switching losses of the two modulation strategies were compared under different output powers. It can be found that the switching losses of the proposed scheme are much smaller than those of DPS_VFM.

[0157] Although preferred embodiments of the invention have been described, those skilled in the art, upon learning the basic inventive concept, can make other changes and modifications to these embodiments. Therefore, the appended claims are intended to be interpreted as including both the preferred embodiments and all changes and modifications falling within the scope of the invention.

[0158] Obviously, those skilled in the art can make various modifications and variations to this invention without departing from its spirit and scope. Therefore, if these modifications and variations fall within the scope of the claims of this invention and their equivalents, this invention also intends to include these modifications and variations.

Claims

1. A modulation method for a series resonant dual active bridge DC-AC microinverter, characterized in that, Includes the following steps: Acquire the basic operating data of the DC-AC micro-inverter, and update the angle of the output voltage and current based on the basic operating data to obtain the set values ​​of the output voltage and output current under steady state. Based on the basic operating data and the set values ​​of output voltage and output current under steady-state conditions, the operating quadrant is determined and the switching frequency is calculated. The switching frequency is compared with the variable frequency modulation range. If the switching frequency is within the variable frequency modulation range, the variable frequency modulation method is used for modulation. If the switching frequency is greater than the maximum value of the variable frequency modulation range, the pulse width modulation method is used for modulation. If the switching frequency is less than the minimum value of the variable frequency modulation range, the dual phase shift control modulation method is used for modulation. The voltage across the resonant cavity in the operating quadrant is controlled according to the modulation process to achieve resonant current regulation, and the output current is modulated to the set value based on the resonant current, thus completing the modulation process of the micro-inverter.

2. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 1, characterized in that, The operating parameters of the DC-AC microinverter include the DC-side voltage, AC-side voltage, transformer cross ratio, and the values ​​of resonant inductance and capacitance during operation.

3. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 1, characterized in that, The expression for calculating the switching frequency is: In the formula, f sw U is the switching frequency. A U B These are the voltages for segment A and segment B, respectively. ref For the required output current, L r It is a resonant inductor.

4. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 1, characterized in that, The specific steps for determining the operating quadrant include: If the DC side voltage is greater than the AC side voltage and the set value of the output current is greater than 0, then the operating quadrant is the buck-positive quadrant. If the DC side voltage is greater than the AC side voltage and the set value of the output current is less than 0, then the operating quadrant is the buck-reverse quadrant. If the DC side voltage is less than the AC side voltage and the set value of the output current is greater than 0, then the operating quadrant is the boost-positive quadrant. If the DC side voltage is less than the AC side voltage and the set value of the output current is less than 0, then the operating quadrant is the boost-reverse quadrant.

5. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 4, characterized in that, The process of achieving resonant current regulation in the step-down positive quadrant using variable frequency modulation is as follows: Phase 1: During the resonant cavity charging phase t∈(0,t1) in the positive half-cycle, according to the buck-positive quadrant operation mode, the voltage across the resonant cavity is obtained as U1-U2 through the DC side voltage U1 and the AC side voltage U2. The voltage across the resonant cavity U1-U2 acts on the resonant cavity, and the resonant current starts to increase from 0. At time t1, the resonant current increases to the positive peak value. Phase 2: During the resonant cavity discharge phase t∈(t1,T / 2) of the positive half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U2, and the resonant current starts to decay from the positive peak value. At time T / 2, the resonant current decays to 0. Phase 3: During the resonant cavity charging phase t∈(T / 2,T / 2+t1) in the negative half-cycle, the DC side voltage is -U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1+U2. The resonant current starts to decrease from 0 and decreases to the negative peak value at time T / 2+t1. Phase 4: During the resonant cavity discharge phase t∈(T / 2+t1,T) in the negative half-cycle, the DC side voltage U1 is 0 and the AC side voltage is U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U2, and the resonant current starts to increase from the negative peak value. At time T, the resonant current increases to 0. Then proceed to the next cycle and repeat the modulation process of the above six stages.

6. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 5, characterized in that, When the variable frequency modulation method achieves resonant current regulation in the step-down positive quadrant, the expression for the resonant current is: In the formula, I L For the resonant current, C r For resonant capacitor, L r For a resonant inductor, U1 and U2 are the input DC voltage and output DC voltage, respectively. A_Buck_Forward U B_Buck_Forward U represents the voltage across the resonant cavity, specifically the voltage in segment A during the forward energy transmission of the stepped-down energy and the voltage in segment B during the forward energy transmission of the stepped-down energy. cmax For the peak value of the resonant capacitance, ω r t1 is the resonant angular frequency, t2 is the resonant cavity charging time, t2 is the resonant cavity discharging time, and T is the period.

7. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 4, characterized in that, The process of achieving resonant current regulation in the buck-positive quadrant using pulse width modulation is as follows: Phase 1: During the resonant cavity charging phase t∈(0,t1) in the positive half-cycle, according to the buck-positive quadrant operation mode, the voltage across the resonant cavity is obtained as U1-U2 through the DC side voltage U1 and the AC side voltage U2. The voltage across the resonant cavity U1-U2 acts on the resonant cavity, and the resonant current starts to increase from 0. At time t1, the resonant current increases to the positive peak value. Phase 2: During the resonant cavity discharge phase t∈(t1,t2) of the positive half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U2, and the resonant current starts to decay from the positive peak value. At time t2, the resonant current decays to 0. Phase 3: During the positive half-cycle phase t∈(t2,T / 2), the DC side voltage U1 is 0 and the AC side voltage is 0. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is 0 and the resonant current remains 0 until the negative half-cycle begins. Phase 4: During the resonant cavity charging phase t∈(T / 2,T / 2+t1) in the negative half-cycle, the DC side voltage is -U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1+U2. The resonant current starts to decrease from 0 and decreases to the negative peak value at time T / 2+t1. Phase 5: During the resonant cavity discharge phase t∈(T / 2+t1,T / 2+t2) in the negative half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U2, and the resonant current starts to increase from the negative peak value. At time T / 2+t2, the resonant current increases to 0. Phase 6: During the resonant cavity discharge phase t∈(T / 2+t2,T) of the negative half-cycle, the DC side voltage U1 is 0 and the AC side voltage is 0. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is 0 and the resonant current remains 0 until the end of this cycle. Then proceed to the next cycle and repeat the modulation process of the above six stages.

8. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 7, characterized in that, When using pulse width modulation (PWM) to regulate resonant current in the buck-positive quadrant, the relationship between resonant current, charging time, and discharging time is expressed as follows: U A_Buck_Forward =U1-U2 U B_Buck_Forward =U2 In the formula, I L For the resonant current, C r For resonant capacitor, L r For a resonant inductor, U1 and U2 are the input DC voltage and output DC voltage, respectively. A_Buck_Forward U B_Buck_Forward U represents the voltage across the resonant cavity, specifically the voltage in segment A during the forward energy transmission of the stepped-down energy and the voltage in segment B during the forward energy transmission of the stepped-down energy. cmax For the peak value of the resonant capacitance, ω r t1 is the resonant angular frequency, t2 is the resonant cavity charging time, t2 is the resonant cavity discharging time, T is the period, f is the switching frequency of the switching transistor, and I is the resonant angular frequency. out This is the output current.

9. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 4, characterized in that, The process of achieving resonant current regulation in the step-down-positive quadrant using the dual-phase-shift control modulation method is as follows: Phase 1: During the resonant cavity charging phase t∈(0,t1) of the positive half-cycle, according to the buck-positive quadrant operation mode, the voltage across the resonant cavity is obtained as U1-U2 through the DC side voltage U1 and the AC side voltage U2. The voltage across the resonant cavity U1-U2 acts on the resonant cavity, and the resonant current begins to increase. At time t1, the resonant current increases to the positive peak value. Phase 2: During the resonant cavity discharge phase t∈(t1,t2) of the positive half-cycle, the DC side voltage U1 is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U2, and the resonant current begins to decay from the positive peak value. Phase 3: During the positive half-cycle phase t∈(t2,T / 2), the DC side voltage is -U1 and the AC side voltage is U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1-U2, and the resonant current continues to decay to the negative value of the resonant current at the beginning of the phase. Phase 4: During the resonant cavity charging phase t∈(T / 2,T / 2+t1) in the negative half-cycle, the DC side voltage is -U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is -U1+U2, and the resonant current begins to decrease. At time T / 2+t1, the resonant current decreases to the negative peak value. Phase 5: During the resonant cavity discharge phase t∈(T / 2+t1,T / 2+t2) in the negative half-cycle, the DC side voltage is 0 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U2, and the resonant current starts to increase from the negative peak value. Phase Six: During the resonant cavity discharge phase t∈(T / 2+t2,T) of the negative half-cycle, the DC side voltage is U1 and the AC side voltage is -U2. According to the buck-positive quadrant operation mode, the voltage across the resonant cavity is U1+U2. The resonant current continues to increase to the initial value of the resonant current, and this cycle ends. Then proceed to the next cycle and repeat the modulation process of the above six stages.

10. The modulation method for a series resonant dual active bridge DC-AC microinverter according to claim 9, characterized in that, When the dual-phase-shift control modulation method achieves resonant current regulation through voltage reduction and positive quadrant adjustment, the expression for the voltage across the resonant cavity is as follows: In the formula, U LC U is the voltage across the resonant cavity. A_Buck_Forward U B_Buck_Forward and U C_Buck_Forward U1 and U2 represent the voltages in the forward energy transmission phases A, B, and C of the resonant cavity, respectively. U1 and U2 are the input DC voltage and output DC voltage, respectively. t1 is the charging time of the resonant cavity, t2 is the discharging time of the resonant cavity, and T is the period.

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