Circuit for improving on-off performance of LED driving chip

Abstract

The invention relates to the technical field of LED drive circuits, and discloses a circuit for improving on-off performance of an LED drive chip, which comprises a voltage transient detection module, a current-limiting charging module, a protection trigger module, a clamping control module, an accumulation charging module and a shutdown protection module. The voltage transient detection module detects whether the voltage of the VOUT pin exceeds a first voltage threshold value or not through a first voltage stabilizing diode, and establishes a current bypass channel; the current-limiting charging module controls the magnitude of bypass current through a current-limiting resistor; the protection trigger module utilizes bypass current to enable ISET voltage to rise to trigger overcurrent protection in a chip, and the ultrahigh-voltage MOSFET is turned off; the clamping control module clamps the ISET voltage in a safe range through a second voltage stabilizing diode; and the accumulative charging module realizes progressive charging of the electrolytic capacitor in a plurality of alternating current periods. Chip thermal breakdown caused by high-voltage and large-current impact at the starting moment can be effectively prevented, the reliability of an LED driving system is improved, and the service life of the LED driving system is prolonged.

Description

Technical Field

[0001] This invention relates to the field of LED driver circuit technology, and more specifically, to a circuit for improving the switching performance of LED driver chips. Background Technology

[0002] In LED lighting systems, linear LED driver chips are widely used in commercial and residential lighting due to their simple structure, low cost, and lack of electromagnetic interference. A typical linear LED driver scheme uses AC mains power rectified by a rectifier bridge to directly drive the LED string, with constant current control achieved through the ultra-high voltage MOSFET inside the driver chip. This scheme features high power factor characteristics, meets the energy efficiency requirements of lighting systems, and holds a significant position in the market.

[0003] However, linear LED driver chips face severe reliability challenges at power-on. When the system powers on, the electrolytic capacitor connected in parallel with the LED has an initial voltage of zero, while the peak rectified input voltage can reach 311V. In the initial stage before the LED is turned on, the electrolytic capacitor is charged through the ultra-high voltage MOSFET inside the driver chip, with a charging current of 2A or even higher. At this time, the MOSFET is simultaneously subjected to high voltage and high current, with instantaneous power consumption exceeding 600W, far exceeding the chip's safe operating area limit. This high-voltage, high-current surge causes the MOSFET junction temperature to rise sharply, exceeding the maximum junction temperature limit within milliseconds, leading to thermal breakdown failure. In applications with frequent switching in smart lighting systems, the driver chip fails after only a few hundred switching cycles, with a failure rate as high as 15%, seriously affecting the reliability of the lighting system and user satisfaction.

[0004] In existing technologies, solutions to this problem mainly include two approaches: internal chip improvement and external current limiting. Internal chip improvement involves adding complex protection circuits and control logic to shut down the MOSFET upon detecting overcurrent or overtemperature. However, this increases chip design complexity and cost, and the response speed is limited by the delay of the detection circuit. External current limiting methods limit the charging current through a large-value series resistor, but this leads to increased power consumption and decreased efficiency during normal operation. Therefore, a simple, reliable, low-cost protection solution that does not affect normal operating performance is needed to address the technical problem of thermal breakdown of linear LED driver chips at power-on. Summary of the Invention

[0005] This invention provides a circuit to improve the power-on and power-off performance of LED driver chips, solving the technical problems of linear LED driver chips experiencing thermal breakdown failure of ultra-high voltage MOSFETs due to high voltage and high current surges at power-on, as well as existing protection schemes having slow response speed, high cost, and impact on normal operation performance.

[0006] This invention provides a circuit for improving the switching performance of an LED driver chip, comprising: The voltage transient detection module acquires the power-on transient input voltage signal and uses a voltage threshold detection method to obtain the high voltage status determination result of the output pin; The current-limiting charging module, based on the high-voltage state determination result, adopts a current bypass channel establishment and current-limiting control method to obtain the controlled electrolytic capacitor charging current; The protection trigger module, based on the rise characteristics of the ISET pin voltage, adopts an internal overcurrent protection trigger method to obtain the turn-off state of the ultra-high voltage MOSFET. The clamping control module, based on the current bypass state after the MOSFET is turned off, uses a second Zener diode clamping method to limit the voltage of the ISET pin and stabilize the bypass current. The cumulative charging module, based on the gradual charging process of electrolytic capacitors, uses the AC periodic cumulative effect to obtain a charging curve in which the capacitor voltage gradually rises to the LED turn-on voltage. The shutdown protection module, based on the voltage and current characteristics of the system shutdown process, uses energy release analysis to obtain the complete working cycle of the safety confirmation and protection circuit during the shutdown process.

[0007] In a preferred embodiment, the voltage transient detection module includes: Based on the safe operating area characteristics of LED driver chips and the operating voltage range of LED strings, a first voltage threshold is determined. A voltage comparison method is used to compare the real-time monitored voltage of the VOUT pin with the first voltage threshold to obtain a high-voltage status determination signal; The first voltage threshold is achieved by the first Zener diode. The cathode of the first Zener diode is connected to the VOUT pin, and the anode is connected to the ISET pin through a current-limiting resistor. When the VOUT voltage exceeds the first voltage threshold, the first Zener diode is turned on, establishing a current bypass path.

[0008] In a preferred embodiment, the voltage transient detection module further includes: Based on the timing characteristics of the high-voltage state determination signal, the timing and frequency of protection triggering during the start-up process are analyzed. In an AC mains-powered LED system, the input voltage fluctuates periodically. When the voltage rises above the first voltage threshold, the first Zener diode turns on, and the protection circuit is activated. When the voltage drops below the first voltage threshold, the first Zener diode turns off, and the protection circuit stops. The protection circuit operates near the voltage peak in each cycle.

[0009] In a preferred embodiment, the current-limiting charging module includes: Based on the conduction characteristics of the first Zener diode, a current bypass path is established from the VOUT pin to the ISET pin; When the first Zener diode is turned on, a new current path is formed between VOUT and ISET. The current flows from the cathode to the anode of the first Zener diode, from the anode to one end of the current-limiting resistor, and then through the current-limiting resistor to the other end. The other end of the current-limiting resistor is connected to the ISET pin. The current flows from the ISET pin through the external setting resistor to ground. This new path bypasses the ultra-high voltage MOSFET inside the chip.

[0010] In a preferred embodiment, the current-limiting charging module further includes: Based on the relationship between the resistance value of the current-limiting resistor and the current flowing through it, the current limiting parameters of the bypass channel are determined. The functional objectives of the current-limiting resistor include limiting the magnitude of the bypass current to prevent overload of the first Zener diode and the current-limiting resistor itself, generating a sufficient voltage drop to raise the ISET voltage to the threshold for triggering chip protection, and carrying the full charging current of the electrolytic capacitor after the chip protection is triggered to achieve effective current limiting. When the ISET voltage exceeds the preset current control threshold, the chip's internal control logic determines it to be an overcurrent state and immediately shuts down the MOSFET for protection.

[0011] In a preferred embodiment, the protection trigger module includes: Calculate the real-time voltage value of the ISET pin based on the voltage drop relationship between the bypass current and the current limiting resistor; Once the bypass channel is established, the bypass current is injected into the ISET node, changing the voltage of ISET. The voltage drop from VOUT through the first Zener diode to ISET is equal to the voltage of VOUT minus the breakdown voltage of the first Zener diode, which equals the voltage of ISET plus the voltage drop of the current-limiting resistor. The voltage drop generated by the bypass current through the current-limiting resistor causes the voltage of ISET to rise significantly.

[0012] In a preferred embodiment, the protection triggering module further includes: A voltage comparison method is used to compare the voltage at the ISET pin with the chip's internal current control threshold to generate an overcurrent detection signal; The current control circuit integrated inside the driver chip includes a voltage comparator, which compares the voltage at the ISET pin with the internal reference voltage. When the ISET voltage is higher than the reference voltage, the comparator outputs a high level, indicating that the output current exceeds the set value or an abnormality has occurred. The control logic immediately reduces the conduction level of the MOSFET or completely turns off the MOSFET to achieve overcurrent protection.

[0013] In a preferred embodiment, the protection triggering module further includes: Based on the overcurrent determination signal, a control logic driving method is adopted to output the MOSFET gate turn-off control signal; The control logic unit inside the chip receives the overcurrent judgment signal output by the comparator and generates the MOSFET gate drive signal based on the overcurrent judgment signal. When the overcurrent judgment signal is received, the control logic immediately enters the protection mode and outputs a turn-off signal to the MOSFET gate drive circuit. The drive circuit pulls the gate voltage low, forcing the MOSFET to turn off. After the MOSFET is turned off, the current between the drain and source is blocked, and the total power consumption of the chip is greatly reduced.

[0014] In a preferred embodiment, the clamping control module includes: The second voltage threshold is determined based on the voltage withstand limit of the ISET pin and the clamping characteristics of the second Zener diode. The second voltage threshold needs to be set below the maximum withstand voltage of ISET, while being higher than the current control threshold of the chip, to ensure that the protection function is not affected. The second voltage threshold is achieved through a second Zener diode. The cathode of the second Zener diode is connected to the ISET pin, and the anode is connected to ground. When the ISET voltage rises to the breakdown voltage of the second Zener diode, the second Zener diode turns on, clamping the ISET voltage at the breakdown voltage value to prevent it from rising further.

[0015] In a preferred embodiment, the cumulative charging module includes: A calculation model for capacitor voltage increment is established based on the capacitor charging characteristics within a single AC cycle. During the operation of the protection circuit, the electrolytic capacitor is charged with a limited current. When the voltage exceeds the first voltage threshold, the first Zener diode is turned on, the protection circuit is activated, and the charging current is limited by the current-limiting resistor as it flows through the bypass channel. The basic relationship of capacitor charging is that the change in voltage across the capacitor is equal to the amount of charge divided by the capacitance value, and the amount of charge is equal to the charging current multiplied by the charging time. Using a recursive accumulation method, the time-series variation curve of the capacitor voltage over multiple AC cycles is calculated. The capacitor voltage shows a step-like increase, and through multiple cycles of gradual charging, the capacitor voltage steadily rises to the LED turn-on voltage.

[0016] The beneficial effects of this invention are as follows: Effective control of high voltage and high current at startup is achieved through an external passive protection circuit, without requiring modification to the chip's internal design. This offers advantages such as high versatility and low cost. The protection circuit uses a first Zener diode to establish a bypass channel, utilizes a current-limiting resistor to control the charging current, and triggers the chip's internal overcurrent protection mechanism to actively turn off the ultra-high voltage MOSFET, reducing the chip's instantaneous power consumption from hundreds of watts to below 30W, completely eliminating the risk of thermal breakdown. Test results show that after adding the protection circuit, the driver chip can withstand more than 50,000 switching cycles without failure, compared to hundreds of cycles without protection, improving reliability by more than 100 times and meeting the long-term use requirements of commercial lighting systems. The protection circuit completely deactivates during normal operation, consuming no power and affecting system performance. It operates only briefly during the power-on phase, accumulating approximately 40 milliseconds of operation. After the LED illuminates, the protection circuit automatically deactivates, the bypass channel closes, and the system's output current accuracy, stability, and efficiency remain identical to when the protection circuit is not active. The electrolytic capacitor is gradually charged through the AC cycle accumulation effect, with an LED illumination delay of approximately 50 milliseconds, falling within the persistence of vision and imperceptible in actual use. The protection circuit's state transitions are entirely driven by natural voltage and current changes, requiring no external control signals or complex logic, demonstrating the advantages of a simple, reliable, and easy-to-implement passive protection scheme. Attached Figure Description

[0017] Figure 1 This is a block diagram of a circuit for improving the switching performance of an LED driver chip according to the present invention; Figure 2 This is a flowchart of a circuit for improving the switching performance of an LED driver chip according to the present invention. Detailed Implementation

[0018] The subject matter described herein will now be discussed with reference to exemplary embodiments. It should be understood that these embodiments are discussed only to enable those skilled in the art to better understand and implement the subject matter described herein, and changes may be made to the function and arrangement of the elements discussed without departing from the scope of this specification. Various processes or components may be omitted, substituted, or added as needed in the examples. Furthermore, some features described in the examples may be combined in other examples.

[0019] At least one embodiment of the present invention discloses a circuit for improving the switching performance of an LED driver chip, such as... Figures 1 to 2 As shown, it includes the following steps: The voltage transient detection module acquires the power-on transient input voltage signal and uses a voltage threshold detection method to obtain the high voltage status determination result of the output pin; The core function of this step is to monitor the voltage status of the driver chip's output pins in real time during the LED driver system's power-on process, and accurately determine whether it is in a high-voltage danger state requiring protection by comparing it with a preset threshold, thus providing a trigger signal for subsequent protection actions; specifically, it includes the following steps: S11, based on the output characteristics of the AC power supply and rectifier circuit, obtain the instantaneous value sequence of the system input voltage; In an LED lighting system, AC mains power is rectified by a rectifier bridge to output a fluctuating DC voltage. For a 220V AC input, the rectified voltage waveform is a full-wave rectified waveform, with the instantaneous value changing from 0V to the peak voltage. The peak voltage equals the AC effective value multiplied by the square root of 2, which is approximately 311V. This voltage is connected to the positive terminal of the LED string through a current-limiting resistor R1. At the moment the system is turned on, the rectified voltage rises from zero and reaches its first peak value after several milliseconds. The input voltage signal is obtained through the natural connection of the circuit; the output of the rectifier bridge is connected to the positive terminal of the LED string through a fuse and the current-limiting resistor R1, and the voltage at this node is the system input voltage. Since the LED string consists of multiple LEDs connected in series, the forward conduction voltage of each LED is approximately 3V, and the total forward voltage of the entire string of LEDs is determined by the number of LEDs connected in series. In the initial stage of power-on, before the input voltage reaches the total forward voltage of the LED string, the LEDs are not conducting and exhibit a high-impedance state. At this time, the input voltage is directly applied to the electrolytic capacitor C1 connected in parallel with the LEDs. The capacitance of electrolytic capacitor C1 is typically chosen to be between 47μF and 220μF for filtering and power factor correction. Initially, the capacitor's voltage is zero. As the input voltage rises, the capacitor begins to charge. Due to its large capacitance and initial zero voltage, the initial charging current is very large, according to the capacitor's charging characteristics. The charging current path is as follows: from the positive terminal of the rectifier bridge output, through the current-limiting resistor R1, it flows into the positive terminal of electrolytic capacitor C1, and out from the negative terminal of C1. This negative terminal connects to the VOUT pin of the driver chip, and the current enters the chip internally, passing through the ultra-high voltage MOSFET, flowing out from the ISET pin, and then through the external setting resistor R2 to ground. In this charging circuit, the voltage at the VOUT pin is equal to the voltage at the negative terminal of capacitor C1. Because the capacitor is in the initial charging stage, the voltage difference across its terminals is very small, approximately zero. Therefore, the voltage at the negative terminal of capacitor C1 is approximately equal to the voltage at the positive terminal, which is approximately equal to the input voltage. Thus, the VOUT pin voltage follows the input voltage change; when the input voltage reaches its peak value of 311V, the VOUT voltage also approaches 311V. By setting a voltage detection point in the circuit and connecting it to the VOUT pin, the instantaneous value sequence of the input voltage can be obtained. This sequence reflects the voltage fluctuation characteristics of the AC mains power after rectification, providing a data basis for subsequent threshold comparisons.

[0020] S12, Based on the safe operating area characteristics of the LED driver chip and the operating voltage range of the LED string, determine the first voltage threshold;The first voltage threshold is a key parameter of the protection circuit, and its setting requires comprehensive consideration of multiple factors. First, let's analyze the operating voltage range of the LED string. Assume the LED string consists of several LEDs connected in series, with each LED's forward voltage between 2.8V and 3.3V, typically 3V. For a string containing 30 LEDs, the total forward voltage range is 84V to 99V, typically 90V. During normal operation, the VOUT pin voltage equals the forward voltage of the LED string, falling within this range. To ensure the protection circuit does not malfunction during normal operation, the first voltage threshold must be higher than the maximum operating voltage of the LED string. Considering the differences in the number of LEDs in different applications and the margin for voltage fluctuation, the first voltage threshold is set to 100V. Next, let's analyze the safe operating area characteristics of the driver chip. The safe operating area of ​​the ultra-high voltage MOSFET inside the chip is determined by three parameters: voltage, current, and power consumption. The maximum withstand voltage of the MOSFET is typically 600V to 700V, providing ample margin in terms of voltage. However, the limitation of the safe operating area mainly comes from instantaneous power consumption. The safe power consumption of the MOSFET is related to factors such as heat dissipation conditions, junction temperature, and operating time. For a typical linear LED driver chip, when the voltage across the MOSFET is 300V and the current flowing through it is 0.5A, the instantaneous power consumption is 150W. If this continues for more than a few milliseconds, the junction temperature will rise sharply, exceeding the maximum junction temperature limit and causing thermal breakdown. At power-on, if the VOUT voltage reaches 311V, the charging current flowing through the MOSFET can reach 2A or even higher, resulting in an instantaneous power consumption exceeding 600W, far exceeding safety limits. Therefore, protection must be activated before the voltage reaches a dangerous level. The first voltage threshold is set to 100V. When the VOUT voltage exceeds this threshold, even if the LED is not yet turned on, protective measures should be taken immediately to prevent the voltage from rising further to a more dangerous area. The first voltage threshold is achieved through the first Zener diode D1. A Zener diode is a special type of diode that is in the off state when the reverse voltage is below its breakdown voltage, with almost no current flowing through it; when the reverse voltage reaches the breakdown voltage, the diode turns on, the current increases rapidly, but the voltage across it remains basically at the breakdown voltage value, exhibiting voltage regulation characteristics. Select a Zener diode with a breakdown voltage of 100V as D1 and connect it in the circuit so that D1 conducts when the VOUT voltage exceeds 100V, establishing a current bypass path. The power rating of the Zener diode must also be considered. During the operation of the protection circuit, bypass current will flow through D1, resulting in power dissipation. The power dissipation of D1 is equal to the voltage across its terminals multiplied by the current flowing through it. Assuming the bypass current is limited to 200mA and the forward voltage drop of D1 is 100V, the power dissipation is 20W. Select a Zener diode with a rated power of 30W or higher to ensure sufficient power margin. Also, choose a device with a good thermal package, such as a TO-220 or TO-247 package, for easy heat dissipation via a heatsink or PCB copper foil.

[0021] S13, using a voltage comparison method, compares the real-time monitored VOUT pin voltage with the first voltage threshold to obtain a high-voltage state determination signal; Voltage comparison is achieved naturally through the conduction characteristics of Zener diode D1. The cathode of D1 is connected to the VOUT pin, and the anode is connected to the ISET pin through a current-limiting resistor R3. During normal operation, the VOUT voltage equals the forward voltage of the LED string, approximately 90V, which is lower than the breakdown voltage of D1 (100V). At this time, D1 is in the off state, with only a tiny reverse leakage current between the cathode and anode, typically in the microamp level and negligible. D1's off state is equivalent to an open circuit; there is no electrical connection between the VOUT and ISET pins, and the protection circuit does not operate. As the VOUT voltage rises with the input voltage at power-on, its change is continuously monitored. When the VOUT voltage rises to 100V, the reverse voltage across D1 reaches the breakdown voltage, and D1 begins to conduct. After conduction, D1 exhibits a low-resistance state, allowing current to flow from the cathode to the anode. The conduction of D1 establishes a low-resistance path between VOUT and ISET, enabling current to flow from VOUT to ISET. The conduction state of D1 serves as the high-voltage condition determination signal. This signal is not a digital or level signal, but rather manifests as the establishment of a current path. When D1 is on, it means the system determines that VOUT is in a high-voltage dangerous state, requiring the activation of protective measures. The accuracy of this determination depends on the precision of D1's breakdown voltage. The breakdown voltage of a Zener diode has a certain error range, typically between ±5 and ±10% of its nominal value. Selecting a high-precision Zener diode can improve the consistency of the determination. Simultaneously, the breakdown voltage is affected by temperature, changing slightly with increasing temperature. For a 100V Zener diode, the temperature coefficient is approximately 0.1V per degree Celsius. In practical applications, the range of ambient temperature variation is limited, and the effect of temperature on the breakdown voltage is within acceptable limits. Furthermore, D1 has a very fast response speed, with a transition time from cutoff to conduction on the order of nanoseconds, far faster than the rate of change of the input voltage. This ensures that when the VOUT voltage exceeds the threshold, protection can be activated immediately without significant delay. The acquisition of the high-voltage condition determination signal provides the triggering condition for subsequent steps, enabling the protection circuit to intervene at the appropriate time, so as not to malfunction during normal operation or delay protection when a dangerous condition occurs.

[0022] S14, based on the timing characteristics of the high-voltage state determination signal, analyze the timing and frequency of protection triggering during the start-up process; In an AC mains-powered LED system, the input voltage exhibits periodic fluctuations. For 50Hz mains power, each cycle is 20 milliseconds. After full-wave rectification, the voltage waveform frequency doubles to 100Hz, with a cycle of 10 milliseconds. Within each 10-millisecond cycle, the voltage rises from zero to a peak value and then falls back to zero. When the voltage rises above 100V, D1 conducts, and the protection circuit activates. When the voltage drops below 100V, D1 cuts off, and the protection circuit stops. Therefore, within each cycle, the protection circuit only operates for a short period near the voltage peak. Calculate the percentage of time the protection circuit operates. Assume the rectified voltage waveform is the absolute value of a sine wave, i.e., the voltage versus time relationship is V equal to Vpeak multiplied by the sine value, where Vpeak is the peak value of 311V. When V equals 100V, the sine value equals 100 divided by 311, which equals 0.321. Calculate the corresponding phase angle using the arcsine function. The phase angle equals the arcsine 0.321, which is approximately 18.7 degrees. The voltage rises to 100V and then falls back to 100V, corresponding to a phase range of 18.7 degrees to 161.3 degrees, spanning 142.6 degrees. This represents 142.6 divided by 180, or 0.79, of the entire 180-degree cycle. Therefore, the protection circuit is active for approximately 79% of each half-cycle. In the initial few cycles after system startup, the protection is triggered in each cycle because the electrolytic capacitor voltage has not yet risen to the LED turn-on voltage. As the capacitor voltage gradually increases, the charging current gradually decreases, but the protection circuit will still operate during the peak period of each cycle as long as the LED is not turned on. The protection triggering time precisely corresponds to the instant the input voltage waveform exceeds 100V. Due to the stable frequency of the AC power supply, the protection triggering exhibits a periodic pattern. This periodic protection action is a key feature of this invention, allowing the electrolytic capacitor to charge gradually over multiple cycles, avoiding a single large current surge. The details of the protection action within a single cycle are analyzed. When the voltage just exceeds 100V, D1 conducts, and the ISET voltage begins to rise. The chip's internal control logic detects that the ISET voltage exceeds the threshold and outputs a turn-off signal to the MOSFET gate. The MOSFET's turn-off time is typically on the order of microseconds, extremely rapid. After the MOSFET turns off, the internal current of the chip is blocked, and the capacitor charging current is diverted to the bypass channel. At this time, the capacitor charges slowly with a current-limited limit. As the input voltage continues to rise to its peak and then falls, when the voltage drops below 100V, D1 is cut off, the ISET voltage drops back, and the chip's MOSFET turns on again. However, by this time, the input voltage has already decreased, and no large current is generated. Throughout the entire cycle, the chip's internal MOSFET is only turned off near the voltage peak, remaining on for the rest of the time or naturally entering a low-power state as the input voltage decreases. Through this periodic protection action, the system achieves precise protection against each voltage peak during the power-on phase, ensuring that the chip always operates within a safe range.The protection frequency is equal to the frequency of the rectified voltage, i.e., 100Hz, triggering 100 protection actions per second. This high-frequency protection response is a significant advantage of external protection circuits compared to slow protection methods, enabling real-time tracking of input voltage changes and providing immediate protection.

[0023] The current-limiting charging module, based on the high-voltage state determination result, adopts a current bypass channel establishment and current-limiting control method to obtain the controlled electrolytic capacitor charging current; The core function of this step is to establish a current bypass path from the electrolytic capacitor to the ISET pin through the first Zener diode D1 after detecting that the VOUT pin is in a high-voltage state. At the same time, the bypass current is limited by the current-limiting resistor R3, thereby effectively controlling the charging current of the electrolytic capacitor and providing conditions for triggering chip protection and reducing system stress. Specifically, it includes the following steps: S21, based on the conduction characteristics of the first Zener diode D1, establish a current bypass path from the VOUT pin to the ISET pin; When D1 is turned on due to VOUT voltage exceeding 100V, a new current path is formed between VOUT and ISET. Analyze the current flow and path characteristics of this path. Before D1 is turned on, the charging current path of electrolytic capacitor C1 is as follows: the capacitor's negative terminal is connected to the VOUT pin, the current enters the chip, passes through the drain and source of the internal high-voltage MOSFET, flows from the chip's internal node to the sampling circuit inside the ISET pin, and finally flows to ground through the external setting resistor R2. In this path, the current flows entirely through the high-voltage MOSFET inside the chip, which is the root cause of thermal breakdown. When D1 is turned on, a new path appears: the capacitor's negative terminal is connected to the VOUT pin, the current flows through the cathode to the anode of D1, from the anode of D1 to one end of the current-limiting resistor R3, through R3 to the other end, the other end of R3 is connected to the ISET pin, and the current flows from the ISET pin through the external setting resistor R2 to ground. This new path bypasses the high-voltage MOSFET inside the chip and forms a loop directly from external devices. Because D1 exhibits low resistance after conduction, and the internal MOSFET of the chip has a certain on-resistance even when fully turned on, the impedance of the new path may be lower than that of the original path. However, it should be noted that a current-limiting resistor R3 is connected in series in the new path, and the large resistance of R3 makes the total impedance of the new path significantly higher than that of the original path. Therefore, at the instant D1 turns on, if the chip's MOSFET is still in the on state, most of the current will still flow through the chip's internal circuitry, with only a small portion flowing through the bypass channel. The true function of the bypass channel is not simply to shunt current, but to trigger the chip's protection mechanism by increasing the voltage at the ISET pin, which will be explained in detail in the next sub-step. The establishment of the bypass channel is instantaneous, and the response speed depends on the switching characteristics of D1. The transition time of the Zener diode from cutoff to conduction is on the order of nanoseconds, almost instantaneous. Once the VOUT voltage reaches 100V, D1 immediately turns on, and the bypass channel is established immediately. This rapid response ensures that the protection circuit can intervene promptly without significant protection delay. The electrical characteristics of the bypass channel are determined by both D1 and R3. After D1 is turned on, the voltage across it is clamped at approximately 100V, exhibiting constant voltage source characteristics. The larger the current through D1, the slightly higher its actual on-state voltage drop, but the change is very small, typically within a few volts. For power Zener diodes, this allows them to withstand relatively large currents while maintaining voltage stability. The selection of the resistance value of R3 is a critical parameter, which will be analyzed in detail in the next sub-step. Establishing the bypass channel lays the foundation for subsequent current limiting control and chip protection triggering, and is a key step in the entire protection process.

[0024] S22, based on the relationship between the resistance value of the current-limiting resistor R3 and the current flowing through it, determine the current limiting parameters of the bypass channel; The current-limiting resistor R3 is a core control component in the bypass channel, and its resistance value needs to be selected by considering multiple factors. First, the functional objectives of R3 are clear: 1) to limit the magnitude of the bypass current, preventing overload of D1 and R3 itself; 2) to generate a sufficient voltage drop so that the ISET voltage rises to the threshold triggering the chip protection; and 3) to carry the entire charging current of the electrolytic capacitor after the chip protection is triggered, achieving effective current limiting. Analyzing the voltage requirements of the ISET pin: For a typical linear LED driver chip, the ISET pin is connected to ground through an external setting resistor R2, and the resistance value of R2 determines the output current. The chip has an internal reference voltage source, generating a reference voltage of approximately 0.6V. During normal operation, the internal operational amplifier adjusts the conduction level of the MOSFET, stabilizing the ISET voltage at 0.6V. At this time, the current flowing through R2 is equal to 0.6V divided by the resistance value of R2. This current is mirrored to the output terminal, forming a constant LED current. When the ISET voltage exceeds 0.6V, the internal control logic determines it as an overcurrent state and immediately turns off the MOSFET for protection. Therefore, the ISET voltage threshold for triggering protection is 0.6V. After the bypass channel is established, the current flowing through R3 creates a voltage drop across R3. This voltage drop, combined with the voltage drop across R2, determines the ISET voltage. Let the current flowing through the bypass channel be I, and the current flowing through R2 be I2. Then the ISET voltage equals I2 multiplied by R2. Simultaneously, the voltage relationship from VOUT through D1 to ISET is: VOUT voltage minus the voltage drop across D1 equals the ISET voltage plus the voltage drop across R3. That is, VOUT minus 100V equals I multiplied by R3 plus I2 multiplied by R2. In the initial stage, the MOSFET is not yet turned off, and the ISET voltage is close to 0.6V. I2 equals 0.6V divided by R2. Assuming R2 is 10 ohms, then I2 is 60mA. If VOUT is 200V, then 100V equals I multiplied by R3 plus 0.6V, that is, I multiplied by R3 equals 99.4V. If R3 is 500 ohms, then I is approximately 199mA. At this point, the ISET voltage begins to rise due to the injection of bypass current. The bypass current I and the R2 current I2 converge at the ISET node and flow together to ground. The ISET voltage is equal to the sum of the two currents multiplied by the equivalent impedance of R2. However, in reality, due to the chip's internal current control mechanism, when the bypass current causes the ISET voltage to exceed the 0.6V threshold, the chip immediately turns off the MOSFET. After the MOSFET is turned off, no current flows into the chip from VOUT, and all charging current must pass through the bypass channel. At this time, the bypass current I is equal to the charging current of the electrolytic capacitor. The magnitude of the charging current is determined by the capacitor characteristics and the total impedance of the circuit. After the protection is triggered, the total impedance of the charging circuit is the current-limiting resistor R1 plus the voltage drop across D1 plus the series connection of R3 and R2.Assuming a peak input voltage of 311V, a voltage drop of 100V from D1, a resistor R1 of 100 ohms, a resistor R3 of 500 ohms, and a resistor R2 of 10 ohms, the total impedance is 610 ohms. The voltage drop is 311V minus 100V, which equals 211V. The current is 211V divided by 610 ohms, which equals approximately 346mA. This current is the charging current after current limiting. Compared to the charging current of over 2A without protection, this is significantly reduced. The resistance value of R3 needs to be selected to ensure that this current is within a safe range, while not being too large, which would lead to an excessively long charging time. Typically, R3 is chosen in the range of 200 ohms to 1000 ohms. The larger the resistance value, the stronger the current limiting effect, but the longer the charging time. Choosing 500 ohms is a typical value, balancing protection effect and charging speed. The power rating of R3 is also important. The current flowing through R3 appears repeatedly in multiple cycles during power-on, although each duration is only a few milliseconds, the peak power consumption is relatively high. Assuming a current of 346mA and R3 of 500 ohms, the instantaneous power consumption is the square of the current multiplied by the resistance, which equals approximately 60W. Considering a duty cycle of approximately 79%, the average power consumption is approximately 47W. However, since the power-on process is short, typically completed within 1 second, the accumulated energy is limited. Selecting a metal film resistor or wire-wound resistor with a rated power of 2W to 5W, utilizing its instantaneous overload capacity and heat capacity, can safely withstand the power consumption pulse during the power-on phase. In the PCB design, sufficient heat dissipation space should be reserved for R3, or multiple resistors should be connected in parallel to share the power. By properly selecting the parameters of R3, the bypass current can be effectively limited, ensuring the reliability of the protection circuit itself.

[0025] S23 uses the principle of current shunting to analyze the distribution and change of the total current in the system after the bypass channel is established; During the brief period between the establishment of the bypass channel and the turn-off of the chip MOSFET, two parallel current paths exist in the system: one through the internal MOSFET and the other through the external bypass channel D1 and R3. Analyze the current distribution during this transition phase. According to circuit principles, the current distribution in parallel paths is inversely proportional to the impedance of each path. The on-resistance of the internal MOSFET is typically in the range of a few ohms to tens of ohms, assumed to be 20 ohms. The impedance of the bypass channel is mainly determined by R3, assumed to be 500 ohms. The impedance ratio of the two paths is 1:25, therefore the current mainly flows through the internal chip, with only a small portion shunted by the bypass channel. Assuming a total charging current of 2A, the current flowing through the MOSFET is approximately 1.92A, and the current flowing through the bypass channel is approximately 0.08A, or 80mA. Although the bypass current is small, it is sufficient to generate a significant voltage drop across R3. 80mA flowing through 500 ohms produces a 40V voltage drop. This voltage drop causes the ISET node voltage to rise. The ISET node is connected to resistors R3 and R2 and the chip's internal sampling circuitry. Before bypass current injection, the ISET voltage is 0.6V, and the current flowing through R2 is 60mA. When the bypass current of 80mA is injected into the ISET node, the total current is 140mA. However, the ISET node is not a simple resistor node but is connected to the chip's internal active control circuitry. The chip's internal current control logic monitors the ISET voltage, and when it detects that the voltage exceeds the 0.6V threshold, it immediately outputs a shutdown signal. Therefore, the injection of bypass current causes the ISET voltage to rise above the threshold, triggering chip protection. The time interval from bypass channel establishment to chip protection triggering is very short, typically on the order of microseconds. The chip's internal comparators and control logic have a very fast response speed, capable of completing detection and shutdown actions within a few microseconds. Once the MOSFET is turned off, the internal path of the chip is cut off, and the total charging current is forced to transfer to the single bypass channel. At this time, the bypass current equals the total charging current, and its magnitude is determined by the input voltage and the total impedance of the circuit. The total impedance includes the current-limiting resistor R1, the dynamic impedance of the Zener diode D1, the current-limiting resistor R3, and the setting resistor R2. Since the resistances of R1 and R3 are much larger than those of D1 and R2, the total impedance is mainly determined by R1 and R3. At the peak input voltage, assuming the input is 311V, VOUT is approximately 311V, and after passing through D1, it is approximately 211V. This voltage is applied across the series connection of R3 and R2. R3 plus R2 is approximately 510 ohms, and the current is approximately 414mA. This current flows from the positive terminal of the power supply through R1, the parallel branch of the LED load, into the positive terminal of C1, and from the negative terminal of C1 through D1, R3, and R2 to ground, forming a closed loop. At this time, the current no longer flows through the internal MOSFET of the chip, and the instantaneous power consumption of the chip decreases significantly. Although the MOSFET is subjected to a high voltage, the current flowing through it drops to the leakage current level, typically below the milliampere level. The instantaneous power consumption is reduced from hundreds of watts to below a few watts, which is completely within the safe range. Electrolytic capacitors are charged with a current that is limited, which slows down the charging speed but avoids overcurrent surges.By employing current shunting and path transfer, the system achieves a transition from a dangerous state to a safe state. The entire transition process is automatic, requiring no external control signals, demonstrating the ingenuity of the circuit design in this invention.

[0026] S24, based on the charging characteristics of the electrolytic capacitor and the magnitude of the current limiting current, calculate the increment of the capacitor voltage within a single AC cycle; During the operation of the protection circuit, the electrolytic capacitor is charged with a limited current, and its voltage gradually increases. The change in capacitor voltage within a single AC cycle is analyzed. The basic relationship for capacitor charging is: the change in voltage across the capacitor equals the amount of charge charged divided by the capacitance value. The amount of charge charged equals the charging current multiplied by the charging time. Within one AC cycle, the protection circuit operates for approximately 79% of the time; assuming a cycle of 10 milliseconds, the operating time is approximately 7.9 milliseconds. However, it should be noted that the charging current is not constant but varies with the input voltage. The input voltage waveform is an absolute sine wave, and the charging current exhibits a similar waveform. To simplify the calculation, the average current method is used to estimate the charging amount. Assuming the peak charging current is 414 mA, the average current is approximately 2 times the peak current divided by π, i.e., approximately 264 mA. Within 7.9 milliseconds, the amount of charge charged is 264 mA multiplied by 7.9 ms, which equals 2.09 millicoulombs. Assuming the capacitance value is 100 μF, the voltage increment is 2.09 mC divided by 100 μF, which equals 20.9 V. This means the capacitor voltage rises approximately 21V per cycle. If the LED string's on-state voltage is 90V, it takes about 4 to 5 cycles, or 40 to 50 milliseconds, to rise from zero to 90V. This time corresponds to the LED's lighting delay, which is usually imperceptible. In practice, as the capacitor voltage rises, the charging current gradually decreases, and the voltage increment also gradually decreases, exhibiting an exponential trend in the charging process. In the initial few cycles, the capacitor voltage is low, the charging current is high, and the voltage rises rapidly. As the capacitor voltage approaches the LED's on-state voltage, the difference between the capacitor and the input voltage decreases, the charging current decreases, and the voltage rise slows down. When the capacitor voltage reaches the LED's on-state voltage, the LED begins to conduct, the capacitor charging current bypasses the LED, and the charging process ends naturally. Through gradual charging over multiple cycles, the capacitor voltage rises smoothly to the target value, avoiding instantaneous large current surges and effectively protecting the driver chip. The calculation results provide a quantitative basis for subsequent analysis of the system's dynamic response and optimization of circuit parameters. Different capacitance values ​​and different current-limiting resistor values ​​will affect the charging time and voltage increment, and can be adjusted according to specific application requirements.

[0027] The protection trigger module, based on the rise characteristics of the ISET pin voltage, adopts an internal overcurrent protection trigger method to obtain the turn-off state of the ultra-high voltage MOSFET. The core function of this step is to utilize the voltage drop generated across the current-limiting resistor by the bypass current to cause the ISET pin voltage to exceed the chip's internal current control threshold, triggering the chip's own overcurrent protection mechanism. This enables the active turn-off of the ultra-high voltage MOSFET, thereby blocking the path of large current flowing through the chip and achieving the purpose of protecting the chip. Specifically, it includes the following steps: S31, based on the relationship between the bypass current and the voltage drop of the current-limiting resistor R3, calculate the real-time voltage value of the ISET pin; The ISET pin is the current sampling pin of the driver chip, and its voltage reflects the magnitude of the output current. During normal operation, ISET is connected to ground through an external setting resistor R2. Internally, the chip adjusts the MOSFET to stabilize the ISET voltage at a reference voltage of 0.6V, thus achieving constant current output. When the bypass channel is established, bypass current is injected into the ISET node, changing the ISET voltage. Analyzing the electrical connections of the ISET node, it is connected to three terminals: one end of the current-limiting resistor R3, one end of the setting resistor R2, and the chip's internal current sampling circuit. From the circuit topology, ISET is the convergence point of three branches. When the bypass channel is operating, current flows into ISET from R3 and out from R2 and the chip's internal sampling circuit. The chip's internal sampling circuit can be represented as a voltage source with internal resistance. During normal operation, the internal voltage source is 0.6V, which stabilizes the ISET voltage through a high-gain operational amplifier. When external current is injected into ISET, if the injected current is small, the internal amplifier can absorb the current by adjusting the MOSFET, maintaining the ISET voltage stability. However, when the injected current exceeds the internal regulation capability, the ISET voltage is forcibly raised. In this invention, the voltage drop generated by the bypass current through R3 is much greater than the 0.6V reference, which is sufficient to overcome the internal regulation capability and cause the ISET voltage to rise significantly. The method for calculating the ISET voltage is as follows. The voltage drop relationship from VOUT through D1 to ISET is: VOUT voltage minus D1 breakdown voltage equals ISET voltage plus the voltage drop across R3. Let VOUT be 200V and D1 breakdown voltage be 100V, then ISET voltage plus the voltage drop across R3 equals 100V. Let the bypass current be I and R3 be 500 ohms, then the voltage drop across R3 is 500I. The ISET voltage equals 100V minus 500I. At the same time, the ISET voltage equals the current flowing through R2 multiplied by the resistance of R2. At the ISET node, according to Kirchhoff's current law, the inflow current equals the outflow current. The bypass current I flows in, and the current I2 in R2 and the internal chip current Ic flow out, so I equals I2 plus Ic. Before the chip protection is triggered, Ic equals the LED current, which is set to tens to hundreds of milliamps. After the protection is triggered, the MOSFET is turned off, and Ic drops to zero. At this time, I equals I². The ISET voltage V_ISET equals I² multiplied by R2. Solving the system of equations: V_ISET equals 100 minus 500I, and I equals V_ISET divided by R2. Solving for V_ISET, we get V_ISET equals 100R² divided by 500 plus R2. Assuming R2 is 10 ohms, V_ISET equals 1000 divided by 510, which is approximately 1.96V. This voltage is much higher than the 0.6V threshold, sufficient to trigger the chip protection. If R2 is 5 ohms, V_ISET is approximately 0.99V, still higher than the threshold. Therefore, within a reasonable range of R2 resistance, the bypass current can make the ISET voltage exceed the protection threshold. The rise rate of the ISET voltage depends on the speed at which the bypass channel is established and the parasitic parameters of the circuit.The switching time of Zener diode D1 is in the nanosecond range, the parasitic capacitance in the circuit is very small, and the rise time of the ISET voltage is in the microsecond range. This speed is much faster than the change rate of the input voltage, ensuring timely protection. By accurately calculating the ISET voltage, the triggering reliability of the protection circuit can be verified, providing a theoretical basis for circuit design and parameter selection.

[0028] S32 uses a voltage comparison method to compare the voltage of the ISET pin with the chip's internal current control threshold to generate an overcurrent judgment signal; The driver chip integrates a current control circuit for constant current output and overcurrent protection. The core of this circuit is a voltage comparator that compares the ISET pin voltage with an internal reference voltage. The reference voltage is typically generated by a bandgap reference source, with a typical value of 0.6V, offering good temperature stability and high accuracy. One input of the comparator is connected to the ISET pin, and the other input is connected to the reference voltage. When the ISET voltage is lower than the reference voltage, the comparator outputs a low level, indicating that the output current is lower than the set value. The control logic then increases the MOSFET's conduction level, increasing the output current. When the ISET voltage equals the reference voltage, the comparator output is at the switching point, and the control logic maintains the MOSFET's current state, stabilizing the output current at the set value. When the ISET voltage is higher than the reference voltage, the comparator outputs a high level, indicating that the output current exceeds the set value or an anomaly has occurred. The control logic immediately decreases the MOSFET's conduction level or completely turns it off, achieving overcurrent protection. The comparator's response time is typically in the microsecond range, enabling rapid detection of changes in the ISET voltage and the output of a corresponding signal. In this invention, once the bypass channel is established, the ISET voltage rises from the normal 0.6V to approximately 2V, far exceeding the reference voltage. The comparator immediately detects this change and outputs a high-level overcurrent determination signal. This signal is transmitted to the control logic unit inside the chip, which makes a protection decision based on the overcurrent signal. The generation of the overcurrent determination signal is completely automatic, requiring no external control, and utilizes the chip's own protection function, which is a significant innovation of this invention. No external comparator or control chip is needed; the chip's internal protection mechanism can be triggered simply through clever circuit connections. This greatly simplifies circuit design and reduces cost and complexity. The reliability of the overcurrent determination signal depends on the comparator's accuracy and anti-interference capability. Modern analog chip comparator designs are very mature, featuring high gain, low offset, and fast response. The comparator's offset voltage is typically in the millivolt range, and its effect is negligible compared to the 0.6V reference and 2V overcurrent voltage. The comparator also has a certain hysteresis characteristic to prevent repeated jumps near the threshold, improving the stability of the determination. The overcurrent detection signal provides the trigger condition for the subsequent MOSFET turn-off and is a key link in the protection process.

[0029] S33, based on the overcurrent judgment signal, uses a control logic driving method to output the MOSFET gate turn-off control signal; The chip's internal control logic unit receives the overcurrent detection signal from the comparator and generates a drive signal for the MOSFET gate based on this signal. The control logic is typically implemented using digital or analog logic circuits, including logic gates, flip-flops, and state machines. When an overcurrent detection signal is received, the control logic immediately enters protection mode and outputs a turn-off signal to the MOSFET gate drive circuit. The MOSFET's turn-on and turn-off are controlled by its gate voltage. For an N-channel enhancement-mode MOSFET, when the gate voltage is higher than a certain threshold of the source voltage, the MOSFET turns on, forming a low-resistance channel between the drain and source. When the gate voltage is lower than the threshold, the MOSFET turns off, and a high-resistance state exists between the drain and source. The function of the gate drive circuit is to adjust the gate voltage according to the instructions of the control logic. During normal operation, the drive circuit outputs an appropriate gate voltage, allowing the MOSFET to operate in the linear or saturation region, achieving current regulation. In protection mode, the drive circuit pulls the gate voltage down to the source voltage or lower, forcing the MOSFET to turn off. The design of the gate drive circuit needs to consider the MOSFET's gate capacitance and drive capability. Ultra-high voltage MOSFETs typically have large gate capacitances, requiring sufficient drive current for rapid charging and discharging to achieve fast switching. The drive circuit typically employs a push-pull output structure, with the upper MOSFET charging the gate and the lower MOSFET discharging. During turn-off, the lower MOSFET conducts, rapidly pulling the gate voltage down, and the MOSFET quickly turns off. The turn-off speed depends on the ratio of the drive current to the gate capacitance, typically on the order of microseconds. In the application of this invention, after the control logic outputs a turn-off signal, the drive circuit completely turns off the MOSFET within a few microseconds. After the MOSFET is turned off, the current between the drain and source is blocked, leaving only a tiny leakage current, typically on the order of microamps to milliamps, negligible compared to the ampere-level charging current. The MOSFET remains off until the ISET voltage falls below the threshold. During the AC cycle, when the input voltage drops, causing the VOUT voltage to fall below the D1 breakdown voltage, D1 turns off, the bypass current disappears, the ISET voltage falls back to its normal value, the control logic exits the protection mode, and the MOSFET turns back on. This periodic turn-off and turn-on ensures that the MOSFET is in a protected state during the high-voltage period of each cycle, avoiding simultaneous high voltage and high current, effectively protecting the device. The design of the control logic and drive circuit fully utilizes the existing functional modules inside the chip, eliminating the need for additional circuitry and demonstrating the efficiency and economy of the solution.

[0030] S34, based on the MOSFET's off state, analyzes the blocking of internal current paths and the decrease in instantaneous power consumption; After the MOSFET is turned off, the current path from VOUT to ISET inside the chip is cut off. Let's analyze the changes in current and power consumption before and after turn-off. Before turn-off, the charging current of the electrolytic capacitor enters the chip from the VOUT pin, flows through the drain of the MOSFET to the source, and from the internal node of the source to the ISET sampling circuit. Assume the charging current is 2A, the MOSFET's on-resistance is 20 ohms, and the voltage drop across the MOSFET is 40V. The VOUT voltage is 200V, and the source voltage is 160V. The MOSFET's power consumption is current multiplied by voltage, i.e., 2A multiplied by 40V equals 80W. Although this is only the power consumption due to the voltage drop, the power consumption is still relatively high due to the large current. In reality, the MOSFET should withstand the voltage from the drain to ground, i.e., 200V. If the current is 2A, the instantaneous power consumption is 400W, which is close to or exceeds the safety limit. After turn-off, only leakage current exists between the MOSFET's drain and source. The magnitude of the leakage current is related to the MOSFET's manufacturing process, temperature, and voltage. For high-voltage MOSFETs, the leakage current is typically in the microamplitude range. Assuming a leakage current of 100μA, a MOSFET voltage of 200V, and a power consumption of 0.02W, this represents a decrease of several orders of magnitude compared to the hundreds of watts before shutdown, well within safe limits. The power consumption of other circuits within the chip primarily comes from control logic, the reference source, comparators, etc., typically ranging from tens to hundreds of milliwatts, negligible compared to the MOSFET's power consumption. Therefore, after the MOSFET is turned off, the chip's total power consumption drops from hundreds of watts to less than 1W, the junction temperature decreases significantly, and the risk of thermal breakdown is completely eliminated. This reduction in power consumption not only protects the MOSFET but also other parts of the chip. Excessive power consumption would cause the overall chip temperature to rise, affecting the performance and reliability of all circuits. By timely shutting down the MOSFET, power consumption is limited to a safe range, ensuring the chip maintains its normal operating temperature throughout the entire power-on process. Analyzing the voltage stress on the MOSFET: After shutdown, the voltage across the MOSFET equals the VOUT voltage minus the source voltage. Since the source is connected to ISET via internal circuitry, and the ISET voltage is approximately 2V, the source voltage is also on the same order of magnitude. The VOUT voltage can reach 311V at peak power-on, therefore the voltage across the MOSFET is approximately 309V. While this voltage is relatively high, it is far below the MOSFET's rated withstand voltage of 600V or 700V, providing sufficient safety margin. High-voltage MOSFETs are designed to withstand such high voltages; as long as the current is limited, the high voltage alone will not cause failure. By actively turning off the MOSFET, time decoupling of voltage and current is achieved, avoiding the dangerous state of simultaneous high voltage and high current. This is the core idea of ​​the protection mechanism of this invention. Power consumption analysis provides quantitative evidence for evaluating the protection effect, proving the effectiveness of the protection circuit.

[0031] The clamping control module, based on the current bypass state after the MOSFET is turned off, uses a second Zener diode clamping method to limit the voltage of the ISET pin and stabilize the bypass current. The core function of this step is to transfer all charging current to the bypass channel after the MOSFET is turned off. To prevent excessive bypass current from causing the ISET pin voltage to exceed its withstand voltage limit and resulting in damage, the ISET voltage is clamped and protected by the second Zener diode D2. This also achieves the final limitation of the bypass current, ensuring the safe and reliable operation of the protection circuit itself. Specifically, it includes the following steps: S41. Based on the increasing bypass current characteristic after MOSFET turn-off, analyze the rising trend of ISET pin voltage and overvoltage risk. Before the MOSFET is turned off, the charging current mainly flows through the inside of the chip, with only a small portion being shunted by the bypass current, resulting in a limited rise in the ISET voltage. After the MOSFET is turned off, the internal current path is blocked, and all charging current must pass through the bypass channel. At this time, the bypass current increases significantly, from tens of milliamps to hundreds of milliamps. The bypass current flowing through the current-limiting resistor R3 generates a larger voltage drop, causing the ISET voltage to rise further. Let's analyze the calculation of the ISET voltage. The voltage relationship from VOUT through D1 to ISET is: VOUT minus the breakdown voltage of D1 (100V) equals the ISET voltage plus the voltage drop across R3. Assuming VOUT is 311V, the bypass current is I, and R3 is 500 ohms, then the ISET voltage equals 211V minus 500I. The magnitude of the bypass current is determined by the total impedance of the circuit. The total impedance of the charging circuit includes the current-limiting resistor R1, the dynamic impedance of the Zener diode D1, the current-limiting resistor R3, and the setting resistor R2. R1 is typically tens to hundreds of ohms; let's assume it's 100 ohms. D1 has a very small dynamic impedance, let's assume it's 5 ohms. R3 is 500 ohms, and R2 is 10 ohms. The total impedance is approximately 615 ohms. Subtracting the D1 breakdown voltage of 100V from the peak input voltage of 311V leaves 211V applied across the total impedance. The current is approximately 343mA (211 divided by 615). This current flowing through R3 creates a voltage drop of 343mA multiplied by 500 ohms, equal to 171.5V. The ISET voltage is 211V minus 171.5V, equal to 39.5V. This voltage is very high, far exceeding the ISET pin's withstand voltage limit. The ISET pin is typically part of the chip's internal low-voltage circuitry, with a withstand voltage limit between 5V and 10V. If the ISET voltage reaches 39.5V, it can cause the internal ESD protection diode to break down or damage the internal circuitry due to overvoltage. Therefore, the ISET voltage must be limited to prevent it from exceeding safe limits. The upward trend of the ISET voltage is inversely proportional to the bypass current. The higher the current, the greater the voltage drop across R3, and the lower the ISET voltage. However, even with a large current, the ISET voltage may still exceed the safety limit. For example, if the current is 400mA, the voltage drop across R3 is 200V, and the ISET voltage is 11V, which is still higher than the 5.5V withstand voltage limit. Therefore, specialized clamping protection measures are required.

[0032] S42 determines the second voltage threshold based on the voltage withstand limit of the ISET pin and the clamping characteristics of the second Zener diode D2; The second voltage threshold is used to protect the ISET pin from overvoltage damage. This threshold needs to be set below the maximum withstand voltage of ISET while being above the chip's current control threshold to ensure that the protection function is not affected. The maximum withstand voltage of the ISET pin should be referenced from the chip's datasheet. For typical linear LED driver chips, the absolute maximum rated voltage of the ISET pin is usually 6V or lower. To ensure a safety margin, the actual operating voltage should be limited to 80% below the maximum rated value, i.e., approximately 5V or lower. Considering that the chip's current control threshold is 0.6V, the ISET voltage needs to be maintained above 0.6V after protection triggering to keep the MOSFET in the off state. Therefore, the second voltage threshold should be set between 0.6V and 5V. Choosing 3V as the second voltage threshold ensures both sufficient protection trigger margin and a sufficient safety margin to prevent overvoltage. The second voltage threshold is implemented through the second Zener diode D2. The cathode of D2 is connected to the ISET pin, and the anode is connected to ground. When the ISET voltage is lower than the breakdown voltage of D2, D2 is in the off state and does not affect the circuit operation. When the ISET voltage rises to the breakdown voltage of D2, D2 conducts, clamping the ISET voltage at the breakdown voltage value to prevent it from rising further. A Zener diode with a breakdown voltage of 3V is selected as D2. Zener diodes come in various breakdown voltage specifications, from a few volts to several hundred volts, which can be selected according to needs. 3V Zener diodes are common and readily available. The power rating of D2 needs to consider the current flowing through it. When D2 is clamped, the ISET voltage is clamped at 3V. According to the previous voltage relationship, VOUT minus 100V equals 3V plus the voltage drop across R3. Assuming VOUT is 311V, the voltage drop across R3 is 211V minus 3V equals 208V. The bypass current is 208V divided by 500 ohms, which equals 416mA. At this time, the current distribution at the ISET node is: the 416mA bypass current flows in, partly through R2 and partly through D2. The current in R2 is 3V divided by 10 ohms, which equals 300mA. The current of diode D2 is 416mA minus 300mA, which equals 116mA. The power dissipation of D2 is 3V multiplied by 116mA, which equals 348mW, less than 0.5W. Choosing a 1W rated power Zener diode provides sufficient power margin. The clamping characteristic of D2 ensures that the ISET voltage will not exceed 3V, remaining completely within the safe range. Simultaneously, 3V is far higher than the 0.6V control threshold, ensuring that the MOSFET remains off and the protection function is unaffected. The setting of the second voltage threshold is a critical parameter in the protection circuit design, directly affecting the circuit's safety and reliability.

[0033] S43 uses the Zener diode clamping principle to achieve precise limiting of the ISET pin voltage and overcurrent protection; When the ISET voltage rises to 3V, D2 conducts, exhibiting a low-resistance state. The volt-ampere characteristic of D2 is: below the breakdown voltage, the current is very small, exhibiting high resistance; at the breakdown voltage, the current increases rapidly, but the voltage remains essentially constant, exhibiting a constant voltage characteristic. After D2 conducts, the ISET node is forcibly clamped at 3V, and the ISET voltage will not continue to rise regardless of the bypass current. Analyzing the circuit state after clamping: The ISET voltage is 3V, the voltage across R2 is 3V, and the current flowing through R2 is 3V divided by the resistance of R2. The bypass current flows into ISET from R3, with part flowing to ground through R2 and the other part flowing to ground through D2. D2 acts as a shunt, absorbing excess current and maintaining the stability of the ISET voltage. The magnitude of the bypass current is determined by the VOUT voltage and the total circuit impedance. The total impedance includes the equivalent impedance of R1, D1, R3, and the ISET branch. The ISET branch consists of R2 and D2 connected in parallel. The resistance of R2 is 10 ohms. The dynamic impedance of D2 after it is turned on is very small, usually below 1 ohm. The equivalent impedance of the two connected in parallel is about 1 ohm. Therefore, the total impedance of the circuit is approximately the sum of the equivalent impedances of R1, R3, and ISET, i.e., 100 + 500 + 1 = 601 ohms. The peak input voltage of 311V minus the breakdown voltage of D1 of 100V leaves 211V applied across the total impedance. The bypass current is 211 divided by 601, approximately 351mA. This current flowing through R3 produces a voltage drop of 351mA multiplied by 500 ohms, which equals 175.5V. VOUT minus 100V minus 175.5V equals 35.5V, which should be equal to the ISET voltage. However, due to the clamping effect of D2, the ISET voltage is limited to 3V. This contradiction indicates that a re-analysis is needed. In fact, when D2 is clamped, the circuit enters a new equilibrium state. The ISET voltage is fixed at 3V. The voltage drop from VOUT to ground is distributed as follows: VOUT to the cathode of D1, the voltage drop across D1 is 100V; the anode of D1 to ISET is the voltage drop across R3; and ISET to ground is 3V. Therefore, VOUT minus 100V minus the voltage drop across R3 equals 3V. The voltage drop across R3 equals VOUT minus 103V. The bypass current equals the voltage drop across R3 divided by the resistance of R3. Assuming VOUT is 311V and the voltage drop across R3 is 208V, the bypass current is 416mA. This current is consistent with the previous calculation. Through the clamping effect of D2, the ISET voltage is precisely limited to 3V, avoiding the risk of overvoltage. Simultaneously, because the ISET voltage is clamped to a lower value, the voltage drop across R3 increases, and the bypass current increases slightly, but remains within a controllable range. The bypass current after clamping is mainly determined by the input voltage and the resistance of R3. The functions of D2 and R2 are to stabilize the ISET voltage and absorb excess current. The entire clamping process is automatic and requires no external control, demonstrating the ingenuity and reliability of the circuit design.

[0034] S44, based on the final limiting effect of bypass current, evaluates the stress and reliability of each component in the protection circuit; When the protection circuit is working, the electrical and thermal stresses borne by each component must be within safe limits to ensure the reliability of the protection circuit itself. The stress state of each component is analyzed one by one. First Zener diode D1: It withstands a voltage equal to its breakdown voltage of 100V and carries a bypass current of approximately 416mA. The power dissipation is 100V multiplied by 416mA, equaling 41.6W. This is the instantaneous power dissipation, lasting approximately 7 to 8 milliseconds in each AC cycle. For the remainder of a cycle, D1 is off, and the power dissipation is zero. The average power dissipation is approximately the instantaneous power dissipation multiplied by the duty cycle, i.e., 41.6W multiplied by 0.79, equaling approximately 33W. However, it should be noted that the power-on process is usually completed within 1 second, and the total energy is limited. A Zener diode with a rated power of 50W is selected, along with a heatsink, to safely withstand the power pulse during the power-on phase. Current-limiting resistor R3: It withstands a voltage of 208V and carries a current of 416mA. The power dissipation is 208V multiplied by 416mA, equaling approximately 86.5W. This is also the instantaneous power consumption, with an average power consumption of approximately 68W. For power resistors, the instantaneous overload capacity can typically reach several times the rated power. Choosing a 5W metal film resistor or a 10W wire-wound resistor, utilizing their heat capacity and instantaneous overload capacity, can withstand short-duration high-power pulses. In PCB design, reserve sufficient heat dissipation space for R3, or use two resistors in parallel to share the power. Second Zener diode D2: It withstands a voltage of 3V, and the current flowing through it, based on the previous analysis, is approximately 116mA. The power consumption is 3V multiplied by 116mA, equal to 348mW. This is very low power consumption, easily handled by a typical 1W Zener diode. Resistor R2: It withstands a voltage of 3V, and the current flowing through it is 300mA. The power consumption is 3V multiplied by 300mA, equal to 900mW. R2 is typically chosen with a rated power of 1W or 2W for safe operation. Electrolytic capacitor C1: During the protection circuit's operation, it is charged with a current-limiting current, significantly reducing current stress and preventing damage to the capacitor. The rated voltage of the capacitor is typically chosen to be higher than the system's maximum voltage; for example, a 400V or 450V capacitor is selected to provide sufficient voltage margin. LED string: No current flows through the LEDs before they are turned on, resulting in no stress. After the LEDs are turned on, the protection circuit deactivates, and the LEDs operate in normal constant current mode. Driver chip: After the MOSFET is turned off, internal power consumption is extremely low, and the chip operates within a safe temperature range. The ISET pin voltage is clamped at 3V, below the withstand voltage limit, eliminating overvoltage risk. Overall, the stress on each component of the protection circuit is within a safe range, providing ample margin. Through reasonable component selection and heat dissipation design, the protection circuit can operate reliably for a long period. In practical applications, extensive power-on / off cycle tests have shown no failures in the protection circuit, fully verifying the design's reliability.

[0035] The cumulative charging module, based on the gradual charging process of electrolytic capacitors, uses the AC periodic cumulative effect to obtain a charging curve in which the capacitor voltage gradually rises to the LED turn-on voltage. The core function of this step is to analyze the charging process of the electrolytic capacitor over multiple AC cycles under the action of the protection circuit, reveal the gradual increase pattern of the capacitor voltage, explain how the system smoothly transitions from a dangerous high-voltage state at startup to a normal operating state, and how the protection circuit automatically deactivates; specifically, it includes the following steps: S51, based on the capacitor charging characteristics within a single AC cycle, establishes a calculation model for the capacitor voltage increment; During the operation of the protection circuit, the electrolytic capacitor is charged with a limited current. The charging process of the capacitor within a single AC cycle is analyzed. After full-wave rectification, the AC mains voltage waveform is the absolute value of a sine wave with a frequency of 100Hz and a period of 10 milliseconds. Within each cycle, the voltage rises from zero to a peak value and then falls back to zero. When the voltage exceeds 100V, D1 conducts, and the protection circuit operates. When the voltage is below 100V, D1 is cut off, and the protection circuit stops. Therefore, the operating time of the protection circuit occupies a portion of each cycle. Based on previous calculations, the operating time accounts for approximately 79%, or about 7.9 milliseconds. During operation, the charging current flows through the bypass channel and is limited by the current-limiting resistor. The charging current is not constant but varies with the input voltage. The input voltage waveform is sinusoidal, and the charging current also exhibits a similar waveform. Let the rectified voltage be V, which is equal to Vpeak multiplied by the sine wave omega multiplied by t, where Vpeak is the peak value (311V) and omega is the angular frequency. The charging current I is equal to (V - 100V - Vcap) divided by R_total, where Vcap is the current capacitor voltage and R_total is the total circuit impedance of approximately 600 ohms. Initially, Vcap is close to zero, and the charging current is mainly determined by V - 100V. As Vcap increases, the charging current gradually decreases. Within a single cycle, assuming Vcap does not change significantly and can be considered constant, the charging current varies with V. The charging charge Q is equal to the integral of the current over time. Due to the complexity of current variation over time, the average value method is used to simplify the calculation. Let the average charging current within the cycle be Iavg, and the operating time be T_on. Then the charging charge Q is equal to Iavg multiplied by T_on. The average current is approximately 2 times the peak current divided by π. Assuming the peak current is 400mA, the average current is approximately 255mA. T_on is 7.9 milliseconds, and the charging charge is 255mA multiplied by 7.9ms, which equals approximately 2.01 millicoulombs. The capacitor voltage increment Delta V is equal to the charging charge divided by the capacitor value. Assuming a capacitor of 100μF, Delta V equals 2.01mC divided by 100μF, which equals 20.1V. This means the capacitor voltage rises approximately 20V per cycle. This result is consistent with previous calculations. The magnitude of the voltage increment is inversely proportional to the capacitance value and directly proportional to the charging current. The voltage increment varies for capacitors of different values. Larger capacitance values ​​result in smaller voltage increments and longer charging times. For example, for a 220μF capacitor, the voltage increment is approximately 9V, requiring more cycles to reach the LED's turn-on voltage. The voltage increment is also related to the charging current, which is controlled by the current-limiting resistor. Adjusting the value of R3 can change the charging speed. Larger resistance results in smaller charging current, smaller voltage increments, and longer charging times, but better protection. Smaller resistance results in larger charging current and shorter charging times, but weaker protection; a trade-off must be struck. The established calculation model provides a theoretical tool for predicting charging time and optimizing circuit parameters.

[0036] S52 uses a recursive accumulation method to calculate the time-series variation curve of capacitor voltage over multiple AC cycles. Based on the single-cycle voltage increment model, the capacitor voltage change over multiple cycles can be calculated recursively. Let the initial capacitor voltage Vcap_0 be 0V. After the first cycle, Vcap_1 equals Vcap_0 plus Delta V_1, where Delta V_1 is the voltage increment of the first cycle. Since the initial capacitor voltage is zero, the charging current is large, and Delta V_1 reaches its maximum value of approximately 20V. At the start of the second cycle, the capacitor voltage is 20V. During the second cycle, due to the increase in capacitor voltage, the charging current decreases slightly, and Delta V_2 is slightly less than Delta V_1, approximately 19V. Vcap_2 equals 20V plus 19V, which equals 39V. In the third cycle, Vcap_3 equals 39V plus 18V, which equals 57V. In the fourth cycle, Vcap_4 equals 57V plus 17V, which equals 74V. In the fifth cycle, Vcap_5 equals 74V plus 15V, which equals 89V. When the capacitor voltage approaches the LED turn-on voltage, assuming the total forward voltage of the LED is 90V, the capacitor voltage is 89V at the end of the 5th cycle, very close to the turn-on voltage. At the beginning of the 6th cycle, the capacitor voltage continues to rise, and when it reaches 90V, the LED begins to conduct. After the LED turns on, current flows through the LED bypass, and the capacitor charging process ends. From power-on to LED turn-on, approximately 5 to 6 cycles are completed, taking about 50 to 60 milliseconds. This time corresponds to the LED lighting delay, which is usually imperceptible because the human eye's visual persistence time is approximately 100 milliseconds, making a 50-millisecond delay almost imperceptible. Plotting the capacitor voltage change over time shows a stepped increase. Each cycle corresponds to a step, with the step height gradually decreasing, reflecting the gradual decrease in charging current. The slope of the curve reflects the charging speed; the initial slope is large, and the slope decreases later, conforming to the general rules of RC charging. Compared to the charging curve without protection, the charging curve under the protection circuit is smoother, avoiding instantaneous voltage jumps and large current surges. Through timing curve analysis, the working process and effect of the protection circuit can be intuitively understood. The curve can also be used to guide the optimization of circuit parameters, such as adjusting the current-limiting resistor value to change the charging speed, or adjusting the first voltage threshold to change the protection activation timing.

[0037] S53, based on the state where the capacitor voltage is close to the LED turn-on voltage, analyze the gradual decrease of the operating frequency of the protection circuit and the automatic exit process; As the capacitor voltage rises, the operating state of the protection circuit gradually changes. Initially, the capacitor voltage is zero, and the peak voltage VOUT is close to the input peak voltage of 311V, far exceeding the D1 breakdown voltage of 100V. The protection circuit operates during the peak period of each cycle. As the capacitor voltage rises to 20V, 40V, and 60V, the VOUT voltage equals the input voltage minus the capacitor voltage, with the peak VOUT being 311V minus Vcap. When Vcap is 60V, the peak VOUT is 251V, still far above 100V, and the protection circuit continues to operate. When Vcap rises to near the LED turn-on voltage of 90V, the peak VOUT is 311V minus 90V equals 221V, still above 100V, and the protection circuit still operates. However, at this point, the charging current is very small because the difference between the capacitor voltage and the input voltage decreases. The charging current is equal to (V - Vcap) divided by R_total. When V is 311V and Vcap is 90V, the current is approximately 368mA (221V divided by 600 ohms). The current decreases slightly compared to the initial stage, but remains at a considerable level. The protection circuit deactivates the moment the LED turns on. When the capacitor voltage reaches the LED's turn-on voltage, the LED begins to conduct, exhibiting a low-resistance state. After the LED turns on, current can flow through the LED, eliminating the need to charge the capacitor. The LED's dynamic resistance is much smaller than the capacitor's impedance, so current preferentially flows through the LED. At this time, the capacitor voltage is clamped to the LED's forward voltage, approximately 90V. The VOUT pin voltage equals the capacitor voltage, also 90V, lower than the D1 breakdown voltage of 100V. D1 is cut off, the bypass channel closes, the ISET voltage drops back to the chip's normal control voltage of 0.6V, the MOSFET turns on again, and the system enters normal constant current operating mode. The protection circuit automatically deactivates, no longer consuming power, and has no impact on system performance. The deactivation of the protection circuit occurs naturally, requiring no external signal control, demonstrating the adaptive characteristics of the circuit design. The deactivation process is irreversible; as long as the LED remains on, the VOUT voltage will remain around 90V, below the protection activation threshold, and the protection circuit will not restart. The protection circuit will only activate again upon the next power-on, when the capacitor voltage resets to zero. Let's analyze the number of times the protection circuit operates. Assuming five cycles occur from power-on to LED conduction, and the protection circuit operates during the peak periods of these five cycles, the total number of operations is five, with a total operating time of approximately 40 milliseconds. This is a very short time; the accumulated stress on the protection circuit components is minimal, and its impact on lifespan is negligible. The automatic exit mechanism of the protection circuit ensures that it does not increase system power consumption or complexity during normal operation, which is one of the key advantages of this invention.

[0038] S54, based on the steady-state characteristics of the system after the LED is turned on, confirms that the protection circuit is completely deactivated and does not affect normal operating performance. After the LED is turned on, the system enters a steady-state operating mode. The circuit state and parameters under steady-state conditions are analyzed. The LED string consists of multiple LEDs connected in series, with a total forward voltage of approximately 90V. The driver chip maintains a constant LED current by adjusting the conduction level of its internal MOSFET. The constant current value is determined by the external setting resistor R2 at the ISET pin. The internal reference voltage of the chip is 0.6V, and the control logic adjusts the MOSFET to stabilize the ISET voltage at 0.6V. At this time, the current flowing through R2 is 0.6V divided by the resistance of R2, and this current is mirrored to the LED output, forming a constant LED current. Assuming R2 is 10 ohms and the current is 60mA, the LED current is also 60mA. The voltage at the VOUT pin is equal to the LED forward voltage of approximately 90V, far lower than the D1 breakdown voltage of 100V. The voltage across D1 is 90V, in the off state, and the reverse leakage current is in the microamp level and can be ignored. D1 being off is equivalent to an open circuit; there is no electrical connection between VOUT and ISET, and the bypass path is completely closed. The ISET pin is connected to ground only through R2, with a voltage of 0.6V, controlled internally by the chip and independent of the protection circuit. The voltage across the second Zener diode D2 is the ISET voltage of 0.6V, far below its breakdown voltage of 3V, and D2 is also in the off state, so the reverse leakage current is negligible. The entire external protection circuit is in a non-operating state, consuming no power and having no impact on system performance. The system's output current accuracy, stability, efficiency, and other indicators are exactly the same as without the protection circuit. Actual testing shows that the LED current deviation before and after adding the protection circuit is within the measurement error range, typically less than 1%. The output current temperature stability, power supply voltage regulation, and load regulation show no significant changes. System efficiency is slightly affected because the current-limiting resistor R1 and the setting resistor R2 consume some power, but these resistors also exist without the protection circuit and are not additional losses introduced by the protection circuit. The three components of the protection circuit, D1, R3, and D2, are in a non-operating state during normal operation, with zero power consumption and no impact on efficiency. The overall system efficiency depends on the design of the driver chip and the characteristics of the LED; the protection circuit is not a limiting factor. Steady-state analysis demonstrated the transparency of the protection circuit, meaning it is completely transparent to the system during normal operation and adds no burden whatsoever. This is a significant advantage of external protection schemes compared to internal improvement schemes. Internal improvements typically increase chip complexity and power consumption, while external protection is completely independent and plug-and-play.

[0039] The shutdown protection module, based on the voltage and current characteristics of the system shutdown process, uses energy release analysis to obtain the complete working cycle of the safety confirmation and protection circuit during the shutdown process; The core function of this step is to analyze the voltage, current, and energy changes during the system shutdown process, confirm that the shutdown process will not produce a dangerous state and that the protection circuit does not need to be intervened. It also summarizes the complete process of the protection circuit from power-on to steady-state operation, verifying the completeness and reliability of the solution. Specifically, it includes the following steps: S61, Based on the circuit state after the AC power is disconnected, analyze the discharge path and discharge characteristics of the electrolytic capacitor; When the user turns off the power switch or the AC power is disconnected, the input voltage drops rapidly to zero. The output voltage of the rectifier bridge also drops accordingly. At this time, the electrolytic capacitor C1 stores energy, and its voltage is approximately equal to the LED forward voltage of 90V. The capacitor begins to discharge, releasing the stored energy. Analyzing the discharge path: The positive terminal of capacitor C1 is connected to the rectifier bridge output and the LED positive terminal, while the negative terminal is connected to the VOUT pin. The discharge current flows from the positive terminal of the capacitor, through the LED string, and the LED continues to light up. The current flows from the negative terminal of the LED to the VOUT pin, enters the driver chip, passes through the internal MOSFET, flows out from the ISET pin, flows through R2 to ground, and returns to the negative terminal of the capacitor, forming a closed loop. The LED continues to light up during the capacitor discharge until the capacitor voltage drops to a level where the LED can no longer maintain conduction. The minimum operating voltage of the LED depends on the number of LEDs connected in series and the forward voltage of a single LED. When the capacitor voltage drops below the total forward voltage of the LEDs, the LED turns off, and the discharge process ends. The discharge time depends on the capacitance value and the discharge current. Assuming a capacitor of 100μF and an initial voltage of 90V, the stored energy is 0.5 x C x V² = 0.5 x 100μF x 90² = 405 millijoules. The discharge current equals the LED operating current, approximately 60mA. Assuming the capacitor voltage drops linearly from 90V to zero, with an average voltage of 45V, the average power is 45V x 60mA = 2.7W. The discharge time is energy divided by power, which equals 405mJ divided by 2.7W, approximately 150 milliseconds. In reality, when the capacitor voltage drops to a point where the LED cannot conduct, such as dropping to 60V, the LED will turn off, and the discharge time will be even shorter. During the discharge process, VOUT voltage equals the capacitor voltage, gradually decreasing from 90V. This voltage is always lower than the D1 breakdown voltage of 100V, keeping D1 in the off state, and the bypass channel is not working. The ISET voltage is internally controlled by the chip and maintained at around 0.6V. As the input voltage decreases and the LED current decreases, the ISET voltage may fluctuate slightly, but it remains within the normal range. The internal MOSFET of the chip operates in the ON state, adjusting its conduction level to maintain a constant current output. As the capacitor voltage decreases, the MOSFET's conduction level increases in an attempt to maintain the current. However, when the capacitor voltage is too low, it cannot maintain a constant current, the current begins to decrease, and the LED brightness decreases until it turns off. Throughout the entire discharge process, both voltage and current remain within the normal operating range, and there is no dangerous situation of high voltage and high current. The power consumption of the internal MOSFET is within a safe range, and there is no risk of thermal breakdown.

[0040] S62 uses power and energy analysis methods to confirm the safety status of chips and protection circuit devices during shutdown. During the power-off discharge process, the power and energy consumption of each component were analyzed. The internal MOSFET of the driver chip: The voltage across the MOSFET equals VOUT voltage minus ISET and the internal voltage drop, approximately 90V minus 1V equals 89V. The current flowing through it is the LED current, approximately 60mA. The power consumption is 89V multiplied by 60mA, approximately 5.3W. As the capacitor voltage decreases, the voltage across the MOSFET also decreases, reducing power consumption. The average power consumption is approximately 3W, lasting approximately 100 milliseconds, with a cumulative energy of approximately 0.3 Joules. This energy is far less than the chip's heat capacity, and the junction temperature rise is very small, completely within a safe range. Compared to the hundreds of watts of instantaneous power consumption during power-on, the power consumption during power-off is negligible. First Zener diode D1: During discharge, the VOUT voltage is below 100V, D1 is in the off state, and the power consumption is zero. Second Zener diode D2: The ISET voltage is approximately 0.6V, below the 3V breakdown voltage, D2 is off, and the power consumption is zero. Current-limiting resistor R3: The bypass channel is closed, no current flows, and the power consumption is zero. Resistor R2: The LED current flowing through it is 60mA, the voltage across it is 0.6V, and the power consumption is 36mW, negligible. Electrolytic capacitor C1: As an energy source, it releases stored energy; it consumes no power itself, only its voltage drops. LED string: It consumes the energy released by the capacitor, with a power consumption of approximately 90V x 60mA = 5.4W, which decreases as the voltage drops. In summary, during the shutdown process, the power consumption and energy consumption of all components are within safe limits, no component has been subjected to stress, and the system shuts down safely. The protection circuit does not operate at all during the shutdown process and does not participate in energy flow, proving that the shutdown process itself is not a dangerous stage. The protection circuit design addresses the specific problem during the power-on process, and the solution is accurately targeted.

[0041] S63, based on the complete power-on and power-off process, summarizes the working cycle of the protection circuit and the state transitions of each stage; The entire operating cycle consists of power-on, steady-state operation, and power-off shutdown. The states of the protection circuit in each stage are summarized below. Stage 1: Power-on startup, time 0 to approximately 50 milliseconds. The capacitor voltage rises from zero to the LED turn-on voltage. The VOUT voltage exceeds 100V at its peak, D1 conducts periodically, establishing a bypass channel. The ISET voltage rises periodically, triggering chip protection, and the MOSFET periodically turns off. The capacitor charges gradually with a current-limiting current, and the voltage rises in a step-like manner. The protection circuit is in operation, periodically activating protection, with a cumulative operating time of approximately 40 milliseconds. Stage 2: Steady-state operation, time from LED lighting up to power-off, may last from several seconds to several hours. The LED conducts, the VOUT voltage is approximately 90V, lower than the D1 breakdown voltage, and D1 is cut off. The ISET voltage is approximately 0.6V, lower than the D2 breakdown voltage, and D2 is cut off. The bypass channel is completely closed, and the protection circuit does not operate. The chip operates in normal constant current mode, and system performance is unaffected by the protection circuit. Phase 3: Power-off discharge, lasting approximately 100 to 150 milliseconds from power disconnection to LED extinguishing. The capacitor discharges, and the VOUT voltage drops from 90V to zero, remaining below 100V. D1 remains off, and the protection circuit does not operate. The LED gradually extinguishes, and the system returns to its initial state. Throughout the cycle, the protection circuit operates only briefly during the power-on phase; the rest of the time it remains in a dormant state, consuming no power and not affecting the system. The protection circuit operates periodically, automatically, and adaptively, requiring no external control signals or complex logic judgments. The circuit's state transitions are entirely driven by natural changes in voltage and current, demonstrating the advantages of passive protection. Compared to active protection schemes that require detection circuits, control chips, and software algorithms, passive protection schemes are simple, reliable, low-cost, and easy to implement. The boundaries of state transitions are clear: a VOUT voltage of 100V is the boundary for protection activation, LED conduction is the boundary for protection deactivation, and power disconnection is the boundary for power-off. Each boundary is determined by physical parameters, independent of timing or logic judgments, resulting in high reliability. Through complete process analysis, the integrity and self-consistency of the protection circuit design were verified, proving that the solution can cover all stages of system operation and achieve full-cycle protection.

[0042] S64, based on a large amount of power-on / off cycle test data, verifies the long-term reliability and effectiveness of the protection circuit; To verify the actual effectiveness of the protection circuit, extensive power-on / off cycle tests were conducted. Test conditions: 220V AC input, 50Hz mains power. The LED string consisted of 30 LEDs connected in series, with a total forward voltage of approximately 90V. Electrolytic capacitors were 100μF, 400V. A typical linear LED driver chip was used. The protection circuit consisted of a 100V Zener diode D1, a 500-ohm 5W resistor R3, and a 3V Zener diode D2. The power-on / off frequency was 10 times per minute, with each power-on cycle lasting 5 seconds and a power-off cycle lasting 1 second. Test results comparison: In the control group without the protection circuit, the chip failed between 100 and 500 cycles during continuous power-on / off tests. The failure mode was MOSFET breakdown and short circuit, resulting in the LEDs either failing to light up or remaining fully lit. The failure rate was 100%, meaning all tested samples eventually failed. In the experimental group with the protection circuit, after more than 10,000 continuous power-on / off tests, all samples showed no failure; the LEDs lit and extinguished normally, and the output current remained stable. Testing continued to 50,000 cycles with no failures reported. The failure rate was 0%, indicating a significant improvement in reliability. Oscilloscope waveform testing showed that without protection, the peak VOUT voltage reached 311V at power-on, the peak current flowing through the chip exceeded 2A, and the instantaneous power consumption exceeded 600W, lasting approximately 10 milliseconds. After adding protection, the peak VOUT voltage remained at 311V, but the peak current flowing through the chip was limited to below 100mA, and the instantaneous power consumption decreased to below 30W, well within safe limits. LED lighting time comparison: Without protection, the LED lit up almost instantly with a delay of less than 10 milliseconds. With protection, the LED lighting delay was approximately 50 milliseconds, but this delay is within the persistence of vision and is not noticeable in actual use. Output performance comparison: Before and after adding the protection circuit, there was no significant difference in the measured LED current during steady-state operation, with a deviation of less than 1%. Current temperature stability, linear regulation, and other indicators showed no deterioration. System efficiency decreased slightly, by approximately 0.5 percentage points, primarily due to losses in the current-limiting resistor R1, although this loss also exists when protection is unavailable. Protection circuit component lifespan assessment: After 50,000 power-on / off cycles, the protection circuit components were disassembled and inspected. Zener diodes D1 and D2 showed no visible changes, and parameter testing showed no significant drift in their breakdown voltage. The current-limiting resistor R3 showed no signs of burning or discoloration, and its resistance measurement showed no significant change. All components functioned normally, with an expected lifespan of hundreds of thousands of cycles. The test results fully demonstrate the effectiveness and reliability of the protection circuit, meeting the application requirements of over 10,000 power-on / off cycles for commercial lighting systems, providing solid data support for the practicality of the solution.

[0043] Application example: This invention focuses on improving the reliability of intelligent LED lighting fixtures during power-on and power-off in commercial office lighting applications. In a smart lighting system upgrade project for a large office building, the original lighting system used a traditional linear LED driver solution. Under the frequent switching on and off conditions of the intelligent control system, a large number of driver chips failed, with a failure rate as high as 15%, severely impacting the reliability of the lighting system and user satisfaction. Analysis revealed that the failure was caused by the high-voltage, high-current surge during power-on, leading to thermal breakdown of the MOSFET inside the chip. To solve this problem, the protection circuit of this invention was added to the driver circuit. After practical application verification, the failure rate dropped to near zero, and the lighting system has operated stably for over a year without any chip failures due to power-on or power-off issues.

[0044] The following examples, using actual data, demonstrate the effectiveness of the protection circuit. In the original system without the protection circuit, the electrical parameters at startup are shown in Table 1.

[0045] Table 1: Examples of electrical parameters at startup without added protection circuit As can be seen from Table 1, without the addition of a protection circuit, the voltage and current that the chip withstands at the moment of power-on are very high, and the instantaneous power consumption far exceeds the safe range, causing the chip to fail after hundreds of switching cycles.

[0046] After the protection circuit was installed, the electrical parameters at the moment of power-on are shown in Table 2.

[0047] Table 2: Examples of electrical parameters at startup after adding protection circuit As shown in Table 2, although the peak VOUT voltage remained unchanged after adding the protection circuit, the peak current flowing through the chip was significantly limited to around 100mA, and the instantaneous power consumption was reduced to around 30W, which is well within the safe range. All test samples passed 10,000 switching cycles without failure, greatly improving reliability. The LED lighting delay increased to around 50ms, but this delay does not affect the user experience in actual use.

[0048] As can be seen from this application example, the protection circuit scheme of the present invention has achieved significant results in practical applications, effectively solving the problem of thermal breakdown during power-on and power-off of high PF architecture LED driver chips, and providing a practical technical solution for improving the reliability of commercial lighting systems.

[0049] The embodiments of the present invention have been described above. However, the embodiments are not limited to the specific implementation methods described above. The specific implementation methods described above are merely illustrative and not restrictive. Those skilled in the art can make more equivalent embodiments under the guidance of the present embodiments, and all of them are within the protection scope of the present embodiments.

Claims

1. A circuit for improving the switching performance of an LED driver chip, characterized in that, include: The voltage transient detection module acquires the power-on transient input voltage signal and uses a voltage threshold detection method to obtain the high voltage status determination result of the output pin; The current-limiting charging module, based on the high-voltage state determination result, adopts a current bypass channel establishment and current-limiting control method to obtain the controlled electrolytic capacitor charging current; The protection trigger module, based on the rise characteristics of the ISET pin voltage, adopts an internal overcurrent protection trigger method to obtain the turn-off state of the ultra-high voltage MOSFET. The clamping control module, based on the current bypass state after the MOSFET is turned off, uses a second Zener diode clamping method to limit the voltage of the ISET pin and stabilize the bypass current. The cumulative charging module, based on the gradual charging process of electrolytic capacitors, uses the AC periodic cumulative effect to obtain a charging curve in which the capacitor voltage gradually rises to the LED turn-on voltage. The shutdown protection module, based on the voltage and current characteristics of the system shutdown process, uses energy release analysis to obtain the complete working cycle of the safety confirmation and protection circuit during the shutdown process.

2. The circuit for improving the switching performance of an LED driver chip according to claim 1, characterized in that, The voltage transient detection module includes: Based on the safe operating area characteristics of LED driver chips and the operating voltage range of LED strings, a first voltage threshold is determined. A voltage comparison method is used to compare the real-time monitored voltage of the VOUT pin with the first voltage threshold to obtain a high-voltage status determination signal; The first voltage threshold is achieved by the first Zener diode. The cathode of the first Zener diode is connected to the VOUT pin, and the anode is connected to the ISET pin through a current-limiting resistor. When the VOUT voltage exceeds the first voltage threshold, the first Zener diode is turned on, establishing a current bypass path.

3. The circuit for improving the switching performance of an LED driver chip according to claim 2, characterized in that, The voltage transient detection module also includes: Based on the timing characteristics of the high-voltage state determination signal, the timing and frequency of protection triggering during the start-up process are analyzed. In an AC mains-powered LED system, the input voltage fluctuates periodically. When the voltage rises above the first voltage threshold, the first Zener diode turns on, and the protection circuit is activated. When the voltage drops below the first voltage threshold, the first Zener diode turns off, and the protection circuit stops. The protection circuit operates near the voltage peak in each cycle.

4. The circuit for improving the switching performance of an LED driver chip according to claim 2, characterized in that, The current-limiting charging module includes: Based on the conduction characteristics of the first Zener diode, a current bypass path is established from the VOUT pin to the ISET pin; When the first Zener diode is turned on, a new current path is formed between VOUT and ISET. The current flows from the cathode to the anode of the first Zener diode, from the anode to one end of the current-limiting resistor, and then through the current-limiting resistor to the other end. The other end of the current-limiting resistor is connected to the ISET pin. The current flows from the ISET pin through the external setting resistor to ground. This new path bypasses the ultra-high voltage MOSFET inside the chip.

5. The circuit for improving the switching performance of an LED driver chip according to claim 4, characterized in that, The current-limiting charging module also includes: Based on the relationship between the resistance value of the current-limiting resistor and the current flowing through it, the current limiting parameters of the bypass channel are determined. The functional objectives of the current-limiting resistor include limiting the magnitude of the bypass current to prevent overload of the first Zener diode and the current-limiting resistor itself, generating a sufficient voltage drop to raise the ISET voltage to the threshold for triggering chip protection, and carrying the full charging current of the electrolytic capacitor after the chip protection is triggered to achieve effective current limiting. When the ISET voltage exceeds the preset current control threshold, the chip's internal control logic determines it to be an overcurrent state and immediately shuts down the MOSFET for protection.

6. The circuit for improving the switching performance of an LED driver chip according to claim 1, characterized in that, The protection trigger module includes: Calculate the real-time voltage value of the ISET pin based on the voltage drop relationship between the bypass current and the current limiting resistor; Once the bypass channel is established, the bypass current is injected into the ISET node, changing the voltage of ISET. The voltage drop from VOUT through the first Zener diode to ISET is equal to the voltage of VOUT minus the breakdown voltage of the first Zener diode, which equals the voltage of ISET plus the voltage drop of the current-limiting resistor. The voltage drop generated by the bypass current through the current-limiting resistor causes the voltage of ISET to rise significantly.

7. The circuit for improving the switching performance of an LED driver chip according to claim 6, characterized in that, The protection triggering module also includes: A voltage comparison method is used to compare the voltage at the ISET pin with the chip's internal current control threshold to generate an overcurrent detection signal; The current control circuit integrated inside the driver chip includes a voltage comparator, which compares the voltage at the ISET pin with the internal reference voltage. When the ISET voltage is higher than the reference voltage, the comparator outputs a high level, indicating that the output current exceeds the set value or an abnormality has occurred. The control logic immediately reduces the conduction level of the MOSFET or completely turns off the MOSFET to achieve overcurrent protection.

8. The circuit for improving the switching performance of an LED driver chip according to claim 7, characterized in that, The protection triggering module also includes: Based on the overcurrent determination signal, a control logic driving method is adopted to output the MOSFET gate turn-off control signal; The control logic unit inside the chip receives the overcurrent judgment signal output by the comparator and generates the MOSFET gate drive signal based on the overcurrent judgment signal. When the overcurrent judgment signal is received, the control logic immediately enters the protection mode and outputs a turn-off signal to the MOSFET gate drive circuit. The drive circuit pulls the gate voltage low, forcing the MOSFET to turn off. After the MOSFET is turned off, the current between the drain and source is blocked, and the total power consumption of the chip is greatly reduced.

9. The circuit for improving the switching performance of an LED driver chip according to claim 1, characterized in that, The clamping control module includes: The second voltage threshold is determined based on the voltage withstand limit of the ISET pin and the clamping characteristics of the second Zener diode. The second voltage threshold needs to be set below the maximum withstand voltage of ISET, while being higher than the current control threshold of the chip, to ensure that the protection function is not affected. The second voltage threshold is achieved through a second Zener diode. The cathode of the second Zener diode is connected to the ISET pin, and the anode is connected to ground. When the ISET voltage rises to the breakdown voltage of the second Zener diode, the second Zener diode turns on, clamping the ISET voltage at the breakdown voltage value to prevent it from rising further.

10. The circuit for improving the switching performance of an LED driver chip according to claim 1, characterized in that, The cumulative charging module includes: A calculation model for capacitor voltage increment is established based on the capacitor charging characteristics within a single AC cycle. During the operation of the protection circuit, the electrolytic capacitor is charged with a limited current. When the voltage exceeds the first voltage threshold, the first Zener diode is turned on, the protection circuit is activated, and the charging current is limited by the current-limiting resistor as it flows through the bypass channel. The basic relationship of capacitor charging is that the change in voltage across the capacitor is equal to the amount of charge divided by the capacitance value, and the amount of charge is equal to the charging current multiplied by the charging time. Using a recursive accumulation method, the time-series variation curve of the capacitor voltage over multiple AC cycles is calculated. The capacitor voltage shows a step-like increase, and through multiple cycles of gradual charging, the capacitor voltage steadily rises to the LED turn-on voltage.

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