System and method for a high-frequency filter

Abstract

Circuit diagram, including: a filter circuit with a first port (①) configured to be coupled to an antenna, a second port (②) and a third port (③), wherein the filter circuit comprises the following: a first passive filter (302), a second passive filter (304), a first power amplifier (346) with an output coupled to the first passive filter (302), a second power amplifier (348) with an output coupled to the second passive filter (304), a first coupler (300) with an input port coupled to the first port (①), an isolated port coupled to the second port (②), a first phase-shifted port coupled to the first passive filter (302), and a second phase-shifted port coupled to the second passive filter (304), wherein the first coupler (300) comprises a -3dB hybrid with quadrature output, and a division network (312) with a first output coupled to an input of the first power amplifier (346), a second output coupled to an input of the second power amplifier (348), and an input coupled to the third port(③).

Description

[0001] The present application claims the benefits of the preliminary US patent application No. 62 / 066,201 filed on October 20, 2014, which is hereby incorporated into the present application by reference. AREA OF INVENTION

[0002] The present disclosure relates generally to an electronic device and more specifically to a system and method for a high-frequency filter. GENERAL STATE OF THE ART

[0003] In many RF systems, such as portable wireless devices, more than one receive or transmit frequency can be active simultaneously in a single radio. If the relevant frequency bands are far apart and / or if the frequency bands are processed with different gains, different frequency channels are separated in the frequency domain and processed in separate signal paths. Many systems today require flexible frequency planning and the simultaneous processing of more than two channels, which makes the design of a fixed-frequency demultiplexer filter (with n frequency bands) challenging. For example, LTE, the 4th generation mobile communications standard, uses carrier aggregation techniques that utilize multiple channels, processing up to three receive (RX) channels and one transmit (TX) channel simultaneously.In the next-generation LTE standard, and very likely in other standards as well, multiple RX paths of different frequencies and multiple TX paths of different frequencies will operate simultaneously. Accordingly, assuming that the TX is already filtered out, the user equipment (UE) separates up to three RX channels, which are divided within the frequency domain.

[0004] Well-known filter circuits and RF transmit / receive systems are used, for example, in the US 2013 / 0 321 097 A1, the US 2013 / 0 201 882 A1, the US 2014 / 0 038 532 A1, the described in US 2012 / 0 281 597 A1, US 2012 / 0 256 702 A1 or US 8 537 723 B2. SUMMARY

[0005] The present invention is based on the objective of creating improved circuits for filters and RF transmit / receive systems whose complexity is reduced.

[0006] This problem is solved by the subject matter of the independent patent claims. BRIEF DESCRIPTION OF THE DRAWINGS

[0007] For the purpose of a more comprehensive understanding of the present invention and its advantages, reference is now made to the following descriptions in conjunction with the accompanying drawings. These show: Fig. 1a a conventional RF receiver that uses switched clusters of diplexer and triplexer filters at the antenna to the LNA interface and switched clusters of diplexer and triplexer filters after the LNA, and Fig. 1b an RF system that uses switched clusters of diplexer and triplexer filters at the antenna to the LNA interface and a triplexer after the LNA; Fig. 2a and Fig. 2b a single insulated filter structure according to the invention and Fig. 2c and Fig. 2d a single tuned isolated filter structure according to the invention, followed by a conventional tuned diplexer; Fig. 3a-3b two cascaded, matched insulated filter structures according to the invention, Fig. 3c n - 1 series-connected, tuned, isolated filter structures according to the invention and Fig. 3d a circuit utilizing a tuned diplexer together with tuned isolated filter structures according to the invention; Fig. 4a to 4j different implementations of tuned isolated filter structures according to the invention; Fig. 5a-5g different implementations of tuned isolated filter structures according to the invention with a bridging mode and a corresponding switching diagram; Fig. 6a and Fig. 6b Simulation schemes for isolated filter structures according to the invention; Fig. 7a-7d s-parameters for an isolated filter structure according to the invention in a low-impedance reflection mode; Fig. 8a-8d s-parameters for an isolated filter structure according to the invention in a high-impedance reflection mode; Fig. 9a-9c a block diagram of an RF system according to the invention and corresponding s-parameter diagram curves of the isolated filter structure according to LNA B; Fig. 10a-10c a scheme of an isolated filter structure according to the invention; Fig. 11 a block diagram of an RF system; Fig. 12 a block diagram of another RF system; Fig. 13a-13d s-parameter diagram curves of the RF system of Fig. 12; Fig. 14a-14e a scheme of an isolated filter structure according to the invention and corresponding s-parameter diagram curves; Fig. 15a-15i a scheme of a SAW filter based on a bridgeable isolated filter structure according to the invention and corresponding s-parameter diagram curves; Fig. 16a-16b exemplary layout diagrams of isolated filter structures according to the invention; Fig. 17a an RF system that uses switched clusters of diplexer and triplexer filters at the antenna to the LNA interface and an isolated bandpass / bandstop filter as well as a diplexer after the LNA; and Fig. 17b an RF system that uses switched clusters of diplexer and triplexer filters at the antenna to the LNA interface and a tuned isolated bandpass / bandstop filter as well as a diplexer after the LNA; Fig. 18 an RF system according to the invention; Fig. 19a and Fig. 19b RF systems which may be directed towards transmitting and receiving for carrier aggregation; Fig. 20a-20c RF systems according to the invention, configured to implement carrier aggregation; Fig. 21a-21c a further RF system according to the invention, configured to implement carrier aggregation; and Fig. 22a-22d Frequency combiners according to the invention.

[0008] Matching reference numerals in different figures generally refer to matching parts unless otherwise indicated. The figures serve to clearly illustrate the relevant aspects of the preferred embodiments and are not necessarily to scale. To further clarify certain embodiments, a number in a figure may be followed by a letter indicating variants of the same structure, material, or process step. DETAILED DESCRIPTION ILLUSTRATIVE FORMS OF EXECUTION

[0009] The manufacture and use of the currently preferred embodiments are explained in detail below. However, it should be understood that the present invention provides many applicable inventive concepts that can be implemented in a wide variety of specific contexts. The specific embodiments described are merely examples of specific ways to manufacture and use the invention and do not limit its scope.

[0010] The present invention is described with regard to preferred embodiments in a specific context, namely a system and method for a flexible high-frequency filter for use in multiband RF systems. Further embodiments can, for example, be applied to other RF systems.

[0011] Tuning the passband of each filter and maintaining the phase-amplitude relationship of all filters, so that no low impedances are provided for the filters within their passbands, is extremely difficult to accomplish.

[0012] Accordingly, conventional solutions such as the one in Fig. Figure 1a illustrates an RF reception path with 100 filter banks, each representing a use case. In each use case, the filters are optimized with respect to phase and size so that the filters do not interfere with each other. For another use case, a new group of filters is combined in phase and amplitude so that they do not interfere with each other. Therefore, an arrangement of use case combinations utilizing a bank of triplexers or diplexers, as shown in Figure 1a, is used. Fig. As shown in Figure 1a, switching is done via RF switches to meet the requirements. This causes hardware redundancy when a single band is used in different applications. As a result, the RF equipment is large and expensive.

[0013] As in Fig. As shown in Figure 1a, a pair of antennas 102 and 104 is coupled to an antenna switch 106, followed by a first filter bank 108 containing diplexers and one or more triplexers. The outputs of the first filter bank 108 are routed to three LNAs: a low-band LNA A, a mid-band LNA B, and a high-band LNA C, via RF switches 110, 112, and 114 at the inputs and RF switches 116, 118, and 120 at the outputs of these LNAs. The RF switches 116, 118, and 120 are followed by a second filter bank 122, which, similar to the first filter bank 108, contains diplexers and triplexers, the outputs of which are selected by another RF switch 124. As shown in Figure 1a, the LNAs are connected to three LNAs: a low-band LNA A, a mid-band LNA B, and a high-band LNA C. Fig. As can be seen in 1a, the first bank of diplexers is similar to the second bank of diplexers, thus causing redundancy.

[0014] Fig. Figure 1b illustrates a system 130 in which a first bank of diplexers / triplexers 132 is coupled to the antennas 102 and 104 via the antenna switch 106 and is routed via RF switches 110, 112 and 114 to a low-band LNA A, a mid-band LNA B and a high-band LNA C, followed by a triplexer 134 instead of a redundant filter bank.

[0015] Although the system avoids Fig. 1b the use of a redundant filter bank, but the design of the triplexer is associated with several design problems, including input loading / mismatch, gain drop in carrier aggregation (CA) mode, and RX noise from active LNAs in CA mode. Ideally, the filter components of the triplexer are phase-shifted relative to each other via the phase ΦA, ΦB, ΦC so that they do not load each other. Furthermore, such multiplexing after the LNAs requires a triplexer with a very high Q factor, the design of which may be particularly challenging if the triplexer is to be tunable to support a flexible frequency scheme. In one embodiment, a tunable bandpass filter (BPF) and a bandstop filter (BSF) with a high Q factor are implemented using an isolated filter core, for which a simple example is given in Fig. Figure 2a illustrates this. As shown, a bandpass / bandstop filter 202 with a first transfer function of H1(s) at a first output and with a second transfer function of 1-H1(s) at a second output can be implemented using a coupling structure 204 such as a -3 dB hybrid with a quadrature output, which is coupled to a BPF filter 206 with a transfer function of H1(s). As shown in Fig. As shown in Figure 2b, due to the reflection ΓH1(s), the transfer function H1(s) from Port 1 to Port 3 exhibits a bandpass characteristic, and the transfer function 1-H1(s) from Port 1 to Port 2 exhibits a bandstop characteristic. To simplify the equations to 1-H1(s), the coupling structure 204 is assumed to be without attenuation, and the reflection coefficient Γ is assumed to be 1 for total internal reflection.

[0016] In one embodiment, a tunable triplexer with a high Q-factor is implemented using a hybrid bandpass filter (BPF) and bandstop filter (BSF) with an isolated filter core and a conventional diplexer with high-pass characteristic H2(s) and low-pass characteristic H3(s), for which in Fig. Figure 2c illustrates a simplified example. As shown, a bandpass / bandstop filter 210 with a first transfer function of H1(s) at a first output and a second transfer function of 1-H1(s) at a second output can be implemented using a coupling structure such as a -3 dB hybrid with a quadrature output 212 coupled to a BPF 214 with a transfer function of H1(s). As shown in Fig. As shown in Figure 2d, due to the reflection ΓH1(s), the transfer function H1(s) from Port 1 to Port 3 exhibits a bandpass characteristic, and the transfer function 1-H1(s) from Port 1 to Port 2 exhibits a bandstop characteristic. To simplify the equations to 1-H1(s), the coupling structure 212 is assumed to be without attenuation, and the reflection coefficient Γ is assumed to be 1 for total internal reflection.

[0017] In one embodiment, port 2 of the tuned filter core is connected to a conventional diplexer with a high-pass characteristic from port 2 to port 4 and a low-pass characteristic from port 2 to port 5. As in Fig. As described in Figure 2d, the entire cascaded transmission from Port 1 to Port 5 follows a low-pass filter with its transfer function [1-H1(s)]*H3(s) and the corresponding high-pass filter characteristic from Port 1 to Port 4 with its transfer function [1-H1(s)]*H2(s). The transmission from Port 1 to Port 3 follows a band-pass function H1(s). The required diplexer filter functions H2(s) and H3(s) can be very relaxed, and yet a very steep transition from LB(S51) to MB(S31) and a very steep transition from MB(S31) to HB(S41) can be achieved due to the contribution of the steep isolated band-stop transfer function 1-H1(s).

[0018] This technique reduces the number of mutually loading filters by one frequency band per isolation bandpass, allowing an n-plexer function to be implemented using an isolated core and an (n-1) plexer solution. Accordingly, the transmission from port 1 to port 3 is primarily defined by the filter transfer function H1(s), whereas the transmission from port 1 to port 2 is primarily defined by the reflection ΓH1(s). Embodiments utilizing such an isolated filter structure offer several advantages. For example, only one type of filter is required, simplifying the tuning of the arrangement, as the bandpass transfer function H1(s) and the bandstop transfer function 1-H1(s) automatically follow each other. Furthermore, the bandstop characteristic is less dependent on correct phase than in a conventional filter due to the isolated nature of the three-port structure in this embodiment.

[0019] In one embodiment, the transfer function H1(s) occurs from port 1 to port 3, and the transfer function (1-H(s)) occurs for the reflected signal from port 1 to port 2. The filter forming the passband / reflection plane can be implemented using a tuned filter. Further filtering can be added to path 2 and path 3 without causing a mismatch at the isolated filter core. This allows for the design of flexible filter banks for the simultaneous operation of multiple frequency bands. An example of such filtering is shown in Fig. Figure 3a shows two cascaded isolated bandpass / bandstop filters 220 and 222 with transfer functions H1(s), 1-H1(s) and H2(s), 1-H2(s) implemented using filter structures 224 and 226 according to the invention based on bandpass filters with transfer functions H1(s) and H2(s).

[0020] As in Fig. As shown in Figure 3a, port 2 of the first filter structure with an insulated core 224 is coupled to the first port of a second filter structure with an insulated core 226 to form a triplexer. Accordingly, the transfer function from port 3 to port 1 is the bandpass transfer function H1(s), and the transfer function from port 5 to port 1 is the filter function [1-H1(s)]*H2(s). However, due to the reflections ΓH1(s) and ΓH2(s) at the input of the two bandpass filters, the transfer function from port 4 to port 1 has two stopbands and a transfer function of [1-(1-H1(s))]*[1-H2(s))]. Amplitude diagrams of these transfer functions are shown in Fig. 3b illustrates this.

[0021] As in Fig. As shown in Figure 3c, additional filter structures with isolated core 230, 232 and 234 can be connected in series to form an (n-2) filter structure using filter structures with isolated core 236, 238 and 240 according to the invention.

[0022] As further in Fig. As shown in 3D, conventional filters can be added to each isolated filter core to form other filter structures (provided complexity and load criteria permit). As shown, port 2 of a first isolated filter structure 250, based on a bandpass filter function H1(s), is coupled to a conventional diplexer 242, based on a transfer function of H2(s) and H3(s), which divides the frequency content at port 2 into two frequency bands. In one example, the first isolated filter structure 250 removes a mid-band using the band-stop characteristic of the isolated filter structure, and the conventional diplexer 242 exhibits a flatter frequency response characteristic. The additional isolated filter structures 254 and 256, based on the transfer functions H2(s) and H3(s), further enhance the filter structure's performance. k-1 (s) and H n-1(s) can be used to isolate further frequency clusters in the high and low bands that are divided by the conventional diplexer. In one embodiment, the mid-band frequency content based on H1(s) is available at port 3 of the isolated filter structure 250.

[0023] Fig. 4a and Fig. Figure 4b illustrates two possibilities for implementing isolated filter structures according to the invention. For example, in Fig. 4a An isolated filter structure (A0) according to the invention is implemented using a coupler, such as a -3dB hybrid 300 with a quadrature output port and an isolation port, followed by phase-shifting elements φ2, two identical filters 302 and 304 or tuned filters with characteristics Hi(s), and then followed by the phase shifters 306 and 308 and a power divider, such as a 3dB combiner 310. In one embodiment, the phase shifters 306 and 308 and the 3dB combiner 310 can be implemented using a dual circuit, a multiphase filter network, or another circuit known in the field. The phase shifters φ2 represent additional phase shifts enclosed in the isolated filter structure.

[0024] In another embodiment, as in Fig. Figure 4b shows an isolated filter structure (B0) according to the invention using a first coupler, such as a -3dB hybrid 300 with a quadrature output port and an isolation port, followed by phase shift elements φ2, two identical filters 302 and 304 or tuned filters with the characteristics Hi(s) and then followed by a second coupler, such as a -3dB hybrid 300 with a quadrature output port and a closed isolation port.

[0025] In further embodiments, the insulated filter structures (A0) and (B0) according to the invention, which are in Fig. 4a and Fig. 4b are shown, furthermore as in Fig. The variants shown in Figures 4c-j can be modified. Variants (Ai) utilize dual-mode, multi-phase filter, or phase-shift networks, followed by a 3dB combiner (RX) or a 3dB divider (TX) to recombine the quadrature signals to provide a filtered signal path from port (1) to port (3) at RX or from port (3) to port (1) at TX; and variants (Bi) utilize second -3dB hybrids to provide a filtered signal path from port (1) to port (3) at RX or from port (3) to port (1) at TX.

[0026] For example, the input of an amplifier, such as an LNA 320, can be compared with reference to the in Fig. 4c illustrated isolated filter structure A1 and the one in Fig. Figure 4d illustrates the isolated filter structure B1 being coupled to port 3. These variants A1 and B1 provide an RX path and are suitable for use as receivers. However, in some cases, port 1 may exhibit some sensitivity to source pulling.

[0027] In further embodiments, the output of an amplifier 330 is, with reference to the in Fig. 4e illustrated isolated filter structure A2 and the one in Fig. Figure 4f illustrates an isolated filter structure B2 coupled to Port 3. These variants A2 and B2 provide a TX path and an amplifier with a good noise figure. Such structures can be used when blocker signals from an antenna coupled to Port 1 cause intermodulation products, which are generated by the TX amplifier and fall within the RX band, returning to Port (1). Accordingly, the corresponding RX path (bandpass filter + LNA) for one band can be located at Port 2.

[0028] In one embodiment, the inputs of amplifiers 342 and 344 can be configured with reference to the inputs in Fig. 4g illustrated isolated filter structure A3 and the in Fig. Figure 4h illustrates the isolated filter structure B3 coupled to the bandpass filters 302 and 304. These variants A3 and B3 prevent source pulling of the RX amplifier, but due to the uncorrelated nature of the noise from both parallel amplifiers, they can introduce a certain degradation in the noise figure.

[0029] In a further embodiment, the outputs of amplifiers 346 and 348 can be configured with reference to the Fig. 4i illustrated isolated filter structure A4 and the one in Fig. Figure 4j illustrates the isolated filter structure B4 coupled to the bandpass filters 302 and 304. These variants A4 and B4 prevent load-pulling of the RX amplifier, but due to the uncorrelated nature of the noise from both parallel amplifiers, they can introduce some degradation of the noise figure. Such an embodiment can be used, for example, in systems where the effect of an antenna coupled to Port 1 on the standing wave ratio (SWR) is attenuated by the two TX amplifiers, so that the summed output power at Port 1 and the supply current of the TX amplifier do not vary as much with respect to the SWR as other solutions, such as variants A2 and B2. In one embodiment, the relative power supplied by the two power amplifiers changes with the change in the SWR at Port 1.The power delivered to Port 1 remains relatively or substantially constant. In some embodiments, a receiving circuit may be coupled to Port 2. For example, a duplexer or diplexer could be coupled to Port 2, followed by additional filtering and one or more LNA circuits. In one embodiment, the duplexer or diplexer divides the frequency content at the port into a band with higher frequencies than the passband of Hi(s) and another band with lower frequencies than the passband of Hi(s). It is understood that the reference to... Fig. The variants of isolated filter structures described in sections 4a-4j are merely examples of variants of isolated filter structures according to the invention. Other variants can be implemented in alternative embodiments.

[0030] In one embodiment, the above-described and in Fig. The variants illustrated in 4a-4j may be further modified to include a deactivation mode, such as a tristate or branch mode, as described in reference to Fig. 5a-5g illustrates this. Although the deactivation mode variants are applied to the variants A0 and B0 described above, deactivation modes can be applied to all variants described above, as well as to other execution variants not explicitly described here.

[0031] In one embodiment, the isolated bandpass / bandstop filter cores 302 and 304 can be bypassed when port 3 is unused, thereby improving the performance of the transmission from port 1 to port 2. In other words, bypassing or disconnecting the isolated filter cores 302 and 304 reduces the load on the transmission path from port 1 to port 2. For example, the isolated filter can be disconnected with the switches 2, and port 1 can be directly connected to port 2 via the switches SW1, as described in variant A0sw1. Fig. 5a and variant B0sw1 in Fig. Figure 5b shows that using the switch SW1 between Port 1 and Port 2 provides a low-loss path when Port 3 is not used.

[0032] In a further embodiment, the isolated filter core can be decoupled via the switches SW2, which are coupled between the input hybrids 300 and the isolated filter structures 302 and 304, as described with reference to variant A0sw2 in Fig. 5c and variant B0sw2 in Fig. Figure 5d illustrates this. In one embodiment, opening the switches SW2 creates a highly reflective impedance plane (i.e., an RF open) at the output of the hybrid structure 300, thereby generating a low-loss path over a wide frequency range. Variants A0sw2 and B0sw2 are particularly suitable for embodiments utilizing inductors, as opening the switches SW2 can prevent DC losses in these inductors.

[0033] The isolated filter core can also be decoupled via the SW1 switches, which are coupled to ground between the input hybrid and the isolated filter structures, as described in variant A0sw3. Fig. 5e and variant B0sw3 in Fig. Figure 5f shows that in one embodiment, closing the switches SW1 creates a highly reflective impedance plane (i.e., an RF short) at the output of the hybrid structure 300, thereby generating a low-loss path over a wide frequency.

[0034] Fig. 5g illustrates a table that shows the setting of the various switches with reference to the mode setting for the in Fig. 5a-5f illustrates and describes the embodiments described above.

[0035] Fig. 6a and Fig. Figure 6b illustrates simulation diagrams for variants A0 and B0. As shown, the first signal with frequency f1 has a frequency that falls within the passband of the bandpass filter Hi(s), and a second signal with frequency f2 has a frequency that falls within the stopband of the bandpass filter Hi(s). Accordingly, the frequency f1 is transmitted from Port 1 to Port 3 via the bandpass filter Hi(s). In contrast, the frequency f2 is reflected by the filter Hi(s) and transmitted to Port 2. In other words, Hi(s) is passband for frequency f1 and stopband for frequency f2.

[0036] During operation, the signal f1 applied to Port 1 is split into two quadrature signals by the left quadrature hybrid 300. These signals pass through the bandpass filters 302 and 304 and are recombined by the phase recombination block 310 or 312 (second quadrature hybrid on the right, or dual-phase filter / multiphase filter, or phase-shifter-splitter combination on the right) before being returned to Port 3. All non-f1 frequencies applied to Port 1 that lie outside the passband of filter Hi(s) are also split into quadrature signals. However, these signals are reflected at filter Hi(s) and reconstructed in phase at the left quadrature hybrid on Port 2. Accordingly, frequencies can be divided into a bandpass transition from Port 1 to Port 3 and an inverse passband / stopband transition from Port 1 to Port 2.

[0037] In various embodiments, Hi(s) can be implemented using a fixed-frequency filter or a tunable filter. Hi(s) can be a low-pass filter, such that the transition from Port 1 to Port 3 follows a low-pass characteristic curve, and the transition from Port 1 to Port 2 follows an inverse low-pass / high-pass characteristic curve. In another embodiment, Hi(s) can be a high-pass filter, such that the transition from Port 1 to Port 3 follows a high-pass characteristic curve, and the transition from Port 1 to Port 2 follows an inverse high-pass / low-pass characteristic curve. In yet another embodiment, Hi(s) can be a band-pass filter, such that the transition from Port 1 to Port 3 follows a band-pass characteristic curve, and the transition from Port 1 to Port 2 follows an inverse passband / stopband characteristic curve.In another embodiment, Hi(s) can be a bandstop filter, such that the transition from Port 1 to Port 3 follows a bandstop characteristic and the transition from Port 1 to Port 2 follows an inverse bandstop / bandpass characteristic. In all the above scenarios, the filter function Hi(s) and its inverse filter function follow each other by manipulating Hi(s).

[0038] In some embodiments, Port 1 and Port 2 are well-matched compared to traditional filters. Furthermore, Port 3 exhibits good matching compared to a traditional filter, for example, when a second hybrid is used instead of dual-circuit, multi-phase filter, or phase-delay elements and power dividers. In one embodiment, the bypass mode described above can also be added to achieve very good performance when sending to Port 3 is not desired or necessary.

[0039] Fig. Figures 7a-7d and 8a-8d illustrate transfer and reflection diagram curves of an isolated filter structure according to the invention, in order to illustrate its dependence on the common-mode phase rotation φ2. Fig. Figures 9a-9d illustrate the case of a low-impedance reflection mode when the hybrid is terminated by a very low impedance at the reflected frequencies f2, and Fig. Figures 8a-8d illustrate the case of a high-impedance reflection mode when the hybrid is terminated by a very high impedance at the reflected frequencies f2.

[0040] Fig. Figure 7a illustrates S21 and S31 in a low-impedance reflection mode. As shown, S21 exhibits a loss of approximately 1.5 dB at both high and low frequencies. Fig. 7b is a version of Fig. 7a with enlarged scale. Fig. Figure 7c illustrates a Smith chart showing the input reflection coefficient of the filter itself (Sin_Iin) and the input reflection coefficient of Port 3 (S33_lin). As can be seen from the Smith chart, the input reflection coefficient for Port 3 remains closer to the center of the Smith chart than the input reflection coefficient of the filter itself. Fig. Figure 7d illustrates the return loss at the filter itself (Sin_dB) and at Port 3 (S33_dB). As shown, the return loss for Port 3 is better than approximately 12 dB across all frequencies, indicating good broadband input matching.

[0041] Fig. Figure 8a illustrates S21 and S31 in a high-impedance reflection mode. As shown, S21 exhibits an attenuation of approximately 1.0 dB at most high and low frequencies. Fig. 8b is a version of Fig. 8a with enlarged scale. Fig. Figure 8c illustrates a Smith chart showing the input reflection coefficient of the filter itself (Sin_Iin) and the input reflection coefficient of Port 3 (S33_Iin). As can be seen from the Smith chart, the input reflection coefficient for Port 3 remains closer to the center of the Smith chart than the input reflection coefficient of the filter itself. Fig. Figure 8d illustrates the return loss shown at the filter itself (Sin_dB) and at Port 3 (S33_dB). As shown, the return loss for Port 3 is better than about 15 dB over most frequencies, indicating good broadband input matching. The two variations illustrate that the left-hand hybrid exhibits higher attenuation when terminated by low impedances. This is related to the relatively low Q-factor, which causes resistive attenuation of the series inductors and their mutual coupling within the hybrid when loaded with low impedance. This results in large inductor currents compared to the relatively small currents / low attenuation of the inductor under high impedance loads. Such coupling is maintained by the hybrid's capacitors, which typically have higher Q-factors than the inductors.Therefore, in some embodiments, the insertion loss can be refined and / or optimized by adjusting the phase φ2 of the dominant coupling mechanism so that the hybrid operates in a low-loss region. Theoretically, this could be due to the low-impedance load if the Q-factor of the inductor is much better than the Q-factor of the capacitor.

[0042] Fig. Figures 9a-9c illustrate an RF system according to the invention and corresponding power curves. Fig. Figure 9a illustrates an RF system 400 in which an isolated filter structure 408 according to the invention is coupled to the antennas 402 and 404 via a crossover 406 at port 1. The mid-band frequencies are passed through the passband of the isolated filter structure 408 via port 3 and amplified by LNA B. High and low frequencies taken from port 4 of the isolated filter structure 408 are split into a high band and a low band by a conventional diplexer 410, selected and filtered by various high-frequency crossovers 412, 414, 418, and 420 and the filter bank 416, amplified by LNAs A and C, and then recombined using a second conventional diplexer 422. Alternatively, the two conventional diplexers can be implemented using isolated filter structures according to the invention.The reassembled outputs of LNA A and C are combined with the mid-band output of LNA B using the isolated filter structure 424 according to the invention.

[0043] Fig. Figure 9b illustrates transmission diagram curves for S41 and S31, showing how the mid-band output of LNA B is selectively filtered with respect to the outputs of LNA A and B. Fig. 9c is a version of Fig. Figure 9b with an enlarged scale, showing that the mid-band response S13 has less than 1 dB attenuation compared to the high-band and low-band responses S14. It is understood that the transmit diagram curves of Fig. 9b and Fig. Figure 9c is only a specific example of the performance of a system according to the invention. The transfer responses of other circuits according to the invention may differ in form and magnitude.

[0044] Fig. Figure 10a illustrates a scheme of an implementation according to the invention of the in Fig. Figure 9a shows the first isolated filter structure 408. The first isolated filter structure 408 includes a first hybrid transformer 430 coupled to the input of the filter 432 and a second hybrid transformer 434 coupled to the output of the filter 432. In one embodiment, the filter 432 is implemented using resonators 440, 442, 444, 446, 448, 450, 452, and is coupled to the hybrid transformer 430 via the magnetic transformers 456 and 458, and to the hybrid transformer 434 via the magnetic transformers 460 and 462.

[0045] Fig. Figure 10b illustrates a schematic of an exemplary LC resonator 470, which serves as a resonator element for the in Fig. The resonators 440, 442, 444, 446, 448, 450, 452, and 454 shown in Figure 10a can be used. Resonator 470 includes capacitors 472 and 474 and inductor 476. Alternatively, resonator 470 can be implemented using structures such as SAW filters, BAW filters, FBAR filters, loosely coupled resonators, LC filters, tunable LC filters, microstrip filters, or other filter structures.

[0046] Fig. Figure 10c illustrates a schematic of a transformer-based circuit used to implement one or both of the features described in Fig. The isolated filter structures 408 and 424 illustrated in Figure 10a can be used. The hybrid 480 contains capacitors 482 and 484 and inductors 486 and 488. In some embodiments, the inductors 486 and 488 can be coupled to each other, for example, using a common magnetic core, and / or they can be coupled due to close proximity between the inductors 486 and 488. In some embodiments, the inductors 486 and 488 can be implemented on an integrated circuit using a spiral inductor transformer. In alternative embodiments, the isolated filter structures 408 and 424 can also be implemented using various -3dB hybrid structures known in the field as Fisher couplers, Collins couplers, or other circuits that provide quadrature signals of the same signal level and provide an isolation port. These include, for example, branch quadrature couplers or ring couplers.

[0047] Fig. Figure 11 illustrates an RF system 500 that includes a conventional pre-LNA filter cluster circuit 502, a conventional LNA cluster 504 and the diplexer 506 configured to couple with a radio. Fig. Figure 12 illustrates an RF system 510 according to the invention, comprising a conventional pre-LNA filter cluster circuit 502, a conventional LNA cluster 504, and a diplexer implemented using an isolated filter structure 512 according to the invention, wherein the output of the mid-band LNA is coupled to port 1 of the isolated filter structure 512 according to the invention, and the output of the diplexer of the conventional LNA cluster is coupled to port 2 of the isolated filter structure 512 according to the invention. Port 3 of the isolated filter structure 512 according to the invention is configured to be coupled to a radio device. In the embodiment of Fig. In section 12, the isolated filter structure is configured to have a narrow bandpass function, allowing the mid-band frequencies amplified by the mid-band LNA to pass through. The reflective ports of the isolated filter structure allow all remaining bands (i.e., the low band and the high band) to pass from port 2 to port 3, but cut the mid-band out of the output of the conventional LNA cluster 504. This results in less stringent requirements for the diplexer of the conventional LNA cluster 504.

[0048] Fig. Figures 13a-13b represent graph curves illustrating the performance of the isolated filter structure 512. Fig. Figure 12 illustrates how the filter resonators are implemented with grid SAW resonators. Fig. Figure 13a illustrates the insertion loss S31, which represents the passband signal path of the filter structure according to the invention, the insertion loss S32, which represents the inverse function of the passband signal path, and the insertion loss S65, which represents the transfer function of the filter element within the isolated filter structure. As shown, S31 exhibits only an attenuation of about 1.3 dB in addition to the filter response itself. Fig. Figure 13b illustrates the insertion losses S32 and S31 with an extended scale. Fig. Figure 13c illustrates a graph of the return loss S22 for the filter with the hybrid and the return loss S55 for the filter without the hybrid. As shown by Fig. As can be seen in Figure 13c, the SAW filter with the hybrid has a better return loss characteristic than the SAW filter without the hybrid. Fig. Figure 13d illustrates a Smith diagram showing the reflection coefficients S22 and S33 for Port 2 and Port 3 respectively.

[0049] Fig. Figure 14a illustrates an isolated filter 550 according to the invention, in which the filter is implemented using standard SAW filters 552 and 554 in series with transmission lines coupled between two hybrids 556 and 558. Fig. Figure 14b illustrates the insertion loss S31, which represents the passband signal path of the isolated filter structure 550 according to the invention, and the insertion loss S32, which represents the inverse function of the passband signal path. Fig. Figure 14c illustrates the insertion losses S32 and S31 with an extended scale. Fig. Figure 14d illustrates a diagram curve of the return loss S33 for the filter with the hybrid and the return loss S55 for the filter without the hybrid. Fig. Figure 14e illustrates a Smith chart showing the reflection coefficient S33 for port 3. In alternative embodiments of the present invention, filters 552 and 554 can be implemented using any piezoacoustic, LC, or cavity filter known in the field.

[0050] Fig. Figure 15a illustrates an isolated filter structure 570 according to the invention, which includes RF switches for use when bridging the filter path of the isolated filter structure 570. In bridging mode, RF switch 1 and RF switch 2 are open and RF switch 3 is closed. When the filter is in operation, RF switch 1 and RF switch 2 are closed and RF switch 3 is open. Fig. Figures 15b-15e illustrate parameter diagram curves S for the isolated filter structure according to the invention of Figure 15a when the filter is in operation, i.e., when RF switch 1 and RF switch 2 are closed and RF switch 3 is open. In contrast, Figures 15b-15e illustrate parameter diagram curves S for the insulated filter structure according to the invention of Figure 15a when the filter is in operation, i.e., when RF switch 1 and RF switch 2 are closed and RF switch 3 is open. Fig. 15f-15i Parameter diagram curves S for the isolated filter structure according to the invention of Fig. 15a, when the filter is bypassed, i.e., when RF switch 1 and RF switch 2 are open and RF switch 3 is closed. As in Fig. As can be seen in diagram 15g, the insertion loss S31 is less than 3 dB when the filter is engaged, but more than 20 dB when the filter is bypassed. Port 3 remains matched regardless of whether the filter is active or bypassed, as shown in diagram 15g. Fig. 15d, Fig. 15e, Fig. 15h and Fig. 15i can be seen.

[0051] Fig. Figure 16a illustrates an exemplary layout of an isolated filter according to the invention, and Fig. Figure 16b illustrates an exemplary layout of an isolated filter according to the invention, which includes bypass switches. In one example, each structure with port numbers (1), (2), (3), (4) corresponds to the corresponding scheme in Figure 16b. Fig. 10c describes a Fischer hybrid core with its corresponding ports P1, P2, P3, P4, where P1 corresponds to (1), P2 to (2), P3 to (3), and P4 to (4). In some embodiments, a certain amount of additional electrical capacitance is inserted between port (1) and port (2) as well as between port (3) and port (4) to create a quadrature coupler for the corresponding frequency.

[0052] Fig. Figure 17a illustrates a system 600 according to the invention, in which a first bank of diplexers / triplexers 632 is coupled to the antennas 602 and 604 via the antenna switch 606 and routed via RF switches 610, 612 and 614 to the low-band LNA A, mid-band LNA B and a high-band LNA C, followed by the diplexer 603 and the isolated filter core 605 instead of a redundant filter bank which is in Fig. 1a is shown. The System 600 of Fig. 17a avoids the use of a redundant filter bank at the LNA output, and the diplexer design, thanks to the support of the isolated filter core, can be performed with fewer design constraints for parameters affecting input loading / mismatch, gain drop in carrier aggregation (CA) mode, and RX noise from active LNAs in CA mode. The diplexer filter components in the 603 diplexer are phase-shifted relative to each other via phase ΦD, so they do not load each other. The 605 isolated filter core appears as a counterpart to the 603 diplexer, simplifying its design.

[0053] Fig. Figure 17b illustrates a system 620 according to the invention, which corresponds to the system of Fig. 17a is similar, but the diplexer 622 and the isolated filter core 624 are represented as tunable according to the LNA. This tunability allows the system to adjust the diplexer and the isolated filter core to follow the corresponding frequency combination usage case, resulting in a less complex filter topology and thus lower attenuation.

[0054] Fig. Figure 18 illustrates a system 640 according to the invention, in which a tuned insulated filter core 642 and a diplexer 644 are coupled to the antennas 602 and 604 via the antenna switch and routed to the low-band LNA A, mid-band LNA B, and a high-band LNA C, followed by a diplexer 648 and an insulated filter structure 650 instead of a redundant filter bank. The components of the input diplexer filter 644 are phase-shifted relative to each other with respect to phase ΦE, so that they do not interfere with each other. The insulated filter core 642 appears as a counterpart to the diplexer 644, which simplifies the diplexer design. The components of the diplexer filter 648 after the LNAs are phase-shifted relative to each other with respect to phase ΦD, so that they do not interfere with each other. The insulated filter core 650 appears to be a counterpart to the diplexer 648, which simplifies diplexer design. With this RF system, size and cost are significantly reduced compared to the... Fig. 1a and Fig. The conventional approaches described in 1b are significantly reduced. In various embodiments, the diplexers 644 and 648 can be LB / HB, LB / BP, BP, HB or BP / BP implementations.

[0055] Fig. Figure 19a illustrates a system 660 according to the invention, which can be used in systems such as LTE to implement carrier aggregation. As shown, the system comprises two power amplifiers PA1 and PA3, which are coupled to the antenna ANT1 and the antenna ANT2 via hybrid bandpass / bandstop filters 662 and 664 according to the invention with insulated filter cores, the structure of which has been explained above, as well as conventional RF switches and diplexers. Several receive paths are shown, each coupled to a power amplifier and utilizing additional hybrid bandpass / bandstop filters 666, 668, 670 and 672 according to the invention with insulated filter cores, as well as low-noise amplifiers and tunable filters. The embodiment of Fig. 19a can be used, for example, to support carrier aggregation in an LTE system with three RF receive paths and a transmit or low-band, mid-band and high-band transmit carrier aggregation.

[0056] Fig. Figure 19b illustrates another system 680 according to the invention, which can also be used in systems such as LTE to implement carrier aggregation. The system of Fig. 19b is similar to the system of Fig. 19a, with the addition of two further power amplifiers PA2 and PA4 and their associated hybrid bandpass / bandstop filters 666 and 668 according to the invention. The embodiment of Fig. 19b can be used, for example, to support carrier aggregation in an LTE system with radio frequencies for up to five receive paths and up to four transmit paths.

[0057] Fig. Figure 20a illustrates an RF system 700 that provides reception and transmission to and from mid-band and high-band switching logic 702 and low-band switching logic 704 via the single-feed antennas Antenna 1 and Antenna 2. As shown, the high-band and mid-band switching logic 702 includes power amplifiers PA1 and PA1, which are selectively coupled to Antenna 1 and Antenna 2 via filters 708 and 710, frequency combiners 1 and 2, and a DPDT RF switch 706. The high-band and mid-band switching logic 702 further includes a first receive path with LNAs LNA1A, LNA2A, and LNA3A, which are coupled to the switch 706 via filters F1A, F2A, and F3A, and frequency combiner 1. Similarly, the high-band and mid-band switching logic 702 also includes a second receive path with the LNAs LNA1B, LNA2B and LNA3B, which are coupled to the crossover 706 via the filters F1B, F2B and F3B and the frequency combiner 1.

[0058] The low-band switching logic 704 contains power amplifiers PA3 and PA4, which are selectively coupled to antennas 1 and 2 via filters 714 and 716, frequency combiners 3 and 4, and the DPDT RF switch 712. The low-band switching logic 704 further contains a third receive path with LNAs LNA4A and LNA5A, which are coupled to the switch 712 via filters F4A and F5A and frequency combiner 3. Similarly, the low-band switching logic 704 further contains a fourth receive path with LNAs LNA4B and LNA5B, which are coupled to the switch 712 via filters F4B and F5B and frequency combiner 4.

[0059] In operation, a transmission path of the high-band and mid-band switching logic 702 and a transmission path of the low-band switching logic 704 can be activated simultaneously. For example, the power amplifier PA1 can be coupled to the antenna 1 via the switch 706, and the power amplifier TX3 can be coupled to the antenna 2 via the switch 712. In one embodiment, the transmission filters 708, 710, 714, and 716 and the frequency combiners 1, 2, 3, and 4 can be implemented using the isolated filter structures according to the invention described herein. Furthermore, the mid-band and high-band filters 752, 754, and 756 coupled to the frequency combiner 1, as well as the mid-band and high-band filters 58, 760, and 762 coupled to the frequency combiner 2, can be implemented using the diplexer 648 and the components described herein. Fig. The insulated filter structure 650 shown in Figure 18 is implemented according to the invention. For example, insulated filter structures according to the invention can be used by employing a tuned bandpass / bandstop filter according to the invention, followed by a tuned diplexer filter with flat guard band transitions.

[0060] Fig. Figure 20b illustrates an RF system 720 that provides reception and transmission to and from mid-band and high-band switching logic 702 and low-band switching logic 704 via a dual-feed antenna 1 and a dual-feed antenna 2. As shown, the dual-feed antenna 1 includes antenna 1 MB / HB for the mid-band and high-band signal path and antenna 1 LB for the low-band signal path. Similarly, the dual-feed antenna 2 includes antenna 2 MB / HB for the mid-band and high-band signal path and antenna 2 LB for the low-band signal path. In one embodiment, the high-band and mid-band switching logic 702 and the low-band switching logic 704 are similar to those shown in Figure 20b. Fig. Figure 20a shows the high-band and mid-band switching logic 702 and the low-band switching logic 704. By using dual-feed antennas, all four power amplifiers PA1, PA2, PA3, and PA4 can transmit simultaneously. For example, the output of power amplifier PA1 can be coupled to antenna 1 MB / HB via the switch 706, the output of power amplifier PA2 can be coupled to antenna 2 MB / HB via the switch 706, the output of power amplifier PA3 can be coupled to antenna 1 LB via the switch 712, and the output of power amplifier PA4 can be coupled to antenna 2 LB via the switch 712. Alternatively, the outputs of power amplifiers PA1, PA2, PA3, and PA4 can be routed to different antennas. An advantageous aspect of the 720 system is its ability to provide multiple transmission channels without the need for highly selective filters.This is due to the fact that in different embodiments each antenna exhibits frequency selectivity and isolation due to spatial separation, and in some implementations isolation due to the orthogonal polarization of the antennas.

[0061] Fig. Figure 20c illustrates the RF system 770, which includes multiple transmit and receive paths coupled to a first dual-feed antenna containing the high-band antenna 1 HB and the low-band and mid-band antenna 1 LB / MB, and to a second dual-feed antenna containing the high-band antenna 2 HB and the low-band and mid-band antenna 2 LB / MB.

[0062] In one embodiment, the high-band RF switch 776 selectively couples the high-band receiver circuit 790 and the high-band transmit / receive circuit 784 to antenna 1 HB and antenna 2 HB. The switch 776 either connects the high-band receiver circuit 790 to antenna 1 HB and the high-band transmit / receive circuit 784 to antenna 2 HB, or it connects the high-band receiver circuit 790 to antenna 2 HB and the high-band transmit / receive circuit 784 to antenna 1 HB.

[0063] The mid-band RF switch 772 selectively couples the mid-band transmit / receive circuit 780 and the mid-band receive circuit 782 to antenna 1 LB / MB and antenna 2 LB / MB via the diplexers 792 and 794, respectively. The switch 772 either routes the mid-band transmit / receive circuit 780 to antenna 1 LB / MB and the mid-band receive circuit to antenna 2 LB / MB, or it routes the mid-band transmit / receive circuit 780 to antenna 2 LB / MB and the mid-band receive circuit to antenna 1 LB / MB.

[0064] Similarly, the low-band RF switch 774 selectively couples the low-band transmit / receive circuit 788 and the low-band transmit / receive circuit 786 to antenna 1 LB / MB and antenna 2 LB / MB via the diplexers 792 and 794, respectively. The switch 774 either routes the low-band transmit / receive circuit 788 to antenna 1 LB / MB and the low-band transmit / receive circuit to antenna 2 LB / MB, or it routes the low-band transmit / receive circuit 788 to antenna 2 LB / MB and the low-band transmit / receive circuit to antenna 1 LB / MB.

[0065] The 770 system can be used, for example, to route various combinations of signal paths to the antennas. For instance, with respect to the high band, the power amplifier PA2 of the transmit / receive circuit 784 can be selectively routed to either antenna 1 HB or antenna 2 HB, while the high-band receive circuit 790 is routed to the other HB antenna. Similarly, with respect to the mid-band, the power amplifier PA1 of the transmit / receive circuit 780 can be selectively routed to either antenna 1 LB / MB or antenna 2 LB / MB, while the mid-band receive circuit 782 is routed to the other LB / MB antenna. On the other hand, both low-band power amplifiers PA3 and PA4 can be routed via the diplexers 792 and 794 to different LB / MB antennas 1 and 2 and transmit simultaneously.

[0066] The embodiments of Fig. 20a-20c can be used, for example, in mobile communication standards such as LTE Advanced, which provides a receive-carrier aggregation function for up to 3 independent RX bands that can vary in frequency.

[0067] Fig. Figure 21a illustrates an RF system 800 that can be configured to transmit in two carrier aggregation modes. The RF system 800 includes a first mid-band / high-band transmit / receive circuit 802, a second mid-band / high-band transmit / receive circuit 804, a first low-band transmit / receive circuit 806, and a second low-band transmit / receive circuit 808. In one embodiment, the first mid-band / high-band transmit / receive circuit 802 can be connected to one of the antennas 1 and antenna 2, and the second mid-band / high-band transmit / receive circuit 804 can be connected to the other of antenna 1 and antenna 2. Such a configuration is activated by closing switch 826 to route the first mid-band / high-band transmission / receiving circuit 802 to the upper port of switch 810, and by opening switch 824.In such a configuration, each mid-band / high-band signal path is coupled to a separate antenna, enabling high linearity and low attenuation.

[0068] Fig. Figure 21b illustrates a diagram representing the guide configuration of the mid-band / high-band circuits 802 and 804 of the RF system 800 when the mid-band / high-band circuits 802 and 804 are of Fig. 21a in this configuration exhibits high linearity and low attenuation. As shown, the first mid-band / high-band circuit 802 is coupled to one port of the crossover 810, and the second mid-band / high-band circuit 804 is coupled to another port of the crossover 810.

[0069] In one embodiment, the switches 820, 822, 824 and 826 can be reconfigured to put the RF system 800 into a blocked-path mode in which the same antenna is used for low-band transmit and receive signals and / or mid-band / high-band transmit and receive signals if the other antenna is blocked or overloaded. For example, switch 824 can be closed to direct the receive path of the mid-band / high-band transmit / receive circuit 804 to a port of switch 810, switch 826 can be opened, switch 822 can be configured to couple the mid-band / high-band transmit / receive circuit 802 with the hybrid of the mid-band / high-band transmit / receive circuit 804, and switch 820 can be configured to couple the mid-band / high-band transmit / receive circuit 804 with another port of switch 810. Fig. Figure 21c illustrates a diagram representing the guide configuration of the mid-band / high-band circuits 802 and 804 of the RF system 800 when the mid-band / high-band circuits 802 and 804 are of Fig. 21a is in this mode with a blocked path.

[0070] Similarly, the low-band transmit / receive circuits 806 and 808 can be configured to be coupled to antennas 1 and antennas 2 in a high linearity, low attenuation mode, and they can be configured in a blocked-path mode similar to the mid-band and high-band transmit / receive circuits 802 and 804 described above. Such configurations can be set via the diplexers 830, 832, 834, 836, and 812. Since antennas 1 and 2 are coupled to the diplexers 810 and 812 via the diplexers 814 and 816, the high linearity, low attenuation configuration and the blocked-path mode can be selected independently for the mid-band / high-band paths and for the low-band paths.For example, the mid-band and high-band paths can be configured in blocked-path mode, while the low-band paths are configured in high linearity, low attenuation mode. In another example, the mid-band and high-band paths can be configured in high linearity, low attenuation mode, while the low-band paths are configured in blocked-path mode. In some embodiments, both the mid-band / high-band paths and the low-band paths can be configured in high linearity, low attenuation mode, or they can both be configured in blocked-path mode. It should be understood that the embodiments of... Fig. 21a-21c are merely specific examples of a system with signal paths that can be selected between modes with high linearity and low attenuation and modes with a blocked path.

[0071] Fig. Figures 22a-22d illustrate frequency combination circuits according to the invention, which are shown connected to various LNAs and PAs, as shown in the figures in Fig. The systems illustrated in 20a-20c and 21a-21c can be used to implement frequency combiners. Fig. Figure 22a illustrates a system 900 in which the frequency combiner 902 is connected via the adjustable filter 920 to the low-noise amplifier LNA3A or LNA3B, via the adjustable filter 922 to the low-noise amplifier LNA2A or LNA2B, via the filter structure section 924 according to the invention to the low-noise amplifier LNA1A or LNA1B, and via the isolated filter structure section 926 according to the invention to the power amplifier PA1 or PA2. The isolation filter sections 924 and 926 each contain a 3dB divider, a +45-degree phase shifter, a -45-degree phase shifter, and adjustable filters, so that, by combination with the -3dB hybrids 914 and 916 and the phase shifters 906, 908, 910, and 912 in the frequency combiner 902, isolated filters according to the invention are formed.During receive operation, the -3dB hybrid 916, in conjunction with the isolated filter section 926, acts as a band-stop filter in a transmission band for signals routed from the -45-degree port of the hybrid 916 to the +45-degree port of the hybrid 916. Energy within the receive band, defined by the frequency response of the filters in the isolation filter section 924 and the hybrid 916, is then sent to the inputs of LNA1A or LNA1B, while energy outside this receive band is sent via the filter 920 to the inputs of LNA3A or LNA3B, or via the phase shifter 904 and the filter 922 to the inputs of LNA2A or LNA2B. During transmission operation, energy within a frequency band defined by the isolated filter section 926 is sent to the -45 degree port of the hybrid 916 and, in some embodiments, to an antenna.

[0072] System 930 from Fig. 22b is similar to the System 900 from Fig. 22a, with the addition of the inventive isolated filter section 940, the phase shifters 936 and 938, and the hybrid 934, which provides an additional degree of filtering and isolation for LNA2A or LNA2B. For example, during operation, the frequency content of a received signal with an operating frequency assigned to LNA1A or LNA2B is rejected by the isolated filter formed from the isolated filter section 940, the phase shifters 936 and 938, and the hybrid 934.

[0073] Fig. Figure 22c illustrates a system 950 in which the frequency combiner 952 is connected via the adjustable filter 950 to a low-noise amplifier LNA5A or LNA5B, via the adjustable filter 953 to the low-noise amplifier LNA4A or LNA4B, and via the filter structure section 954 according to the invention to the power amplifier PA3 or PA4. The isolation filter section 954 includes a 3dB divider, a +45-degree phase shifter, a -45-degree phase shifter, and adjustable filters, such that, by combination with the -3dB hybrid 95816 and the phase shifters 960 and 962 in the frequency combiner 952, isolated filters according to the invention are formed. During receive operation, the -3dB hybrid 958, in conjunction with the isolated filter section 954, serves as a bandstop filter in a transmission band for signals that are forwarded from the -45 degree port of the hybrid 916 to the +45 degree port of the hybrid 958.Energy outside this band is then sent via filter 951 to the inputs of LNA 5A or LNA 5B, or via phase shifter 956 and filter 953 to the inputs of LNA 4A or LNA 4B. During transmission operation, energy within a frequency band defined by the filter of the isolated filter section 954 is sent to the -45-degree port of hybrid 958 and, in some embodiments, to an antenna.

[0074] System 970 from Fig. 22b is similar to the System 900 from Fig.22a, with the addition of the inventive isolated filter section 940, the phase shifters 936 and 938, and the hybrid 934, which provides an additional degree of filtering and isolation for LNA2A or LNA2B. For example, during operation, the frequency content of a received signal with an operating frequency assigned to LNA1A or LNA2B is rejected by the isolated filter formed from the isolated filter section 940, the phase shifters 936 and 938, and the hybrid 934.

[0075] The present document summarizes embodiments of the present invention. Other embodiments are also understood to constitute the entirety of the description and the claims filed herein. A general aspect includes a circuit comprising: several filter circuits, each comprising a first port, a second port, and a third port, wherein a second port is coupled to a first port of a second of the several filter circuits, each of the several filter circuits comprising: a first passive filter, a second passive filter, a first coupler with an input port coupled to the first port, an isolated port coupled to the second port, a first phase-shifted port connected to the first passive filter, and a second phase-shifted port connected to the second passive filter.and a combination network with a first input coupled to the first passive filter, a second input coupled to the second passive filter, and an output coupled to the third port.

[0076] Implementations may include one or more of the following features. The circuit, wherein the first coupler includes a first -3dB hybrid with a quadrature output. The circuit, wherein the combination network includes a second -3dB hybrid with a quadrature output and an isolation port. In some embodiments, the combination network includes: a power divider, a first phase shifter coupled between the first passive filter and the power divider, and a second phase shifter coupled between the second passive filter and the power divider. The first passive filter and the second passive filter may be tunable filters. In some embodiments, a second port of the second of the multiple filter circuits is coupled to a first port of a third of the multiple filter circuits.

[0077] Another general aspect includes a circuit comprising: a filter circuit with a first port, a second port, and a third port, wherein the filter circuit includes a first passive filter, a second passive filter, a first coupler with an input port coupled to the first port, an isolated port coupled to the second port, a first phase-shifted port connected to the first passive filter, and a second phase-shifted port connected to the second passive filter, wherein the first coupler includes a -3dB hybrid with quadrature output, and a combination network comprising a first input coupled to the first passive filter, a second input coupled to the second passive filter, and an output coupled to the third port, wherein the combination network includes a power divider, a first phase shifter,which is coupled between the first passive filter and the power divider, and contains a second phase shifter coupled between the second passive filter and the power divider, and an amplifier having an output coupled to the third port.

[0078] Implementations may include one or more of the following features. In some embodiments, the filter circuit is configured to: pass a first frequency band from the third port to the first port using a passband transfer function; pass a second and a third frequency band from the first port to the second port; and filter the first frequency band from the first port to the second port, wherein the second frequency band is lower than the first frequency band and the third frequency band is higher than the first frequency band. The first frequency band may cover a frequency range between approximately 2110 MHz and approximately 2170 MHz. In some embodiments, the circuit further includes a diplexer coupled to the second port, the diplexer being configured to separate the second frequency band from the third frequency band.The diplexer may contain a transfer function that has a flatter crossover response than the filter circuit.

[0079] In some embodiments, the amplifier may include a power amplifier, the first port may be coupled to an antenna, and / or the first and second passive filters may be tunable filters. In some embodiments, the amplifier includes a low-noise amplifier (LNA).

[0080] Another general aspect includes a circuit comprising: a first filter circuit with a first port configured to be coupled to an antenna, a second port, and a third port, wherein the first filter circuit includes a first passive filter, a second passive filter, a first coupler comprising an input port coupled to the first port, an isolated port coupled to the second port, a first phase-shifted port connected to the first passive filter, and a second phase-shifted port connected to the second passive filter, wherein the first coupler includes a -3dB hybrid with quadrature output, and a first combination network comprising a first input coupled to the first passive filter, a second input coupled to the second passive filter, and an output coupled to the third port.wherein the first filter circuit is configured to pass a first frequency band from the first port to the third port using a passband transfer function, and is configured to pass a second frequency band and a third frequency band from the first port to the second port and to reject the first frequency band from the first port to the second port, wherein the first frequency band contains higher frequencies than the first frequency band and the third frequency band contains lower frequencies than the first frequency band, and a first LNA having an input coupled to the third port, a first frequency division circuit coupled to the first port of the first filter circuit, wherein the first frequency division circuit is configured to divide the third frequency band from the second frequency band, a second LNA having an input,which is coupled to a second frequency band output of the first frequency division circuit, and includes a third LNA which has an input that is coupled to a third frequency band output of the first frequency division circuit.

[0081] Implementations may include one or more of the following features. The circuit further includes a frequency combination circuit with a first input coupled to an output of the second LNA and a second input coupled to an output of the third LNA. In some embodiments, the circuit further includes a second filter circuit with a first port configured to provide a combined frequency output, a second port coupled to an output of the frequency combination circuit, and a third port coupled to an output of the first LNA. The second filter circuit includes a third passive filter, a fourth passive filter, a second coupler with an input port coupled to the first port, an isolated port coupled to the second port, and a first phase-shifted port coupled to the third passive filter.and a second phase-shifted port coupled to the fourth passive filter, wherein the first coupler includes a -3dB hybrid with quadrature output, and a second combination network with a first input coupled to the third passive filter, a second input coupled to the fourth passive filter, and an output coupled to the third port, wherein the second filter circuit is configured to pass a first frequency band from the first port to the third port using the passband function, and is configured to pass the second and third frequency bands from the first port to the second port and to reject the first frequency band from the first port to the second port.

[0082] In some embodiments, the first passive filter, the second passive filter, the third passive filter and the fourth passive filter contain tunable filters and / or the first frequency division circuit and the frequency combination circuit are tunable.

[0083] In one embodiment, the first combination network comprises a first power divider, a first phase shifter coupled between the first passive filter and the first power divider, and a second phase shifter coupled between the second passive filter and the first power divider. The second combination network comprises a second power divider, a third phase shifter coupled between the third passive filter and the second power divider, and a fourth phase shifter coupled between the fourth passive filter and the second power divider. In some embodiments, the first frequency division circuit and the frequency combination circuit each comprise a duplexer or a diplexer. The circuit may further comprise a parallel filter bank coupled between the first frequency division circuit and the second and third LNAs.In some embodiments, the parallel filter bank includes a tunable filter bank.

[0084] Another general aspect involves a circuit with a filter circuit comprising a first port configured to be coupled to an antenna, a second port, and a third port. The filter circuit includes a first passive filter, a second passive filter, a first power amplifier with an output coupled to the first passive filter, a second power amplifier with an output coupled to the second passive filter, a first coupler with an input port coupled to the first port, an isolated port coupled to the second port, a first phase-shifted port coupled to the first passive filter, and a second phase-shifted port coupled to the second passive filter.The first coupler contains a -3db hybrid with quadrature output and a division network with a first output coupled to an input of the first power amplifier, a second output coupled to an input of the second power amplifier, and an input coupled to the third port.

[0085] Implementations may include one or more of the following features. The circuit, wherein the division network includes a power divider, a first phase shifter coupled between the first power amplifier and the power divider, and a second phase shifter coupled between the second power amplifier and the power divider. In some embodiments, the circuit is configured to stabilize the output power at the first port with respect to changes in the standing wave ratio (VSWR) at the first port. The filter circuit may be configured to provide transmission filtering for the first and second power amplifiers and to filter TX noise in a receive band to the antenna at the first port and to a passband of the first and second passive filters at the second port.

[0086] Another general aspect includes a circuit that contains the following: a first RF front end of a first band, a second RF front end of a first band, a first multi-feed antenna, a second multi-feed antenna, and a first RF switch configured to couple the first RF front end to a first element of the first multi-feed antenna and the second RF front end to a first element of the second multi-feed antenna in a first configuration, and configured to couple the first RF front end to the first element of the second multi-feed antenna and the second RF front end to the first element of the first multi-feed antenna in a second configuration.

[0087] Implementations may include one or more of the following features. The circuit, wherein: the first RF front end includes a first power amplifier and the second RF front end includes a second power amplifier. In some embodiments, the first RF front end further includes a first receive path and the second RF front end further includes a second receive path. The first receive path may include multiple first filters coupled in series with corresponding multiple first low-noise amplifiers, and the second receive path may include multiple first filters coupled in series with corresponding multiple first low-noise amplifiers.The circuit can further include a third RF front end of a second band, a fourth RF front end of the second band, and a second RF switch configured, in a third configuration, to couple the third RF front end to a second element of the first multi-feed antenna and the fourth RF front end to a second element of the second multi-feed antenna, and, in a fourth configuration, to couple the third RF front end to the second element of the second multi-feed antenna and the fourth RF front end to the second element of the first multi-feed antenna. In some embodiments, the first band has a higher frequency than the second band.

[0088] In some embodiments, the circuit further includes: a third RF front end of a second band, a fourth RF front end of the second band, a second RF switch configured in a third configuration to couple the third RF front end to the first element of the first multifeed antenna and to couple the fourth RF front end to the first element of the second multifeed antenna, and configured in a fourth configuration to couple the third RF front end to the first element of the second multifeed antenna and to couple the fourth RF front end to the first element of the first multifeed antenna;a first diplexer with inputs coupled to the first and second RF diplexers, and an output coupled to the first element of the first multifeed antenna, and a second diplexer with inputs coupled to the first and second RF diplexers, and an output coupled to the first element of the second multifeed antenna.

[0089] The circuit may further include: a fifth RF front end of a third band, a sixth RF front end of the third band, and a third RF switch configured, in a fifth configuration, to couple the fifth RF front end to a second first element of the first multifeed antenna and the sixth RF front end to a second element of the second multifeed antenna, and configured, in a sixth configuration, to couple the fifth RF front end to the second element of the second multifeed antenna and the sixth RF front end to the second element of the first multifeed antenna. In some embodiments, the first band has a higher frequency than the second band, and the third band has a higher frequency than both the first and second bands.

[0090] In some embodiments, the first RF front end includes a first receive path, a first transmit path, and a first frequency combiner configured to couple an input of the first receive path and an output of the first transmit path to a first port, and the second RF front end includes a second receive path, a second transmit path, and a second frequency combiner configured to couple, in a first mode, an input of the second receive path and an output of the second transmit path to a second port, and, in a second mode, to couple the input of the second receive path, the output of the second transmit path, and the first port of the first RF front end to the second port, wherein the first port is coupled to a first input of the first RF switch and the second port is coupled to a second input of the first RF switch.

[0091] In various embodiments, the first frequency combiner includes: a first power divider with an input coupled to the first transmission path, a first phase shifter coupled to a first output of the first power divider, a second phase shifter coupled to a second output of the first power divider, a first filter coupled to the first phase shifter, a second filter coupled to the second phase shifter, and a hybrid circuit with an input port coupled to the first filter, an isolated power output coupled to the second filter, a first phase-shift output coupled to the first receive path, and a second phase-shift output coupled to the first port. The circuit may further include: a third second-band RF front end, a fourth second-band RF front end,a second RF diplexer configured to couple, in a third configuration, the third RF front end to the first element of the first multi-feed antenna and the fourth RF front end to the first element of the second multi-feed antenna, and configured to couple, in a fourth configuration, the third RF front end to the first element of the second multi-feed antenna and the fourth RF front end to the first element of the first multi-feed antenna, a first diplexer with inputs coupled to the first and second RF diplexers and an output coupled to the first element of the first multi-feed antenna, and a second diplexer with inputs coupled to the first and second RF diplexers and an output coupled to the first element of the second multi-feed antenna.

[0092] In one embodiment, the third RF front end includes a third receive path, a third transmit path, and a third frequency combiner configured to couple an input of the third receive path and an output of the third transmit path to a third port; the fourth RF front end includes a fourth receive path, a fourth transmit path, and a fourth frequency combiner configured to couple, in a third mode, an input of the fourth receive path and an output of the fourth transmit path to a fourth port, and, in a fourth mode, to couple the input of the fourth receive path, the output of the fourth transmit path, and the third port of the second RF front end to the fourth port, wherein the third port is coupled to a first input of the second RF switch and the fourth port is coupled to a second input of the second RF switch.

[0093] Advantages of embodiments employing such an isolated filter structure include, among others, the possibility of performing selective filtering using only one type of filter, since the passband transfer function H1(s) and the stopband transfer function 1-H1(s) automatically follow one another. Furthermore, the stopband characteristic is less dependent on the correct phase than in a conventional filter, due to the isolated nature of the three-port structure according to the invention.

[0094] Another advantage of these embodiments is, among others, the ability to implement multiband filters in RF front ends without using redundant filter banks. This reduces the amount of hardware required to implement multiband RF front ends, resulting in smaller form factors and lower system implementation costs.

[0095] Another advantage of these embodiments is, among others, the ability to implement flexible carrier aggregation schemes. For example, in some embodiments, two transmission paths can be selected and routed to two separate antennas to achieve higher linearity and lower attenuation. However, if one path is blocked, the system can be reconfigured to transmit two bands on the same antenna.

[0096] Although the present invention has been described with reference to illustrative embodiments, this description should not be interpreted in a restrictive sense. Various modifications and combinations of the illustrative embodiments, as well as further embodiments of the invention, will be apparent to those skilled in the art from the description.

Claims

[1] Circuit, comprising: a filter circuit with a first port (①) configured to be coupled to an antenna, a second port (②) and a third port (③), wherein the filter circuit comprises the following: a first passive filter (302), a second passive filter (304), a first power amplifier (346) with an output coupled to the first passive filter (302), a second power amplifier (348) with an output coupled to the second passive filter (304), a first coupler (300) with an input port coupled to the first port (①), an isolated port coupled to the second port (②), a first phase-shifted port coupled to the first passive filter (302), and a second phase-shifted port coupled to the second passive filter (304), wherein the first coupler (300) comprises a -3dB hybrid with quadrature output, and a division network (312) with a first output coupled to an input of the first power amplifier (346), a second output coupled to an input of the second power amplifier (348), and an input coupled to the third port(③). [2] Circuit according to claim 1, wherein the division network (312) comprises a power divider (310), a first phase shifter (306) coupled between the first power amplifiers (346) and the power divider (310), and a second phase shifter (308) coupled between the second power amplifiers (348) and the power divider (310). [3] Circuit according to claim 1 or 2, wherein the circuit is configured to stabilize an output power at the first port (①) with respect to changes in a standing wave ratio (VSWR) at the first port (①). [4] Circuit according to one of claims 1 to 3, wherein the filter circuit is configured to provide transmit filtering for the first and second power amplifiers (346, 348) and to filter TX noise in a receive band to the antenna in the first port (①) and to a passband of the first and second passive filters (302, 304) at the second port (②). [5] Circuit, comprising: a filter circuit with a first port (①) configured to be coupled to an antenna, a second port (②) and a third port (③), wherein the filter circuit comprises the following: a first passive filter (302), a second passive filter (304), a first power amplifier (342) with an input coupled to the first passive filter (302), a second power amplifier (342) with an input coupled to the second passive filter (304), a first coupler (300) with an input port coupled to the first port (①), an isolated port coupled to the second port (②), a first phase-shifted port coupled to the first passive filter (302), and a second phase-shifted port coupled to the second passive filter (304), wherein the first coupler (300) comprises a -3dB hybrid with quadrature output, and a combination network (312) with a first output coupled to an output of the first power amplifier (346), a second output coupled to an output of the second power amplifier (348), and an input coupled to the third port(③). [6] Circuit according to claim 5, wherein the combination network (312) comprises a combiner (310), a first phase shifter (306) coupled between the first power amplifier (342) and the combiner (310), and a second phase shifter (308) coupled between the second power amplifier (344) and the combiner (310). [7] Circuit, comprising: a plurality of filter circuits according to one of claims 1 to 6, where the second port of a first of the several filter circuits is connected to a first port of a second of the several filter circuits. [8] Circuit, comprising: a first RF frontend of a first band, a second HF frontend of the first band, a first multi-feed antenna, a second multi-feed antenna and a first RF switch (706, 774, which is configured in a first configuration to couple the first RF frontend with a first element of the first multifeed antenna and to couple the second RF frontend with a first element of the second multifeed antenna, and is configured in a second configuration to couple the first RF frontend with the first element of the second multifeed antenna and to couple the second RF frontend with the first element of the first multifeed antenna, the first RF frontend includes the following: a first receive path (RX1A-RX3A, RX4A-RX5A), a first transmission path (TX1, TX3), and a first frequency combiner configured to couple an input of the first receive path (RX1A-RX3A, RX4A-RX5A) and an output of the first transmit path (TX1, TX3) to a first port, and the second RF frontend includes the following: a second receive path (RX1B-RX3B, RX4B-RX5B), a second transmission path (TX2, TX4), and a second frequency combiner configured for this purpose, In a first mode, an input of the second receive path (RX1B-RX3B, RX4B-RX5B) and an output of the second transmit path (TX2, TX4) are coupled to a second port and in a second mode, to couple the input of the second receive path (RX1B-RX3B, RX4B-RX5B), the output of the second transmit path (TX2, TX4) and the first port of the first RF front end with the second port, wherein the first port is coupled to a first input of the first RF switch (706, 774) and the second port is coupled to a second input of the first RF switch (706, 774). [9] Circuit according to claim 8, wherein: the first RF front end includes a first power amplifier (PA1, PA3) and The second RF front end includes a second power amplifier (PA2, PA4). [10] Circuit according to claim 8 or 9, wherein: the first receive path (RX1A-RX3A, RX4A-RX5A) comprises several first filters (F1A-FSA) which are coupled in series with corresponding several first low-noise amplifiers (LNA1A-LNA5A), and the second receive path (RX1B-RX3B, RX4B-RX5B) includes several first filters (F1B-F5B) which are coupled in series with corresponding several first low-noise amplifiers (LNA1B-LNASB). [11] Circuit according to any one of claims 8 to 10, further comprising: a third HF frontend of a second band, a fourth HF frontend of the second band and a second RF switch (712, 772, 776) configured to couple the third RF frontend with a second element of the first multifeed antenna in a third configuration and to couple the fourth RF frontend with a second element of the second multifeed antenna, and configured to couple the third RF frontend with the second element of the second multifeed antenna in a fourth configuration and to couple the fourth RF frontend with the second element of the first multifeed antenna. [12] Circuit according to claim 11, wherein the first band is of a higher frequency than the second band. [13] Circuit according to claim 11 or 12, further comprising: a first diplexer (792) with inputs coupled to the first and second RF diplexers (772, 774) and an output coupled to the first element of the first multifeed antenna, and a second diplexer (794) with inputs coupled to the first and second RF diplexers (772, 774) and an output coupled to the first element of the second multifeed antenna. [14] Circuit according to one of claims 11 to 13, wherein The third RF frontend includes the following: a third receive path (RX4A-RX5A), a third transmission path (TX3), a third frequency combiner configured to couple an input of the third receive path (RX4A-RX5A) and an output of the third transmit path (TX3) to a third port, The fourth RF frontend includes the following: a fourth receive path (RX4B-RX5B), a fourth transmission path (TX4), a fourth frequency combiner configured for this purpose: in a third mode to couple an input of the fourth receive path (RX4B-RX5B) and an output of the fourth transmit path (TX4) with a fourth port and in a fourth mode, to couple the input of the fourth receive path (RX4B-RX5B), the output of the fourth transmit path (TX4) and the third port of the second RF front end with the fourth port, wherein the third port is coupled to a first input of the second RF switch (712) and the fourth port is coupled to a second input of the second RF switch (712). [15] Circuit according to any one of claims 8 to 14, further comprising: a fifth HF frontend of a third band, a sixth HF frontend of the third band and a third RF switch (776) configured to couple the fifth RF front end with a second first element of the first multifeed antenna in a fifth configuration and to couple the sixth RF front end with a second element of the second multifeed antenna, and configured to couple the fifth RF front end with the second element of the second multifeed antenna in a sixth configuration and to couple the sixth RF front end with the second element of the first multifeed antenna. [16] Circuit according to claim 15, wherein: the first band is of a higher frequency than the second band and The third band has a higher frequency than the first and second bands. [17] Circuit according to any one of claims 8 to 16, wherein the first frequency combiner comprises: a first power divider with one input coupled to the first transmit path, a first phase shifter coupled to a first output of the first power divider, a second phase shifter coupled to a second output of the first power divider, a first filter coupled with the first phase shifter, a second filter coupled to the second phase shifter, and a hybrid circuit with an input port coupled to the first filter, an isolated power output coupled to the second filter, a first phase-shift output coupled to the first receive path, and a second phase-shift output coupled to the first port.

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