Electronic module, motor vehicle, and method for limiting an input current during a switch-on process of the module

Abstract

Electronic module (17) with a source terminal (11) for receiving an input current (Is) from an electrical voltage source (12) and with an electrical input capacitance (Cin) effective with respect to the source terminal (11), wherein the input capacitance (Cin) is connected to the source terminal (11) via a transistor circuit (18), wherein the transistor circuit (18) is configured to guide the input current (Is) through a respective switching path (19) of at least one transistor (M1, M2), wherein a control device (20) is configured to switch a control voltage (Vgs) at a respective control terminal (g) of the at least one transistor (M1, M2) in several stages (24) during a switch-on process (15) to limit the input current (Is), from a blocking value (25), at which each switching path (19) is blocked, to a conducting value (26), at which a through-resistance of each switching path (19) is minimized,wherein, without feedback, the same control voltage (Vgs) profile is always provided for each switch-on process (15), characterized in that a respective time duration (Ts) of each of the stages (24) has a value corresponding to a time constant of the input capacitance (Cin) such that during one stage (24) or during two or three stages (24) the at least one transistor (M1, M2) transitions from a blocking state to an electrically conducting state and the input current (Is) rises to a maximum (16) and then falls again.

Description

[0001] The invention relates to an electronic module, such as one that may be used, for example, to operate a light-emitting diode headlight of a motor vehicle. The module can be connected to an electrical voltage source via a source terminal to receive an input current. An input capacitance of the module is effective with respect to the source terminal, which is why the inrush current of the input current must be limited when the module is switched on. For this purpose, the input capacitance is connected to the source terminal via a transistor circuit. The invention also includes a motor vehicle with the module according to the invention and a method for limiting the input current during a switch-on process.

[0002] The problem is based on Fig. 1 illustrated once again. Fig. Figure 1 shows a module 10 with a source terminal 11 to which an electrical voltage source 12, for example, the electrical system of a motor vehicle, is connected. A supply voltage Vs of the voltage source 12 can drive or generate an input current Is at the source terminal 11, which, when the module 10 is switched on, must charge an input capacitance Cin, which can be formed, for example, by a capacitor. The actual load or the actual electrical components to be operated in a functional unit 13 of the module 10 are shown in Fig. 1 is represented by a load resistor Rload. The circuit can be closed via a ground potential 14. Fig. Figure 1 further illustrates the time course of the input current Is over time in a diagram. During a switch-on process 15, the discharged input capacitance Cin causes a high inrush current 16, limited only by the internal resistance Rs of the supplying voltage source 12, the internal resistance of the input capacitance Cin itself, the resistance of the conductors, and the rise time of the switch-on pulse. The diagram shows that the current can rise to over 180 A.

[0003] In mechanical switching components used to switch the input current (e.g., relays, mechanical switches), the high inrush current promotes the formation of arcs, which can destroy the contact surfaces. In electronic switching components (e.g., MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) or thyristors), there is a risk that the high inrush current will exceed their maximum permissible values ​​according to the datasheet and (depending on the amount of energy contained in the inrush pulse) can also lead to the destruction of the junction or the bonding wires.

[0004] Therefore, in many applications in the field of automotive technology, but also in the consumer or industrial sector, it is essential to limit the inrush current 16 with regard to the current intensity.

[0005] A module of the type described above, in which the input capacitance is connected to the source terminal via a transistor circuit to limit the input current, is known, for example, from US 6 094 036 A. In this case, a capacitor is directly connected in parallel to a pre-charge resistor and a field-effect transistor bridging the resistor. To limit the input current to the capacitor, the field-effect transistor must be operated by a current regulator via a closed current control loop. This is technically complex and susceptible to the influence of parameter fluctuations, such as those caused by aging and / or heating of the components.

[0006] A method for reducing inrush current is also known from US 5,122,724 A. Here, a field-effect transistor (FET) is connected between an input capacitance and ground potential, and the electrical resistance of a switching path of the FET is gradually reduced by means of an RC circuit. The rate at which the resistance of the switching path is reduced is determined by the values ​​of the RC circuit and must be adapted to the switching behavior of the FET. If the switching behavior changes, for example due to heating, this is not taken into account by the RC circuit.

[0007] In DE 11 2014 003 904 T5 it is described that an inverter capacitor of a circuit can be gradually charged via a switching element by setting a current command value for the switching element to a value V1 for a predetermined period of time after switching on a relay switch and to the value V2 after the period of time has elapsed and holding it there, whereby for the value V1 a smaller current value than the maximum current value results and for the value V2 the switching element is then fully switched on.

[0008] From DE 199 40 579 B4, a device is known for operating an electrical load, by means of which a load current is ramped up from 0 when the load is to be switched on. For this purpose, a control system is implemented using an operational amplifier, by which an actual value of the load current is regulated to a setpoint value, which is specified by a trapezoidal wave signal. A current mirror can be provided in the operational amplifier.

[0009] In DE 39 13 446 A1 a circuit for a current mirror is described which additionally has a differential amplifier.

[0010] From DE 695 18 795 T2 a stepper motor control is known in which a pulse width modulated control signal is converted into an analog, stepless control signal for a constant current driver circuit by means of a low-pass filter.

[0011] From DE 10 2014 110 263 A1, a power supply circuit for LEDs is known, which includes a buck converter designed as a switching converter and whose switch is controlled depending on the output voltage of the power supply circuit. When the LEDs are switched on, an anomaly detection current is first generated for a duration T1 in a first stage before the LEDs are put into operation. The anomaly detection current can also be pulsed.

[0012] US 2012 / 0 063 043 A1 describes a current limiting device for an electrical load which, depending on a fault sensor, controls an activator to activate a current limiter.

[0013] JP 2003-244936A discloses a power supply unit that, in a circuit using a battery, prevents mismatches when starting the voltage supply to a load and can extend battery life. The power supply applies a stepped voltage to a drive voltage to control it into a conduction state, which, when the load is supplied with the supply voltage, outputs a supply voltage to the gate terminal of a FETQ4 at the drain terminal.

[0014] US 9,935,480 B2 discloses a power switching device that enables and interrupts the power supply to a load. A MOSFET is provided between the power supply lines and the DC power source. This MOSFET, located upstream of the power supply lines, controls the connection and disconnection of the DC power source. The power supply lines include a ripple capacitor to stabilize the supply voltage against fluctuations in the load current supplied by the DC power source. The power switch is operated by gradually increasing the MOSFET's output voltage from open to closed, thereby minimizing the current charging the ripple capacitor.

[0015] The invention is based on the objective of providing a technically simple way to limit the input current during the switching-on process of an electronic module, whereby a change in the switching behavior of the transistor circuit used (caused, for example, by a temperature change) should also be taken into account.

[0016] The problem is solved by the subject matter of the independent patent claims. Advantageous embodiments of the invention are described by the dependent patent claims, the following description, and the figures.

[0017] The electronic module described above, with the input capacitor connected to the source terminal via a transistor circuit, is extended according to the invention by the fact that the transistor circuit is configured to guide the input current through a respective switching path of at least one transistor, wherein a control device is configured to switch a control voltage at a respective control terminal of the at least one transistor in several stages during a switch-on process to limit the input current, from a blocking value, at which each switching path is blocked or electrically non-conductive, to a conducting value, at which the through-resistance of each switching path is minimized. During the switch-on process, the input capacitor is initially charged.In this process, the resistance of each switching path is gradually reduced from a non-conducting state until it is minimized. The minimized resistance is also referred to as Rdson. In a field-effect transistor (FET), this switching path is the drain-source path. The control terminal is the gate of such a FET. The input current is the current received at the source terminal. The control voltage is specifically the gate-source voltage of the transistor.

[0018] The invention offers the advantage that the control voltage for controlling the at least one transistor is changed in steps or in several steps. A transistor switches from an electrically blocking state to an electrically conducting state only at a threshold voltage, which is temperature-dependent. Accordingly, the at least one transistor only becomes electrically conductive at a specific stage of the stepwise switched control voltage. However, the at least one transistor does not switch fully on (to Rdson), but its on-resistance will be greater than the minimum possible value. By further stepwise changes of the control voltage, the on-resistance is then gradually reduced, so that the input current remains limited.If at least one transistor changes its switching behavior, for example due to a temperature change, this does not need to be explicitly considered when setting the control voltage. Rather, the at least one transistor will simply become electrically conductive at a different stage and then, starting from this stage, its on-resistance will decrease stepwise with each subsequent stage. Thus, the input current limit is maintained. Therefore, no explicit adjustment of the control voltage to a changed electrical behavior of the at least one transistor is necessary. Without feedback (i.e., in an open-loop system), the same control voltage profile is always provided for each switch-on process. This allows the electronic module according to the invention to be implemented with minimal circuitry and thus cost-effectively.

[0019] The turn-on process, i.e., the charging of the input capacitance with a limited input current, is carried out over a multitude of stages. This has the advantage that if the turn-on behavior of at least one transistor in the transistor circuit changes, and the threshold voltage of that transistor shifts to another stage, the charging of the input capacitance does not extend beyond the last stage or take longer, so that an unlimited input current (at Rdson) would flow to the input capacitance.To achieve this, the duration of each stage preferably has a value corresponding to a time constant of the input capacitance such that during one stage, or during only two or three stages, at least one transistor transitions from an electrically off state to an electrically conducting state, and subsequently the input current rises to a maximum and then falls again. In other words, switching on the input current until it rises to its maximum takes only one stage, or at most two or three stages.

[0020] The invention also includes further developments that result in additional advantages.

[0021] To avoid a feedback loop or feedback, i.e., to provide pure control for the at least one transistor, the control device is preferably configured to set the stages independently of the current operating state of the at least one transistor. This results in the same control voltage profile for every switch-on process.

[0022] To ensure a stable control voltage at each control terminal of the at least one transistor, the control device preferably includes at least one current mirror circuit. This current mirror circuit is configured to set a control current in a resistive element as a function of a control signal that controls the current mirror circuit. The resistive element can be connected between the gate and source terminals of the at least one transistor. The control voltage for the at least one transistor drops across this resistive element. This offers the advantage that the at least one transistor does not need to be directly controlled by the control signal; instead, the control signal is only required to specify the control current, which then generates the control voltage across the at least one resistive element, for example, regardless of the current temperature.This is particularly temperature-stable.

[0023] It may be provided that only a single current mirror circuit is used. To enable this, the at least one transistor is a P-channel FET (P-channel field-effect transistor). This advantageously allows the control voltage to have a value lower than the supply voltage of the voltage source. For example, the voltage source can provide a supply voltage of 12 V, 24 V, or 48 V, while the control voltage can be less than 10 V.

[0024] Alternatively, at least one transistor can be designed as an N-channel FET (N-channel field-effect transistor). To generate a suitable control voltage, two current mirror circuits are used, connected in a cascade. The first current mirror circuit generates a control signal from the drive signal for the second current mirror circuit, which then sets the control current to generate the control voltage for at least one transistor.

[0025] The control signal for controlling the at least one current control circuit can be generated by means of switching logic. This is preferably provided by a microcontroller. This allows the timing of the control signal to be set and thus preferably modified by programming or specifying a program module for the microcontroller. The control device therefore preferably comprises switching logic coupled to the at least one current mirror circuit and configured to specify the control signal for setting the control voltage by means of a control program. If the switching logic is not a microcontroller, the control program can also be implemented as a fixed circuit arrangement, for example as an ASIC (Application Specific Integrated Circuit) or by means of a flip-flop circuit.By providing software control via a control program, the current limiting method can be parameterized or adjusted.

[0026] The control signal can be a pulse-width modulated output signal of the switching logic, passed through a low-pass filter circuit. This low-pass filter circuit can, for example, include a capacitor whose capacitance is matched to a switching frequency of the pulse-width modulation such that the resulting control signal remains constant over several switching cycles of the pulse-width modulation. Alternatively, instead of pulse-width modulation, the output signal of a digital-to-analog converter (DAC) can be used directly as the control signal for the switching logic. While pulse-width modulation is technically simpler to implement than a DAC, the latter can adapt the timing of the control signal more quickly due to the absence of a low-pass filter circuit.Thus, for example, a shutdown or shutdown process for the module can be carried out by means of the digital-to-analog converter by blocking at least one transistor.

[0027] For such a rapid shutdown of at least one transistor, it can be additionally or alternatively provided that one electrode of a capacitor, which is used to reduce ripple in the drive signal, is connected to ground potential via a switching element. The switching element can, for example, be based on a transistor, particularly a field-effect transistor. The capacitor can be, for example, the capacitor of the low-pass filter circuit. The switching element is configured to electrically connect the capacitor's electrode to ground potential, discharging the capacitor, depending on a shutdown signal. This also causes the drive signal to drop to ground potential, thus switching the at least one transistor off.

[0028] The input current can be routed through the switching path of just a single transistor. This makes limiting the input current feasible with minimal circuitry and therefore cost-effective. However, such a transistor, in the form of a field-effect transistor, can incorporate a so-called body diode. To prevent reverse polarity protection in this case, preventing the voltage source from being connected with reversed poles (ground or negative at the source terminal and positive at the module's ground terminal), two transistors are preferably used, their switching paths connected in series. The forward bias of each transistor's body diode is thus oppositely connected. With the switching paths reverse-biased, the body diodes therefore also block each other.

[0029] In the manner described, the input capacitance can be provided predominantly or entirely by an additional capacitor or by several capacitors connected in parallel. An input capacitor can be used as an energy buffer and / or as a filter for a voltage and / or current waveform, for example, to ensure electromagnetic compatibility and / or robustness against fluctuations in the supply voltage of the voltage source.

[0030] The at least one transistor is preferably connected in a positive line (i.e., not at ground potential) upstream of both the input capacitor itself and the other functional components of the module, i.e., other electronic functional components or electronic circuits of the module, as viewed from the source connection. This allows the at least one transistor to electrically isolate not only the input capacitor but also the other electronic functional components from the source connection.

[0031] The electronic module is preferably intended for use in a motor vehicle. Accordingly, the invention also provides a motor vehicle with an electrical voltage source and with an embodiment of the electronic module according to the invention. The voltage source can, for example, be the vehicle's electrical system, such as a 12 V, 24 V, or 48 V system. The electronic module can, for example, be used to operate the LEDs of a headlight in the motor vehicle, for instance, to act as a current source for the LEDs.Limiting the input current during the power-up process of the electronic module has the advantage that an emergency shutdown of the voltage source or the module due to exceeding a threshold value for the input current is prevented, since this threshold value is not exceeded by limiting the input current.

[0032] The operation of the module according to the invention results in the method according to the invention, by which the input current is limited during a switch-on process of the electronic module. The method assumes that an electrical input capacitance of the module is connected via a transistor circuit to a source terminal for a voltage source. During the switch-on process, the input current from the electrical voltage source is received via the source terminal. The transistor circuit guides the input current through a respective switching path of at least one transistor. Here, a control device for limiting the input current switches a control voltage at a respective control terminal of the at least one transistor in several stages from a blocking value, at which every switching path is blocked, to a conducting value, at which the through-resistance of each switching path is minimized.

[0033] The invention also includes further developments of the method according to the invention, which have features already described in connection with the further developments of the module according to the invention. For this reason, the corresponding further developments of the method according to the invention are not described again here.

[0034] The following are exemplary embodiments of the invention described. This is illustrated by: Fig. 1 a schematic representation of an electronic module without power-on limit; Fig. 2 a schematic representation of an embodiment of the module according to the invention; Fig. 3 a diagram with schematic curves of a control voltage, a control signal and an input current over time; Fig. 4 a flowchart to illustrate an embodiment of the method according to the invention; Fig. 5 a diagram with measurement signals as they result from the procedure for different operating states of transistors; Fig. 6 a schematic circuit diagram of the module of Fig. 2; Fig. 7 a schematic circuit diagram with a module that has two current mirrors; Fig. 8 a schematic circuit diagram with a module that has only one transistor; Fig. 9 a schematic circuit diagram with a module that has two current mirrors and only one transistor for current limiting; Fig. 10 a schematic circuit diagram of a module with a rapid shutdown; and Fig. 11 a schematic circuit diagram of a module with a digital-to-analog converter.

[0035] The embodiment described below is a preferred embodiment of the invention. In this embodiment, the described components each represent individual features of the invention, which can be considered independently of one another. Each of these features further develops the invention independently and can therefore be considered part of the invention individually or in a combination other than that shown. Furthermore, the described embodiment can also be supplemented by other features of the invention already described. The scope of protection of the invention is determined solely by the claims.

[0036] In the figures, functionally identical elements are each provided with the same reference symbols.

[0037] Fig. 2 shows in accordance with Fig. 1. A module 17, to whose source terminal 11 a voltage source 12 of the described type can be connected. The module 17 can be installed in a motor vehicle. A module can be, for example, a control unit or a component of a control unit. The motor vehicle can be a motor vehicle, e.g., a passenger car or a truck.

[0038] An input capacitor Cin and the other electronic functional components (represented by Rload) are connected upstream of a transistor circuit 18. This means that the transistor circuit 18 is electrically connected between the source terminal 11 on the one hand and the input capacitor Cin and the functional components Rload on the other, such that the input current Is is guided through switching paths 19 of transistors M1 and M2 of the transistor circuit 18. Thus, the transistor circuit 18 is connected between the source terminal 11 and all the electrical functional components 13 of the module 17 to be switched. Transistors M1 and M2 are field-effect transistors with a drain d, a source s, and a control terminal / gate g. A control voltage Vgs (gate-source voltage) is generated by a control current Ic across a resistive element Rg.

[0039] The control current Ic is set by a control device 20, which includes a current mirror circuit 21 for setting the control current Ic. A logic circuit 22 can generate a pulse-width modulated output signal PWM using a switchable voltage source Vdd. The pulse-width modulated output signal PWM can be smoothed or its ripple reduced by a low-pass filter circuit 23. This provides a control signal Vpwm at the output of the low-pass filter circuit 23, which controls the current mirror circuit 21. The low-pass filter circuit 23 can include a capacitor Cpwm and, to provide an RC circuit, also an electrical resistor Rpwm.

[0040] The control signal Vpwm can be provided at the respective control terminals of two transistors Q1, Q2 of the current mirror circuit 21. The same control current Ic then flows to ground potential 14 through the resistors Re1, Re2 connected downstream of each transistor Q1, Q2.

[0041] Fig. Figure 2 illustrates how, by setting a stepped control signal Vpwm, the input current Is during a switch-on process 15 is set with a maximum 16 that is smaller than without the transistor circuit 18 (compare Fig. 1) . For example, the maximum 16 is less than 10% of the maximum that results without the transistor circuit 18.

[0042] Fig. Figure 3 illustrates how, by adjusting the duty cycle or duty ratio D of the pulse width modulation PWM, the control signal Vpwm is set in several steps 24' from a blocking value 25' (D = 0 %) to a switching value 26' (D = 100 %) and thereby the control voltage Vgs is set from a blocking value 25 (0 V) to a switching value 26 in several steps 24.

[0043] Once a stage 24 is reached, at which a switching threshold or threshold voltage Vgsthr is reached, the turn-on process 15 begins, i.e., transistors M1 and M2 become electrically conductive upon reaching the threshold voltage Vgsthr. Within this stage 24, or within two stages 24, the input current Is reaches its maximum 16 and then falls again. This means that a limiting phase 27 for the input current Is lasts at most one, two, or three times the stage duration Ts.

[0044] To set the stage duration Ts using the logic circuit 22, as described in Fig. Figure 4 illustrates how a control program 28 is executed by the switching logic 22. The switching logic 22 can, for example, include a microcontroller.

[0045] Starting from a start S10, the number of stages Dnum 24' and the stage duration Ts can be set in step S11. In step S12, the step size Ds = 100% / Dnum is set. In step S13, the control of the transistor circuit begins with the off state 24' (D = 0%). In step S14, the current stage is maintained for the stage duration Ts. In step S15, it is checked whether the on-state 26' has been reached. If this is the case (represented by a "+" symbol), the process is stopped in step S16, i.e., the on-resistance is maintained by keeping the on-state 26'. If the switching value 27' has not yet been reached in step S15 (represented by a "-" symbol), the next stage is set in step S17 by increasing the pulse width modulation PWM by increasing the duty cycle D by the step size Ds (D = D + DS) and continuing at step S14.

[0046] Fig. Figure 5 illustrates how the switch-on process 15 begins at different stages 24' for three different temperatures T1, T2, T3 (T1 > T2 > T3). The temperatures T1, T2, T3 were measured at transistors M1, M2. The respective temperature T1, T2, T3 thus describes a current operating state of at least one transistor M1, M2.

[0047] Fig. Figure 6 illustrates module 17 again for further description. Fig. 2. The electronic “resistance”, as provided by the transistor circuit 18 for the input current Is, is in the embodiment of Fig. Circuit 6 uses two P-channel MOSFETs as transistors M1 and M2 to eliminate the conductivity of their body diodes. For this purpose, the two source contacts of transistors M1 and M2 are electrically connected. A gate voltage lower than their source voltage is required to drive the two transistors M1 and M2. In other words, the gate potential (g) must be lower than the source contact potential (s). Therefore, this control voltage (Vgs) has a negative sign when the source contact (s) is used as the reference.

[0048] The control voltage Vgs is generated using an electrical resistance element Rg and a variable control current Ic, adjustable via software. For this purpose, the current mirror 21, consisting of transistors Q1 and Q2, is used. Transistors Q1 and Q2 are preferably provided by a dual transistor to eliminate temperature and transistor parameter fluctuations. Resistors Re1 and Re2 can also have the same value to act as local negative feedback elements, thus symmetryly balancing the current mirror 21.

[0049] Thus, the control current Ic, which flows through Re1 in the left leg and is reflected through Re2 in the right leg, is given by Ic=(Vpwn−Vbe) / Re1, where Vbe is the base-emitter voltage of transistor Q1.

[0050] The control signal Vpwm is generated from the pulsed PWM voltage source Vdd (typically 5 V or 3.3 V from a microcontroller output) using the low-pass filter circuit 23 (Rpwm / Cpwm) by varying the duty cycle D. It is calculated as: Vpwm=D⋅Vdd.

[0051] The control voltage Vgs for transistors M1, M2 is therefore: Vgs=((D⋅Vdd−Vbe)−Re1)⋅Rg and can be specified using the duty cycle D.

[0052] A time constant Tpwm of the low-pass filter or low-pass filter circuit 23 can be chosen such that the control voltage Vgs stabilizes after the change of the duty cycle step (D = D + DS) within the step duration Ts: Tset=6⋅Tpwm=6⋅Rpwm⋅Cpwm<0.5 Ts. Tset is the time until the steady state is reached after a change in the duty cycle D.

[0053] Ts is the duration of time spent in the respective duty cycle, i.e., the stage duration or the operating point.

[0054] Through this sweep or sequence of stages 24, a suitable operating point for the transistors of the transistor circuit 18 is always and under all circumstances found, at which the field-effect transistors are actively operated, i.e., with a through-resistance greater than the minimum through-resistance Rdson. Subsequently, upon reaching the switching value, they are guaranteed to operate with the minimum through-resistance, i.e., switched to saturated, low-impedance switching operation.

[0055] To limit a current surge during the switch-on process 15, the following diagram results as an idealized curve: Fig. 3.

[0056] Fig. Figure 5 illustrates a simulation with the three temperatures T1 = 150°C, T2 = 25°C, T3 = -40°C.

[0057] It is evident that thermal stability is also guaranteed thanks to the current imprinting and the control of the control current Ic by means of the current mirror 21. For the respective maximum 16, a deviation of a maximum of 2 A results.

[0058] The control program 28 allows the current limiting to be implemented compactly and in an unregulated, open-loop method, and can be adapted to different transistor types by simple modification or parameterization in step S11.

[0059] Fig. Figure 7 shows an embodiment of module 17 in which the transistors M1 and M2 of transistor circuit 18 are N-channel field-effect transistors. The N-channel field-effect transistors M1 and M2 have the advantage of a lower minimum on-resistance Rdson (channel resistance) at the same purchase price compared to P-channel field-effect transistors.

[0060] To enable transistors M1 and M2 to switch on, they require a gate voltage at gate g that is greater than the source voltage at source s. For this purpose, a second supply voltage Vhigh is provided in module 17, which is greater than the supply voltage Vs of voltage source 12 for module 17. The supply voltage Vhigh can be provided, for example, by a charge pump (H-bridge driver) or a boost converter, such as those used in LED lighting devices for vehicle headlights. Suitable integrated circuits (ICs) for this purpose are available in the state of the art.

[0061] Reverse polarity protection for the source terminal 11 can be provided as described by connecting the switching paths 19 of transistors M1 and M2 in series with the body diodes of transistors M1 and M2 connected in opposite directions. For this purpose, the sources s of the two transistors M1 and M2 can be connected together.

[0062] To drive the N-channel field-effect transistors M1 and M2, an additional current mirror 21' is provided with transistors Q3 and Q4, which transfer the control current Ic into the voltage range above the supply voltage Vs. The current mirror circuits 21 and 21' are connected in a cascade, meaning that the control current Ic of current mirror circuit 21 drives the current mirror circuit 21'.

[0063] The two transistors Q3 and Q4 are advantageously implemented in a single package, i.e., as a dual transistor, which results in the described advantages. The symmetry of resistors Re3 and Re4 enables local negative feedback to balance the current mirror 21'.

[0064] The rest of the circuit is analogous to the circuit of module 17 from Fig. 6 is structured, which is why the corresponding explanation of the reference symbols refers to Fig. 6 is referred to.

[0065] Fig. Figure 8 illustrates an embodiment of module 17, which corresponds to the embodiment of Fig. 6 corresponds, but the transistor circuit 18 has only a single transistor M1, through whose switching path 19 the input current Is is passed.

[0066] Fig. 9 shows one for Fig. 8 corresponding circuit with only one transistor M2, through whose switching path 19 the input current Is is passed.

[0067] Fig. Figure 10 shows a section of a module 17, which additionally features a switch element Moff for rapid shutdown. This switch element allows the capacitor Cpwm of the low-pass filter circuit 23 to be electrically connected to ground potential 14. When the switch element Moff is conducting, the capacitor Cpwm can be discharged via a discharge resistor Roff. This sets the control signal Vpwm to the blocking value 25', thus switching transistors M1 and M2 of the transistor circuit 18 (and, in the case of a single transistor, either transistor M1 or M2 of the transistor circuit 18) off. In the event of reverse polarity, this blocking switching should occur as quickly as possible; that is, the disconnection of the source connection 11 from the functional components 13 should occur with a time constant that is smaller than the time constant of the low-pass filter circuit 23 itself.Without the switching element Moff, transistors M1 and M2 of transistor circuit 18 remain electrically conductive until their gate voltage falls below the threshold voltage Vgsh, which can take more than 10 ms depending on the component values. To accelerate this turn-off process, the switching element Moff is provided, which is optionally connected to the positive electrode of capacitor Cpwm via a protective or discharge resistor Roff. This allows the discharge of capacitor Cpwm to be accelerated. The switching element Moff can be implemented as a transistor. For example, when using a microcontroller, which can also be used for logic circuit 22, it can be implemented as an open-drain microcontroller pin or open-drain output of the microcontroller.

[0068] Fig.Figure 11 illustrates a section of an embodiment of module 17 in which, instead of a logic circuit 22 with a pulse-width modulated output signal, an output signal Vdac of a digital-to-analog converter (DAC) can be used. Such an output signal Vdac can be generated by a microcontroller's digital-to-analog converter. The low-pass filter circuit 23 is then unnecessary. A protective resistor Rdac can be provided instead of the low-pass filter circuit 23. Using a digital-to-analog converter (DAC) also reduces the switch-off time compared to using a pulse-width modulated signal with a downstream low-pass filter circuit 23, since there is no time constant of an RC circuit (Cpwm).

[0069] The module offers the following advantages: Flexible adjustment of the inrush current amplitude of the input current Is via software: When manufacturing multiple modules, field-effect transistors with different FET parameters can be used (for example, with different threshold voltages Vgsh). Software parameterization using control program 28 is then possible and, in particular, less time-consuming than adapting an electronic circuit.

[0070] The circuit can also be used as reverse polarity protection: If the controlling microcontroller is supplied by its own diode (which is usually the case in practice), then the proposed circuit in the version with the transistor circuit with two transistors M1, M2 blocks in the case of reverse polarity and thus protects the downstream electronics of the functional components 13.

[0071] The circuit is also a nearly ideal switching switch with very low leakage current. This is important, for example, in automotive applications where leakage current would drain the vehicle battery. A transistor circuit with two transistors M1 and M2 completely isolates the voltage source 12 from the functional components 13, i.e., a load.

[0072] The gate-source voltage Vgs of transistors M1 and M2 is always limited: No protective elements (Zener diode, suppressor diode, also known as TVS diode (transient voltage suppression) or transorb) are required for this limitation, as are usually necessary to protect the gate g and source s of transistors M1 and M2 in a circuit. Gate g and source s are electrically connected via the resistor Rg (typically less than 2 kΩ), and the maximum voltage drop is that caused by the control current Ic.

[0073] Precise control current imprinting and transition thanks to the transistors in current mirror circuits: The current mirror 21, 21' is stable thanks to the use of a dual transistor and the regulated Vpm voltage source, which can be stably provided by a microcontroller. Consequently, the imprinted control current Ic is also stable, which ultimately leads to reproducible maxima 16 of the turn-on value of the input current Is.

[0074] The circuit is cost-efficient and economical thanks to the few components required and the use of an existing microcontroller.

[0075] Overall, the example shows how the invention can provide a method and circuit for limiting the inrush current of an electronic module. Reference symbol list 10 Module 11 Source connection 12 Voltage source 13 functional components 14 Mass potential 15 Switch-on process 16 Maximum Module 17 18 transistor circuit 19 Switching section 20 Control unit 21 Current Mirror 21' Current mirror circuit 22 Logic circuit 23 Low-pass filter circuit Level 24 24' step 25 blocking value 25' Locking value 26 switching value 26' switching value 27 Limitation phase 28 Tax program Is input current M1 Transistor M2 Transistor Q1 Transistor Q2 Transistor PWM output signal Vpwm control signal IC control current Vgs control voltage Cpwm capacitor Ts stage duration S10 step S11 step S12 step S13 Step S14 step S15 step S16 step S17 step Duty cycle T1 temperature T2 temperature T3 temperature DAC Digital-to-Analog Converter Cin input capacity g control connection Vdac control signal Rg resistance element Moff switch element Rload electronic functional components motor vehicle

Claims

[1] Electronic module (17) with a source terminal (11) for receiving an input current (Is) from an electrical voltage source (12) and with an electrical input capacitance (Cin) effective with respect to the source terminal (11), wherein the input capacitance (Cin) is connected to the source terminal (11) via a transistor circuit (18), wherein the transistor circuit (18) is configured to guide the input current (Is) through a respective switching path (19) of at least one transistor (M1, M2), wherein a control device (20) is configured to switch a control voltage (Vgs) at a respective control terminal (g) of the at least one transistor (M1, M2) in several stages (24) during a switch-on process (15) to limit the input current (Is), from a blocking value (25), at which each switching path (19) is blocked, to a conducting value (26), at which a through-resistance of each switching path (19) is minimized,where, without feedback, the same control voltage (Vgs) curve is always provided for each switch-on process (15), characterized by , that a respective time duration (Ts) of each of the stages (24) has a value corresponding to a time constant of the input capacitance (Cin) such that during one stage (24) or during two or three stages (24) the at least one transistor (M1, M2) transitions from a blocking state to an electrically conducting state and the input current (Is) rises to a maximum (16) and then falls again. [2] Module (17) according to claim 1, wherein the control device (20) is configured to set the stages (24) independently of a current operating state (T1, T2, T3) of the at least one transistor (M1, M2). [3] Module (17) according to one of the preceding claims, wherein the control device (20) comprises at least one current mirror circuit (21, 21') and the at least one current mirror circuit (21, 21') is configured to set a control current (Ic) in an electrical resistance element (Rg) depending on a control signal (Vpwm, Vdac), such that the control voltage (Vgs) for the at least one transistor (M1, M2) drops across the resistance element (Rg). [4] Module (17) according to claim 3, wherein a single current mirror circuit (21) is provided and the at least one transistor (M1, M2) is each a P-channel field-effect transistor. [5] Module (17) according to claim 3, wherein the at least one transistor (M1, M2) is each an N-channel field-effect transistor and two current mirror circuits (21, 21') are provided which are connected to form a cascade. [6] Module (17) according to one of claims 3 to 5, wherein the control device (20) comprises a switching logic (22) coupled to the at least one current mirror circuit (21, 21') and configured to specify the control signal (Vpwm, Vdac) for setting the control voltage (Vgs) by means of a control program (28). [7] Module (17) according to claim 6, wherein the control signal (Vpwm) is a pulse width modulated output signal (PWM) of the switching logic (22) guided via a low-pass filter circuit (23) or an output signal (Vdac) of a digital-to-analog converter (DAC) of the switching logic (22). [8] Module (17) according to one of claims 3 to 7, wherein for a rapid shutdown of the at least one transistor (M1, M2) an electrode of a capacitor (Cpwm), which is provided to reduce a ripple of the control signal (Vpwm), is connected to a ground potential (14) via a switching element (Moff) and the switching element (Moff) is configured to electrically connect the electrode to the ground potential (14) for a discharge of the capacitor (Cpwm) depending on a shutdown signal. [9] Module (17) according to one of the preceding claims, wherein in the transistor circuit (18) two transistors (M1, M2) are provided, the switching sections (19) of which are connected in series, wherein the forward directions of a respective body diode of the transistors (M1, M2) are connected in opposite directions. [10] Module (17) according to one of the preceding claims, wherein the input capacitance (Cin) is provided to a predominant part or completely by a further capacitor or a parallel connection of several capacitors. [11] Module (17) according to one of the preceding claims, wherein the at least one transistor (M1, M2) is connected in a positive line upstream of both the input capacitance (Cin) and other electronic functional components (Rload) of the module (17) as seen from the source terminal (11). [12] Motor vehicle (MV) with an electrical voltage source (12) and with an electronic module (17) according to one of the preceding claims. [13] Method for limiting an input current (Is) during a switch-on process (15) of an electronic module (17), wherein an electrical input capacitance (Cin) of the module (17) is connected via a transistor circuit (18) to a source terminal (11) of the module (17), wherein during the switch-on process (15) the input current (Is) is received via the source terminal (11) from an electrical voltage source (12) and the transistor circuit (18) guides the input current (Is) through a respective switching path (19) of at least one transistor (M1, M2), wherein a control device (20) for limiting the input current (Is) applies a control voltage (Vgs) to a respective control terminal (g) of the at least one transistor (M1, M2) in several stages (24) from a blocking value (25), at which each switching path (19) is blocked, to a conducting value (26), at which a through resistance of each switching path (19) minimized, switches,where, without feedback, the same control voltage (Vgs) curve is always provided for each switch-on process (15), characterized by , that a respective time duration (Ts) of each of the stages (24) has a value corresponding to a time constant of the input capacitance (Cin) such that during one stage (24) or during two or three stages (24) the at least one transistor (M1, M2) transitions from a blocking state to an electrically conducting state and the input current (Is) rises to a maximum (16) and then falls again.

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