Driving power switches with variable currents

Abstract

A current-source gate driver includes circuitry that is configured to determine a rate of change of a current flowing through a power switch and generate a signal indicative of the rate of change of the current flowing through a power switch and a variable current source configured to output a drive current to a control terminal of a power switch, wherein the variable current source is configured to decrease magnitude of the drive current in response to the signal indicating that a rate of change in the current flowing through the power switch is positive.

Description

[0001] DRIVING POWER SWITCHES WITH VARIABLE CURRENTS

[0002] This application claims the priority of US provisional application serial number 63 / 612,452 filed 20 Dec. 2023 and entitled “Driving Power Switches with Variable Currents,” the contents of which are incorporated herein by reference.

[0003] TECHNICAL FIELD

[0004] This invention relates to driving the control terminals (e.g., gates) of power transistors with variable currents.

[0005] BACKGROUND

[0006] Power switches are switching devices designed to switch relatively high voltages and currents. Examples of power switches include insulated gate bipolar transistors (IGBTs), metal-oxide-semiconductor field-effect transistors (MOSFETs), high-electron- mobility transistors (HEMTs), and the like. These power switches are commonly fabricated using semiconductor materials such as silicon, silicon carbide, gallium nitride, or other semiconductor materials. In some cases, individual power switches will include more than one physical transistor. For example, two transistors in a cascode configuration can form a single power switch.

[0007] In practice, power switches are generally associated with dedicated control, communication, and fault protection circuitry. For example, the provision of drive signals to a power switch can be handled by gate driver circuitry. Gate drivers provide an interface between the higher-power power switches and lower-power control circuitry. For example, gate drivers can receive lower power control, data, or other signals from a controller or microprocessor and provide drive signals to control terminals of power switches.

[0008] Other examples of circuitry that can be associated with power switches include desaturation (DESAT) protection circuitry, clamping circuitry, over current protection circuitry, and the like. In many cases, such circuitry is integrated with gate driver circuitry and they may be sold as a single commercial unit.

[0009] Gate drivers can generally be classified as voltage source gate drivers or current source gate drivers. In idealized form, voltage source gate drivers maintain a fixed output voltage regardless of load and the current provided to the control terminal of the power switch is a function of, e.g., that output voltage, the voltage at the control terminal of the power switch, and the impedance between the control terminal and the output of the voltage source gate driver. In the context of insulated gate bipolar transistor power switches, the control terminal is a gate and the impedance is referred to as a gate resistor, a gate turn-on or turn-off resistor, or the like. As an aside, although ideal voltage source gate drivers do not exist, real-world devices are generally capable of mimicking idealized behavior over a range of operational conditions.

[0010] In idealized form, current source gate drivers maintain a fixed output current that is — in theory — independent of the voltage at the control terminal of the power switch. Once again, although ideal current source gate drivers do not exist, real-world devices are generally capable of mimicking idealized behavior over a range of operational conditions.

[0011] DESCRIPTION OF DRAWINGS

[0012] FIG. 1 is a schematic representation of circuitry in which the control terminal of an IGBT power switch is driven by a current-source driver that includes a variable current source.

[0013] FIG. 2 includes schematic representations of time traces of various waveforms that arise during driving of the gate of an IGBT power switch with a variable current source during a transition from the OFF state to the ON state.

[0014] FIG. 3 is a schematic representation of circuitry in which the control terminal of an IGBT power switch is driven by a current-source driver that includes a variable current source.

[0015] FIG. 4 is a schematic representation of one circuit in which the driving of IGBT switch could damage another component.

[0016] Like reference symbols in the various drawings indicate like elements.

[0017] For didactic purposes, the detailed description is cast in terms of insulated gate bipolar transistor (IGBT) power switches and their terminals (i.e., gates, collectors, and emitters). However, corresponding teachings can be applied to a variety different enhancement- or depletion-mode devices (e.g., bipolar junction transistors (BJTs), metal- oxide-semiconductor field-effect transistors (MOSFETs), or high-electron-mobility transistors (HEMTs)) with electron or hole charge carriers, alone or in combination with other transistors. The devices can be implemented in silicon, silicon carbide, gallium nitride, or other semiconductor materials.

[0018] DETAILED DESCRIPTION

[0019] FIG. l is a schematic representation of circuitry 100 in which the control terminal of an IGBT power switch 105 is driven by a current-source driver 150 that includes a variable current source 110. The current with which variable current source 110 drives IGBT power transistor 105 is varied in response to a detected rate of change of the emitter current IE 115. In particular, as discussed further below, the current that drives IGBT switch 105 is varied to ensure that rate of change of the emitter current IE 115 is slowed to reduce the “hardness” of switching or commutation of components that are coupled to circuitry 100.

[0020] In more detail, diode 120 is coupled between the collector and emitter of IGBT switch 105. Diode 120 can be integrated with IGBT switch 105 (i.e., in a “co-pack”) or diode 120 can be a discrete component that is coupled across IGBT switch 105. In either case, diode 120 imitates the body diode of a MOSFET and provides a freewheeling path for reverse current flow.

[0021] The schematically-represented implementation of IGBT power switch 105 includes both a main emitter 125 and an auxiliary emitter 130. Main emitter 125 is configured to output emitter current IE 115 from IGBT power switch 105 to a load. In power switching applications, emitter current IE 115 is quite large and even small parasitics can give rise to relatively large differences between the voltage at the main emitter 125 and the voltage at the emitter of the IGBT semiconductor chip. In the schematic representation of FIG. 1, some of these parasitics are represented as a parasitic inductance LE 135.

[0022] In contrast with main emitter 125, auxiliary emitter 130 is not intended to output emitter current IE 115 to a load. Rather, auxiliary emitter 130 is coupled closer to the semiconductor material of the IGBT chip than main emitter 125 and thus presents a relatively lower distortion connection to the emitter region of the IGBT chip. Auxiliary emitter 130 can be used by driver and measurement circuitry to “see” the voltage at the emitter region in the semiconductor material of the IGBT chip without the distortion caused by emitter current IE 115 on main emitter 125.

[0023] In general, main emitter 125 and auxiliary emitter 130 will be separate terminals of an IGBT module. IGBT modules generally mount one or more IGBT chips over a baseplate that is molded to connection terminals. The metallization on the IGBT chips — including the base, emitter, and collector of the semiconductor device— is wired (typically, bond-wired) to the terminals. Dimensions are kept small to reduce internal wiring inductance and other parasitics. In implementations where main emitter 125 and auxiliary emitter 130 are terminals of an IGBT module, parasitic inductance LE 135 includes this internal wiring inductance.

[0024] In addition to variable current source 110, current-source driver 150 also includes circuitry 140 that is configured to detect the rate of change of emitter current IE 115. Circuitry 140 is coupled to main emitter 125 and auxiliary emitter 130. As discussed further below, circuitry 140 is configured to detect the changes in emitter current IE that occur as IGBT power switch 105 switches into and out of conduction. In response to detection of these changes, circuitry 140 varies the current with which variable current source 110 drives IGBT power switch 105. By varying the current that drives IGBT power switch 105, circuitry 140 protects components from large transients that can arise due to “hard” switching of emitter current IE. Although circuitry 140 is illustrated as entirely contained within current-source driver 150, in some implementations, all or a portion of circuitry 140 can be external to current-source driver 150.

[0025] FIG. 2 includes schematic representations of time traces of various waveforms 205, 210, 215, 220 that arise during driving of the gate of an IGBT power switch with a variable current source during a transition from the OFF state to the ON state. As discussed further below, waveforms 205, 210, 215, 220 illustrate switching in which the forward current flow in at least one other diode is interrupted at the same time that this IGBT power switch turns on. For example, when IGBT power switches are assembled in a half-bridge, the anti-parallel or freewheeling diode of another IGBT power switch could be interrupted at the same time that this IGBT power switch turns on.

[0026] Regardless of the nature and number of other diodes, the reverse recovery current may flow in the same direction as the current in the IGBT power switch that is turning on. The IGBT power switch that is turning on could transiently conduct both the current that is intended for the load as well as the reverse recovery current of the other diode. Without additional protective measures (e.g., large turn-on gate resistor), the rise time (i.e., the change) of the reverse recovery current could damage the other diode(s). However, by driving the control terminal of the IGBT power switch that is turning on with a variable current source, the IGBT power switch can be turned on relatively quickly while also avoiding damage to the other diode(s) without additional protective measures.

[0027] In more detail, waveform 205 is the current with which a variable current source drives the gate of the IGBT power switch during the transition. Vertical position of waveform 205 represents the magnitude of that current with respect to a zero current at dashed line 206.

[0028] Waveform 210 is the voltage that arises between the gate and the emitter of the IGBT power switch, i.e., the gate-to- emitter voltage VGE. Vertical position of waveform 210 represents the magnitude of that voltage with respect to an OFF state gate-to- emitter voltage at dashed line 211. OFF state gate-to-emitter voltages in IGBT devices can be zero or negative (for example, -5 to -10 volts is common) and ON state gate-to-emitter voltages are typically below 30 volts (for example, 15-20 volts is common).

[0029] Waveform 215 is the emitter current IE that exits the (main) emitter of the IGBT power switch. Vertical position of waveform 215 represents the magnitude of emitter current IE with respect to a zero current at dashed line 216. In the OFF state, the emitter current IE is essentially zero.

[0030] Waveform 220 is the voltage across the IGBT power switch, i.e., the collector-to- emitter voltage VCE. Vertical position of waveform 220 represents the magnitude of that voltage with respect to a zero voltage value at dashed line 221. In general, in power switching applications, the collector-to-emitter voltage can range between several hundreds of volts in the OFF state to essentially zero volts in the ON state. For example, in IGBT devices, the collector-to-emitter voltage in the ON state will be a diode-drop plus any voltage drop due the physical limitations of the device.

[0031] The vertical scales of waveforms 205, 210, 215, 220 are thus quite different.

[0032] For didactic purposes, the transition from the OFF state to the ON state can be divided into four different stages in time. In the schematic representation of waveforms 205, 210, 215, 220, these stages are demarcated by vertical lines 225, 230, 235, 240, 245 that designate different times.

[0033] In the OFF state (i.e., before the OFF-to-ON transition), the collector-to-emitter voltage VCE is high (e.g., several hundreds of volts) and the emitter current IE remains essentially zero.

[0034] Time 225 represents the start of first stage of the transition from the OFF state to the ON state. At time 225, the variable current source starts to drive the gate of the IGBT power switch with a gate current of magnitude IG1, as represented by waveform 205. The transition from zero gate current before time 225 to the gate current is schematically represented in waveform 205 as a vertical line. In real-world devices, the transition will have a finite slope that changes with time.

[0035] With the injection of the gate current into the gate beginning, the gate-to-emitter voltage VGE will also begin to rise at time 225, as represented by waveform 210. As represented by waveforms 205, 210, injection of gate current will continue throughout the first stage and the gate-to-emitter voltage VGE will continue to increase. In this first stage, the magnitude of the gate current is schematically represented as constant and the slope of the gate-to-emitter voltage VGE in waveform 210 is schematically represented as a line segment with a constant slope. In real-world devices, deviations will generally occur.

[0036] Time 230 represents the end of the first stage and the start of the second stage of the OFF-to-ON transition. Time 230 designates the time when the gate-to-emitter voltage in waveform 210 reaches the threshold voltage VTH of the IGBT device. Threshold voltage VTH is the minimum gate-to-emitter voltage that creates a conductive path between the collector and emitter terminals of the IGBT device such that emitter current IE can flow. In general, the threshold voltage VTH of different IGBT devices will vary somewhat from device to device. As waveform 210 reaches the threshold voltage VTH of the particular IGBT device to which it is coupled, the collector-to-emitter voltage VCE starts to drive the flow of emitter current IE (as represented by waveform 215) and the collector-to-emitter voltage VCE starts dropping (as represented by waveform 220).

[0037] The rate of change in the emitter current IE can be detected and used to trigger a decrease in the gate current with which a variable current source drives the gate of the IGBT power switch during the transition. Referring to FIG. 1, the rate of change in the emitter current IE can be detected by circuitry 140. As represented in waveform 205, the variable current source decreases the gate current from a first magnitude IG1 to a second magnitude IG2. The transition between magnitudes IG1 and IG2 is schematically represented as a vertical line and the magnitudes of both IG1 and IG2 are represented as constant. In real-world devices, deviations will generally occur. Further, in some implementations, the magnitude of one or both of IG1 and IG2 can vary over time, e.g., between discrete values or along a continuous path. However, the magnitude of the gate current will be lowered in the second stage.

[0038] Also in the second stage, represented in waveform 210, the rate of increase in the gate-to-emitter voltage in waveform 210 decreases. This is due at least in part to the variable current source driving the gate of the IGBT device with the lower magnitude IG2 gate current. Other factors will also influence the rate of increase in the gate-to-emitter voltage, including charging of the input capacitance of the IGBT device.

[0039] During the second stage, the magnitude of the emitter current IE reflects both the magnitude of the current that flows to the load and an excess current 250. The total emitter current IE may even rise to a level that is higher than the load current ILOAD with the IGBT device fully conductive. As an aside, for illustrative purposes, the load current ILOAD with the IGBT device in the ON state is shown as having a steady state value, as would be the case with a constant load. This is not necessarily the case and variable loads with variable load currents ILOAD can also be powered.

[0040] This excess 250 is a reverse recovery current that flows through one or more other diodes and is transiently conducted by the IGBT device. However, as the other diode(s) recover, the excess current 250 drops. As illustrated by curve 215, the rate of change in the emitter current IE transitions from a relatively high value to a zero and then negative value as the current flowing to the load approaches its steady state value and the reverse recovery current drops.

[0041] In general, the same circuitry that detects the high rate of change in the emitter current IE and triggers the decrease in the gate current from magnitude IG1 to IG2 can also detect the zero or negative rate of change in the emitter current IE and triggers an increase in the gate current from magnitude IG2 to, e.g., a magnitude IG3, thus ending the second stage at time 235. For example, the circuitry can detect the high rate of change in the emitter current IE is zero, i.e., that the peak emitter current IE has been reached. Referring to FIG. 1 , the rate of change in the emitter current IE can be detected by circuitry 140. In other implementations, the end of the second stage is defined, e.g., by measuring a passage of time.

[0042] Regardless of how the second stage is ended, the third stage of the OFF-to-ON transition starts at time 235. The gate current is increased from magnitude IG2 to IG3 by the variable current source, as shown in waveform 205. As shown in waveform 210, the gate voltage resumes its rise and approaches its maximum value VGMAX as the input capacitance of the IGBT device charges. In some cases, IG3 can equal IG1, but this is not necessarily the case. Further, in general and as shown in waveform 210, the gate voltage will approach a maximum value that reflects, e.g., the magnitude of the voltage that supplies the variable current source 110. In many implementations, the magnitude of the gate current will fall below the magnitude of IG2 and approach zero during the third stage and thereafter.

[0043] With the reverse recovery current dropping, the emitter current IE approaches the steady-state load current ILOAD with the IGBT device fully conductive. The rate of decrease in the collector-to-emitter voltage VCE increases, as represented by waveform 220. During the third stage, the collector-to-emitter voltage VCE will eventually fall to a nearly zero voltage value at dashed line 221.

[0044] The fourth stage of the OFF-to-ON transition starts at time 240. As shown in waveform 210, the gate voltage has approached its maximum value VGMAX and the gate current falls to nearly zero, as shown in waveform 205. With the gate fully charged, the IGBT device is fully conductive and the emitter current IE remains at the load current ILOAD. Also, the collector-to-emitter voltage VCE remains at a nearly zero voltage value at dashed line 221.

[0045] FIG. 3 is a schematic representation of circuitry 300 in which the control terminal of an IGBT power switch 105 is driven by a current-source driver 150 that includes a variable current source 110. Circuitry 300 is one possible implementation of circuitry 100 (FIG. 1) and, like circuitry 100, drives IGBT switch 105 with a current that is varied to ensure that rate of change of the emitter current IE 115 is slowed to reduce the “hardness” of switching or commutation of components that are coupled to circuitry 100.

[0046] In addition to current- source driver 150, circuitry 300 includes interface circuitry 305, 310 that is coupled between a sense input Snsl of current-source driver 150 and main emitter 125 of IGBT power switch 105. Circuitry 305 is a voltage divider and low- pass filter stage and configured to pass a low-pass filtered version of the voltage on main emitter 125 of IGBT power switch 105 to a sense input of current- source driver 150. In the illustrated implementation, the low- pass filter stage of circuitry 305 is implemented by resistances R4, R5 and a capacitance C2. At low frequencies, the voltage VC2 across R5 and C2 is a divided version of the voltage on main emitter 125. The particular values of resistances R4, R5 and a capacitance C2 — and the behavior of the low-pass filter — can be adjusted to the application context.

[0047] Circuitry 310 is an input stage of a differentiator 315. Differentiator 315 is circuitry that is configured to produce an output that represents the rate of change of the input. A positive change in the input into differentiator 315 with time results in a positive output. A negative or zero change in the input with time causes the output of differentiator 315 to be zero or negative. In addition to the input stage of circuitry 310, the illustrated implementation of differentiator 315 also includes a feedback network 320 and an operational amplifier 325. Together, input stage circuitry 310 and feedback network 320 define the response of differentiator 315 to changes in the voltage on main emitter 125. In the illustrated implementation, input stage circuitry 310 includes a resistance R6 and a capacitance C3 and feedback network 320 includes a resistance R7 and a capacitance C4. Capacitance C3 isolates differentiator 315 from the DC component of the voltage on main emitter 125. In the illustrated implementation, input stage circuitry 310 is external to current-source driver 150 and the particular values of resistance R6 and a capacitance C3 — as well as the response of differentiator 315 at different frequencies — can be adjusted to the application context.

[0048] In addition to feedback network 320 and amplifier 325, circuitry 140 also includes a second sense input Sns2, a buffer 330, and an AND gate 335. Second sense input Sns2 is coupled to the auxiliary emitter 130 of IGBT power switch 105. In the illustrated implementation, second sense input Sns2 is coupled to a reference node. The voltage at second sense input Sns2 may serve as a reference for some or all of the circuitry in current-source driver 150. For example, in the illustrated implementation, negative voltage VNEG can be defined with respect to the reference voltage at second sense input Sns2. For example, in some implementations, current-source driver 150 includes a voltage source (e.g., a battery or a power converter, not shown) that defines negative voltage VNEG with respect to the reference voltage at second sense input Sns2.

[0049] In the illustrated implementation, negative voltage VNEG is coupled to the noninverting input of amplifier 325 and provides a reference against which the rate of change of the voltage on the inverting input of amplifier 325 is compared. Since negative voltage VNEG itself is referenced to the voltage on the auxiliary emitter 130 and the voltage on the inverting input of amplifier 325 represents the changing component of voltage on the main emitter 135, differentiator 315 differentiates the voltage that arises across parasitic inductance LE 135. Even though parasitic inductance LE 135 is quite small in value, changes in the emitter current IE 115 with time can be isolated — even in the context of a high power switching environment. In particular, voltage swings that are due to factors other than the changes in the emitter current IE 115 will generally impact the voltage on the auxiliary emitter 130 and the voltage on the main emitter 135 to nearly the same extent. However, the voltage that arises across parasitic inductance LE 135 is impacted most by changes in the emitter current IE 115.

[0050] Further, by using parasitic inductance LE 135 as a sense inductor, power losses due to other sense components are avoided.

[0051] Buffer 330 is coupled to the output of differentiator 315 and configured to decouple that output from AND gate 335. The inputs of AND gate 335 are coupled to receive the buffered output of differentiator 315 and a drive state signal 340. Drive state signal 340 can be, e.g., a pulse- width modulation control signal or other signal that indicates when IGBT power switch 105 is to be in the ON state. In the illustrated implementation, a high drive state signal 340 indicates that IGBT power switch 105 is to be in the ON state. In the event that both the IGBT power switch 105 is to be in the ON state and the voltage difference between main emitter 125 of IGBT power switch and the auxiliary emitter 130 increases (i.e., the di / dt of load current 115 is positive), then differentiator 315 and AND gate 335 will output a high state signal that is received by variable current source 110.

[0052] Both drive state signal 340 and the signal that is output from AND gate 335 are coupled to variable current source 110. Variable current source 110 is configured to respond to drive state signal 340 being in a state that indicates that IGBT power switch 105 is to be in the ON state by driving the gate of the IGBT power switch 105 with a gate current that has first magnitude IG1 (FIG. 2). Variable current source 110 is configured to respond to the drive state signal 340 being in a state that indicates that IGBT power switch 105 is to be in the ON state and the signal that is output from AND gate 335 indicating that the di / dt of the load current 115 is positive by driving the gate of the IGBT power switch 105 with a gate current that has second magnitude IG2 (FIG. 2).

[0053] In the implementation illustrated in FIG. 2, variable current source 110 will continue to drive the gate of the IGBT power switch 105 with a gate current that has constant second magnitude IG2 throughout the second stage. As mentioned above, this is not necessarily the case. However, at some point, the rate of change in the emitter current IE will transition from a relatively high value (i.e., positive) to a zero and then negative value as the current flowing to the load approaches its steady state value and the reverse recovery current drops. Differentiator 315 will respond by outputting a low state signal that changes the logic state of the output of AND gate 335. The changed logic state is received by variable current source 110, which responds by driving the gate of the IGBT power switch 105 with a gate current that has a higher magnitude, e.g., IG3.

[0054] FIG. 4 is a schematic representation of one circuit in which the driving of IGBT switch 105 could damage another component, namely, a variable-frequency drive in which an inverter 400 is coupled to provide power to a three-phase motor 405. Inverter 400 includes half-bridge circuits 401, 402, 403 and a collection of current-source drivers 150a, 150b, ... For the sake of illustration, only current-source drivers 150a, 150b that drive the IGBTs in half-bridge circuit 401 are shown. Three-phase motor 405 includes a set of motor winding inductances L1-L3.

[0055] Each half-bridge circuit 401, 402, 403 couples and decouples a respective motor winding inductance L1-L3 to a high or low rail in phase to drive motor 405. In an illustrative example, during a first phase, the high side IGBT power switch 105b in halfbridge circuit 401 and a low side IGBT power switch in half-bridge circuit 402 will be conductive and current will flow from the high rail to the low rail through winding inductances LI, L2. To transition to a second phase, these high side and low side IGBT power switches will be turned off and another high side and low side IGBT power switch will be turned on.

[0056] With these high side and low side IGBT power switches being turned off, the winding inductances LI, L2 will resist demagnetization and the end of current flow by forcing the current through the low-side diode of 401 and the high-side diode of 402. Once, the high side and low side IGBT power switches are turned-on again, a reverse recovery current may flow through diode 120a in half-bridge circuit 401. Further, this reverse recovery current will flow through high side IGBT power switch 105a in halfbridge circuit 401, adding to the emitter current IE in high side IGBT power switch 105b. Such an addition of currents is illustrated in waveform 215 (FIG. 2) and can be sensed by comparing the voltages of the main emitter and auxiliary emitter of high side IGBT power switch 105b, as discussed above.

[0057] A number of implementations have been described. Nevertheless, it will be understood that various modifications may be made. For example, multiple gate current magnitudes may be used. As another example, variable current sources and main emitter and auxiliary emitter voltage sensing may be used to control the transitions of IGBT power switches from an ON state to an OFF state.

[0058] As yet another example, a voltage source gate driver can be used to achieve variable gate drive currents. In particular, instead of switching between currents, different gate turn-on resistances can be used in conjunction with a voltage source gate driver to achieve variable driving currents. For example, in the first stage, the gate turnon resistance could have a value R, leading to a relatively high gate current and fast switching. In the second stage, the gate turn-on resistance could have a value 4R, leading to a relatively lower gate current and slower switching. In the third stage, the gate turnon resistance could have a value R or lower (e.g., zero gate turn-on resistance). As with a current source gate driver, the rate of change in the current flowing through the power switch can be used to trigger the transitions between stages.

[0059] Accordingly, other implementations are within the scope of the following claims.

Claims

WHAT IS CLAIMED IS:

1. A driver for the control terminal of a power switch, the driver comprising: di / dt sense circuitry that is configured to sense a rate of change of a current flowing through the power switch and generate a signal indicative of the sensed rate of change of the current flowing through the power switch; and driver circuitry configured to output a variable drive current to a control terminal of the power switch, wherein the driver is configured to decrease magnitude of the drive current in response to the signal indicating that the sensed rate of change in the current flowing through the power switch is positive.

2. The gate driver of claim 1, wherein the circuitry comprises: a first input coupled to a first sense terminal of the driver circuitry; and a second input coupled to a second sense terminal of the driver circuitry, wherein the di / dt sense circuitry is configured to detect a difference in a rate of change of respective voltages at the first and second sense terminals.

3. The gate driver of claim 2, wherein the di / dt sense circuitry includes a differentiator configured to differentiate the voltage at the first input with reference to the voltage at the second input.

4. The gate driver of claim 1, wherein the driver circuitry is configured to increase magnitude of the drive current in response to the signal indicating that the rate of change in the current flowing through the power switch is zero or negative.

5. The gate driver of any preceding claim, wherein the driver circuitry includes a variable current source.

6. The gate driver of any of claims 1 to 4, wherein the driver circuitry includes a voltage source and a variable gate turn-on resistance.

7. A power converter comprising: the gate driver of any preceding claim; and a power switch module, the power switch module comprising an inductance that impedes main current flow through the power switch module, wherein the gate driver iscoupled to sense a voltage difference across the inductance.

8. The power converter of claim 7, wherein the inductance is a parasitic inductance.

9. The power converter of claim 7, wherein the power switch module comprises: a main output terminal configured to output current to a load, and an auxiliary output terminal coupled closer to semiconductor material of the power switch, wherein the parasitic inductance is coupled between the main output terminal and the auxiliary output terminal.

10. The power converter of claim 7, wherein the power switch module comprises an IGBT power switch.

11. A power converter comprising: an IGBT power switch module comprising a gate, a main emitter configured to output emitter current to a load, an auxiliary emitter coupled closer to semiconductor material of the IGBT chip, and an inductance between the main emitter and the auxiliary emitter; and a current-source gate driver comprising a variable current source coupled to output current the gate of the IGBT power switch, a first sense input coupled to the main emitter, a second sense input coupled to the auxiliary emitter, circuitry configured to differentiate a voltage on the main emitter and output a signal indicative of a result of the differentiation, wherein the variable current source is coupled to receive the signal indicative of the result of the differentiation and configured to vary the output current in response.

12. The power converter of claim 11, wherein the inductance is a parasitic inductance.

13. The power converter of claim 11 or 12, wherein the variable current source is configured to decrease the output current in response to the signal indicating that the di / dt of the emitter (load) current has a positive gradient.

14. The power converter of any one of claims 11 to 13, wherein the variable current source is configured to increase the output current in response to the signal indicating that the di / dt of the emitter (load) current has a zero or negative gradient.

15. An inverter comprising the power converter of any one of claims 7 to 14.

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