Single-stage current feed type bidirectional DC-AC converter and fixed frequency control method thereof

Abstract

The invention belongs to the technical field of DC-AC converters, and relates to a single-stage current feed type bidirectional DC-AC converter and a fixed frequency control method thereof.The single-stage current feed type bidirectional DC-AC converter comprises a DC side composed of two Boost half bridges with inductors, an AC side composed of half bridges and an AC side filter, a transformer and a control device; the control device is used for controlling the direct current side and the alternating current side to describe the power transmission characteristics of the circuit by transconductance Gm, and the current control of the alternating current side is realized by a feedforward method; fixed-frequency linear phase shift control can be carried out in an ultra-wide DC side voltage range, bidirectional real-time power flow control can be realized in a relatively wide DC side voltage range, and a complex control loop design theory is not needed; therefore, the cost and the power consumption of the controller are effectively reduced, and a higher power density upper limit and a higher efficiency upper limit are achieved.

Description

Technical Field

[0001] The invention belongs to the technical field of DC-AC converters and relates to a single-stage current-fed bidirectional DC-AC converter and a fixed-frequency control method thereof. Background Art

[0002] Small DC-AC converters provide a connection method from low-voltage DC voltage (level between 20V-60V) to grid voltage (220VAC), and are widely used in photovoltaic micro-inverters, vehicle battery chargers (below 60V), UPS (uninterruptible) power supply systems and home outdoor emergency power supplies. In order to provide sufficient safety for individual users, electrical isolation is required in this type of DC-AC converter. Since the DC-AC converter needs to have both boost and DC-AC conversion functions, a two-stage conversion architecture was mostly used in early solutions. This architecture is similar to the non-isolated string inverter solution, the difference is that the non-isolated Boost converter is replaced by an isolated DC-DC converter (such as LLC, DAB) converter. The architectural solutions of DC-AC converters in the prior art are classified as follows: Figure 1 shown.

[0003] The quasi-single-stage architecture is based on a two-stage structure, and realizes DC-AC conversion without a high-voltage DC bus by controlling the sinusoidal output voltage of the isolated DC-DC. The single-stage architecture is the focus of current research. The flyback single-stage DC / AC topology was widely used in early micro-inverter solutions. However, due to its low transformer utilization, its power level cannot meet the needs of existing single photovoltaic panels (above 400W). In addition, applications such as household distributed energy storage systems and on-board chargers require bidirectional DC-AC power transmission from battery to grid and AC-DC power transmission from grid to battery. Therefore, a single-stage high-frequency AC chain solution with bidirectional power flow capability has begun to become the solution selected in mainstream research and products. Its characteristic is that energy is transmitted through the phase difference of the high-frequency AC voltage square wave on the high-frequency transformer. Specifically, it realizes AC working condition application by using a bidirectional switch on the full bridge on the AC side based on the DAB converter.

[0004] Under low-voltage DC input, the DAB phase-shift solution, in which the AC side structure is simplified to a half-bridge, is more cost-effective. It is similar to the core control concept of the traditional DAB DC-DC converter, both of which use phase-shift modulation to directly control the flow power.

[0005] However, the DC-AC converter in the prior art has the following defects:

[0006] 1. Although the two-stage solution is mature in technology and simple to control, it has the disadvantages of complex circuit structure, high cost, and a low upper limit in power density and efficiency.

[0007] 2. The quasi-single-stage solution has a certain optimization in efficiency and cost compared with the former because of the reduction in the number of high-frequency switching devices and decoupling capacitors. However, the number of devices has not been reduced, so the power density of the converter has no obvious advantage over the two-stage solution. Moreover, the same wide voltage gain operating range as the single-stage solution makes it significantly more difficult to implement than the two-stage solution.

[0008] 3. Single-stage solution (flyback), although simple in structure and low in cost, is limited by the non-symmetrical transformer structure and prominent high voltage and current stress issues, and the power level can no longer meet current requirements. In addition, bidirectional power flow cannot be achieved in this technical solution, which means that its application scope is limited to unidirectional DC-AC conversion scenarios such as photovoltaic micro inverters.

[0009] 4. The high-frequency AC chain solution characterized by a cycloconverter AC full bridge (12 devices in total) has too many components and faces the same dilemma as the quasi-single-stage solution. In addition, for low-voltage scenarios, the cost is too high and the control is complex.

[0010] 5. The dual active bridge (DAB) scheme (8 devices in total) for real-time power control using phase-shift modulation is currently dominated by the voltage-fed DAB circuit topology. However, the DAB voltage gain, which changes periodically from 0 to 0, makes the control of the converter difficult. In addition, the voltage-fed DAB scheme has a discontinuous input current and a narrow voltage range on the DC side - a disadvantage when photovoltaic panels or batteries are connected.

[0011] Therefore, a DC-AC converter with a simple circuit topology, a simple control scheme, and a wider soft switching operating range and a control method thereof are needed to solve the above technical problems. Summary of the invention

[0012] The purpose of the present invention is to propose a current-fed DAB solution to address the problems of the prior art. The topology of the present invention uses two independent half-bridges instead of a full bridge on the DC side to avoid circulating current on the DC input side under the proposed fixed-frequency high-linearity control strategy. The solution of the present invention can achieve an extremely high DC side voltage application range, which is twice that of the voltage-fed solution.

[0013] The technical solution adopted by the present invention to solve the technical problem is: a single-stage current-fed bidirectional DC-AC converter, comprising:

[0014] DC side, the input end of the DC side is connected to the DC bus v dc ; The DC side is composed of two Boost half-bridges with inductors; the Boost half-bridges include: a first DC side half-bridge and a second DC side half-bridge;

[0015] On the AC side, the output end of the AC side is connected to the AC bus vac ; The AC side is composed of an AC side half bridge and an AC side filter; the AC side half bridge includes: an AC side bidirectional half bridge;

[0016] A transformer, wherein the primary side thereof is connected to the output terminal of the DC side, and the secondary side thereof is connected to the input terminal of the AC side; the transformer is a high frequency transformer, and the transformation ratio of the transformer is 1:n;

[0017] Control device, the control device is used to control the DC side and the AC side to transconductance G m Describe the power transmission characteristics of the circuit, and implement the AC side current control by a feedforward method; the control sampling signals of the control device include: DC side port voltage, DC side inductor current, AC side port current and port voltage.

[0018] Preferably, the DC side includes: switching transistors Q1-Q4, an inductor L dc1 and L dc2 , output capacitor C dc1 and C dc2 , inductance L dc1 is the first DC side filter inductor, inductor L dc2 is the second DC side filter inductor, the first DC side half bridge is composed of switching transistors Q1 and Q2, the second DC side half bridge is composed of switching transistors Q3 and Q4, and the output capacitor C dc1 is the first DC side clamping capacitor, output capacitor C dc2 The second DC side clamp capacitor is connected as follows: DC busbar v dc The positive input terminal is connected to the inductor L dc1 and L dc2 The input terminal, inductor L dc1 The output end is connected to one end of the primary side of the transformer, and the other end of the primary side of the transformer is connected to the inductor L dc2 The output end of the switch transistors Q1 and Q2 are connected in series to the output capacitors C dc1 The connection node of the switching transistors Q1 and Q2 is connected to the inductor L dc1 The output end of the switch transistors Q3 and Q4 are connected in series to the output capacitors C dc2 The connection node of the switching transistors Q3 and Q4 is connected to the inductor L dc2 Output terminal; output capacitor C dc1 and C dc2 The negative pole is connected to the DC bus v dc The negative electrode;

[0019] The AC side includes: switching transistors S1~S4, inductor L ac and L lk , Series capacitor C ac1 and Cac2 , inductance L ac is the AC side filter inductor, inductor L lk is the transformer cascade inductor, capacitor C ac1 and C ac2 The AC side is connected in series with the capacitor, and the AC side bidirectional half bridge is composed of switching transistors S1~S4; the connection method of the AC side is: AC bus v ac The positive input terminal is connected in series with the inductor L ac , Series capacitor C ac1 , Series capacitor C ac2 Then connected to the AC bus v ac The negative input terminal of the switching transistors S1 to S4 is connected in series in sequence, and the two ends are connected to the inductor L ac With the series capacitor C ac1 The connection node, AC busbar v ac The negative input terminal of the switching transistor S2 and the switching transistor S4 are connected in series with the inductor L lk Then connected to one end of the secondary side of the transformer, and the other end of the secondary side of the transformer is connected to the series capacitor C ac1 and C ac2 The mixed frequency band filter circuit filters out DC, 100 Hz and frequency components above 200 kHz.

[0020] The present invention also discloses a fixed frequency control method for a single-stage current-fed bidirectional DC-AC converter, which is used to control the above-mentioned DC-AC converter. The control method includes: dividing the converter into 8 regions according to the DC side duty cycle D and the phase shift ratio φ and mainly operating on the linear forward maximum power output trajectory located in the II region and the IV region; the control quantity relationship is expressed as:

[0021]

[0022] In formula (4), D represents the duty cycle of the DC side device, and φ represents the phase shift angle ratio.

[0023] Preferably, the converter is divided into 8 regions according to the DC side duty cycle D and the phase shift ratio φ and completely operates on a continuous bidirectional trajectory located in regions II, III and IV, so as to obtain a minimum duty cycle D based on the minimum duty cycle D. min The phase-shift control mode is used as a substitute for the PWM mode in low-power transmission to improve the low-power control accuracy.

[0024] More preferably, the control method uses a transconductance G m The complete and unique control trajectory over the wide DC link voltage range described is expressed as:

[0025]

[0026] In formula (6), k m Represents the characteristic coefficient of the circuit.

[0027] More preferably, in the control method, the converter is G m The control trajectory is used as the core to realize the AC current control loop combining feedforward and feedback, which specifically includes the following steps:

[0028] The first step is to use the PLL phase-locked loop to obtain the phase reference of the grid voltage and then generate the control setting current reference i ac,ref ; Then a small correction is made through the PR controller to obtain G m The current reference value required by the controller;

[0029] The second step is to obtain the real-time control transconductance G according to the input current reference value and the sampled DC side voltage value. m value; the polarity of this value is determined by the grid voltage v ac The polarity of

[0030] The third step is to obtain the control quantity D and φ and generate the drive PWM of Q1 to Q4 and S1 to S4.

[0031] Preferably, the control method adopts a control method for the DC side current i effective within a specific frequency range. Ldc The proportional control link is used to suppress possible LC resonance phenomenon.

[0032] The beneficial effects of the present invention are:

[0033] 1. The present invention can perform fixed-frequency linear phase shift control within an ultra-wide DC side voltage range; the present invention can achieve bidirectional real-time power flow control within a relatively wide DC side voltage range. The voltage range of the DC side application (20-60V) can reach about twice that of the traditional voltage-fed solution (30-50V). However, the control method implementation does not become complicated due to the wide DC side voltage range. The unique and fixed transconductance G m The control trajectory correspondence is applied to feedforward predictive control. This control method only requires a small PR feedback for correction, so there is no need for complex control loop design theory. At the same time, the linearization degree of the control trajectory is as high as 98.7% and is applied at a fixed switching frequency, so the controller does not need to have strong computing power and memory space. Therefore, the present invention effectively reduces the cost and power consumption of the controller.

[0034] 2. The present invention has a higher and fixed switching frequency level; the present invention works at a fixed switching frequency of more than 100kHz. The magnetic parts and capacitors of the converter can be designed according to the switching frequency of more than 100kHz; therefore, the present invention has a higher upper limit of power density.

[0035] 3. The present invention has a smaller high-frequency current ripple level at both ends of the DC side and the AC side; the present invention can realize bidirectional real-time power flow control within a wider DC side voltage range. This real-time power control can bring smaller high-frequency current ripple on the AC side, and the DC side staggered parallel input inductor connection method also makes the DC side current i Ldc The high-frequency current ripple is also significantly reduced compared to the voltage-fed solution. Therefore, the continuous DC side current of the present invention can be applied to the charge and discharge management of the battery or the MPPT technology of the photovoltaic module.

[0036] 4. The present invention has a single-stage structure with few switching devices and a wide ZVS operating range; the present invention is a single-stage structure with only 8 switching devices and a simple structure; it has high feasibility under a simple fixed-frequency and high-linearity control scheme; the smaller number of switching devices and the wider soft switching range enable the present invention to have a higher efficiency ceiling. BRIEF DESCRIPTION OF THE DRAWINGS

[0037] Figure 1 It is a technical scheme diagram of an isolated DC / AC converter in the prior art;

[0038] Figure 2 It is a converter technical scheme diagram of a single-stage current-fed bidirectional DC-AC converter and a fixed-frequency control method thereof of the present invention;

[0039] Figure 3 is a specific implementation block diagram of a current controller based on transconductance of the present invention;

[0040] Figure 4 It is the key voltage and current waveform diagram of the present invention in 8 different working areas;

[0041] Figure 5 G of the present invention m Control trajectory diagram, where Figure 5 (a) is the complete control trajectory represented in 8 different regions, Figure 5 (b) is the corresponding G m (from -0.1S to 0.1S) and the corresponding relationship diagram of the control quantity;

[0042] Figure 6 is the time ratio k of the soft switching (ZVS) at the rated voltage of the present invention zvs Analysis diagram, where Figure 6 (a) corresponds to Q1 and Q3, Figure 6 (b) corresponds to Q2 and Q4, Figure 6 (c) corresponds to S1 and S4;

[0043] Figure 7 It is a simulation waveform diagram of the key voltage and current of the converter under the rated working condition of the present invention (bidirectional 400W);

[0044] Figure 8 This is a comparison diagram of the effect verification of the DC side current control under rated working conditions of the present invention (forward 400W);

[0045] Fig. 9 is an experimental verification diagram of realizing zero voltage switching (ZVS) of the switching device Q2 under rated working conditions of the present invention, wherein Fig. 9 (a) Forward 400W, Fig. 9 (b) Reverse 400W;

[0046] Fig.10 is a complete control block diagram of the present invention under the other two usage methods, wherein Fig.10 (a) Off-grid inverter mode, Fig.10 (b) PFC rectification mode.

[0047] In the figure, 11, inductor L dc1 ; 12. Inductance L dc2 ; 21, first DC side half bridge; 22, second DC side half bridge; 31, capacitor C dc1 ; 32. Capacitor C dc1 ; 4. Transformer; 5. Inductor L lk ; 61. Capacitor C ac1 ; 62, capacitor C ac2 ; 7. AC side bidirectional half bridge; 8. Inductor L ac ; 9. Mixed frequency band filter circuit. DETAILED DESCRIPTION

[0048] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the relevant technologies in the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0049] refer to Figures 1 to 10 In this embodiment, Figure 2 The single-stage current-fed bidirectional DC-AC converter topology and control method proposed in the present invention are demonstrated. The circuit is electrically isolated by a high-frequency transformer. The DC side consists of two identical boost half-bridges (Q1, Q2 and Q3, Q4), with an input inductor L dc1 and L dc2 In parallel, the output capacitor C dc1 and C dc2 The two half-bridges work 180 degrees alternately, and the switch nodes a and b are connected to the two ports on the primary side of the transformer. The AC side consists of a half-bridge (S1 to S4) with a bidirectional switch and a series capacitor Cac1 and C ac2 The connection node is connected to the two ports on the secondary side of the transformer. Finally, it passes through the AC filter inductor L ac Connected to the power grid (AC power supply).

[0050] In terms of control, on the one hand, the present invention adopts a hybrid fixed-frequency real-time AC current control method based on PWM control and phase shift control. The two control variables are the DC side device opening duty cycle D (lower tube Q2 and Q4) and the phase shift angle ratio φ (from Q1 to S2). On the AC side, the switching device exhibits the characteristics of high frequency operation in half of the power frequency cycle and direct conduction in the other half of the cycle. When the grid voltage v ac When it is greater than 0, S2 and S3 switch at a 50% duty cycle at a complementary high frequency, while S4 and S1 remain on. ac When it is less than 0, S4 and S1 switch at high frequency while S2 and S3 remain open. It is worth noting that S4 and S2, S3 and S1 work symmetrically.

[0051] The controller determines the phase of the grid voltage through the phase-locked loop (PLL) link and generates a current reference i ac,ref After a slight correction, this reference value and the DC side voltage v dc Determine the equivalent transconductance G required for the converter m , and then generate the corresponding control quantities D and φ. Finally, 4 groups of PWM drive signals corresponding to 8 switching devices are generated. m It is the equivalent transconductance characteristic description of the converter, the unit is S:

[0052]

[0053] where k m is the characteristic coefficient of the circuit, defined as n / (4L lk f s ).

[0054] By G m , can achieve a wide range of DC side voltage v dc Input lower AC side current i ac The positive value represents the forward power transfer under DC-AC conversion, and the negative value represents the reverse power transfer under AC-DC conversion.

[0055] On the other hand, in addition to G m In addition to the current control loop, the DC side current loop in a specific frequency band is also used to suppress the DC side LC resonance phenomenon. The two boost units on the DC side will have the risk of LC resonance, and the resonance center frequency f r for:

[0056]

[0057] Where L dc is the total parallel inductance on the DC side / μH; C dc is the total parallel capacitance on the DC side / μF.

[0058] Table 1: Reference key design parameters (for simulation verification)

[0059] Circuit parameters value Working parameters value <![CDATA[DC side filter inductance L dc1 and L dc2 > 18μH AC voltage RMS <![CDATA[220V rms ]]> <![CDATA[DC-side capacitor C dc1 and C dc2 > 30μF AC side voltage frequency 50Hz Transformer ratio n 4 DC side voltage range 20-60V <![CDATA[Transformer cascaded inductor L 1k > 85μH Rated Power 400W <![CDATA[AC side filter inductor L ac > 18μH Switching frequency 100kHz

[0060] Under the reference design parameters given in Table 1, the frequency range of this resonance is 2-6kHz. Therefore, the total inductor current i Ldc The frequency components that may be included are: DC component, 100Hz, 2-6kHz resonant component and high-frequency component above 200kHz. Based on this, the present invention uses a mixed frequency band filter to extract the resonant component in the DC side current to form a DC side current proportional control loop that is only effective for the resonant frequency band. The DC side input current has a very small high-frequency current ripple, and the calculation formula is:

[0061]

[0062] Figure 3 Demonstrated based on G m The specific block diagram of the AC current control loop. The current control is mainly based on feedforward and supplemented by feedback. The initial current reference value is slightly corrected by the PR (or PI) controller to generate the current real-time reference value, and then G is calculated according to formula (1) m The absolute value of G m The polarity of the grid voltage v ac The polarity of G m The value is set by f(D, φ) with an accuracy of 0.0001S to correspond to the trajectory point set of the control amount. m The range is set to -0.1S to 0.1S. Of the corresponding 2000 trajectory points, 98.7% can be directly calculated according to the linear relationship. Therefore, only 13 trajectory points need to be stored in the processor memory in advance.

[0063] Explanation of the differences from existing similar topologies (in terms of circuit topology)

[0064] The present invention replaces the full bridge in the traditional DAB with a dual-path input interleaved parallel boost circuit on the DC side. dc1 and L dc2 The continuous and low ripple DC input current is achieved, which reduces the demand for DC side filters. The present invention is characterized in that:

[0065] 1) The DC side of the present invention has only one external interface and no high-voltage DC port ( Figure 2 Middle C dc1 and C dc2 At run time, the present invention exhibits a change in C dc1 and C dc2 Voltage.

[0066] 2) The DC side of the present invention is composed of two independent half-bridges, and the outputs are not connected in parallel. The technical purpose of this change in the present invention is to avoid the formation of unnecessary power frequency circulating current (which will increase losses) on the DC side under the control strategy proposed by the present invention.

[0067] Basic principles on which control operations are based

[0068] Table 2: Power transfer partitions and calculations

[0069]

[0070] Figure 4 The key voltage and current waveforms of all eight operating states of the converter in the present invention on the switching frequency time scale are shown. The duty cycle of the two Boost half-bridges on the DC side is exactly the same, so their voltage and current only differ in phase. The operating state depends on the DC side inductor L dc1 , L dc2 The current waveform is divided into A and B. According to the transformer cascade inductance L lk The current waveforms are divided into types I-IV. The power transfer characteristics of different working states are presented in Table 2.

[0071] The fixed frequency high linearity control scheme adopted in the present invention enables it to operate in the IIIA zone, the IIA zone and the IIB zone (DC-AC conversion) or the IIIA zone, the IVA zone and the IVB zone (AC-DC conversion). The main control method is PWM control, which satisfies the following formula:

[0072]

[0073] It shows that the PWM control in the present invention is performed at the maximum phase shift power transfer point under a fixed duty cycle. A linear relationship between the AC output current and the control amount can be obtained:

[0074]

[0075] The linear control relationship enables the current controller to achieve a more accurate output through feedforward control, which effectively reduces the design pressure of the feedback controller. However, when the output current is too low (including the zero-crossing commutation process), the duty cycle D cannot effectively control the power transmission due to the existence of dead time. At the same time, the rate of change of the current is the largest in the entire cycle, so the commutation process often has waveform distortion. G from positive power flow to negative power flowm The trajectory will also show discontinuous changes in the control amount at 0, which is an unstable factor for energy control.

[0076] Figure 5 (a) shows the complete bidirectional current control trajectory after adding the transition section based on phase shift control. min The PWM linear control sections of the IIA and IVA regions are connected. At the same time, the circuit design can also reduce two isolated drive power supplies through bootstrapping. The final full control trajectory can be divided into 5 sections, G m The expression is expressed in (6):

[0077]

[0078] A. Segment 1 - PWM linear control segment (positive);

[0079] B. Segment 2 - Phase-shifted nonlinear control segment (positive);

[0080] C. Section 3 - Phase-shifted linear control section (including power transfer zero point);

[0081] D. Segment 4 - Phase-shift nonlinear control segment (negative);

[0082] E. Segment 5 - PWM linear control segment (negative).

[0083] Segments 1 and 5 are in PWM control mode, and segments 2-4 are in phase-shift control mode.

[0084] refer to Figure 5 (b), the equivalent G in the present invention m The corresponding relationship between the control quantity D and φ is shown, ranging from -0.1S to 0.1S. Among them, only segments 2 and 4 are nonlinear control segments, accounting for only 1.3%. Therefore, the control method adopted by the present invention has a linearity of 98.7%, which greatly reduces the design difficulty of the control loop and the requirements of the microprocessor chip on the hardware.

[0085] The upper and lower limits of the nonlinear segment are defined as G m,12 and G m,23 :

[0086]

[0087] The present invention has a wider ZVS region under the fixed frequency high linearity control method. The ZVS judgment formulas of Q1 and Q3, Q2 and Q4, and AC side devices S1-S4 are equivalent, and the judgment formulas are as follows:

[0088]

[0089] Where ton,1 represents the opening time of Q1, t on,2 represents the opening time of Q2, t on,s Indicates the opening time of S2.

[0090] Figure 6 The ZVS effect diagram in the full DC side voltage range and full power output range is given. zvs It represents the ZVS time ratio in one power frequency cycle, and 1 represents full-time soft switching. The results show that the DC side devices are in full-time soft switching range of more than 75%, and the AC side reaches 100%.

[0091] In order to verify the technical effect of the solution of the present invention, simulation and experiments are carried out according to the parameters given in Table 1.

[0092] Figure 7 The invention demonstrates the rated voltage (40V on the DC side and 220V on the AC side) rms ) and input voltage v at rated power (400W) dc 、Input current i Ldc , capacitor voltage v cdc1 , inductor current i Llk The waveform of the voltage and current on the AC side is v ac and i ac It can be verified that it has very small high-frequency current ripple on both the DC side and the low-voltage side, and the THD of the AC side current is only 2.45% at rated full load.

[0093] Figure 8 The comparison of the voltage and current waveforms of the two ports of the present invention before and after enabling the DC side current (within a specific frequency range) is shown. It can be seen that before the DC side control is enabled, the DC side has an LC oscillation described by equation (2) with a frequency between 2-6kHz. It is reflected in the inductor current i Ldc and capacitor voltage v cdc1 and v cdc2 After enabling DC side control, the resonance is suppressed. This will eliminate the audible noise caused by the DC side inductor and improve the THD level of the AC side current.

[0094] Fig. 9 The present invention shows the gate-source voltage v of device Q2 under rated full load conditions. gs and the drain-source voltage v ds The experimental waveform measurement results show that it has a zero voltage switching (ZVS) effect in half a power frequency cycle, which is consistent with the theoretical analysis.

[0095] Fig.10 The complete control structure of the converter connection method when applied to off-grid inverter and PFC rectifier is shown ( Figure 2 It is a bidirectional DC / AC under grid connection, which only includes current control).

[0096] (1) Off-grid inverter - actively supports AC voltage

[0097] A DC voltage source (20V-60V) is used to supply power to a single-phase AC load, and the converter actively constructs a 220V AC power grid through an AC voltage outer loop. The internal current control is still the control method proposed by the present invention.

[0098] (2) Rectifier (with active power factor correction, or PFC)

[0099] The power grid supplies power to DC loads with a wide voltage range (20V-60V). The converter controls the DC side voltage through the DC voltage outer loop, and the internal current control is still the control method proposed by the present invention. Under this rectification operation, the input current and input voltage are in phase, so it belongs to an active power factor correction circuit (PFC).

[0100] This embodiment provides a fixed frequency linear phase shift control in an ultra-wide DC side voltage range.

[0101] The current-fed DAB converter adopted in this embodiment can realize bidirectional real-time power flow control in a very wide DC side voltage range under the proposed high linear transconductance control strategy. The voltage range of DC side application (20-60V) can reach about twice that of the traditional voltage-fed solution (30-50V). However, the control method implementation does not become complicated due to the wide DC side voltage range.

[0102] The unique and fixed transconductance Gm control trajectory correspondence is applied to feedforward predictive control. This control method only requires a small PR feedback for correction, so there is no need for complex control loop design theory. At the same time, the linearization degree of the control trajectory is as high as 98.7% and is applied at a fixed switching frequency, so the controller does not need to have strong computing power and memory space - this can effectively reduce the cost and power consumption of the controller.

[0103] This implementation has a higher and fixed switching frequency level

[0104] The control method adopted in this embodiment enables the bidirectional DC / AC converter to operate at a fixed switching frequency of more than 100kHz. The magnetic components and capacitors of the converter can be designed according to the switching frequency of more than 100kHz. For example, under the reference design parameters of Table 1, the design frequencies of the transformer and the cascade inductor Llk are both 100kHz, while the remaining inductors and capacitors are all 200kHz. Therefore, the bidirectional DC / AC converter design proposed in this embodiment has a higher power density upper limit.

[0105] This implementation has a smaller high-frequency current ripple level (on both the DC and AC sides)

[0106] The current-fed DAB converter adopted in this embodiment can realize bidirectional real-time power flow control in a very wide DC side voltage range under the proposed fixed-frequency high-linear transconductance control strategy. This real-time power control can bring smaller high-frequency current ripple on the AC side. Both simulation and experiments show that its THD under full load is less than 3%. The DC side staggered parallel input inductor connection method also makes the DC side current i Ldc The high-frequency current ripple is also significantly reduced compared to the voltage-fed solution. The continuous DC side current can be used for battery charge and discharge management or MPPT technology of photovoltaic modules.

[0107] This embodiment has a single-stage structure with few switching devices and a wide ZVS operating range.

[0108] The current-fed dual active bridge bidirectional DC-AC converter adopted in this embodiment is a single-stage structure with only 8 switching devices and a simple structure. It has high feasibility under a simple fixed-frequency and high-linearity control scheme. The small number of switching devices and the wide soft switching range enable the converter to have a higher efficiency ceiling.

[0109] In summary, the present invention can perform fixed-frequency linear phase shift control within an ultra-wide DC side voltage range, and can realize bidirectional real-time power flow control within a relatively wide DC side voltage range, and the present invention does not require complex control loop design theory. Therefore, the present invention effectively reduces the cost and power consumption of the controller, and has a higher power density upper limit and a higher efficiency upper limit.

[0110] It should be emphasized that the above are only preferred embodiments of the present invention and do not limit the present invention in any form. Any simple modifications, equivalent changes and modifications made to the above embodiments based on the technical essence of the present invention are still within the scope of the technical solution of the present invention.

Claims

1. A single-stage current-fed bidirectional DC-AC converter, characterized in that: include: The DC side, the input end of which is connected to the DC bus v dc ; The DC side is composed of two Boost half-bridges with inductors, and the Boost half-bridges include: a first DC side half-bridge (21) and a second DC side half-bridge (22); The AC side, the output end of the AC side is connected to the AC bus v ac ; The AC side is composed of an AC side half bridge and an AC side filter, and the AC side half bridge includes: an AC side bidirectional half bridge (7); A transformer (4), wherein the primary side of the transformer (4) is connected to the output end of the DC side, and the secondary side of the transformer (4) is connected to the input end of the AC side; A control device is used to control the DC side and the AC side to transconductance G m Describe the power transmission characteristics of the circuit, and implement the AC side current control by a feedforward method; the control sampling signal of the control device includes: DC side port voltage, DC side inductor current, AC side port current and port voltage.

2. A single-stage current-fed bidirectional DC-AC converter according to claim 1, characterized in that: The DC side includes: switching transistors Q1-Q4, inductor L dc1 (11) and L dc2 (12) Output capacitor C dc1 (31) and C dc2 (32), the connection mode of the DC side is: DC busbar v dc The positive input terminal is connected to the inductor L dc1 (11) and L dc2 (12) at the input, inductor L dc1 The output end of (11) is connected to one end of the primary side of the transformer (4), and the other end of the primary side of the transformer (4) is connected to the inductor L dc2 (12) The output terminal; the switching transistors Q1 and Q2 are connected in series to the output capacitors C dc1 The connection node of the switching transistors Q1 and Q2 is connected to the inductor L dc1 (11) output terminal; the switching transistors Q3 and Q4 are connected in series to the output capacitors C dc2 The two ends of (32) are connected to the connection node of the switching transistors Q3 and Q4. dc2 (12) output terminal; output capacitor C dc1 (31) and C dc2 The negative pole of (32) is connected to the DC bus v dc The negative electrode; The AC side includes: switch transistors S1-S4, inductor L ac (8) and L lk (5) Series capacitor C ac1 (61) and C ac2 (62), the connection mode of the AC side is: AC busbar v ac The positive input terminal is connected in series with the inductor L ac (8) Series capacitor C ac1 (61) Series capacitor C ac2 (62) and then connected to the AC busbar v ac The negative input terminal of the switching transistors S1 to S4 is connected in series in sequence, and the two ends are connected to the inductor L ac (8) With series capacitor C ac1 (61) connection node, AC busbar v ac The negative input terminal of the switching transistor S2 and the switching transistor S3 are connected in series with the inductor L lk (5) is then connected to one end of the secondary side of the transformer (4), and the other end of the secondary side of the transformer (4) is connected to a series capacitor C ac1 (61) and C ac2 (62) connection node.

3. A fixed frequency control method for a single-stage current-fed bidirectional DC-AC converter, characterized in that: The fixed frequency control method is used to control the DC-AC converter according to any one of claims 1 or 2, and the control method comprises: dividing the converter into 8 regions according to the DC side duty cycle D and the phase shift ratio φ and mainly operating on the linear forward maximum power output trajectory located in the II region and the IV region; the control quantity relationship is expressed as: In formula (4), D represents the duty cycle of the DC side device, and φ represents the phase shift angle ratio.

4. The method for controlling a single-stage current-fed bidirectional DC-AC converter according to claim 3, characterized in that: The converter is divided into 8 regions according to the DC side duty cycle D and the phase shift ratio φ and fully operates on a continuous bidirectional trajectory located in regions II, III and IV to achieve a minimum duty cycle D. min The phase-shift control mode is used as a substitute for the PWM mode in low-power transmission to improve the low-power control accuracy.

5. The method for controlling a fixed frequency of a single-stage current-fed bidirectional DC-AC converter according to claim 4, characterized in that: In the control method, the transconductance G m The complete and unique control trajectory over the wide DC link voltage range described is expressed as: In formula (6), k m Represents the characteristic coefficient of the circuit.

6. The method for controlling a fixed frequency of a single-stage current-fed bidirectional DC-AC converter according to claim 5, characterized in that: In the control method, the converter is based on G m The control trajectory is used as the core to realize the AC current control loop combining feedforward and feedback, which specifically includes the following steps: The first step is to use the PLL phase-locked loop to obtain the phase reference of the grid voltage and then generate the control setting current reference i ac,ref ; Then a small correction is made through the PR controller to obtain G m The current reference value required by the controller; The second step is to obtain the real-time control transconductance G according to the input current reference value and the sampled DC side voltage value. m value; the polarity of this value is determined by the grid voltage v ac The polarity of The third step is to obtain the control quantity D and φ and generate the drive PWM of Q1 to Q4 and S1 to S4.

7. The method for controlling a fixed frequency of a single-stage current-fed bidirectional DC-AC converter according to claim 3, characterized in that: The control method adopts a control method for the DC side current i effective within a specific frequency range. Ldc The proportional control link is used to suppress possible LC resonance phenomenon.

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