Modular DC-DC converter and battery charging device including same

Abstract

The present invention relates to a modular DC-DC converter and a battery charging device including the same. A modular DC-DC converter and a battery charging device are provided. The modular DC-DC converter includes a first converter arranged on an input side, a plurality of second converters arranged on an output side, and a plurality of high-frequency transformers arranged between the first converter and the second converter. The first converter and the high-frequency transformer are connected in series on the input side, and the second converters are connected in parallel on the output side.

Description

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS

[0002] This application claims priority to U.S. Provisional Patent Application No. 62 / 826,268, filed on March 29, 2019, the disclosure of which is incorporated by reference in its entirety into this specification. Background Art

[0003] Low-speed electric vehicles (LS-EVs) are often used in retirement communities and suburbs. LS-EVs operate over short distances and run primarily on batteries. LS-EVs can be effectively used to replace conventional cars for short trips that last less than ten minutes. LS-EVs can be used in airports, stadiums, country clubs, resorts, and golf courses.

[0004] One of the main problems of LS-EV is the long battery charging time. The LS-EV recharging process is usually completed via a standard 120VAC or 240VAC single-phase outlet. Such a process takes more than eight hours and is usually completed at night. Therefore, it is necessary to quickly recharge the battery pack to improve the utilization rate of LS-EV. Summary of the invention

[0005] The present disclosure generally relates to a modular DC-DC converter and a battery charging device including the modular DC-DC converter for fast charging of a low-speed electric vehicle.

[0006] In view of the present disclosure, and without limiting the scope of the present disclosure in any way, in one aspect of the present disclosure that can be combined with any other aspect listed herein unless otherwise specified, a modular DC-DC converter is provided. The modular DC-DC converter includes a first converter arranged on the input side, a plurality of second converters arranged on the output side, and a plurality of high-frequency transformers arranged between the first converter and the second converter. The first converter and the high-frequency transformer are connected in series on the input side, and the second converter is connected in parallel on the output side.

[0007] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the first converter comprises only one full-bridge converter.

[0008] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the second converter includes at least two full-bridge converters.

[0009] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the first converter includes at least one of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

[0010] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the second converter includes two or more of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

[0011] In one aspect of the disclosure, which may be combined with any other aspect listed herein unless otherwise specified, the modular DC-DC converter further includes a first capacitor coupled to the first converter at an input side.

[0012] In one aspect of the present disclosure, which may be combined with any other aspect listed herein unless otherwise specified, the modular DC-DC converter further includes a second capacitor coupled at an output side to one of the second converters.

[0013] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the second converter comprises three full-bridge converters.

[0014] In view of the present disclosure, and without limiting the scope of the present disclosure in any way, in one aspect of the present disclosure that can be combined with any other aspect listed herein unless otherwise specified, a battery charging device is provided. The battery charging device includes a first converter arranged on the input side, a plurality of second converters arranged on the output side, and a plurality of high-frequency transformers arranged between the first converter and the second converter. The first converter and the high-frequency transformer are connected in series on the input side, and the second converters are connected in parallel on the output side.

[0015] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the first converter comprises only one full-bridge converter.

[0016] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the second converter includes at least two full-bridge converters.

[0017] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the first converter includes at least one of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

[0018] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the second converter includes two or more of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

[0019] In one aspect of the present disclosure, which may be combined with any other aspect listed herein unless otherwise specified, the battery charging device further includes a first capacitor coupled to the first converter at an input side.

[0020] In one aspect of the present disclosure, which may be combined with any other aspect listed herein unless otherwise specified, the battery charging device further includes a second capacitor coupled at an output side to one of the second converters.

[0021] In one aspect of the disclosure which may be combined with any other aspect listed herein unless otherwise specified, the second converter comprises three full-bridge converters.

[0022] The reader will appreciate the foregoing details, as well as other details, upon considering the following detailed description of certain non-limiting embodiments of modular DC-DC converters according to the present disclosure. BRIEF DESCRIPTION OF THE DRAWINGS

[0023] The features and advantages of the present technology, including the modular DC-DC converter and battery charging apparatus described herein, may be better understood by referring to the accompanying drawings, in which:

[0024] Figure 1 is a schematic diagram of a DAB converter circuit according to an embodiment of the present disclosure.

[0025] Figure 2A-2E is a schematic diagram of a multi-module DC-DC converter configuration; Figure 2A It is a universal multi-module DC-DC converter; Figure 2B It is the FB circuit configuration; Figure 2C It is HB circuit configuration;

[0026] Figure 2D is a DAB circuit configuration; and Figure 2E It is based on the resonant circuit of the DAB DC-DC converter.

[0027] Figure 3 is a schematic diagram of a fast charger DC-DC converter according to an embodiment of the present disclosure.

[0028] Figure 4 is a schematic diagram of a fast charger DC-DC converter circuit according to an embodiment of the present disclosure.

[0029] Figure 5 is a schematic diagram of a small signal model of an n-module ISIP-OSOP DC-DC converter according to an embodiment of the present disclosure.

[0030] Figure 6 is a schematic diagram of a three-module ISOP fast charger DC-DC converter circuit according to an embodiment of the present disclosure.

[0031] Figure 7 is a schematic diagram of a small signal model of an ISOP fast charger DC-DC converter according to an embodiment of the present disclosure.

[0032] Figure 8 is a schematic diagram of an OCS control scheme for a three-module ISOP fast charger DC-DC converter according to an embodiment of the present disclosure.

[0033] Figures 9A-9D is a graph of simulation results for a control scheme for a three-module ISOP fast charger DC-DC converter according to an embodiment of the present disclosure. Fig. 9A is a graph of modular input voltage; Fig. 9B is a graph of input current; Fig. 9C is a graph of modular output current; and Fig.9D is a graph of the total output current.

[0034] The reader will appreciate the foregoing details as well as other details after considering the following detailed description of certain non-limiting embodiments of the present technology including modular DC-DC converters and battery charging devices according to the present disclosure. The reader may also understand certain such additional details after using the present technology including modular DC-DC converters, battery charging devices, and control methods described herein. DETAILED DESCRIPTION

[0035] The present disclosure generally relates to a modular DC-DC converter and a battery charging device including the modular DC-DC converter.

[0036] LS-EV fast charging is necessary to overcome the downtime caused by the long charging process and allow better utilization of LS-EVs. If the charging process is completed in less than an hour with fast charging, it will require a higher charging rate. High-power three-phase fast charging technology has become a well-established technology in the industrial EV market and has achieved significant reductions in time, cost and labor savings. However, there are many challenges when they begin to be applied to LS-EVs that are primarily charged through single-phase circuits and handled by consumers.

[0037] Generally, there are three charging levels for EVs. In level 1 and level 2, the charging process is carried out over a considerable period of time at the place where the vehicle is charged at home. The main disadvantage of these charging levels is that the charging process takes from 4 hours to 20 hours to reach a full charge, depending on the size and charge state of the battery. These levels are not suitable for long distances. In other words, in level 1 and level 2 charging methods, additional time is required to reach a full charge, which hinders the transition from conventional vehicles powered by gasoline to battery-powered electric vehicles. In this regard, level 3 charging infrastructure has been introduced to reduce the time of the charging process and provide fast refueling. Long charging times are avoided in level 3 fast charging because battery charging can be achieved in less than thirty minutes. In order to achieve fast charging, the power electronic interface of the fast charger should be accurately designed to achieve high power factor and high efficiency, and provide galvanic isolation. In addition, a well-regulated output DC voltage must be obtained.

[0038] Fast charging technologies such as constant current constant voltage (CC-CV) and multi-stage constant current constant voltage (MCC-CV). MCC-CV can be applied to improve battery acceptance. In order to have the ability to deliver high power, some advanced charging technologies such as CC-CV method using negative pulse and variable frequency pulse charging are used to improve charging acceptance.

[0039] According to an embodiment of the present disclosure, a modular DC-DC converter for LS-EV fast charging is developed. Charging at a low voltage of 48V on the output side requires high current, which poses a challenge when using a single converter unit. Typically, two different DC-DC converter approaches are used for EV fast charger applications. The first approach is to use a two-stage voltage source converter in which semiconductor devices are connected in series and / or in parallel to achieve the desired rating. The second approach is to use a modular converter that can have two-stage or multi-stage properties.

[0040] DC-DC converters with modular structure provide easier maintenance, such as hot-swap capability, redundancy, scalability and ride-through capability. In addition, the required rating can be achieved by the modules used, each of which handles a small portion of the total required power. The modularity concept applies to the entire stage, including transformers and power electronics.

[0041] One of the attractive DC-DC converters used in fast charging EV applications is the dual active bridge (DAB) topology based on a multi-module DC-DC converter, because bidirectional power flow can be achieved. In addition, a higher switching frequency in the AC link can be achieved, which can lead to significant weight and size reduction. In addition, soft switching techniques can be applied to avoid high switching losses.

[0042] In the present disclosure, a low-cost and high-efficiency DC-DC converter for LS-EV fast charger is provided. Different DC-DC converters are prepared for EV applications. In addition, according to the system specifications, the configuration of the DC-DC converter power stage is provided. In addition, in order to study the dynamic performance of a multi-module DC-DC converter, a generalized small signal analysis applicable to input series input parallel output series output parallel (ISIP-OSOP) is provided. This is achieved by studying the small signal analysis of a full-bridge phase-shifted (FB-PS) DC-DC converter, a two-module input parallel output series (IPOS), a three-module input series output parallel (ISOP), and a four-module input series input parallel output series (ISIPOS), respectively. Therefore, the generalized model is used to derive a small signal model for the ISOP DC-DC converter. In addition, a control method for the ISOP DC-DC converter is provided to ensure equal distribution of power among the modules.

[0043] It should be understood that as battery sizes and capacities vary (36V–48V; 140Ahr–200Ahr), the resulting power rating of the fast charger will vary between 2.5kW–6.6kW. Therefore, in order to achieve greater versatility in implementing a fast charger for the LS-EV, a modular design will be an effective solution to meet the varying needs of these batteries. In order to meet the varying power needs of the vehicle, modular power stages ranging between 1.5kW–2kW can be used as building blocks. With modular power stages, a variety of charger sizes and models can be implemented using several building blocks according to embodiments of the present disclosure.

[0044] According to an embodiment of the present disclosure, the developed power conversion stage will be in a modular manner, where each modular power stage is rated at 1.5kW.Battery charging via a single-phase power source is demanded by consumers because single-phase charging allows consumers to conveniently charge their EVs through standard single-phase outlets available at their homes.

[0045] Energy flow can be from grid to vehicle (G2V) and also from vehicle to grid (V2G). According to an embodiment, the EV charger includes two stages: the first stage is an AC-DC converter that interfaces with the grid. This converter has two functions: operating with unity power factor and providing low harmonic distortion for both input voltage and current and output voltage and current. The second stage is a DC-DC converter. This converter must control the charging current, ensure that the appropriate voltage is delivered to the battery, and reduce both voltage ripple and current ripple. In order to achieve G2V mode and V2G mode, these two stages must support bidirectional power flow.

[0046] According to an embodiment, the DC-DC converter is a basic isolation converter, a basic resonant converter, a two-stage DAB converter, and a multi-module converter.

[0047] The requirements for designing EV battery chargers can be highlighted in terms of high power density, high efficiency, low cost and galvanic isolation. Increasing the switching frequency reduces the passive component size and cost. However, increasing the switching frequency increases the switching losses. Therefore, soft switching techniques and resonant topologies are used to increase the switching frequency without sacrificing the efficiency of the power stage. Another requirement that is necessary for the EV charger system is to choose a topology with the ability to control high currents on the output side. In addition, in order to electrically disconnect the grid from the vehicle, galvanic isolation is required. This can be accomplished by using a high frequency transformer according to an embodiment.

[0048] Isolation topologies can be divided into two categories: single-ended and double-ended. The advantage of the double-ended topology over the single-ended topology is that it requires a smaller magnetic core and does not require an additional cleanup winding. The order of basic isolated DC-DC converters from low power to high power is flyback, forward, push-pull, half-bridge (HB) and full-bridge (FB).

[0049] In Table 1, basic isolated DC-DC converters are compared in terms of output power range, cost, number of active switches, and electrical stress.

[0050] Table 1. Comparison of basic isolated DC-DC converters

[0051]

[0052] As illustrated in Table 1, FB DC-DC converters can be employed in higher power applications when compared to other topologies. However, FB converters suffer from high switching losses because all switches are operated under hard switching. In order to avoid the high switching losses associated with conventional FB converters, a phase-shifted full-bridge (PS-FB) DC-DC converter is another FB configuration used in EV battery charger applications. This topology differs from conventional topologies in terms of control methodology. Using a PS-FB DC-DC converter allows lower switching losses to be achieved through zero voltage switching.

[0053] Resonant converters are able to achieve very low switching losses, enabling such converters to operate at high switching frequencies. Basic resonant converters are series resonant converters (SRC), parallel resonant converters (PRC), and series-parallel converters denoted (SPRC or LCC resonant converters). In SRC, the DC gain is always less than one. The load impedance is very large compared to the resonant tank impedance under light load conditions, making it difficult to regulate the output voltage under light load conditions. The main disadvantages of SRC can be highlighted in terms of regulation of the output voltage at light load, high energy cycles, and high currents during the off period as the voltage on the input side increases.

[0054] Unlike SRC, light load regulation problems are avoided in PRC. However, PRC operates at higher switching frequencies as the voltage on the input side increases. Moreover, the cycle energy is much higher even under light load conditions when compared to SRC. Therefore, the main disadvantages of PRC can be highlighted in terms of high energy cycles and high currents during the off period.

[0055] According to an embodiment of the present disclosure, a multi-module DC-DC converter includes a combination of SRC and PRC. SPRC has the advantages of both SRC and PRC. In this topology, the output voltage can be adjusted under no-load conditions. In addition, the circulating energy is smaller when compared with PRC. In addition, the input current is smaller when compared with PRC, but the input current is higher when compared with SRC. Therefore, it can be said that the circulating energy in SPRC is smaller than PRC.

[0056] Similar to SRC and PRC, SPRC operates at a higher switching frequency as the input voltage increases. In addition, as the input voltage increases, the turn-off current of the switching device also increases. Therefore, it can be observed that SPRC combines the advantages of SRC and PRC, namely, the regulation of the output voltage under no-load conditions and smaller circulating energy. However, these three topologies may not be adjusted at high input voltages because high switching losses and high conduction losses will result. In addition, high circulating energy will occur at high voltages on the input side.

[0057] Another resonant converter configuration is the LLC resonant converter. In this converter, two resonant frequencies are available at the SPRC. The advantages of this configuration can be summarized with the ZVS capability even under no-load conditions, and therefore, the switching losses are reduced, which will lead to high efficiency. In addition, the output voltage can be regulated under all load conditions. In addition, due to the ZVS on the primary side and ZCS on the secondary side, the LLC resonant converter provides high efficiency, making the LLC a promising candidate topology. The HB-LLC resonant converter has been designed and implemented for EV battery charging applications. The DC-DC converter is designed to achieve high efficiency, wide output voltage range, low cost, and low switching losses to maximize energy transfer.

[0058] The LLC topology may lose soft switching in reverse mode. Therefore, the bidirectional power transfer capability is limited and the efficiency is compromised. However, this can be avoided by adopting a bidirectional CLLC resonant converter. The operation of the CLLC resonant converter is similar to that of the LLC resonant converter in forward (G2V) and reverse (V2G) modes. Therefore, ZVS and ZCS in both modes can be obtained, and the switching losses can be reduced, thereby improving the efficiency of the charger.

[0059] According to an embodiment of the present disclosure, the DAB DC-DC converter is a topology that can be used for both low power applications and high power applications. Figure 1 The DAB configuration shown in comprises two active bridges connected via a medium / high frequency AC transformer. The DAB can be constructed using a single phase bridge or a three phase bridge depending on the design criteria. Figure 1 The 2L-DAB shown in is usually operated in square wave mode. The intermediate transformer leakage inductance limits the maximum power flow and is used as an energy transfer element. This topology is capable of bidirectional power flow, which can be achieved by controlling the phase shift between the two bridges and the amplitude of the output voltage of each bridge. The switch can be switched under ZVS and / or zero current switching (ZCS). Therefore, switching losses are reduced and power efficiency is improved. However, ZVS operation (i.e., soft switching characteristics) is not guaranteed at light loads. In other words, the conventional DAB soft switching area is limited to a narrow output voltage range. However, in EV battery charger applications, when the battery is charged, the converter will operate for a period of time under no load, and the load will be absolutely zero. Therefore, in order to achieve the project requirements of achieving high efficiency, ZVS should be maintained under all load conditions. Therefore, several control strategies are proposed to increase the soft switching range and reduce the transformer current. In addition, a high conversion ratio can be obtained by selecting the transformer turns ratio.

[0060] To increase the DAB soft switching range, series R-DAB (SR-DAB) is provided. In this topology, only one additional capacitor is required. SR-DAB is usually modulated by a frequency modulation scheme. However, in order to modulate the power transfer, a large switching frequency range is required, which will cause difficulties in filter and control design.

[0061] According to an embodiment, a frequency modulated CLLC-R-DAB is provided. In this topology, the converter is able to operate over large variations in input voltage while maintaining soft switching capabilities. A smaller switching frequency range is used to modulate the CLLC-R-DAB converter when compared to the SR-DAB. According to an embodiment, a DAB with a tuned LCL network is provided. It is shown that the bridge current amplitude is reduced when operating at the LCL network tuned frequency. This minimizes high frequency link conduction losses and improves the utilization of semiconductor devices in the two full bridges.

[0062] According to an embodiment, a DAB with a tuned CLLC structure is provided to improve the power density of the converter. This topology reduces the number of required magnetic components and transformer magnetizing inductance while maintaining similar operating characteristics to the LCL-DAB topology.

[0063] By means of the modular design of DAB, easier scalability of the system can be achieved. In addition, by inserting redundant modules, system reliability is achieved. The possible architectures for connecting multiple DAB units are divided into four main categories, which are: Input Series Output Series (ISOS), Input Parallel Output Parallel (IPOP), Input Series Output Parallel (ISOP), Input Parallel Output Series (IPOS).

[0064] To meet the voltage and power requirements of EV fast charger applications, two different approaches have been established. The first approach is by integrating semiconductor devices into a two-stage converter topology with series / parallel connection. However, the series connection of the power switches results in an asymmetric sharing of the voltage between the switching devices due to the unequal parameters of the switches (such as switching delay, leakage inductance, and collector-to-emitter capacitance). Therefore, a voltage balancing method is required to avoid any failure that may occur, as the failure of one component will cause the entire converter to fail, resulting in low reliability.

[0065] In the second method, it is usually pursued to build a power electronic converter in a modular manner. The modular converter contains a number of smaller modules. Building the converter in a modular manner is a cost-effective solution. In addition, it is possible to hot-swap smaller modules in the event of a fault, which makes the maintenance of such converters easier. In addition, the number of modules can be proportionally expanded according to the power rating of the system. In addition, by installing more modules, the concept of redundancy can be established. In other words, in a DC-DC converter based on a modular structure, each unit handles a small part of the total input power. Therefore, the selected power switch has a lower voltage rating and therefore has a higher switching frequency capability. Therefore, due to the increase in switching frequency, the transformer size will be reduced. In order to avoid the disadvantages associated with the first method, a modular converter topology such as a multi-module converter is used to provide modularity features and achieve voltage and power requirements, wherein the topology based on the multi-module converter includes a resonant converter and HB and FB converters.

[0066] Figure 2A-2E A multi-module DC-DC converter having a configuration that can be employed in an EV charger application is shown. In this converter, each submodule (SM) operates at a single module rated voltage and contributes to a small portion of the total output power. Therefore, such a converter can operate at a higher switching frequency without facing challenges in its design and without sacrificing conversion efficiency.

[0067] According to an embodiment of the present disclosure, this DC-DC converter is based on a modular structure and is capable of providing unidirectional and bidirectional power flow. In this arrangement, additional control techniques are applied to ensure equal voltage and / or current sharing between SMs.

[0068] Figure 2A It is a general-purpose multi-module DC-DC converter. Figure 2B It is the FB circuit configuration. Figure 2C It is an HB circuit configuration. Figure 2D It is a DAB circuit configuration. Figure 2E It is based on the resonant topology of the DAB DC-DC converter.

[0069] exist Figure 2E In the FB converter, the FB converter adjusts its frequency according to the AC link resonant frequency to produce a voltage waveform with sinusoidal behavior across the primary and secondary windings of the transformer. This will enable the switching device to operate near ZCS.

[0070] DAB based on multi-module DC-DC converters is able to achieve higher switching frequencies in the AC link, which leads to weight and size reduction. In addition, by adopting soft switching technology, the losses of the converter are reduced, thus achieving higher efficiency.

[0071] According to an embodiment, the system specifications are as follows: the input voltage of the module is 110V@1.5kW and the input current of the module is 13A@1.5kW. However, the total output voltage and output current are 48V and 93.75A, respectively. Therefore, the modular power stage connection based on the conventional DAB DC-DC converter is ISOP. However, the input specifications can be handled by only one converter, and due to the high output current on the output side, more than one converter may be required to avoid high power losses and achieve maximum efficiency. Therefore, the selected topology is a multi-module DC-DC converter DAB, and only one bridge is adopted on the primary side, and the modularity concept will be applied on the high-frequency transformer and the second bridge, where the connection of the modules will be as follows Figure 3 ISOP shown in . Due to the high output current not only a parallel connection on the output side is required so that all modules share equal current but also a small ripple current is required, which will thus reduce the capacitor size and avoid any damage to the capacitor during the charging process.

[0072] According to an embodiment of the present disclosure, a generalized small signal analysis applicable to an ISIP-OSOP configuration is provided.

[0073] like Figure 4 The ISIP-OSOP universal DC-DC converter configuration shown in includes n isolated DC-DC modules which are connected in series and / or in parallel on the input side and in series and / or in parallel on the output side.

[0074] By ensuring input current sharing (ICS) and IVS, the input current of each module is reduced to And the input voltage of each module is reduced to Among them, I in is the input current of the ISIP-OSOP DC-DC converter and V in is the input voltage of the ISIP-OSOP DC-DC converter, α is the number of modules connected in parallel on the input side and β is the number of modules connected in series on the input side. Similarly, by ensuring OCS and output voltage sharing (OVS), the output current of each module is reduced to And the output voltage of each module is reduced to Among them I o is the output current of the ISIP-OSOP DC-DC converter and V o is the output voltage of the ISIP-OSOP DC-DC converter, a is the number of modules connected in parallel on the output side and b is the number of modules connected in series on the output side.

[0075] According to an embodiment, a Figure 5 The small signal model for the ISIP-OSOP converter is shown in Figure 2. The input current and input voltage of each module are and And the output current and output voltage of each module are and Therefore, the load resistance of each module is Therefore, it is possible to use the effect of changing the filter inductor current on the duty cycle modulation and the effect of changing the input voltage on the duty cycle modulation as the and as well as Figure 5 I eq The expression is as follows:

[0076]

[0077] Equation (1) can be rewritten as:

[0078]

[0079] in

[0080]

[0081] Equation (3) can be rewritten as:

[0082]

[0083]

[0084] Based on the small signal model, the n-module ISIP-OSOP DC-DC converter small signal model is constructed. The input voltage disturbance is expressed as However, the input voltage disturbance per module is expressed as where j = 1, 2, ..., β. The capacitor voltage disturbance is expressed as Where j = 1, 2, ..., b.

[0085] Based on the modularity feature and to reduce the small signal transfer function in terms of complexity, the modules are assumed to have equal effective duty cycle, transformer turns ratio, capacitor and inductor values.

[0086] Therefore, K1=K2=…=K n =K, C1=C2=…=C n =C, C d1 =C d2 =…=C dn =C d , and L1=L2=…=L n = L. In addition, it is assumed that all modules share the same input voltage. Therefore, the DC input voltage of each module is Although each module has a different duty cycle perturbation, all modules are assumed to have the same normalized time shift. In addition, the ESR of the output capacitor is considered in this model.

[0087] The following small signal equation is obtained by applying KVL and KCL:

[0088]

[0089]

[0090] Adding equation (7) together:

[0091]

[0092] Equation (8) can be written as:

[0093]

[0094] Define the summation term of the input and output voltages of the module that appears after summing the equations in (6):

[0095]

[0096] in:

[0097] If all modules on the input side are connected in series, γ=1.

[0098] If all modules on the input side are connected in parallel, then γ = α.

[0099] If the modules on the input side are connected in series and in parallel, then

[0100]

[0101] in:

[0102] If all modules on the output side are connected in series, then c=1.

[0103] If all modules on the output side are connected in parallel, then c = a.

[0104] If the modules on the output side are connected in series and in parallel, then

[0105] Control to output voltage transfer function: The relationship between output voltage and duty cycle is obtained by adding the equations in (6), assuming and Where k = 1, 2, ..., n and k ≠ j, and substitute into (2), (4), (9), (10) and (11).

[0106] Adding equation (6) together:

[0107]

[0108]

[0109]

[0110] Further simplification of (14) yields (15)

[0111]

[0112] Control to filter inductor current transfer function: The relationship between the filter inductor current and the duty cycle is obtained by using (9) to obtain the following equation:

[0113]

[0114] Substituting (16) into (14), and assuming and Wherein k=1, 2, ..., n and k≠j.

[0115]

[0116] Further simplification of (17) will yield (18).

[0117]

[0118] Output Impedance

[0119] Similarly, as derived in the small signal modeling of the ISOP DC-DC converter based FB topology

[12] , the output impedance of the ISIP-OSOP converter can be obtained by rewriting the KCL equation in (7) such that:

[0120]

[0121] Summing the equation in (19):

[0122]

[0123] Therefore, (7) is modified as follows:

[0124]

[0125] The relationship between output voltage and output current is obtained by assuming and j=1, 2, ..., n, obtained by adding all equations in (6-6) and substituting into (2), (4), (10), (11) and (21).

[0126]

[0127]

[0128] Further simplification of (23) will yield (24).

[0129]

[0130] Converter Gain

[0131] The relationship between the output voltage and the input voltage is assumed by j=1, 2, ..., n, obtained by adding all equations in (6) and substituting (2), (4), (9), (10) and (11) in the added equations.

[0132]

[0133]

[0134] Rearranging (26) will produce (27).

[0135]

[0136] The same analysis is performed in a FB-PS DC-DC converter based on a modular structure ISOP. However, the modular approach will be applied only on the high frequency transformer and the second FB converter. According to an embodiment, the analysis is not limited to unidirectional power flow and can be applied to bidirectional power flow.

[0137] According to an embodiment of the present disclosure, a Figure 6 The ISOP converter is shown in FIG. The ISOP converter includes a plurality of high frequency transformers and FB converters connected in series and in parallel at the input and output ends, respectively. By ensuring equal IVS and OCS, the input voltage of each module is reduced to And the output current of each module is reduced to Among them, V in and I o They are the ISOP input voltage and ISOP output current respectively.

[0138] The power rating of each module will be 1.5kW and in order to achieve the desired power rating of 4.5kW, three modules need to be connected in series on the input side and in parallel on the output side. Based on the mentioned specifications on the input side, it was decided to adopt only one FB converter with the ability to handle the input current 13.26A. However, due to the high output current 93.75A, a modular approach is required to avoid high losses and achieve the highest efficiency.

[0139] Use the small signal circuit for FB-PS DC-DC converter to provide and Figure 7 The small signal circuit model is shown in .

[0140] Based on the modularity feature and to reduce the small signal transfer function in terms of complexity, it is assumed that all modules have equal effective duty cycle, transformer turns ratio, capacitor and inductor values. Therefore, K1 = K2 = K3 = K, C1 = C2 = C3 = C and L1 = L2 = L3 = L. In addition, it is assumed that the modules share the same input voltage. Therefore, the DC input voltage of each module is In addition, the ESR of the output capacitor is considered in this model, however, it can be eliminated for simplicity. Since the input voltage of each module is So the output current of each module is Therefore, the load resistance for each module is 3R.

[0141] Therefore, the values ​​for the six parameters defined previously are known and are shown in Table 2.

[0142] Table 2. Values ​​of six parameters for ISOP fast charger DC-DC converter

[0143]

[0144] Substituting the six parameters with their values ​​in equations (15), (18), (24) and (27) produces the following transfer functions presented in Table 3.

[0145] Table 3. Generalized model validation for a three-module ISOP DC-DC converter

[0146]

[0147] However, the first two transfer functions as control to output voltage and control to filter inductor current are assumed wk = 1, 2, ..., n and k ≠ j. In other words, the two transfer functions are derived with respect to the effect of a duty cycle disturbance of only one module. However, since only one module is available on the primary side, the duty cycle disturbance will appear 3 times. Therefore, and Therefore, Table 3 is updated to produce Table 4.

[0148] Table 4. Generalized model validation for a three-module ISOP DC-DC converter

[0149]

[0150] Power Sharing in ISOP Fast Charger DC-DC Converters

[0151] The modules in actual applications are not identical and any mismatch in parameter values ​​may cause unequal power distribution between modules. Therefore, the voltage of a single module is unbalanced and, in addition, heavily loaded modules are subjected to overheating stress. Therefore, a control scheme is required to ensure uniform power sharing between modules to achieve reliable operation for the ISIP-OSOP DC-DC converter. For example, if the connection is ISOP, a control scheme is required to ensure IVS and OCS. If the connection is ISIPOS, a control scheme is required to ensure IVS, ICS and OVS.

[0152] According to an embodiment, the converters are connected in series on the input side and in parallel on the output side. Therefore, a control scheme that ensures IVS and OCS is required.

[0153] A control scheme for active power balancing between modules is addressed for a three-module ISOP DC-DC converter according to an embodiment.

[0154] Assume that Figure 6 The modules shown in are lossless, and the input and output power of each module can be expressed as:

[0155]

[0156] Where V cd1 、V cd2 and V cd3 are the DC input voltages across the transformer windings for module 1 and modules 2 and 3, respectively, Iout1 ,I out2 and I out3 are the output currents of modules 1, 2 and 3, I in and V out are the input voltage and output voltage of the three modules.

[0157] If OCS is completed, it means I out1 =I out2 =I out3 Therefore, according to (28), the following equation is obtained.

[0158] V cd1 I in =V cd2 I in =V cd3 I in (29)

[0159] According to (29), it can be inferred that V cd1 =V cd2 =V cd3 , which means that if OCS is implemented among the three modules, IVS sharing is also achieved.

[0160] Alternatively, if IVS is implemented, it means Therefore, according to (28), according to (30), we can infer that I out1 =I out2 =I out3 , which means that if IVS is achieved between the three modules, OCS is also achieved. Therefore, in order to identify the output side control, the output voltage control for the overall converter as well as the output current controller is dedicated to ensure stable operation of the parallel connected modules. In order to achieve uniform power distribution, the output voltage control using small signal analysis in addition to the overall control scheme is described and will be further described.

[0161] V out I out1 =V out I out2 =V out I out3 (30)

[0162] According to the parameters listed in Table 2, the update is and The effect of changing the filter inductor current on the duty cycle modulation and the effect of changing the input voltage on the duty cycle modulation and I eq .

[0163]

[0164] can be obtained by substituting into the R defined previously in Section V d Let's write equation (31) as follows.

[0165]

[0166]

[0167] Similar steps can be performed according to (33):

[0168]

[0169]

[0170] The following small signal equation for the proposed ISOP DC-DC converter is obtained by applying KVL and KCL:

[0171]

[0172]

[0173]

[0174]

[0175]

[0176] One goal of ISOP control is to ensure equal power sharing between modules so that IVS and OCS are achieved. According to an embodiment of the present disclosure, a direct OCS control scheme with a closed-loop controller is provided. The OCS control scheme includes an outer output current loop and three inner current loops. The output current loop generates a common reference for the three inner current loops, where the current feedback for each module is its own output current rather than the sum of the other two output currents. The ISOPDC-DC converter configuration will result in a stable control scheme through direct OCS control.

[0177] The present disclosure provides a DC-DC converter for a fast charger, it should be noted that as the charging rate increases, the battery can be exposed to overcharging or overheating. This threatens the service life of the battery. Therefore, the charging process should be terminated once the battery is fully charged to avoid overcharging the battery. Therefore, considering the reflective charging method including hiccup charging or negative pulse charging, the control scheme designed for the three-module ISOP DC-DC converter is current controlled. This charging method is based on applying a short negative pulse or a short discharge pulse during the charging cycle. This algorithm provides great advantages that can be highlighted in the following aspects: shortening the charging time and reducing the temperature rise. Generally, the reflective charging method includes three charging sequences, which are: a positive charging pulse, a quiescent period where no charging occurs, and a negative charging pulse or a discharge pulse. Therefore, the designed control scheme is based on controlling the output current of the ISOP DC-DC converter so that the output current profile is based on such as Figure 8 Reflected charging shown in .

[0178] Figure 8 The figure shows the OCS control scheme for three-module ISOP DC-DC converters. Figure 8 It can be seen that the control scheme includes an outer current loop and three inner current loops. The current feedback for module 1 is its own output current and this also applies to modules 2 and 3. Under steady-state conditions, all current feedbacks follow a common reference provided by the output current loop with zero static error. Therefore, the following equation is obtained:

[0179]

[0180] According to the KVL equation presented in equation (36), the output current can be controlled. This is obtained by substituting (32), (34) and pressing Substitution And settings By doing so, the following equation is obtained:

[0181]

[0182] Equation (42) can be expressed in matrix form such that:

[0183]

[0184] in and

[0185] According to (43), the control to output current transfer function is obtained as follows:

[0186]

[0187] in:

[0188] A(s)=s2LRC+s(L+RdRC)+Rd+2R (45)

[0190]

[0191] The relationship between the input voltage and output current of each module can be derived using the KCL equation presented in (37). This is done by equating the first two equations in (37) and substituting into (32), (34) and (35), resulting in:

[0192]

[0193] Further simplification of (47) will yield (48).

[0194]

[0195] According to (48), the following equation is obtained:

[0196]

[0197] in: and

[0198] According to (49), we can conclude So that:

[0199]

[0200] Generalizing equation (50) yields:

[0201]

[0202] set up Substituting into (51) in (38) gives:

[0203]

[0204]

[0205] therefore,

[0206]

[0207] Substituting (54) into (51) yields:

[0208]

[0209] Equation (55) can be expressed in matrix form such that:

[0210]

[0211] In order to study the disturbance in the input voltage of each module due to interference, it is assumed that The analysis is due to the disturbance in the load current Perturbations in j=1, 2, and 3.

[0212] according to Figure 8 The OCS control scheme block diagram shown in Figure 2 allows the disturbance in the load current to be written as:

[0213]

[0214] Where: G iLd is the control to output current transfer function and G PI is the PI controller transfer function.

[0215] Substituting (57) into (56) gives:

[0216]

[0217] Among them, the items Written as Therefore, the following generalized equation is obtained.

[0218]

[0219] According to the OCS control scheme block diagram, the internal feedback tracks the common reference signal provided by the external output voltage loop. So that: And according to equation (59), the following equation is obtained.

[0220]

[0221] Therefore, it can be inferred from (60) that even when the load current changes, the fluctuation in the input voltage per module is not affected and remains equal to zero.

[0222] Simulation Results

[0223] Dedicated power balancing control provides uniform power distribution among the modules, thus ensuring reliable, safe and stable operation in the presence of external transients and parameter mismatches.The overall control scheme (controller) for the ISOP DC-DC converter is examined.

[0224] The controller is tested considering a three-module ISOP DC-DC converter with a rated power of 4.5 kW and an input voltage of 340 V. The effectiveness of the power balancing controller is evaluated considering parameter mismatch as presented in Table 5. In other words, in order to test the power balancing capability of the controller while handling uncertainties, the component parameters for each module are assumed to be different.

[0225] Table 5. System parameters used in the simulation

[0226]

[0227] The simulation results are shown in Figure 9, testing the control scheme by reflectively charging the output current reference signal. A charging pulse is applied so that the charging cycle starts at 0.2s and ends at 0.6s. Then, a quiescent period lasting 0.1s is applied to the output current reference signal.

[0228] As can be seen from Figure 9, the control scheme for the three-module ISOP DC-DC converter is able to compensate for the mismatch introduced into each module parameter. The results shown in Figure 9 demonstrate that Figure 8 The power sharing controller shown in compensates for the negative effects caused by system parameter mismatch. The modular input voltage and modular output current are equally shared among the four modules. In addition, the output current of the DC-DC converter follows the reference signal applied based on the reflected charging. Therefore, it can be inferred that the control scheme (power sharing controller) is reliable and achieves equal power parts among the modules.

[0229] The present technology including modular DC-DC converters as described herein provides many advantages. For example, the charging process can be completed from a single-phase outlet. The configuration according to an embodiment is an input series output parallel (ISOP) DC-DC converter, wherein control of the ISOP DC-DC converter can be achieved via direct output current sharing (OCS) control and input voltage sharing without the need for an IVS control loop. In other words, control of the ISOP DC-DC converter can be achieved via OCS, and the instability issues associated with conventional ISOP converters controlled via direct OCS are avoided. This is because according to an embodiment, the configuration of the ISOP DC-DC converter includes only a single drive capacitor on the input side.

[0230] The ISOP DC-DC converter also utilizes a smaller number of semiconductor devices, since only one converter is used on the input side according to an embodiment. Furthermore, the use of modular converters allows for low power rating switches and reduces the current stress per module. The parallel connection on the output side also allows for small ripple content, which will reduce the capacitor size at the output and avoid battery degradation.

[0231] According to the embodiments as described herein, possible applications of the present technology including modular DC-DC converters include, for example, low speed electric vehicles (LS-EV), golf carts, and utility electric vehicles. The modular DC-DC converter can also be used as a fast charger for EV applications, where it can provide output current with low ripple content, reduce output filter capacitor size and improve battery life by avoiding significant temperature increase in the battery.

[0232] It should be understood that various changes and modifications to the currently preferred embodiments described herein will be apparent to those skilled in the art. Such changes and modifications can be made without departing from the spirit and scope of the present subject matter and without diminishing its intended advantages. It is therefore intended that such changes and modifications be covered by the appended claims.

Claims

1. A modular DC-DC converter, comprising: a first converter, the first converter being arranged on the input side; a plurality of second converters, the plurality of second converters being arranged on the output side; a plurality of high frequency transformers, wherein the plurality of high frequency transformers are arranged between the first converter and the second converter; External output current loop; as well as a plurality of inner current loops, each of which is provided for each of the plurality of second converters; wherein the external output current loop generates a common reference for the plurality of internal current loops, wherein the first converter and the high frequency transformer are connected in series at the input side, and Therein, the second converters are connected in parallel on the output side.

2. The modular DC-DC converter according to claim 1, wherein: The first converter comprises only one full-bridge converter.

3. The modular DC-DC converter according to claim 1, wherein: The second converter includes at least two full-bridge converters.

4. The modular DC-DC converter according to claim 1, wherein: The first converter includes at least one of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

5. The modular DC-DC converter according to claim 1, wherein: The second converter includes two or more of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

6. The modular DC-DC converter of claim 1, further comprising: A first capacitor is coupled to the first converter at the input side.

7. The modular DC-DC converter of claim 1, further comprising: A second capacitor is coupled to a second one of the second converters at the output side.

8. The modular DC-DC converter according to claim 1, wherein: The second converter includes three full-bridge converters.

9. A battery charging device, comprising: a first converter, the first converter being arranged on the input side; a plurality of second converters, the plurality of second converters being arranged on the output side; a plurality of high frequency transformers, wherein the plurality of high frequency transformers are arranged between the first converter and the second converter; External output current loop; as well as a plurality of inner current loops, each of which is provided for each of the plurality of second converters; wherein the external output current loop generates a common reference for the plurality of internal current loops, wherein the first converter and the high frequency transformer are connected in series at the input side, and Therein, the second converters are connected in parallel on the output side.

10. The battery charging device according to claim 9, wherein: The first converter comprises only one full-bridge converter.

11. The battery charging device according to claim 9, wherein: The second converter includes at least two full-bridge converters.

12. The battery charging device according to claim 9, wherein: The first converter includes at least one of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

13. The battery charging device according to claim 9, wherein: The second converter includes two or more of a flyback converter, a forward converter, a push-pull converter, a half-bridge converter, and a full-bridge converter.

14. The battery charging device according to claim 9, further comprising: A first capacitor is coupled to the first converter at the input side.

15. The battery charging device according to claim 9, further comprising: A second capacitor is coupled to a second one of the second converters at the output side.

16. The battery charging device according to claim 9, wherein: The second converter includes three full-bridge converters.

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