Power conversion device

Abstract

In the power conversion device, there is a semiconductor switch element, which has a control electrode terminal and two main electrode terminals, and the current flowing between the two main electrode terminals is controlled by a drive signal applied to the control electrode terminal; and a drive circuit, which generates the drive signal synchronously with the input signal, and turns on and off the semiconductor switch element by the drive signal, and the drive circuit detects the current flowing between the two main electrodes when the semiconductor switch element is turned off, and if the detected current becomes less than a fixed value, the drive capacity is reduced. Thus, the dv / dt noise of the semiconductor switch element operating in a relatively small current region when switching is reduced.

Description

Technical Field

[0001] The present invention relates to a power conversion device including a semiconductor switching element, and more particularly to a power conversion device that adjusts the driving capability of the semiconductor switching element according to an output current of the power conversion device. Background Art

[0002] Conventionally, there has been proposed a technique for reducing noise generated by switching operations of semiconductor switching elements such as IGBTs constituting a power conversion device.

[0003] For example, Patent Document 1 discloses a switching element driving device, which includes the following steps: after the MOS transistor of the upper arm is turned on with a high driving capability, if the output current of the MOS transistor reaches the output current at the time of the previous switch, the driving capability of the MOS transistor is reduced to mitigate the rate of change of the reverse recovery current of the freewheeling diode of the lower arm, thereby reducing the noise during switching.

[0004] However, there is a problem of how to reduce the dv / dt noise of a power converter that operates in a relatively small output current region. This problem will be described in more detail below with reference to the accompanying drawings.

[0005] For example, in Fig. 9 In the power conversion device shown in FIG. 1 that supplies power to a load having an inductance L, consider the dv / dt at the moment when the IGBT 81a of the upper arm constituting the half-bridge circuit changes from the on state to the off state, causing both the IGBTs of the upper and lower arms to be in the off state, and then the IGBT 82a of the lower arm changes from the off state to the on state. In this case, the waveforms of the collector-emitter voltage (VCE) of the turned-on IGBT 82a of the lower arm and the current (I) flowing in the forward direction of the freewheeling diode 81b of the upper arm are as follows: Fig.10 shown. Fig.10 It is a diagram showing the action starting from the initial state in which both the IGBTs of the upper and lower arms are turned off. The solid line a is the waveform when the power conversion device operates in a region where the output current is relatively large (large current region), and the dotted line b is the waveform when the power conversion device operates in a region where the output current is relatively small (small current region).

[0006] As shown in the figure, in the solid line a, the reverse recovery current of the freewheeling diode 81b flowing through the upper arm recovers smoothly, so the dv / dt of the collector-emitter voltage (VCE) of the IGBT82a of the lower arm is relatively small. In contrast, in the dotted line b, the reverse recovery current of the freewheeling diode 81b flowing through the upper arm recovers sharply, so the dv / dt of the collector-emitter voltage (VCE) of the IGBT82a of the lower arm becomes relatively large. That is, as the current of the power conversion device decreases, the dv / dt noise tends to increase due to the reverse recovery characteristics of the freewheeling diode.

[0007] Therefore, when the current of the power converter is monitored and the power converter is operated in a low current region, it is desirable to take measures to mitigate dv / dt. However, the above-mentioned Patent Document 1 does not reduce dv / dt noise when the power converter is operated in a low current region.

[0008] Prior art literature

[0009] Patent Literature

[0010] Patent Document 1: Japanese Patent Application Publication No. 2018-182818 Summary of the invention

[0011] Technical issues

[0012] The present invention is made in view of the above situation, and its purpose is to provide a power conversion device that can detect the situation where the power conversion device with a semiconductor switching element operates in a small current region, and reduce the dv / dt noise during switching of the semiconductor switching element when operating in the small current region.

[0013] Technical Solution

[0014] In order to achieve the above-mentioned purpose, the power conversion device of the present invention is characterized in that the power conversion device comprises: a semiconductor switching element, which has a control electrode terminal and two main electrode terminals, and the current flowing between the two main electrode terminals is controlled by a driving signal applied to the control electrode terminal; and a driving circuit, which generates the driving signal synchronously with the input signal and turns on and off the semiconductor switching element by the driving signal, and when the semiconductor switching element is turned off, the driving circuit detects the current flowing between the two main electrode terminals of the semiconductor switching element and adjusts the driving capacity.

[0015] In the power converter, the current (main current) flowing between the main electrodes is detected when the semiconductor switching element is turned off, thereby determining the current region (large current region or small current region, etc.) in which the power converter operates.

[0016] Then, the driving capability of turning on and off the semiconductor switch element is adjusted based on the magnitude of the main current detected at the moment of turning off the semiconductor switch element, that is, the magnitude of the current that can be supplied to the control electrode terminal is adjusted. Thus, the dv / dt noise that varies according to the magnitude of the main current can be effectively reduced.

[0017] In particular, when the main current becomes less than a fixed value, it is preferable to reduce the driving capability. This can reduce the rate of change of the voltage between the main electrodes when the semiconductor switch element is turned on in the next cycle, thereby effectively reducing dv / dt noise.

[0018] It should be noted that, if a semiconductor switch element including a sense terminal that outputs a sense current corresponding to the main current is used, the main current can be easily detected using the sense current.

[0019] Specifically, the driving circuit can be composed of a current detection unit that outputs a selection signal based on the size of the sensed current, a switch unit that outputs a voltage signal for determining the driving capability based on the selection signal, and a driving unit that generates the driving signal using the voltage signal and the input signal.

[0020] The current detection unit includes: a sense voltage generating circuit that converts a sense current output from a sense terminal into a voltage and outputs the voltage as a sense voltage; a comparator that compares the sense voltage with a reference voltage and outputs the comparison result; a latch circuit that holds a signal indicating the comparison result in synchronization with the input signal; and a selection circuit that outputs the selection signal based on the output of the latch circuit.

[0021] The switch unit includes: a plurality of voltage-dividing resistor elements inserted in series between a voltage source and a reference potential; and a switch element that selects any one of at least two different voltages generated in the plurality of voltage-dividing resistor elements according to the selection signal and outputs it as the voltage signal.

[0022] The driving unit outputs a driving signal of a current corresponding to the voltage signal in synchronization with the input signal to control the semiconductor switch element to be turned on, and extracts the charge stored in the control electrode of the semiconductor switch element when the semiconductor switch element is turned off.

[0023] When the semiconductor switch element is turned off, the comparator of the current detection unit determines whether the sense voltage is equal to or higher than the reference voltage, and the determination result is held by the latch circuit until the next cycle is turned off.

[0024] The sense voltage generating circuit of the current detection unit can be configured by inserting a resistance element between the sense terminal and the reference potential.

[0025] In the configuration of the current detection unit, it is preferred that a constant current source is also provided, and the sensing voltage is increased by a predetermined voltage by passing the current output from the constant current source through the resistor element. In the current region where the main current is relatively small and dv / dt noise becomes a problem, if the sensing current is converted into a sensing voltage for detection in this region by using a configuration of a general resistor element, the sensing voltage corresponding to this region becomes an extremely small value of, for example, about 0.1V. Therefore, by superimposing the constant current output from the constant current source on the resistor element that generates the sensing voltage, the sensing voltage can be simply increased to the operation guarantee range of the comparator. At this time, the reference voltage input to the comparator is also set higher than the increase amount.

[0026] It should be noted that the value of the constant current and the resistance value of the resistor element are determined by the value of the sense current serving as a threshold value and the operation guarantee range of the comparator.

[0027] Technical Effects

[0028] According to the present invention, since the driving current of the semiconductor switch element is suppressed by detecting the operation in the low current region, the dv / dt noise can be effectively reduced. In addition, by suppressing the driving current according to the current region, the power loss and noise of the semiconductor switch element can be optimized compared with the past. BRIEF DESCRIPTION OF THE DRAWINGS

[0029] Figure 1 It is a block diagram showing the overall configuration of the power conversion device according to the first embodiment of the present invention.

[0030] Figure 2 It is shown Figure 1 A circuit diagram of an example of a filter circuit.

[0031] Figure 3 It is shown Figure 1 A circuit diagram of an example of a latch circuit.

[0032] Figure 4 It is shown Figure 1 A circuit diagram of an example of a selection circuit.

[0033] Figure 5 It is a waveform diagram showing the effect of the embodiment of the present invention.

[0034] Figure 6 It is a block diagram showing the overall configuration of a power conversion device according to a second embodiment of the present invention.

[0035] Figure 7 yes Figure 6 Other embodiments of the present invention.

[0036] Figure 8It is a block diagram showing the overall configuration of a power conversion device according to a third embodiment of the present invention.

[0037] Fig. 9 This is a diagram for explaining problems of conventional power conversion devices.

[0038] Fig.10 This is a waveform diagram for explaining problems of a conventional power conversion device.

[0039] Explanation of symbols

[0040] 1 IPM (power conversion unit)

[0041] 2 Switching elements

[0042] 3 IC chip (drive circuit)

[0043] 21, 81a, 82a IGBT (semiconductor switching element)

[0044] 21a Current sensing IGBT

[0045] 21b Sense terminal

[0046] 22, 81b, 82b freewheeling diode

[0047] 30 Sensing voltage generation circuit

[0048] 31 Current detection unit

[0049] 32, 338 Comparator

[0050] 33 Filter circuit

[0051] 34 Latch circuit

[0052] 35 Selection Circuit

[0053] 36 Switch

[0054] 37 Drive unit

[0055] 38, 39, 40, 332 Constant current source

[0056] 50, 51 Bias circuit

[0057] 331, 351 NOT elements

[0058] 337 Capacitor

[0059] 341 D-type flip-flop

[0060] 371 Operational Amplifier

[0061] 333, 334, 335, 375, 376, 377, 378 P-type MOSFET

[0062] 336, 372, 373, 374 N-type MOSFET

[0063] R1~R6 resistance element

[0064] S signal line

[0065] SW1, SW2 switch elements DETAILED DESCRIPTION

[0066] Hereinafter, the power conversion device of each embodiment of the present invention will be described in detail with reference to the accompanying drawings. The power conversion device of the present invention is characterized in that the following circuit configuration is adopted, which makes it possible to adjust the driving capability (capability of providing current to charge the gate of the switching element) according to the main current of the semiconductor switching element, and reduce the switching noise of the semiconductor switching element, specifically, the influence of dv / dt noise when the semiconductor switching element is turned on. Hereinafter, each embodiment will be described.

[0067] (First Embodiment)

[0068] (constitute)

[0069] The IPM (Intelligent Power Module) 1 of this embodiment is a power conversion device suitable for, for example, joints of industrial robots, elevators, or commercial air conditioners. Figure 1 As shown, the structure mainly includes a switching element 2 and an IC chip 3.

[0070] The switching element 2 is sometimes composed of an IGBT 21 and a freewheeling diode 22 connected in reverse parallel to the IGBT 21, or it can be a single chip RC-IGBT. The IGBT 21 is turned on according to a drive signal applied to the gate (control electrode terminal) from the IC chip 3, and a current (main current) flows from the collector to the emitter, supplying a fixed power to a load not shown. The emitter of the IGBT 21 is connected to a reference potential of the IC chip 3 described later. In addition, a current sensing IGBT 21a is connected in parallel to the IGBT 21, and a sensing current is output at a fixed ratio according to the main current.

[0071] The IC chip 3 is a driving circuit that changes the magnitude of the current driving the IGBT 21 according to the magnitude of the sensed current, and mainly includes a current detection unit 31, a switch unit 36, and a driving unit 37. The current detection unit 31 outputs a selection signal to the switch unit 36 ​​based on the magnitude of the sensed current. The switch unit 36 ​​outputs a reference voltage signal for determining the magnitude of the driving current of the IGBT 21 based on the selection signal. The driving unit 37 generates a driving current of a predetermined magnitude based on the reference voltage signal and outputs it to the gate of the IGBT 21.

[0072] The current detection unit 31 includes: a comparator 32; a filter circuit 33 connected to the output terminal of the comparator 32; a latch circuit 34 that latches the output signal of the filter circuit 33; and a selection circuit 35 that receives the output signal of the latch circuit 34 and outputs a selection signal to the switch unit 36.

[0073] The non-inverting input terminal of the comparator 32 is connected to the sensing terminal 21b of the IGBT 21 and is connected to the reference potential via the resistor R1. The sensing current output from the sensing terminal 21b is converted into a voltage (sense voltage) by the resistor R1. The inverting input terminal of the comparator 32 is connected to the positive side of the reference voltage source Vref1. The negative side of the reference voltage source Vref1 is connected to the reference potential.

[0074] The comparator 32 compares the voltage across the resistor R1 with the reference voltage Vref1 and outputs an H-level signal if the voltage across the resistor R1 is larger, or outputs an L-level signal if the voltage value of the reference voltage source Vref1 is larger.

[0075] The filter circuit 33 is a circuit that delays the rise of the output signal of the comparator 32 by a predetermined time. Figure 2 The circuit configuration shown is used to achieve this.

[0076] like Figure 2As shown, the filter circuit 33 includes a P-type MOSFET 333 and a P-type MOSFET 334 connected to the power supply VCC and forming a current mirror circuit, and a constant current source 332 inserted between the drain of the P-type MOSFET 333 and the reference potential. In addition, a P-type MOSFET 335 and an N-type MOSFET 336 are inserted in series between the drain of the P-type MOSFET 334 and the reference potential. The connection point of the P-type MOSFET 335 and the N-type MOSFET 336 is connected to the non-inverting input terminal of the comparator 338 and one end of the capacitor 337. The inverting input terminal of the comparator 338 is connected to the positive side of the reference voltage source Vref2. The other end of the capacitor 337 and the other end of the reference voltage source Vref2 are connected to the reference potential. The input end of the NOT element 331 becomes the input end of the filter circuit 33 and is connected to the output end of the comparator 32. The output end of the NOT element 331 is connected to the gate of the P-type MOSFET 335 and the gate of the N-type MOSFET 336. The output terminal of the comparator 338 becomes the output terminal of the filter circuit 33 and is connected to the input terminal of the latch circuit 34 .

[0077] In the filter circuit 33 configured in this way, when the input to the filter circuit 33 changes from the L level to the H level, the output of the NOT element 331 becomes the L level. Then, the N-type MOSFET 336 becomes OFF, the P-type MOSFET 335 becomes ON, and the capacitor 337 is charged with the current specified by the constant current source 332. Then, when the charging voltage of the capacitor 337 exceeds the reference voltage Vref10, the output of the comparator 338 becomes the H level.

[0078] Thereafter, when the input to the filter circuit 33 becomes L level, the output of the NOT element 331 becomes H level. Then, the N-type MOSFET 336 becomes on, the P-type MOSFET 335 becomes off, and the charge charged in the capacitor 337 is rapidly discharged. As a result, the voltage at the non-inverting input terminal of the comparator 338 becomes lower than the reference voltage Vref2, and the output of the comparator 338 becomes L level.

[0079] Therefore, the filter circuit 33 delays the output for a fixed time only when the input signal becomes H level. The delay time can be adjusted according to the charging time of the capacitor 337 and the reference voltage Vref2. The charging time of the capacitor 337 is determined by the output current value of the constant current source 332, the mirror ratio of the current mirror circuit, and the electrostatic capacitance value of the capacitor 337.

[0080] exist Figure 3, an example of the structure of the latch circuit 34 is shown. As shown in the figure, the latch circuit 34 can be implemented by a D-type flip-flop 341. The D terminal (data input terminal) of the D-type flip-flop 341 is connected to the output terminal of the filter circuit 33, and the C terminal (clock input terminal) is connected to the signal line S, and the input signal S1 is input. In addition, the Q terminal (output terminal) of the D-type flip-flop 341 is connected to the input terminal of the selection circuit 35.

[0081] The latch circuit 34 having such a configuration latches the output signal TL (L level or H level) of the filter circuit 33 according to the rise of the input signal S1 supplied from the external control circuit (not shown) via the signal line S, and outputs its state to the Q terminal. The output state of the Q terminal is maintained until the next rise of the input signal S1. Although it will be described in detail later, the IGBT is turned on when the input signal S1 is at the L level, and the IGBT is turned off when the input signal S1 is at the H level.

[0082] The selection circuit 35 outputs selection signals VL and VM generated based on the output signal from the latch circuit 34 to the switch section 36. Figure 4 An example of the configuration of the selection circuit 35 is shown in FIG.

[0083] The input terminal of the NOT element 351 is connected to the output terminal of the latch circuit 34. The output terminal of the NOT element 351 becomes the output terminal of the selection circuit 35, and the selection signal VL is output from the output terminal. In addition, the input terminal of the NOT element 351 also becomes another output terminal of the selection circuit 35, and the selection signal VM is output from the output terminal.

[0084] When the main current of the IGBT 21 is small, the sense voltage generated based on the sense current becomes lower than the reference voltage Vref1, and the output of the comparator 32 becomes L level. As a result, the output of the latch circuit 34 latched in response to the rise of the input signal S1 becomes L level. Therefore, since the input of the NOT element 351 is L level and the output becomes H level, the selection signal VL of the selection circuit 35 becomes H level, and the selection signal VM becomes L level.

[0085] When the main current of the IGBT 21 is large, the sense voltage becomes larger than the reference voltage Vref1, so the output of the comparator 32 becomes H level. As a result, the output of the latch circuit 34 latched in accordance with the rise of the input signal S1 becomes H level. Therefore, since the input of the NOT element 351 is H level and the output becomes L level, the selection signal VL of the selection circuit 35 becomes L level, and the selection signal VM becomes H level.

[0086] The switch section 36 includes a switch element SW1 and a switch element SW2, resistor elements R2 to R4 connected in series to form a resistor voltage divider circuit, and a reference voltage Vref2. One end (resistance element R4 side) of the resistor voltage divider circuit formed by the resistor elements R2 to R4 is connected to the reference voltage Vref2, and the other end (resistance element R2 side) is connected to the reference potential.

[0087] The switch element SW1 and the switch element SW2 can be respectively constituted by, for example, a CMOS switch (transmission gate) formed by connecting an N-type MOSFET and a P-type MOSFET in parallel, and become conductive when the gate signal is at an H level. The switch element SW1 or the switch element SW2 is selectively turned on according to the output from the selection circuit 35. Specifically, a selection signal VL is input to the gate of the switch element SW1, and the switch element SW1 is turned on when the output signal of the comparator 32 is at an L level. A selection signal VH is input to the gate of the switch element SW2, and the switch element SW2 is turned on when the output signal of the comparator 32 is at an H level.

[0088] The input side of the switch element SW1 is connected to a connection point between the resistor element R2 and the resistor element R3 , and the output side is connected to a non-inverting input terminal of an operational amplifier (Operational amplifier) ​​371 described later.

[0089] The input side of the switch element SW2 is connected to the connection point between the resistor element R3 and the resistor element R4 , and the output side is similarly connected to the non-inverting input terminal of the operational amplifier 371 .

[0090] By means of the resistor voltage divider circuit composed of the resistor elements R2 to R4, a relatively low voltage Vref4 is applied to the non-inverting input terminal of the operational amplifier 371 when the switch element SW1 is turned on, and a relatively high voltage Vref3 (> Vref4) is applied to the non-inverting input terminal of the operational amplifier 371 when the switch element SW2 is turned on. These correspond to the case where the main current of the IGBT 21 is small and the case where the main current of the IGBT 21 is large, respectively. It should be noted that the resistance values ​​of the resistor elements R2 to R4 are appropriately set so as to obtain the desired resistor voltage divider effect.

[0091] The driving unit 37 changes the driving capability of the IGBT 21 according to the magnitude of the main current of the IGBT 21. The driving unit 37 includes an operational amplifier 371, an N-type MOSFET 372, turn-off MOSFETs 373 and 374, a power supply VCC, P-type MOSFETs 375 and 376 connected to the power supply VCC to form a current mirror circuit, and a resistor R5. The basic principle of the driving unit 37 is described in, for example, Japanese Patent Publication No. 2013-219633, so the detailed description of the whole is omitted here, but the driving unit 37 is a component that supplies the current obtained by applying the voltage Vref3 or Vref4 to the resistor R5 to the gate of the IGBT 21 according to the virtual short circuit of the operational amplifier 371. The following is a description of the points related to this embodiment.

[0092] As a point related to the present embodiment, the shutdown MOSFET 373 (switching circuit) is described. The shutdown MOSFET 373 is composed of, for example, an N-type MOSFET, and its drain is connected to the drain of the P-type MOSFET 376 constituting the current mirror circuit, and is connected to the gate of the IGBT 21. The source of the shutdown MOSFET 373 is connected to the reference potential. In addition, the gate of the shutdown MOSFET 373 is connected to the signal line S. The shutdown MOSFET 373 is turned on by supplying an H-level input signal S1 to the gate via the signal line S, and the gate of the IGBT 21 becomes conductive with the reference potential. As a result, the charge accumulated in the gate of the IGBT 21 is extracted and the IGBT 21 is turned off. It should be noted that at this time, the shutdown MOSFET 374 is also turned on, and as a result, the N-type MOSFET 372 is turned off, so that the current output from the drain of the P-type MOSFET 376 as the output of the current mirror circuit becomes zero.

[0093] On the other hand, the turn-off MOSFET 373 is turned off when an L-level input signal S1 is applied to the gate via the signal line S, and conduction between the gate of the IGBT 21 and the reference potential is cut off. Then, a predetermined drive signal is supplied from the P-type MOSFET 376 to the gate of the IGBT 21, turning on the IGBT 21.

[0094] As described above, when the input signal S1 is at the L level, the IGBT 21 is turned on, and when the input signal S1 is at the H level, the IGBT 21 is turned off.

[0095] (action)

[0096] Next, the operation of the IPM 1 having the above configuration will be described.

[0097] In the current detection unit 31 , a sense voltage proportional to the magnitude of the main current of the IGBT 21 is applied to the non-inverting input terminal of the comparator 32 .

[0098] When the magnitude of the main current of the IGBT 21 is larger than the fixed value, the sense voltage becomes larger than the reference voltage Vref1, so the output signal of the comparator 32 becomes H level. Then, the output of the filter circuit 33 and the output of the latch circuit 34 latched in response to the rise of the input signal S1 both become H level. As a result, the selection signal VL output from the selection circuit 35 becomes L level, and the selection signal VM becomes H level.

[0099] According to the selection signal, the switch element SW1 of the switch section 36 is placed in a non-conductive state, the switch element SW2 is placed in a conductive state, and a relatively high reference voltage Vref3 is output to the operational amplifier 371 of the drive section 37 .

[0100] While the input signal S1 is at the H level, the N-type MOSFET 374 is turned on and the N-type MOSFET 372 is turned off. Therefore, no current flows through the P-type MOSFET 375 and the P-type MOSFET 376 constituting the current mirror circuit. On the other hand, since the N-type MOSFET 373 is turned on, the gate of the IGBT 21 becomes the L level, and the IGBT 21 is turned off.

[0101] Thereafter, when the input signal S1 changes to the L level, the N-type MOSFET 374 is turned off, and the output of the operational amplifier 371 becomes an output corresponding to the reference voltage Vref3. As a result, the N-type MOSFET 372 is turned on, and the voltage across the resistor R5 becomes Vref3. If r5 represents the resistance value of the resistor R5, a current of [voltage value Vref3 / resistance value r5] flows through the resistor R5. Since the N-type MOSFET 373 is turned off, a current proportional to [voltage value Vref3 / resistance value r5] is supplied to the gate of the IGBT 21 from the P-type MOSFET 376 constituting the current mirror circuit, turning on the IGBT 21. In this way, when the power conversion device operates in a large current region, IGBT21 is turned off when the input signal S1 is at an H level, and a drive signal (gate charging current) proportional to [voltage value Vref3 / resistance value r5] (the proportional coefficient is the mirror ratio of the current mirror circuit) is supplied to the gate of IGBT21 when the input signal S1 is at an L level, thereby turning on the IGBT21.

[0102] When the magnitude of the main current of the IGBT 21 is less than the fixed value, the sense voltage becomes less than the reference voltage Vref1, and therefore the output signal of the comparator 32 becomes L level. Then, the output of the filter circuit 33 and the output of the latch circuit 34 latched in response to the rise of the input signal S1 also become L level. As a result, the selection signal VL outputted from the selection circuit 35 becomes H level, and the selection signal VM becomes L level.

[0103] According to the selection signal, the switch element SW1 of the switch section 36 is turned on, the switch element SW2 is turned off, and a relatively low reference voltage Vref4 is output to the operational amplifier 371 of the drive section 37 .

[0104] During the period when the input signal S1 is at the H level, the IGBT21 is turned off in the same manner as described above. Thereafter, when the input signal S1 changes to the L level, the N-type MOSFET374 is turned off, and the output of the operational amplifier 371 becomes an output corresponding to the reference voltage Vref4. As a result, the N-type MOSFET372 is turned on, and the voltage across the resistor R5 becomes Vref4. Therefore, a current of [voltage value Vref4 / resistance value r5] flows through the resistor R5. The N-type MOSFET373 is turned off, and a current proportional to [voltage value Vref4 / resistance value r5] is supplied to the gate of the IGBT21 from the P-type MOSFET376 constituting the current mirror circuit, turning on the IGBT21. In this way, when the power conversion device operates in the low current region, the IGBT21 is turned off when the input signal S1 is at the H level, and when the input signal S1 is at the L level, the IGBT21 is supplied with a drive signal (gate charging current) proportional to [voltage value Vref4 / resistance value r5] and becomes conductive.

[0105] Since [voltage value Vref3 / resistance value r5 ]>[voltage value Vref4 / resistance value r5 ], when the power conversion device operates in a small current region, the driving capability for the gate of the IGBT 21 decreases compared to when the power conversion device operates in a large current region.

[0106] (Function, Effect)

[0107] As described above, according to this embodiment, the driving capability of the gate of the IGBT 21 is adjusted according to the main current of the IGBT 21 constituting the power conversion device. Specifically, the magnitude of the main current of the IGBT is monitored by sensing the current, and if the value becomes less than a fixed value, it is considered that the power conversion device is operating in a low current region, and the driving capability of the gate of the IGBT is reduced. As a result, if Figure 5As shown, dv / dt is relaxed. In this figure, the single-point dashed line c is the action waveform when the driving capability is reduced in the small current region. It can be seen that compared with the case where the driving capability is not adjusted even in the small current region (the previous case) as shown by the dotted line b, the change rate of the collector-emitter voltage (VCE) becomes gentle.

[0108] In this way, by increasing the driving capability in the large current region where loss is more of a problem because the influence of noise is small, and reducing the driving capability in the small current region where switching noise is a problem (because the current is small, the loss is relatively small), the power loss and noise of the semiconductor switching element can be optimized compared to the past.

[0109] (Second Embodiment)

[0110] Next, a second embodiment will be described.

[0111] Figure 6 2 is a block diagram of a power conversion device according to a second embodiment. Figure 1 The difference is that a constant current source 38 is provided on the connection point of the resistor R1 that converts the sense current into a voltage and the sense terminal 21b. Figure 1 Since the same elements are the same, the same reference numerals are given to the same elements and the description thereof is omitted.

[0112] The constant current output from the constant current source 38 flows to the reference potential through the resistor R1. As a result, the sense voltage value increases by the voltage generated by the constant current value and the resistance value of the resistor R1.

[0113] In fact, sometimes about 10% of the rated current of the IGBT is defined as the low current area. Figure 1 As shown, when the low current region is determined by using the sense current, the reference voltage Vref1 of the comparator 32 becomes a low value of about 0.1 V. Such a low value may be out of the operating range of the comparator, and the comparator may not be able to compare input signals with high accuracy.

[0114] According to the present embodiment, since a fixed voltage generated by a constant current is added to a sense voltage generated by passing the sense current of the IGBT 21 through the resistor R1, and the added voltage is compared with the reference voltage Vref1, the reference voltage Vref1 can be set within the operating range of the comparator. Thus, whether the power conversion device is operating in a low current region can be detected with high accuracy.

[0115] (Other embodiments)

[0116] exist Figure 7Another embodiment for detecting a low current region with high accuracy is shown in FIG. Figure 6 The difference is that a resistor R6 is inserted between the constant current source and the resistor R1 . The connection point between the resistor R1 and the resistor R6 is connected to the sensing terminal 21 b , and the connection point between the resistor R6 and the constant current source 38 is connected to the non-inverting input terminal of the comparator 32 .

[0117] In this embodiment, a fixed voltage generated by passing a constant current through resistors R1 and R6 is added to the sense voltage generated across resistor R1 by the sense current. This allows the IGBT to be detected in the low current region within the guaranteed operation range of the comparator.

[0118] (Third Embodiment)

[0119] Next, a third embodiment will be described.

[0120] Figure 8 2 is a diagram showing the configuration of a power conversion device according to a third embodiment. Figure 1 The difference is that a series circuit of a constant current source and a P-type MOSFET is inserted between the inverting input terminal of the comparator 32 and the reference voltage source Vref1 , and between the non-inverting input terminal of the comparator 32 and the output terminal of the sensing voltage generating circuit 30 .

[0121] Hereinafter, the configuration of this embodiment will be described in detail.

[0122] exist Figure 8 In the embodiment, the source of the P-type MOSFET 377 is connected to the output terminal of the constant current source 39 and the non-inverting input terminal of the comparator 32. The drain of the P-type MOSFET 377 is connected to the reference potential. The gate of the P-type MOSFET 377 is connected to one end of the resistor element R1 and the sensing terminal 21b. The other end of the resistor element R1 is connected to the reference potential.

[0123] In addition, the source of the P-type MOSFET 378 is connected to the output terminal of the constant current source 40 and the inverting input terminal of the comparator 32. The drain of the P-type MOSFET 378 is connected to the reference potential. The gate of the P-type MOSFET 378 is connected to the reference voltage source Vref1. The constant current source 39 and the constant current source 40 have the same temperature characteristics.

[0124] In addition to Figure 1 Since the same elements are the same, the same reference numerals are given to the same elements and the description thereof is omitted.

[0125] In the IPM1 having the above-mentioned structure, a voltage value obtained by increasing the sense voltage by a voltage (bias voltage) mainly determined by the output current value of the constant current source 39 and the on-resistance value of the P-type MOSFET 377 is input to the non-inverting input terminal of the comparator 32. On the other hand, a voltage value obtained by increasing the voltage value of the reference voltage source Vref1 by a voltage (bias voltage) mainly determined by the output current value of the constant current source 40 and the on-resistance value of the P-type MOSFET 378 is input to the inverting input terminal of the comparator 32. Therefore, the series circuit of the constant current source 39 and the P-type MOSFET 377, and the series circuit of the constant current source 40 and the P-type MOSFET 378 function as bias circuits (50, 51) for increasing the voltage, respectively.

[0126] The output current values ​​of the constant current sources 39 and 40 are adjusted in advance so that the voltages input to the non-inverting input terminal and the inverting input terminal of the comparator 32 become predetermined voltage values ​​when the IPM 1 is at a certain temperature.

[0127] In the operating state of the IPM1 adjusted in this way, if the temperature rises, the output current values ​​of the constant current sources 39 and 40 vary due to the influence of temperature. However, even in such a case, since the bias voltage also varies in the same manner due to the use of constant current sources with the same temperature characteristics, the difference in the voltage values ​​input to the non-inverting input terminal and the inverting input terminal of the comparator 32 respectively becomes a value that cancels the voltage variation caused by the temperature characteristics of the constant current sources 39 and 40.

[0128] Therefore, in e.g. Figure 6 , Figure 7 In the circuit shown, the detection accuracy of the comparator 32 may decrease due to the temperature characteristics of the constant current source 38, but the circuit of this embodiment can suppress the decrease in detection accuracy. That is, according to this embodiment, in addition to the effects of the first and second embodiments, the magnitude of the current of the IGBT can be detected with high accuracy even when the temperature in the device changes.

[0129] The present invention is not limited to the above-mentioned embodiments, and can be implemented in various modifications without departing from the scope of the main purpose. For example, although in the above-mentioned embodiments, the operation area is divided into a large current area and a small current area, and the driving ability of the IGBT is reduced when the operation in the small current area is detected, the current area can be further subdivided, and the driving ability can be adjusted in multiple stages according to the size of the output current.

Claims

1. A power conversion device, characterized in that: The power conversion device comprises: A semiconductor switch element having a control electrode terminal and two main electrode terminals, wherein a current flowing between the two main electrode terminals is controlled by a drive signal applied to the control electrode terminal; as well as a drive circuit that generates the drive signal in synchronization with an input signal and turns the semiconductor switch element on and off by the drive signal, The drive circuit detects the current flowing between the two main electrode terminals of the semiconductor switch element and adjusts the drive capability when the semiconductor switch element is turned off. The drive circuit reduces the drive capability when the current flowing between the two main electrode terminals of the semiconductor switch element detected at the time of turning off the semiconductor switch element becomes equal to or less than a fixed value.

2. The power conversion device according to claim 1, characterized in that: The semiconductor switch element includes a sense terminal that outputs a sense current corresponding to a current flowing between the two main electrode terminals. The drive circuit detects a current flowing between the two main electrode terminals using the sense current output from the sense terminal.

3. The power conversion device according to claim 2, characterized in that: The driving circuit comprises: a current detection unit that outputs a selection signal based on the magnitude of the sensed current; a switch section that outputs a voltage signal for determining the driving capability based on the selection signal; as well as The driving unit generates the driving signal by using the voltage signal and the input signal.

4. The power conversion device according to claim 3, characterized in that: The current detection unit comprises: a sensing voltage generating circuit that converts the sensing current output from the sensing terminal into a voltage and outputs it as a sensing voltage; a comparator that compares the sensed voltage with a reference voltage and outputs a result of the comparison; a latch circuit for holding a signal indicating a result of the comparison in synchronization with the input signal; as well as A selection circuit outputs the selection signal based on the output of the latch circuit.

5. The power conversion device according to claim 4, characterized in that: The sense voltage generating circuit is constituted by a resistance element inserted between the sense terminal and a reference potential, and outputs a voltage generated by a sense current output from the sense terminal flowing through the resistance element.

6. The power conversion device according to claim 5, characterized in that: The sense voltage generating circuit further includes a constant current source, and causes a current output from the constant current source to flow through the resistor element, thereby increasing the sense voltage by a predetermined voltage.

7. The power conversion device according to claim 6, characterized in that: inserting a second resistance element between the constant current source and the resistance element, The sensing voltage is separated from a connection point between the constant current source and the second resistance element.

8. The power conversion device according to claim 7, characterized in that: The switch unit comprises: a plurality of voltage-dividing resistor elements inserted in series between a voltage source and a reference potential; and A switch element selects any one of at least two different voltages generated in the plurality of voltage-dividing resistor elements according to the selection signal, and outputs the selected voltage as the voltage signal.

9. The power conversion device according to claim 8, characterized in that: The driving unit comprises: a current mirror circuit that supplies a driving signal of a current corresponding to the voltage signal to a control electrode terminal of the semiconductor switch element in synchronization with the input signal to turn on the semiconductor switch element; as well as A switch circuit extracts the charge accumulated in the control electrode of the semiconductor switch element to turn off the semiconductor switch element.

10. The power conversion device according to claim 4, characterized in that: The power conversion device comprises: A first bias circuit is inserted between the comparator and the sensing voltage generating circuit, and connects a first constant current source and a first MOSFET in series to generate a first bias voltage; as well as a second bias circuit, which is inserted between the comparator and the voltage source generating the reference voltage, and connects a second constant current source having the same temperature characteristics as the first constant current source in series with a second MOSFET to generate a second bias voltage; The first bias circuit adds a first bias voltage to the sensing voltage and supplies the first bias voltage to the non-inverting input terminal of the comparator. The second bias circuit adds a second bias voltage to the reference voltage and supplies the resultant voltage to the inverting input terminal of the comparator.

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