System and method for current control of boost converter with dual counter-wound inductors

Abstract

A system may include a power converter including at least one stage having: a double counter-wound inductor configured such that its windings generate opposing magnetic fields in its magnetic core; and a current control subsystem for controlling current through the double counter-wound inductor. The current control subsystem may be configured to minimize the magnitude of the magnetizing current of the double counter-wound inductor to prevent core saturation of the double counter-wound inductor and to adjust the amount of output current delivered by the power converter to a load according to a reference input signal.

Description

Technical Field

[0001] The present disclosure relates generally to circuits for audio devices, piezoelectric devices, haptic feedback devices, and / or other devices, including but not limited to personal audio devices such as wireless telephones and media players, and more particularly to an enhanced multi-level boost converter that may be used in such devices. Background Art

[0002] Personal audio devices (including wireless phones, such as mobile / cellular phones, cordless phones, mp3 players and other consumer audio devices) are widely used. Such personal audio devices may include circuit systems for driving a pair of headphones, one or more speakers, piezoelectric transducers, tactile feedback transducers and / or other transducers. Such circuit systems generally include a driver, which includes a power amplifier for driving a transducer output signal to the transducer. Typically, a power converter can be used to provide a supply voltage to the power amplifier so as to amplify the signal driven to the speaker, earphone, piezoelectric transducer, tactile feedback transducer or other transducers. A switching power converter is an electronic circuit that converts a power supply from a direct current (DC) voltage level to another DC voltage level. Examples of such switching DC-DC converters include, but are not limited to, boost converters, buck converters, buck-boost converters, inverting buck-boost converters and other types of switching DC-DC converters. Therefore, using a power converter, a DC voltage such as a DC voltage provided by a battery can be converted into another DC voltage for powering a power amplifier.

[0003] A battery powered system may use a boost converter to generate a power supply for an audio amplifier that is greater than the voltage of the battery. For example, the motivation for using a boost converter in a battery powered transducer is to generate a larger signal swing at the output of the transducer amplifier than can be achieved by powering the amplifier directly from the battery.

[0004] Traditionally, while a boost converter and an amplifier powered from the boost converter are typically fabricated on the same integrated circuit, the boost converter typically requires a boost inductor external to the integrated circuit, which requires a large amount of space. However, recent advances in manufacturing have enabled inductors with magnetic cores to be integrated into the integrated circuit die. Advantages of integrated inductors can include smaller overall circuit area, a significant reduction in height in a direction perpendicular to the surface of the integrated circuit, lower electromagnetic interference emissions, and less variation in the physical properties of the inductor.

[0005] Despite advances in inductor manufacturing, designing a boost converter with an integrated inductor can be challenging. External boost converter inductors for audio applications typically have an inductance between 1μH and 2μH and saturate at currents between 2.5A and 4A. However, a typical integrated inductor may have an inductance in the range of tens to hundreds of nanohenries, with a current saturation limit at or below 1A. A typical boost converter for audio may supply 12V to a 10W load from a 4V battery supply. Therefore, even assuming 100% efficiency, a standard boost converter design would draw an input current of 2.5A, which is well beyond the saturation point of the integrated inductor. A multiphase converter can be used to distribute the current to multiple inductors, but the smaller inductors result in larger current ripple, which may still exceed the saturation limit.

[0006] In order to use an integrated inductor, the design of the power converter must overcome its limitations of low inductance and low saturation current. One solution to this problem is to use a multi-wound inductor with an improved boost converter architecture.

[0007] Multi-wound inductors can be used to weaken the magnetic field in the core and prevent premature saturation. Figure 1A A multi-wound inductor 100 is depicted having two coils 102a and 102b wound around a common magnetic core 104 . Figure 1B A cross-sectional side view of inductor 100 is depicted, depicting the current in each of coils 102a and 102b, where "·" depicts current I1 flowing out of the page in a direction perpendicular to the plane of the page, and at the same time "X" depicts current I2 flowing into the page in a direction perpendicular to the plane of the page. Coils 102a and 102b can be wound in opposite directions so that positive currents generate opposite fields in each coil. Therefore, the total magnetic flux through core 104 is can be equal to the magnetic flux from coil 102a and the magnetic flux from coil 102b The difference between the magnetic flux in coils 102a and 102b The currents I1 and I2 in the coils 102a and 102b may be respectively proportional.

[0008] When the magnetic field in the magnetic core 104 exceeds the threshold value B sat When the inductor 100 is saturated, the magnetic field can be related to the total magnetic flux in the magnetic core 104. is proportional to, and therefore can be proportional to the difference in current (e.g., I1-I2). As a result, the saturation limit of inductor 100 can be given as:

[0009]

[0010] in is the difference between current I1 and the current I2 that saturates the inductor 100, and for an integrated inductor this difference may typically be around 0.5 A to 1.0 A. The above equation (1) may only be valid for low to moderate levels of current. Figure 1C The saturation curve of current I2 versus current I1 is shown. The dashed line depicts the saturation boundary 108 from equation (1), while the shaded area depicts the true saturation region 110 defined by the boundary ABCDE. For low currents, the unsaturated region 112 is a strip along the main diagonal, as described in equation (1). However, at larger currents, the unsaturated region 112 shrinks in width until, at very large currents, the inductor 100 is always saturated. This effect may occur because the field cancellation between coils 102a and 102b may not be perfect, particularly at their respective ends. Moreover, some inductor designs may use an additional one of coils 102a, 102b to control the coupling coefficient, which may further reduce the field cancellation. As a result, the inductor 100 may saturate even if the current difference |I1–I2| is within its limits. Therefore, the condition of equation (1) may represent a necessary (but not sufficient) condition for saturation. Alternatively, a sufficient condition for the inductor 100 to be unsaturated is that the currents I1 and I2 must be located in the unsaturated region 112 defined by points ABCDE.

[0011] Multi-wound inductors can extend the range of winding currents before the device saturates. For example, if current I2 is zero, current I1 may extend only to Figure 1C , and remain unsaturated. However, with appropriately chosen values ​​of current I2, the range of current I1 can be extended to point D or even point C and remain unsaturated due to the field cancellation of currents I1 and I2. This range extension can be used to help solve the saturation problem of the integrated boost inductor. However, the boost architecture must also be designed to take advantage of the benefits of the multi-wound inductor.

[0012] Figure 2 An example of a single-stage boost converter 200 that can be used with a multi-wound inductor 100 and having a load 202 is depicted. The single-stage boost converter 200 can use a capacitor 204 to stabilize its output voltage V out The battery 206 can supply the input voltage V to the single-stage boost converter 200. in The single-stage boost converter 200 may include a plurality of switches 210, 212, 214, and 216, each of which has a gate G to receive a control signal to control the conductivity of the switch (eg, selectively open and close the switch). Such a control signal may include Figure 2 The pulse width modulated control signals marked as P1 and P2 in Figure 2 Marked as and The switches 210 and 212 may switch the top coil 102a of the inductor 100 between a charging state (where the coil 102a is coupled between the battery 206 and ground) and a transfer state (where the coil 102a is coupled between the power source 206 and the load 202). Similarly, the switches 214 and 216 may switch the bottom coil 102b of the inductor 100 between a charging state (where the coil 102b is coupled between the battery 206 and ground) and a transfer state (where the coil 102b is coupled between the power source 206 and the load 202). The boost voltage ratio V out / V in is related to the pulse width modulation duty cycle D of the control signals P1 and P2 by a formula very similar to that of a standard boost converter:

[0013]

[0014] Assume no inductor losses or switching losses.

[0015] Figure 2 The single-stage boost converter 200 depicted in FIG. 2 may not be able to prevent the multi-winding inductor from saturating at practical boost voltages and output powers. For example, Figure 3A The circuit simulation of the currents I1 and I2 of the single-stage boost converter 200 in one pulse width modulation cycle is depicted, wherein the output voltage V out is 12V, the output power is 10W and the input voltage V in is 4V, which can represent the standard nominal operating conditions for boost converters in audio applications. Figure 3A The plotted simulation results also model the resistive losses in switches 210 , 212 , 214 , and 216 and inductor 100 . Figure 3B The current difference I1–I2 and the saturation level of the inductor 100 are depicted Figure 3C The current I2 and I1 are plotted on the graph and also plotted on Figure 3C The saturation boundary I sat , showing that despite Figure 3B The current difference I1–I2 is kept at the saturation level Below, but in Figure 3C Their respective amplitudes exceed the saturation boundary I sat Therefore, the single-stage boost converter 200 may not be useful for the desired application.

[0016] Figure 4An example of a two-stage boost converter 400 that can be used with a multi-wound inductor 100 and having a load 202 is depicted. Each stage 401a, 401b of the two-stage boost converter 400 can be connected to Figure 2 4 and the stages 401a, 401b may be coupled in series. One disadvantage of the two-stage boost converter 400 is that it requires two capacitors 204 and 205 to stabilize the output of each stage 401, compared to the single capacitor 204 required for the single-stage converter 200. The capacitors 204 and 205 can be large and significantly impact the overall circuit area.

[0017] In the architecture of the two-stage boost converter 400, the boost output of the first stage 401a supplies the input voltage to the second stage 401b. Therefore, the total boost ratio of the two stages 401 is the product of the boost ratios of each stage 401a, 401b. Because the two stages 401a, 401b can be operated at the same duty cycle, the total boost ratio of the two-stage boost converter 400 can be given as:

[0018]

[0019] Assume no inductor losses or switching losses. Comparing equation (3) with equation (2), the two-stage boost converter 400 may require a lower duty cycle than the single-stage boost converter 200 to achieve the same boost voltage ratio. For example, to boost from 4V to 12V, the single-stage boost converter 200 may require a duty cycle of 0.67 compared to 0.42 for the two-stage boost converter 400. A lower duty cycle can reduce the amplitude of the current ripple, which should help prevent saturation.

[0020] Figure 5A The current I of the two-stage boost converter 400 in one pulse width modulation cycle is depicted. 1-STAGE1 ,I 2-STAGE1 ,I 1-STAGE2 and I 2-STAGE2 Circuit simulation. Figure 5A and Figure 3A The comparison shows that the coil current ripple amplitude can be significantly reduced. Figure 5B The current difference I of the inductor 100 is depicted. 1-STAGE1 –I 2-STAGE1 , current difference I 1-STAGE2 –I 2-STAGE2 and saturation level Figure 5C The graph depicts the current I 2-STAGE1 to I 1-STAGE1 , and I 2-STAGE2 to I 1-STAGE2 And also in Figure 5C The saturation boundary I sat . In Figure 5B and Figure 5C Respectively Figure 3A and Figure 3B When compared, it is shown that the oversaturation problem of the single-stage boost converter 200 is greatly improved. The current of the inductor 100 of the first stage 401a will now be within the saturation limit. However, the current of the inductor 100 of the second stage 401b may still exceed the saturation limit. The problem with the two-stage boost converter 400 is that even if the duty cycle is small, the second stage 401b is still derived from the output of the first stage 401a, and the output of the first stage 401a is at a voltage V higher than the battery 206. in Therefore, when the control signal P1 is asserted, the inductor 100 of the second stage 401b may experience a larger voltage drop, and this may cause Figure 5B The larger current difference I 1-STAGE2 –I 2-STAGE2 .

[0021] Therefore, for the desired application, neither the single-stage boost converter 200 nor the two-stage boost converter 400 can satisfy the saturation constraint of the inductor 100 .

[0022] By including this discussion in the background section, applicants assert that none of the contents of this background section are prior art that could be used to support a denial of the recited claims based on prior art. Summary of the invention

[0023] According to the teachings of the present disclosure, one or more disadvantages and problems associated with existing inductor-based power converters may be reduced or eliminated.

[0024] According to an embodiment of the present disclosure, a system may include a power converter including at least one stage having: a dual anti-wound inductor configured so that its windings generate opposite magnetic fields in its magnetic core; and a current control subsystem for controlling the current through the dual anti-wound inductor. The current control subsystem may be configured to minimize the magnitude of the magnetizing current of the dual anti-wound inductor to prevent the core of the dual anti-wound inductor from saturating and to adjust the amount of output current delivered by the power converter to a load according to a reference input signal.

[0025] According to these and other embodiments of the present disclosure, a method for controlling current through a double reverse-wound inductor integral with a power converter may be provided, wherein the double reverse-wound inductor is configured so that its windings generate opposite magnetic fields in its magnetic core. The method may include minimizing the magnitude of the magnetizing current of the double reverse-wound inductor to prevent the core of the double reverse-wound inductor from saturating, and adjusting the amount of output current delivered by the power converter to a load based on a reference input signal.

[0026] According to these and other embodiments of the present disclosure, a system for controlling current through a double counter-wound inductor integral with a power converter, wherein the double counter-wound inductor is constructed so that its windings generate opposite magnetic fields in its magnetic core, may include an input terminal for receiving a reference input signal and a current control subsystem, the current control subsystem being configured to minimize the amplitude of the magnetizing current of the double counter-wound inductor to prevent core saturation of the double counter-wound inductor, and to adjust the amount of output current delivered by the power converter to a load based on the reference input signal.

[0027] From the drawings, descriptions and claims included herein, the technical advantages of the present disclosure are obvious to those skilled in the art. The objects and advantages of the embodiments will be realized and achieved at least by the elements, features and combinations particularly pointed out in the claims.

[0028] It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are not restrictive of the claims set forth in the disclosure. BRIEF DESCRIPTION OF THE DRAWINGS

[0029] A more complete understanding of the present embodiments and their advantages may be obtained by referring to the following description in conjunction with the accompanying drawings, in which like reference numerals represent like features, and in which:

[0030] Figure 1A and Figure 1B depicts a multi-winding integrated inductor according to an embodiment of the present disclosure;

[0031] Figure 1C The embodiment according to the present disclosure is shown Figure 1A and Figure 1B Saturation curve of the current in the multi-winding integrated inductor shown in FIG.

[0032] Figure 2 A single-stage boost converter using a multi-wound integrated inductor according to an embodiment of the present disclosure is shown;

[0033] Figure 3A Describes an embodiment of the present disclosure Figure 2Circuit simulation of current flow in a multi-winding integrated inductor of a single-stage boost converter shown in FIG. 1 over a pulse width modulation cycle;

[0034] Figure 3B Describes an embodiment of the present disclosure Figure 2 Circuit simulation of current difference and current saturation level of a multi-wound integrated inductor of a single-stage boost converter shown in;

[0035] Figure 3C The embodiment according to the present disclosure is shown Figure 2 Saturation curve of the current in the multi-winding integrated inductor of the single-stage boost converter shown in FIG.

[0036] Figure 4 A two-stage boost converter in which each stage uses a multi-wound integrated inductor according to an embodiment of the present disclosure is shown;

[0037] Figure 5A Describes an embodiment of the present disclosure Figure 4 Circuit simulation of the current in a multi-winding integrated inductor of a two-stage boost converter shown in FIG. 1 over a pulse width modulation cycle;

[0038] Figure 5B Describes an embodiment of the present disclosure Figure 4 Circuit simulation of current difference and current saturation level of a multi-wound integrated inductor of a two-stage boost converter shown in;

[0039] Figure 5C The embodiment according to the present disclosure is shown Figure 4 Saturation curve of the current in the multi-winding integrated inductor of the two-stage boost converter shown in FIG.

[0040] Figure 6 illustrates selected components of an example personal mobile device according to an embodiment of the present disclosure;

[0041] Figure 7 A block diagram showing selected components of an example integrated circuit for a personal mobile device driving a transducer according to an embodiment of the present disclosure;

[0042] Figure 8 Block and circuit diagrams showing selected components of an example switch-mode amplifier according to an embodiment of the present disclosure;

[0043] Fig. 9 shows selected components of an enhanced two-stage boost converter in which each stage uses a multi-wound integrated inductor according to an embodiment of the present disclosure;

[0044] Fig. 10A and Fig. 10BDepicted is a diagram showing a value of a switch control signal for an enhanced two-stage boost converter according to an embodiment of the present disclosure. Fig. 9 An equivalent circuit diagram of the connection of selected components of an enhanced two-stage boost converter;

[0045] FIG. 11A to FIG. 11C Describes an embodiment of the present disclosure Fig. 9 Circuit simulation of current in a multi-winding integrated inductor of an enhanced two-stage boost converter shown in FIG. 1 over a pulse width modulation cycle;

[0046] Fig. 12A depicts an example model for modeling the effects of disturbances in the generation of a pulse width modulation control signal in accordance with an embodiment of the present disclosure;

[0047] Fig. 12B depicts an ideal pulse width modulation control signal and an ideal pulse width modulation control signal affected by a disturbance according to an embodiment of the present disclosure;

[0048] Fig.13A and Fig. 13B Describes the use of embodiments according to the present disclosure Fig. 12A Simulation of an example step disturbance in the generation of a pulse width modulation control signal of a disturbance model;

[0049] Fig.14A Describes an embodiment of the present disclosure. Fig. 9 The enhanced two-stage boost converter shown in Fig.13A and Fig. 13B The magnetizing current of the inductor and the magnetizing current saturation limit caused by the response of the step disturbance depicted in;

[0050] Fig. 14B FIG. 2 shows a period of operation prior to a disturbance in the generation of a pulse width modulation control signal according to an embodiment of the present disclosure. Fig. 9 Saturation curve of the current in the multi-winding integrated inductor of the enhanced two-stage boost converter shown in FIG.

[0051] Fig. 14C FIG. 2 shows a period of operation following a disturbance in the generation of a pulse width modulation control signal according to an embodiment of the present disclosure. Fig. 9 Saturation curve of the current in the multi-winding integrated inductor of the enhanced two-stage boost converter shown in FIG.

[0052] Fig.15 An example voltage control loop that may be used in conjunction with a boost converter according to an embodiment of the present disclosure is shown;

[0053] Fig.16Ashows selected components of an example current control scheme that may be used in conjunction with a boost converter according to an embodiment of the present disclosure;

[0054] Fig. 16B It shows that the embodiment according to the present disclosure can be Fig.16A Selected components of an example control subsystem for use in conjunction with a boost converter and current control scheme are depicted;

[0055] Fig.17A Showing selected components (including a cycle average calculator) for performing calculation of a cycle average of an inductor coil current according to an embodiment of the present disclosure;

[0056] Fig. 17B Shown are selected components of a system for generating a pulse width modulated control signal using a triangular carrier signal of a triangular modulator and using such a triangular carrier signal to trigger a midpoint sampler configured to sample a current value at a midpoint of the "ON" time of the pulse width modulated control signal in accordance with an embodiment of the present disclosure;

[0057] Fig. 17C shows selected portions of a boost converter and selected components of a midpoint sampler according to an embodiment of the present disclosure;

[0058] Fig.18A and Fig.18B depicts a simulation of an example step disturbance in the generation of a pulse width modulation control signal according to an embodiment of the present disclosure;

[0059] Fig.18C Describes an embodiment of the present disclosure in response to Fig.18A Example waveform of the cycle average current of the inductor current with a step disturbance;

[0060] FIG. 19A to FIG. 19C Various waveforms and plots of a three-cycle simulation of coordination of a midpoint sampler and a triangle modulator according to an embodiment of the present disclosure Fig. 17C The sampling process;

[0061] Fig. 20A and Fig. 20B depicts waveforms showing the effects of offsets caused by resistive losses and imbalances between boost converter stages according to an embodiment of the present disclosure;

[0062] Fig.21A and Fig. 21B The selected components for coordinate transformation according to an embodiment of the present disclosure are shown, including a forward transformation block and an inverse transformation block;

[0063] Fig. 21Cshows a saturation curve of current within a multi-wound integrated inductor depicting transformed coordinate axes according to an embodiment of the present disclosure;

[0064] FIG. 22A to FIG. 22D depicts a simulation of an example step disturbance in the generation of a pulse width modulation control signal and application of a forward transformation to an inductor coil current according to an embodiment of the present disclosure;

[0065] FIG. 23A to FIG. 23C depicts selected components of an observer for use in a control subsystem of a boost converter according to an embodiment of the present disclosure;

[0066] FIG. 24A to FIG. 24D The pulse width modulation control signal disturbance and the simulated actual value of the coil current according to the embodiment of the present disclosure are depicted as well as the Fig. 23B The estimated version of these parameters for the Kalman filter estimate shown in;

[0067] Fig.25A shows selected components of a control subsystem of a boost converter implementing two independent control loops for magnetization and battery mode of the control subsystem according to an embodiment of the present disclosure; and

[0068] Fig.25B Described is a method for Fig.25A An example implementation of a state-space model of a control block of a control subsystem is shown in FIG. DETAILED DESCRIPTION

[0069] Figure 6 An example personal mobile device 1 according to an embodiment of the present disclosure is shown. Figure 6 The personal mobile device 1 is depicted with a speaker 7. The speaker 7 is merely an example, and it should be understood that the personal mobile device 1 may be used in conjunction with a variety of transducers, including magnetic coil speakers, piezoelectric speakers, tactile feedback transducers, etc. Additionally or alternatively, the personal mobile device 1 may be coupled to headphones 3 in the form of a pair of earbud speakers 8A and 8B. Figure 6 The earphones 3 depicted in the figure are merely examples, and it should be understood that the personal mobile device 1 can be used in conjunction with a variety of audio transducers, including but not limited to headphones, earbuds, in-ear headphones, and external speakers. Plug 4 can provide a connection of the earphones 3 to the electrical terminals of the personal mobile device 1. The personal mobile device 1 can provide a display to the user and receive user input using a touch screen 2, or alternatively, a standard liquid crystal display (LCD) can be combined with various buttons, sliders, and / or dials provided on the face and / or sides of the personal mobile device 1. Similarly, Figure 6As shown, the personal mobile device 1 may include an integrated circuit (IC) 9 for generating an analog signal for transmission to the speaker 7, the earphone 3 and / or another transducer.

[0070] Figure 7 FIG. 1 is a block diagram showing selected components of an example IC 9 of a personal mobile device for driving a transducer according to an embodiment of the present disclosure. Figure 7 As shown, the microcontroller core 18 can supply a digital audio input signal DIG_IN to a digital-to-analog converter (DAC) 14, which can convert the digital input signal into an analog input signal V IN DAC 14 can supply an analog signal V to amplifier 16 I , the amplifier can amplify or attenuate the analog input signal V I To provide a differential audio output signal V O , the audio output signal can operate a speaker, a headphone transducer, a piezoelectric transducer, a tactile feedback transducer, a line level signal output, and / or other suitable output. In some embodiments, DAC 14 can be an integral component of amplifier 16. Power supply 10 can provide a power rail input for amplifier 16. In some embodiments, power supply 10 can include a switch mode power converter, as described in more detail below. Although Figure 6 and Figure 7 It is envisioned that the IC 9 resides in a personal mobile device, but the systems and methods described herein may also be applied to electrical and electronic systems and devices other than personal mobile devices, including transducer systems for use in computing devices, automobiles, buildings, or other structures larger than personal mobile devices.

[0071] Figure 8 FIG. 1 is a block diagram and circuit diagram showing selected components of an example switch mode amplifier 20 according to an embodiment of the present disclosure. In some embodiments, the switch mode amplifier 20 may implement a Figure 7 All or part of the amplifier 16 described. Figure 8 As shown, switch mode amplifier 20 may include a loop filter 22 , a controller 24 , and a power converter 26 .

[0072] The loop filter 22 may include a circuit configured to receive an input signal (eg, an audio input signal V IN or its derivative) and a feedback signal (e.g., an audio output signal V O , its derivative or indicative audio output signal V OThe controller 24 may be any system, device or apparatus that generates a controller input signal to be transmitted to the controller 24 based on such input signal and feedback signal. In some embodiments, such controller input signal may include a signal indicating an integral error between the input signal and the feedback signal. In other embodiments, such controller input signal may include a signal indicating a target current signal or a target voltage signal, the target current signal being used as the output current I OUT The target voltage signal is driven to a load coupled to the output terminal of the second control loop 28 as the output voltage V O is driven to the load.

[0073] The controller 24 may include any system, device, or apparatus configured to generate a signal based on an input signal (eg, an input signal INPUT), an output signal V O and / or other characteristics of the switch-mode amplifier 20, controlling the switching of a switch integrated with the power converter 26 so as to transfer electrical energy from the power supply V to the power supply V according to the input signal. SUPPLY is transferred to the load of the switch mode amplifier 20.

[0074] Power converter 26 may include a power converter configured to receive a voltage V at its input. SUPPLY (for example, provided by power supply 10) and generates its output voltage V at its output O In some embodiments, the voltage V SUPPLY The power may be received through input terminals of the power converter 26, which may include a positive input terminal and a negative input terminal, the negative input terminal being coupled to a ground voltage. As described in more detail in the present disclosure (including, but not limited to, reference below), Fig. 9 14), the power converter 26 may include a power inductor and a plurality of switches, which are controlled by a control signal received from the controller 24 to convert the voltage V SUPPLY Converted into voltage V O , so that the audio output signal V O is a function of the input signal to the loop filter 22 .

[0075] Fig. 9 Depicted are selected components of an example enhanced two-stage boost converter 900 that can be used with a multi-wound inductor 100 and having a load 202 according to an embodiment of the present disclosure. In some embodiments, the enhanced two-stage boost converter 900 can be used to implement Figure 7 In these and other embodiments, the enhanced two-stage boost converter 900 can be used to implement the Figure 8 All or a portion of the power converter 26 depicted in FIG. Fig. 9The enhanced two-stage boost converter 900 shown in FIG. 1 is similar in many respects to the Figure 4 4. The two-stage boost converter 400 depicted in FIG. 4 is a two-stage boost converter 400, and therefore, only the differences between the enhanced two-stage boost converter 900 and the two-stage boost converter 400 are discussed below. In particular, while the first stage 901a of the enhanced two-stage boost converter 900 may be similar to the first stage 401a of the two-stage boost converter 400, the enhanced second stage 901b of the enhanced two-stage boost converter 900 may include a second stage 401b of the two-stage boost converter 400. Fig. 9 Arranged as shown and as Fig. 9 As shown by the control signals P1, P2, and 901a and 901b and capacitor 905 (instead of capacitor 405). As in the two-stage boost converter 400, the inductor 100a, 100b of each of the stages 901a and 901b is a double counter-wound inductor comprising a plurality of coils including coils 102a and 102b and wound in such a manner that when the current through coils 102a and 102b is positive, the magnetic fields generated in the magnetic core 104 by coils 102a and 102b cancel.

[0076] Fig. 10A and Fig. 10B An equivalent circuit diagram showing the connections of selected components of the enhanced two-stage boost converter 900 based on the values ​​of the switch control signals for the enhanced two-stage boost converter 900 according to an embodiment of the present disclosure is depicted. In particular, Fig. 10A Depicts the situation when the control signal P1 is enabled (and the control signal The connection of the top coil 102a of each of the inductors 100a and 100b when the inductor 100a and 100b are invalid, and Fig. 10B Depicts the situation when the control signal P1 is deactivated (and the control signal For clarity of illustration, Fig. 10A and Fig. 10B Any resistive switching losses are neglected.

[0077] like Fig. 10A As can be seen, when the control signal P1 is enabled (and the control signal When the inductor 100a and the top coil 102a of the inductor 100b are connected in parallel with the power supply of the battery 206 and the ground. Fig. 10A The configuration shown in FIG. 1 is the charging phase of the enhanced two-stage boost converter 900, in which energy is stored in the top coil 102a. Fig. 10BAs can be seen, when the control signal P1 is invalid (and the control signal When the inductor 100a and the top coil 102a of the inductor 100b are connected in series with the power supply of the battery 206 and the ground. Fig. 10B The configuration shown in FIG. 9 is the transfer phase of the enhanced two-stage boost converter 900 , in which energy is transferred from the top coil 102 a to the capacitor 204 and the load 202 .

[0078] Therefore, the unique behavior of charging the coils 102a from both stages in parallel and transferring the stored energy from the coils 102a in series can be an advantage of this architecture.The bottom coil 102b of the inductors 100a and 100b can be controlled in a similar manner.

[0079] Because the first stage 901a and the enhanced second stage 901b are charged in parallel and transferred in series, the overall step-up ratio is the sum of the contributions of each stage, as given by:

[0080]

[0081] Assume no resistive losses. Equation 4 shows that the boost action of each stage 901a, 901b is additively combined, in contrast to the two-stage boost converter 400 where the boost action of each stage 401a, 401b is multiplicatively combined. As a result, the enhanced two-stage boost converter 900 may require a smaller duty cycle than the single-stage boost converter 200 in order to achieve the same boost ratio that can minimize current ripple (although to a lesser extent than the two-stage converter 400).

[0082] When the control signal P1 of the enhanced two-stage boost converter 900 changes from valid to invalid (and the control signal When the current I in the coil 102a changes from inactive to active, 1-STAGE1 and I 1-STAGE2102a and 102b. The inductors 100a and 100b may not be completely equal. Such unequal currents may occur because when the control signal P1 is enabled, the conduction path resistances of the coils 102a for the inductors 100a and 100b may be different (e.g., when the control signal P1 is enabled, the inductor 100b may have an additional switch in its conduction path, which may add switch resistance). When two inductors with different currents are connected in series, the current in one (or both) of the inductors must change quickly to meet continuity. However, the rapid change of the current in the inductor may generate a larger potential destructive voltage in the circuit. To solve this problem, the switch 912 can couple the capacitor 905 between the common electrical node of the coil 102a (when the control signal P1 is disabled) and ground, thereby providing an alternative path for any such overcurrent. For the bottom coil 102b of the inductors 100a and 100b, the switch 916 can be used for similar purposes used by the switch 912.

[0083] Capacitor 905 can be much smaller, with minimal impact on overall circuit area, than capacitor 405 of two-stage boost converter 400. In fact, in some cases, such a capacitor can have a capacitance small enough that capacitor 905 can be formed within the integrated circuit of enhanced two-stage power converter 900. A natural consequence of the architecture of enhanced two-stage power converter 900 is that capacitor 905 can balance the current between first stage 901a and enhanced second stage 901b.

[0084] FIG. 11A to FIG. 11C Describes an embodiment of the present disclosure Fig. 9 Circuit simulation of the current in a multi-winding integrated inductor of a two-stage boost converter 900 over one pulse width modulation cycle is shown in FIG. Fig.11A Example control signals P1 and P2 are depicted, and Fig. 11B The currents of the coils 102a and 102b of the inductors 100a and 100b are depicted. During the state where the control signal P1 is enabled and the control signal P2 is disabled or the control signal P1 is disabled and the control signal P2 is enabled, at least one pair of coils 102 is coupled in series to the load 202, such as Fig. 10B As shown. In these conditions, energy can be transferred from the magnetic fields of inductors 100A and 100B, while energy can be simultaneously stored in the magnetic fields. In fact, in this state, energy can be transferred from one coil 102 of inductor 100 to the other coil 102 of inductor 100. This transfer action can prevent excessive energy from accumulating in magnetic core 104, thereby potentially preventing premature saturation.

[0085] The enhanced two-stage power converter 900 can prevent current saturation because it can minimize the total magnetic field in the magnetic core 104, thereby minimizing the amount of magnetic energy stored in the magnetic core 104. The total magnetic field in the magnetic core 104 can be proportional to the magnetizing current I mag Proportional to , the magnetizing current (for each inductor 100) can be defined as:

[0086]

[0087] When the magnetizing current I mag Greater than or equal to the magnetizing current saturation limit , the magnetic core 104 may be saturated.

[0088] Fig. 11C The magnetizing current I of the inductors 100a and 100b are respectively plotted. mag1 and I mag2 When both control signals P1 and P2 are asserted, the current in coil 102 increases because both are coupled in parallel between power and ground, as previously described in Fig. 10A However, during this state, the magnetizing current I mag1 and I mag2 Remains relatively flat because the magnetic flux generated by each coil 102 changes at an equal rate and therefore the difference remains constant. In effect, this flattening of the current can produce a "flat top" to the magnetizing current waveform, such as Fig. 11C As shown, this prevents the magnetizing current I mag1 and I mag2 saturation.

[0089] Fig. 9 The advantage of the two-stage enhanced boost architecture shown in Figure 2 The single-level architecture depicted in Figure 4 Compared to the two-stage architecture depicted in FIG, it can reduce the peak current and can minimize the energy stored in the core of the inductor 100, thereby minimizing the possibility of core saturation. Due to these advantages, the current control method described below considers using Fig. 9 However, the concepts, methods and systems discussed below can be extended to Figure 2 The single-level architecture depicted in Figure 4 The two-stage architecture depicted in or any other boost architecture, including but not limited to those boost architectures disclosed in U.S. Patent Serial No. 16 / 692,072 filed on November 22, 2019 (the entire contents of which are incorporated herein by reference).

[0090] As discussed above, the output of the enhanced two-stage boost converter 900 can be controlled by the duty cycle of the control signals P1 and P2. These duty cycles can be time-varying signals that must be carefully selected to prevent core saturation due to disturbances and to regulate the amount of output current.

[0091] Discussed above FIG. 11A to FIG. 11C Idealized pulse width modulation and current waveforms during steady-state operation are depicted. However, in reality, the enhanced two-stage boost converter 900 may experience some deviations from these idealized waveforms. If the pulse width modulation control signals P1 and P2 are generated by a digital process, clock jitter, quantization effects, and mismatches in the digital components may cause the actual pulse width modulation control signals P1 and P2 to be different from what they command. In addition, non-idealities in the analog circuitry of the enhanced two-stage boost converter 900, such as switching times of transistor switches, gate drive effects (e.g., switch non-overlap times), and thermal effects, may cause additional deviations in the response of the enhanced two-stage boost converter 900.

[0092] For example, according to an embodiment of the present disclosure, Fig. 12A depicts an example model for modeling the effects of disturbances in the generation of pulse width modulated control signals P1 and P2, and Fig. 12B An ideal pulse width modulated control signal (eg, control signal P1) and an ideal pulse width modulated control signal affected by a disturbance are depicted. Fig. 12A As shown, two separate pulse width modulation (PWM) generators 1202 can use two independent duty cycles D1 and D2 to generate control signals P1, P2, and P3 for the enhanced two-stage boost converter 900. as well as Each PWM generator 1202 may have an independent disturbance source that causes a perturbation in the PWM waveform it generates, such as Fig. 12B This disturbance may shift one or more edges of the PWM waveform, effectively changing its duty cycle.

[0093] Such perturbations can be problematic because they can cause the inductor to saturate. Fig.13A and Fig. 13B Describes the use Fig. 12A Example of a step disturbance simulation of the disturbance model. Fig.13A and Fig. 13B In the example of FIG. 1 , a 1% step disturbance is introduced into the duty cycle D1 at time t = 0.2 μs, while the duty cycle D2 remains undisturbed, as shown in FIG. Fig.13A shown. Fig. 13B A simulation of the current through the coils 102a and 102b of the inductor 100 in response to a step disturbance is shown.

[0094] Fig.14A FIG. 1 depicts the magnetizing current I of the inductors 100 a and 100 b according to an embodiment of the present disclosure. mag1 and I mag2 and saturation level The magnetizing current I mag1 and I mag2 It is composed of an enhanced two-stage boost converter 900 Fig.13A and Fig. 13B The response to the step disturbance depicted in . Fig. 14B saturation curve of the current in the multi-winding integrated inductor of the enhanced two-stage boost converter 900 during an operation cycle before a disturbance in the generation of the pulse width modulation control signal according to an embodiment of the present disclosure. Similarly, Fig. 14C A saturation curve of the current within the multi-winding integrated inductor of the enhanced two-stage boost converter 900 during an operation cycle following a disturbance in the generation of a pulse width modulation control signal according to an embodiment of the present disclosure is shown.

[0095] In applying Fig.13A Before the disturbance, Fig. 14B As shown, the current of the inductor 100 can be maintained at the saturation boundary I sat However, if Fig. 14C As shown, after a number of cycles after the disturbance, one or more currents of the inductor 100 may exceed the saturation boundary I sat .

[0096] Therefore, as shown above, the control signals P1, P2, and A 1% disturbance in the control signals P1, P2 may be sufficient to cause a failure of the enhanced two-stage boost converter 900 due to inductor core saturation. A larger disturbance in either or both of the control signals P1, P2 may cause the enhanced two-stage boost converter 900 to exceed its limits. and The disturbance generation of the integrated inductor 100 is very sensitive. This sensitivity may be mainly due to the small inductance value of the integrated inductor 100. In order to ensure normal operation, the control signals P1, P2, and To regulate the current and prevent disturbance-induced inductor saturation.

[0097] In addition to the disturbance suppression problem discussed above, the control signals P1, P2, and To deliver power to the load 202 of the enhanced two-stage boost converter 900 and maintain the regulated output boost voltage V outIn a boost converter for audio applications, the output boost voltage V out It can be tripled or greater to meet the requirements of class H or class G / H amplifiers. The load on the output of the enhanced two-stage boost converter 900 can also vary over a wide range. The audio content may have a high crest factor and may change quickly from mute (0 watts) to full scale (~10 watts) in tens of microseconds. Additionally, in the case where the load 202 is a speaker, the impedance of the load 202 can be doubled or greater over operating frequency and temperature. Therefore, it can be expected that the enhanced two-stage boost converter 900 can respond to rapidly changing output voltage commands and load conditions.

[0098] Voltage control loops are often used to meet time-varying output voltage and load requirements. Fig.15 Selected components of an example voltage control loop 1500 are shown that may be used in conjunction with a boost converter 1506 (eg, which may be implemented using the enhanced two-stage boost converter 900 or any other suitable boost converter) in accordance with embodiments of the present disclosure.

[0099] like Fig.15 As shown, the compensator 1502 can receive a reference voltage V ref , the reference voltage can represent the desired adjusted output boost voltage V out , and may be time-varying. The compensator 1502 may compare the reference voltage V ref and the sensed output boost voltage V out , and generates a control signal CTRL based on the comparison and control algorithm. In many architectures, the control signal CTRL can drive an internal current control loop implemented by the current controller 1504. This current control loop can adjust the amount of average current delivered to the capacitor 204 and the load 202, thereby adjusting the output boost voltage V out However, to operate properly, the current controller 1504 may need to simultaneously prevent the inductor from saturating.

[0100] Thus, it may be desirable that a boost converter (e.g., enhanced two-stage boost converter 900 or other boost converter 1506) using a multi-winding inductor 100 have a current controller 1504 that meets two requirements: (a) the current controller 1504 adjusts the current in the inductor 100 to prevent disturbance-induced saturation; and (b) the current controller 1504 adjusts the output current driven to the load of the boost converter 1506 to meet the requirements of an audio system having a time-varying boost voltage and load.

[0101] Fig.16A 1 shows selected components of an example current control scheme that may be used in conjunction with boost converter 1506 according to an embodiment of the present disclosure. Fig.16A As shown, the current controller 1504 may include a measurement block 1602 configured to receive a sensed inductor current I (e.g., current I1 or I2 of the inductor 100) and perform calculations on the sensed inductor current I to generate a signal indicative of the inductor coil current I1 or I2. The current controller 1504 may have two external signals: (a) a saturation control signal SATCTRL, which may adjust current saturation; and (b) a current control signal ICTRL, which controls a target output current delivered to the load 202. The control block 1604 may receive the output of the measurement block 1602, the saturation control signal SATCTRL, and the current control signal ICTRL to generate a control signal for the switch of the boost converter 1506.

[0102] Fig. 16B It shows that the embodiment according to the present disclosure can be Fig.16A Selected components of an example control subsystem 1600 are depicted for use in conjunction with a boost converter 1506 and a current control scheme. Fig. 16B As shown, the compensator 1502 can be used to ref and the sensed output boost voltage V out The current control signal ICTRL is generated by comparing the current control signal V to the boost converter 1506, which allows the compensator 1502 to adjust the amount of current delivered from the boost converter 1506 and thus adjust the output boost voltage V out .like Fig. 16B As shown, the saturation control signal SATCTRL can be permanently set to zero to minimize the energy stored in the core of the inductor 100 and prevent saturation due to disturbances. The control block 1604 can receive the output of the measurement block 1602 (indicating the measured current inductor coil current I1 or I2), the saturation control signal SATCTRL and the current control signal ICTRL to generate control signals P1, P2, and P3 for the switches of the boost converter 1506. and

[0103] The following discussion outlines several example embodiments for implementing control subsystem 1600, which are grouped into four sections related to the control or measurement of circuit parameters.

[0104] 1. Generate PWM control signal and calculate cycle average value

[0105] The measurement process of the measurement block 1602 may include calculating the cycle averages of the inductor coil currents I1 and I2 of the coils 102 a and 102 b of the inductor 100 in a manner coordinated or associated with the PWM generation process. Fig.17ASelected components for performing calculations of cycle averages of inductor coil currents I1 and I2 are shown, including a cycle average calculator 1704 , according to an embodiment of the present disclosure.

[0106] To further illustrate this functionality, Fig.18A and Fig.18B Depicted is a simulation of an example 1% step disturbance introduced in duty cycle D1 at time t=0.2 μs, while duty cycle D2 is Fig.18A The shown remains undisturbed. Fig.18B A simulation of the current through the coils 102a and 102b of the dual-wound inductor 100 in response to a step disturbance is shown. Fig.18C A graph of average winding currents I1 and I2 of coils 102a and 102b is depicted. Calculating the average value of currents I1 and I2 can eliminate the effect of current ripple within a PWM cycle and can be a clearer indicator of the current dynamic behavior. Therefore, the use of average currents can simplify the current control performed by current controller 1504.

[0107] It is worth noting that Fig.18B and Fig.18C Only the currents I1 and I2 of the coils 102a and 102b of the first-stage inductor 100a of the enhanced two-stage power converter 900 are depicted, and the current of the second-stage inductor 100b is omitted. However, due to the enhanced two-stage architecture, the currents I1 and I2 of the inductor 100a can be similar to the current of 100b, and the current controller 1504 can take advantage of this similarity by measuring the current of only one of the inductors 100 and controlling all currents based on this measurement.

[0108] The cycle average calculator 1704 can calculate the average cycle value of the currents I1 and I2 in any suitable manner, including directly implementing a circuit that automatically integrates the currents I1 and I2 within the cycle by sampling and summing multiple points, or using the triangular carrier signal of the triangular modulator 1702 to generate the pulse width modulation control signals P1, P2, as well as As is known in the art, and using such a triangular carrier signal to trigger a midpoint sampler 1706, the midpoint sampler is configured to sample the values ​​of currents I1 and I2 at the midpoint of the "ON" time of the pulse width modulated control signals P1, P2, as shown in FIG. Fig. 17B As described in more detail below, midpoint sampler 1706 can provide an approximation of the cycle average of currents I1 and I2.

[0109] The midpoint sampler 1706 can sample the two currents I1 and I2 simultaneously, or perform alternating sampling of the currents I1 and I2, such as Fig. 17C shown. Fig. 17C FIG. 9 shows a portion of an enhanced two-stage boost converter 900, wherein switches 210 and 214 are controlled by control signals P1 and P2, respectively. Fig. 17C In the illustrated embodiment, the midpoint sampler 1706 can be implemented by a sense resistor 1708 and an analog-to-digital converter (ADC) 1710. The sense resistor 1708 can be coupled between a ground voltage and a common node of switches 210 and 214. The ADC 1710 can sample the voltage across the sense resistor 1708, which can indicate the current flowing to ground. The analog-to-digital converter 1710 can be triggered by a trigger signal from the triangle modulator 1702 to sample the midpoint current value. Using a single ADC 1710 and sense resistor 1708 can be advantageous because they form a smaller circuit, and if two ADCs and sense resistors are to be used, the smaller circuit can avoid possible mismatches.

[0110] FIG. 19A to FIG. 19C Various waveforms of a three-cycle simulation of the coordination of the midpoint sampler 1706 and the triangle modulator 1702 according to an embodiment of the present disclosure are depicted and Fig. 17C The sampling process. Fig.19A and Fig.19B The generation of pulse width modulated control signals P1 and P2 from a triangular carrier is depicted. Fig.19A A triangular carrier wave CARRIER is described, having a minimum value of -1 and a maximum value of +1 and a period equal to the pulse width modulation period of the pulse width modulation control signals P1 and P2. The reference signals ref1 and ref2 can be related to the desired duty cycles D1 and D2 as follows:

[0111] ref1=2D1-1 (5)

[0112] ref2=1-2D2 (6)

[0113] Fig.19B Pulse width modulated control signals P1 and P2 are depicted, which can be derived from the reference signals ref1 and ref2 and the triangular carrier CARRIER as follows:

[0114]

[0115] Control Signal and They can be the logical complements of the control signals P1 and P2 respectively.

[0116] When the triangular carrier CARRIER is equal to -1 and +1 respectively, the following can be achieved: Fig. 17C The alternating sampling time of current I1 and I2 is FIG. 19A to FIG. 19C, respectively, are depicted by points labeled I1samp and I2samp. By sampling at the maximum and minimum values ​​of the triangular carrier CARRIER, sampling is configured to be performed at the midpoint of the "ON" time of the control signals P1 and P2. Such midpoint sampling can be advantageous because it is maximally away in time from the edges of the control signals P1 and P2, where switching transients may distort the measurement. In addition, midpoint sampling can ensure that the current I1 is sensed when the control signal P1 is enabled and the control signal P2 is disabled, and the current I2 is sensed when the control signal P2 is enabled and the control signal P1 is disabled, regardless of the desired duty cycles D1 and D2.

[0117] Sampling at the midpoint of the PWM waveform has the added benefit that it approximates the cycle average of the inductor currents I1 and I2. Fig.19C The simulated winding currents I1 and I2 (assuming lossless inductors and switches) and the sampled current values ​​are depicted. Due to the symmetry of the switch states and the accompanying antisymmetry of the currents about the midpoint of the control signal "ON" time, the sampled values ​​can represent the cycle averages of the inductor currents I1 and I2.

[0118] During actual operation, resistive losses and the imbalance between the two stages 901a and 901b of the enhanced two-stage boost converter 900 may distort the current waveforms so that they are no longer piecewise linear waveforms that are anti-symmetric about the midpoint of the "ON" time of the control signal. This distortion may cause an offset between the actual cycle averages of the inductor currents I1 and I2 and the values ​​from the midpoint sampling. Fig. 20A and Fig. 20B An example of such a shift from a simulation with resistive losses and imbalance between stages 901a and 901b is shown.

[0119] Despite these offset errors, the midpoint sampled values ​​of the inductor currents I1 and I2 can still be used in the current control loop described above. First, the errors are typically small relative to the values ​​being measured. Second, the outer voltage regulation loop can use an integrator to eliminate any errors in the commanded output current. Third, when the winding current difference approaches zero, the winding current difference estimated from the midpoint sample is also close to zero. Therefore, if the estimated coil current difference |I1-I2| is controlled to be zero, the actual winding current difference |I1-I2| can also be close to zero, which can prevent saturation.

[0120] 2. Apply conversion to current measurement and duty cycle

[0121] In some embodiments, the control subsystem of boost converter 1506 may implement a coordinate transformation to decouple the signals in order to simplify the control of boost converter 1506 . Fig.21A and Fig. 21B2 shows selected components for coordinate transformation according to an embodiment of the present disclosure, including a forward transformation block 2102 and an inverse transformation block 2104. Fig.21A and Fig. 21B As shown, the forward transformation block 2102 can be applied to the measurement of the inductor currents I1 and I2, and the control block 1604 can be used to transform the measured values ​​I1 and I2 of the inductor currents I1 and I2. m and I b Operate to generate a duty cycle control signal D m and D b These duty cycle control signals can be inversely transformed by the inverse transformation block 2104 to generate duty cycle control signals D1 and D2 for driving the PWM generator 1202.

[0122] The forward transformation block 2102 may apply the following transformation to generate the transformed current measurement value I m and I b :

[0123]

[0124] Among them I m can be called the magnetizing current, and I b The magnetizing current may be equal to the difference between the inductor currents I1 and I2, and may therefore be proportional to the net magnetic field in the core of the multi-wound inductor 100. The term "battery current" as used herein is not limited in any way to a current originating from a battery, but may originate from a battery or any suitable power source, or may be a mathematical equivalent / transformed value identifying a battery current or a power source current. For example, the battery current I b can be the sum of coil currents I1 and I2, and for a single-stage converter, the battery current I b can be accurately equal to the actual current drawn from the battery or power source. For a two-stage converter, the battery current I b may no longer be equal to the actual battery or supply current, but it can be called that because it is the same mathematical transformation. Since the coil currents I1 and I2 can be the period average of the winding currents, the magnetizing current I m and the battery current I b It can also represent the period average of the magnetizing current and the battery current.

[0125] The inverse transform of transform block 2102 may be applied by inverse transform block 2104 to generate a duty cycle control signal:

[0126]

[0127] Fig. 21BIn this case, the saturation control signal SATCTRL and the current control signal ICTRL received by the control block 1604 can be respectively determined by the magnetizing current reference and battery current reference Due to this transformation, the control block 1604 can operate in the magnetization / battery coordinate space rather than the coil 101a / coil 102b coordinate space.

[0128] Fig. 21C A saturation curve of the current within the multi-wound integrated inductor 100 depicting transformed coordinate axes according to an embodiment of the present disclosure is shown. Fig. 21C The graph depicts a curve similar to Figure 1C The saturation region of the saturation region 110, where the current I1 is on the horizontal axis and the current I2 is on the vertical axis. The transformation of equation (9) can be interpreted as a rotation transformation, which rotates the (I1, I2) axis by 45° to a new set of axes (I m , I b ),like Fig. 21C This rotational transformation is useful because for most of the unsaturated region 112, the magnetizing current I m The coordinates may be that the boost converter 1506 is away from exceeding the magnetizing saturation limit On the other hand, the orthogonal battery current I b The coordinates do not have this constraint and can represent the current flowing through the boost converter 1506 to its output. At large current values, the battery current I b There will be a maximum current constraint, but the approximation in the transformed coordinate space will be useful for control within most of the operating space.This example embodiment depicts how the transformation decouples saturation protection from the output current requirement, allowing each quantity to be controlled independently.

[0129] The transformation of equations (9) and (10) may also have the added benefit of decoupling the system mode of the converter. The enhanced two-stage boost converter 900 may be modeled using the example state space averaging technique described in Erickson, Robert W. and Dragan Maksimociv, "Fundamentals of Power Electronics: Second Edition", (Spring Science and Business Press, 2001, which is incorporated herein by reference). Such a model may also be linearized around the nominal operating point using a small signal approximation. Assuming that the current dynamics are much faster than the output voltage dynamics, the system including the boost converter 1506 and its control subsystem may be modeled as a second-order ordinary differential equation as follows:

[0130]

[0131] Wherein the lowercase variables i1, i2, d1 and d2 represent the small signal deviations of the currents I1 and I2 and the duty cycles D1 and D2 from the steady state, L is the self-inductance of the inductor 100, M is the mutual inductance of the inductor 100, is the resistance as a function of the switch resistance, inductor resistance, and steady-state duty cycle, and is used as the power supply voltage, output boost voltage V out , switch resistance, inductor resistance, and steady-state current. If the transformation of equations (9) and (10) is applied to equation (11), the result can be:

[0132]

[0133] Among them, the lowercase variable i m 、i b ,d m and d b Indicates current I m and I b And the duty cycle D m and D b is the small signal deviation from steady state, and k is the coupling coefficient defined by M / L. In this case, the matrix is ​​diagonalized so that the magnetizing mode and the battery mode are orthogonal. Therefore, the system of boost converter 1506 and its control subsystem can be decoupled into two independent first-order modes.

[0134] FIG. 22A to FIG. 22D Depicted are the results when the aforementioned transformation is applied to the previous example of a 1% step disturbance in duty cycle D1. Fig.22A depicts a duty cycle D1 with a disturbance occurring at 0.2 μsec, and Fig. 22B The simulated cycle averages of the coil currents I1 and I2 of the inductor 100a are depicted. Although the perturbation occurs only at duty cycle D1, both currents I1 and I2 are affected due to the coupled dynamics of equation (11). The transformation equation (9) can be applied to these currents I1 and I2 to produce Fig. 22C The magnetizing current I m and Fig.22D The battery current I b The currents I1 and I2 can exhibit a second-order response, while the magnetizing current I m and the battery current I b A simple first-order response can be shown. The magnetizing current I m and the battery current I b These first-order decoupled dynamic characteristics will be easier to control in the control subsystem.

[0135] 3. Use Observer to measure current

[0136] The cycle average measurements of currents I1 and I2 may be noisy. Additionally, if the implementation from Fig. 17C If the sampling method is used, the information from one of the coils 101 may be lost in each sampling period. These disadvantages may degrade the performance of the control subsystem. Therefore, the control subsystem may implement observer 2300, such as Fig.23A In operation, the observer 2300 may receive measured current data I meas and generate improved estimates of currents I1 and I2. Observer 2300 may use a model of the control subsystem to filter out noise and fill in any missing information. The system model may require knowledge or information about the desired duty cycles D1 and D2, as well as an indicator of which of the current values ​​I1 and I2 is currently being measured. The latter indicator signal is Fig.23A Marked as I meas The observer 2300 may be implemented using one of several approaches, including but not limited to: a Luenberger filter, a Kalman filter, and a sliding mode observer. Fig. 23B An implementation based on a Kalman filter is shown. If the measured current data I meas From Fig. 17C The current sampling scheme shown in Fig. 23B Kalman filter architecture. Fig. 23B The Kalman filter implementation of 2304 can be operated by using the model 2302 and the known model inputs of the desired duty cycles D1 and D2 and forming estimates of the currents I1 and I2. These estimates can be combined with the measured current data I by the subtractor 2304. meas The error signal e can be multiplied by the Kalman gain K and then used to adjust the model estimate so that the error is minimized. The signal can indicate the measured current data I meas Whether to measure current I1 or current I2.

[0137] The system model can be obtained by discretizing the continuous-time model equation (11) using any standard method (e.g., forward Euler, bilinear transformation, zero-order hold (ZOH), etc.) and rewriting it in standard form:

[0138]

[0139] Where A and B are 2x2 matrices, and I is the sampling time index. This model can be extended to include the effects of noise and disturbances on the generation of control signals P1 and P2:

[0140]

[0141] y i =H i x i +r i (15)

[0142] in is the state vector, dist1 and dist2 are disturbance estimates; is the model input vector; F is a 4×4 block matrix consisting of the 2×2 matrix A from equation (13), the 2×2 matrix B from equation (13), the 2×2 zero matrix, and the 2×2 identity matrix; G is a 4×2 block matrix consisting of the 2×2 matrix B and the 2×2 zero matrix, w is a 4×1 vector of process noise, y is the output I1 or I2; and r is the scalar measurement noise. The matrix H i can vary with time, depending on the signal I meas mode value: If the current I1 is being measured, then H i = [1 0 00], and if current I2 is being measured, then H i =[0 1 0 0]. Model 2302 responds to signal I meas The mode dependency is Fig. 23B Indicated by dotted line.

[0143] The Kalman filter implemented by model 2302 can be implemented in any suitable number of ways using equations (14) and (15) (e.g., according to Simon, Dan, "Optimal State Estimation," Wiley 2006, which is incorporated herein by reference in its entirety). In one possible implementation, the Kalman recursion can be given by:

[0144]

[0145] in is the estimated state vector containing the estimated coil currents and the disturbance, and z i is the internal state vector (eg, a priori state estimate).

[0146] Kalman gain K i Any of the standard techniques (e.g., those disclosed in the Simon reference cited above) may be used to derive the gain. However, the gain calculation must take into account the time-varying H based on the measured current data. i In one embodiment, the Kalman gain can be pre-calculated compared to several samples. In steady state, the Kalman gain will be calculated according to the signal I measmode alternates between two sets of values. These steady-state values ​​can be adjusted based on the signal I meas mode to store and apply the state, such as Fig. 23B As shown by the dotted line in .

[0147] FIG. 24A to FIG. 24D Depicted are simulation results of the enhanced two-stage power converter 900 in response to the previously described 1% step disturbance. Fig. 23B The Kalman filter can be used to estimate the disturbances on the currents I1 and I2 and the control signals P1 and P2. The simulation shown can be used Fig. 17C The current sampling technology disclosed in. Fig.24A and Fig. 24B The actual and estimated disturbances on the control signals P1 and P2 are plotted, while Fig.24C and Fig.24D Actual currents I1 and I2 and estimated currents I1 and I2 are plotted. The plotted currents may be deviations from a steady state. Fig. 23B The Kalman filter can quickly converge to the disturbance of the control signals P1 and P2 and the estimation of both the currents I1 and I2, even if the measured current data I meas The measured value of includes only one of the currents I1 and I2 on each sample.

[0148] Fig.23C An alternative embodiment of the observer 2300 is depicted. Fig.23C In an embodiment of the present invention, two independent observers, each with a differential model 2302a and 2302b and gains K1 and K2, can be used. When current I2 is being measured, the first observer 2300a can form an estimate of current I1. Its input can be the instantaneous measurement of current I1 and the previous measurement of current I2. Similarly, when current I1 is being measured, the second observer 2300b can form an estimate of current I2. Its input can be the instantaneous measurement of current I2 and the previous measurement of current I1. Fig. 23B Different from the example of FIG. 2 , the models 2302a, 2302b and the gains K1, K2 may be fixed. meas mode, an estimate of the currents I1 and I2 can be obtained from the output of the first model 2302a or the second model 2302b.

[0149] 4. Apply independent control to the decoupled current signals

[0150] like Fig.25AAs shown, in some embodiments, two independent control loops with control blocks 1604a and 1604b can be used to control currents I1 and I2, which control blocks separately control the magnetization mode and the battery mode. Control blocks 1604a and 1604b can be implemented using any standard control algorithm (including but not limited to proportional, proportional integral, proportional integral derivative or state space). Fig.25B An example of a state-space control 2500 that can be used to control one or both of the control blocks 1604a and 1604b is depicted. The state-space control 2500 can implement a servo control architecture (e.g., as disclosed in Ogata, Katsuhiko, Discrete-Time Control Systems, Prentice Hall, 1995) that includes an additional integrator 2502 to eliminate steady-state errors. Such control can be based on a decoupled state-space model of the system. The decoupled model can be obtained by applying the magnetization / battery conversion equations (9) and (10) to the discretized model of equation (13):

[0151]

[0152] where a m 、a b 、b m and b b is a scalar coefficient. Applying the magnetization / battery transformation can diagonalize the matrix and decouple the magnetization / battery modes, as discussed previously. Therefore, a separate state-space model can be written for each mode. In a practical implementation, there may be non-zero computational delay times for all blocks of the algorithm. However, in this case, it is assumed that upon receiving the measured current data I meas There is one sample delay between the start of the PWM operation and the calculation of the next PWM command.

[0153] This one sample delay can also be included in the model used for each mode as follows:

[0154]

[0155] where up is the internal state representing the input during the previous sample. The models given by equations (18) and (19) can each be used to construct a servo control using an approach similar to that of the Ogata reference cited above. The result will be Fig.25B The block 2504 labeled "b" may use b from equation (19) or (20) m or b b The gains are calculated using the coefficients, depending on whether the control is used in magnetizing mode or battery mode. The gains K1, K2, and K3 can be calculated using any standard state-space method (e.g., pole placement, quadratic optimal control, etc.).

[0156] As used herein, when two or more elements are referred to as being “coupled” to each other, the term indicates that the two or more elements are in electrical or mechanical communication, as applicable, whether indirectly or directly, with or without intervening elements.

[0157] The present disclosure encompasses all changes, substitutions, changes, alterations and modifications to the example embodiments herein that would be understood by a person of ordinary skill in the art. Similarly, where appropriate, the appended claims encompass all changes, substitutions, changes, alterations and modifications to the example embodiments herein that would be understood by a person of ordinary skill in the art. In addition, references to devices or systems or components of devices or systems adapted to, arranged to, capable of, configured to, enabled to, enabled to operate or operable to perform a specific function in the appended claims encompass the device, system or component, regardless of whether it or the specific function is activated, turned on or unlocked, as long as the device, system or component is so adapted, arranged, capable of, configured to, enabled to, enabled to operate or operable. Therefore, without departing from the scope of the present disclosure, the systems, devices and methods described herein may be modified, added or omitted. For example, the components of the systems and devices may be integrated or separated. Moreover, the operations of the systems and devices disclosed herein may be performed by more, fewer or other components, and the described methods may include more, fewer or other steps. Additionally, the steps may be performed in any suitable order. As used herein, "each" refers to every member of a set or every member of a subset of a set.

[0158] Although exemplary embodiments are shown in the drawings and described below, the principles of the present disclosure may be implemented using any number of techniques, whether currently known or not. The present disclosure should not be limited in any way to the exemplary implementations and techniques shown in the drawings and described above.

[0159] Unless specifically noted otherwise, items depicted in the drawings are not necessarily drawn to scale.

[0160] All examples and conditional language cited herein are for teaching purposes to help readers understand the disclosure and concepts contributed by the inventors to advance the art, and are interpreted as not being limited to these specific cited examples and conditions. Although the embodiments of the present disclosure have been described in detail, it should be understood that various changes, substitutions and modifications may be made thereto without departing from the spirit and scope of the present disclosure.

[0161] Although specific advantages have been listed above, various embodiments may include some, none, or all of the listed advantages. Additionally, other technical advantages will become apparent to those of ordinary skill in the art after reading the foregoing drawings and descriptions.

[0162] To assist the Patent Office and any reader of any patent issuing in this application in interpreting the appended claims, Applicant wishes to note that unless the words "means for" or "step for" are expressly used in a particular claim, they are not intended to trigger 35 U.S.C. §112(f) in any of the appended claims or claim elements.

Claims

1. A system for controlling current through a power converter having a double counter-wound inductor, comprising: A power converter comprising at least one stage having: a double counter-wound inductor configured such that its windings generate opposing magnetic fields in its magnetic core; as well as a current control subsystem for controlling a current through the dual counter-wound inductor, the current control subsystem being configured to: minimizing the magnitude of the magnetizing current of the double counter-wound inductor to prevent core saturation of the double counter-wound inductor; and An amount of output current delivered to a load by the power converter is adjusted according to a reference input signal.

2. The system of claim 1, wherein the power converter is a boost converter. 3 . The system of claim 1 , wherein the reference input signal is a reference voltage indicative of a desired voltage level to be driven to the load. 4 . The system of claim 1 , wherein the current control subsystem is further configured to enable the magnetizing current and the output current to be controlled independently of each other.

5. The system of claim 1 , wherein the current control subsystem is further configured to transform a current parameter from a first coordinate space defined by a first current through a first winding of the dual counter-wound inductor and a second current through a second winding of the dual counter-wound inductor to a second coordinate space defined by the magnetizing current and the output current in order to minimize the magnitude of the magnetizing current to prevent core saturation of the dual counter-wound inductor and adjust the amount of the output current.

6. The system of claim 1 , wherein the current control subsystem includes an observer configured to improve measurement accuracy of the first current and the second current by estimating the first current when the first current through the first winding of the dual counter-wound inductor is measured out of phase with respect to and by estimating the second current when the second current through the second winding of the dual counter-wound inductor is measured out of phase with respect to.

7. The system of claim 1 , wherein the current control subsystem comprises a current control loop for controlling the power converter, and is further configured to measure the following as feedback parameters of the current control loop: a first average current in a first winding of the double counter-wound inductor over a duration of a pulse width modulation cycle of the first winding; and A second average current in a second winding of the dual counter-wound inductor over a duration of a pulse width modulation cycle of the second winding.

8. A method for controlling current through a double counter-wound inductor integral with a power converter, the double counter-wound inductor being configured so that its windings generate opposing magnetic fields in its magnetic core, the method comprising: minimizing the magnitude of the magnetizing current of the double counter-wound inductor to prevent core saturation of the double counter-wound inductor; and An amount of output current delivered to a load by the power converter is adjusted according to a reference input signal.

9. The method of claim 8, wherein the power converter is a boost converter.

10. The method of claim 8, wherein the reference input signal is a reference voltage indicative of a desired voltage level to be driven to the load.

11. The method of claim 8, further comprising controlling the magnetizing current and the output current independently of each other.

12. The method of claim 8 , further comprising, in order to minimize the magnitude of the magnetizing current to prevent core saturation of the dual counter-wound inductor and adjust the amount of the output current, transforming a current parameter from a first coordinate space defined by a first current passing through a first winding of the dual counter-wound inductor and a second current passing through a second winding of the dual counter-wound inductor to a second coordinate space defined by the magnetizing current and the output current.

13. The method of claim 8 further comprising implementing an observer configured to improve accuracy of measurement of the first current and the second current by estimating the first current when the first current through the first winding of the double counter-wound inductor is measured out of phase with respect to and by estimating the second current when the second current through the second winding of the double counter-wound inductor is measured out of phase with respect to.

14. The method of claim 8, further comprising implementing a current control loop for controlling the power converter, and further configured to measure the following as feedback parameters of the current control loop: a first average current in a first winding of the double counter-wound inductor over a duration of a pulse width modulation cycle of the first winding; and A second average current in a second winding of the dual counter-wound inductor over a duration of a pulse width modulation cycle of the second winding.

15. A system for controlling current through a double counter-wound inductor integral with a power converter, the double counter-wound inductor being configured so that its windings generate opposing magnetic fields in its magnetic core, the system comprising: an input for receiving a reference input signal; as well as A current control subsystem, wherein the current control subsystem is configured to: minimizing the magnitude of the magnetizing current of the double counter-wound inductor to prevent core saturation of the double counter-wound inductor; and An amount of output current delivered to a load by the power converter is adjusted according to the reference input signal.

16. The system of claim 15, wherein the power converter is a boost converter.

17. The system of claim 15, wherein the reference input signal is a reference voltage indicative of a desired voltage level to be driven to the load.

18. The system of claim 15, wherein the current control subsystem is further configured to enable the magnetizing current and the output current to be controlled independently of each other.

19. The system of claim 15, wherein the current control subsystem is further configured to transform a current parameter from a first coordinate space defined by a first current through a first winding of the dual counter-wound inductor and a second current through a second winding of the dual counter-wound inductor to a second coordinate space defined by the magnetizing current and the output current in order to minimize the magnitude of the magnetizing current to prevent core saturation of the dual counter-wound inductor and adjust the amount of the output current.

20. The system of claim 15, wherein the current control subsystem includes an observer configured to improve accuracy of measurement of the first current and the second current by estimating the first current when the first current through the first winding of the dual counter-wound inductor is measured to be out of phase and by estimating the second current when the second current through the second winding of the dual counter-wound inductor is measured to be out of phase.

21. The system of claim 15, wherein the current control subsystem comprises a current control loop for controlling the power converter, and is further configured to measure the following as feedback parameters of the current control loop: a first average current in a first winding of the double counter-wound inductor over a duration of a pulse width modulation cycle of the first winding; and A second average current in a second winding of the dual counter-wound inductor over a duration of a pulse width modulation cycle of the second winding.

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