Power conversion device

Abstract

The power conversion device of the present application includes an inverter that converts a direct current voltage into an alternating current voltage to drive a synchronous motor, and a magnetic pole position correction unit that corrects an error between a rotation angle sensor and a rotor magnetic pole position of the synchronous motor. The magnetic pole position correction unit includes an actual current phase operation unit that operates a current phase based on a current when a three-phase line-to-line short circuit is caused in the rotation of the synchronous motor, and an ideal current phase calculation unit that calculates an ideal current phase based on a rotation speed of the rotor and a temperature of a stator, and corrects the magnetic pole position based on a difference between outputs of the actual current phase operation unit and the ideal current phase calculation unit.

Description

TECHNICAL FIELD

[0001] The present application relates to a power conversion device. BACKGROUND

[0002] In the case of controlling a system in which an inverter drives a permanent magnet synchronous motor, information on the magnetic pole position of the rotor with respect to the stator is required. In the case of an electric vehicle, the magnetic pole position information is acquired using a rotation angle sensor (a resolver or the like) mounted to the rotor, and torque control of the motor is performed, but if there is an angle error caused by an installation error of the rotation angle sensor, torque different from the torque command value is generated, and this results in deterioration of the drivability of the vehicle.

[0003] Therefore, the actual rotor position of the permanent magnet synchronous motor must be estimated, and the offset error of the rotation angle sensor is detected based on the estimated actual rotor position. As such a related art, for example, Patent Literature 1 and Patent Literature 2 are described.

[0004] Related Art Documents

[0005] Patent Literature

[0006] Patent Literature 1: Japanese Patent Laid-Open No. 2014-050122

[0007] Patent Literature 2: Japanese Patent Laid-Open No. 2017-212783 SUMMARY

[0008] Problems to be Solved by the Invention

[0009] The invention described in Patent Literature 1 is to find an angle at which the q-axis current becomes 0 by applying only the d-axis voltage in the rotor stopped state. However, this method makes it difficult to calibrate the size of Vd and the energizing time and to determine 0 of the q-axis current, and thus it is difficult to sufficiently improve the precision.

[0010] The invention described in Patent Literature 2 is to calculate the offset error based on the difference between the phase current detected in the state in which the motor terminals are short-circuited and the theoretical value of the phase current. However, if the rotation speed is relatively low (for example, 1000 to 2000 rpm), the error caused by the temperature of the motor, particularly the temperature of the stator, is large, and thus it is difficult to sufficiently improve the precision.

[0011] The problem to be solved by the present application is to detect the offset error of the rotation angle sensor of the rotor with high precision.

[0012] Technical Means for Solving the Problem

[0013] Therefore, the power conversion device of the present application is characterized by comprising: an inverter that converts a direct-current voltage into an alternating-current voltage to drive a synchronous motor; and a magnetic pole position correction section that corrects an error of a rotation angle sensor and a rotor magnetic pole position of the synchronous motor, the magnetic pole position correction section comprising: an actual current phase operation section that operates a current phase from a current when a three-phase interline short circuit is made in rotation of the synchronous motor; and an ideal current phase calculation section that calculates an ideal current phase from a rotation speed of the rotor and a temperature of a stator, the magnetic pole position being corrected from a difference between outputs of the actual current phase operation section and the ideal current phase calculation section.

[0014] Effects of the Invention

[0015] The present application can detect an offset error of a rotation angle sensor of a rotor with high precision. BRIEF DESCRIPTION OF DRAWINGS

[0016] Figure 1 A diagram showing a schematic circuit configuration of the power conversion device.

[0017] Figure 2 A processing block diagram for explaining processing of the controller 4.

[0018] Figure 3 A flowchart showing the order of offset error calculation.

[0019] Figure 4 A diagram showing waveforms of each phase current after the three-phase short circuit is started.

[0020] Figure 5 A block diagram showing the configuration of the magnetic pole position correction section 46 in Embodiment 1.

[0021] Figure 6 A graph showing the relationship between the rotation speed and the ideal current phase.

[0022] Figure 7 A block diagram showing Modification Example 1 of the ideal phase current calculation section 55.

[0023] Figure 8 A block diagram showing Modification Example 2 of the ideal phase current calculation section 55.

[0024] Figure 9 A block diagram showing the configuration of the magnetic pole position correction section 46 in Embodiment 2.

[0025] Figure 10 A block diagram showing Modification Example of the magnetic pole position correction section 46 in Embodiment 2. DETAILED DESCRIPTION

[0026] Next, an embodiment of the power conversion device of the present application will be described with reference to the drawings. Further, the same reference numerals are applied to the same elements in each drawing, and overlapping description will be omitted.

[0027] Embodiment 1

[0028] Figure 1 A diagram showing the outline circuit configuration of the power conversion device. Figure 1 In the present embodiment, the power conversion device of the motor 2 is mainly composed of the inverter 1, the battery 3, the controller 4, and the like. The inverter 1 is a three-phase voltage type two-level inverter.

[0029] The motor 2 is connected to a drive mechanism of an electric vehicle system, and the vehicle is propelled by the rotation of the motor 2. The motor 2 of the present embodiment is an alternating-current motor, and is a three-phase interior permanent magnet synchronous motor (IPMSM). The motor 2 operates by the interaction of the magnetic flux generated by the permanent magnet provided on the rotor, which is not shown, and the magnetic field generated by the current iu, iv, iw flowing through the three-phase windings 5, 6, 7 fixed to the armature. iu, iv, iw are the U-phase current, the V-phase current, and the W-phase current, respectively. Further, in the present embodiment, a permanent magnet synchronous motor is used for the description, but the present application is also established even for other synchronous motors such as a wound-field synchronous motor.

[0030] The motor 2 is provided with a magnetic pole position sensor 8. The magnetic pole position sensor 8 has a function of detecting the magnetic pole position of the rotor of the motor 2 as a rotation angle. The magnetic pole position sensor 8 outputs a magnetic pole position signal 9 (θ) to the controller 4. The processing of the magnetic pole position signal 9 (θ) in the controller 4 will be described later. As the magnetic pole position sensor 8, a resolver, a rotary encoder, an absolute encoder, or the like can be used. Figure 2

[0031] The current sensor 10 detects the current flowing through the windings 5, 6, 7, and outputs a U-phase current sensor signal 11 (a signal indicating the current iu), a V-phase current sensor signal 12 (a signal indicating the current iv), and a W-phase current sensor signal 13 (a signal indicating the current iw) to the controller 4. The processing of the current sensor signals 11, 12, 13 in the controller 4 will also be described later. Figure 2

[0032] ​​The inverter 1 includes switching elements 14, 15, 16, 17, 18, 19 and freewheeling diodes 20, 21, 22, 23, 24, 25. The switching elements 14 to 19 in this embodiment are Si-IGBTs including a gate terminal, a collector terminal, and an emitter terminal. The freewheeling diodes 20 to 25 are connected between the collector terminals and the emitter terminals of the switching elements 14 to 19, respectively. With regard to the freewheeling diodes 20 to 25, in the case where the collector terminal of the switching element 14 to 19 is at a higher potential than the emitter terminal, current flows through the freewheeling diode 20 to 25, and a high reverse voltage is prevented from being applied to the switching element 14 to 19. However, the inverter circuit is not limited to the combination of Si-IGBTs and freewheeling diodes, and can be configured by other semiconductor elements.

[0033] The switching between the on and off of the switching elements 14 to 19 is performed in accordance with gate drive signals 26, 27, 28, 29, 30, 31 connected to the gate terminals of the switching elements 14 to 19, respectively. The six gate signals 32 that become the sources of the gate drive signals 26 to 31 are generated by the controller 4 and output to a gate drive circuit 35. The gate drive circuit 35 converts the gate signals 32 into the potentials required for the switching between the on and off of the switching elements 14, 15, 16, 17, 18, 19 and outputs the gate drive signals 26, 27, 28, 29, 30, 31. The generation of the gate signals 32 in the controller 4 will be described later. Figure 2

[0034] The emitter terminal of the switching element 14 and the collector terminal of the switching element 15 are connected to each other, and the connection point thereof is connected to the winding 5 to flow the current iu. The emitter terminal of the switching element 16 and the collector terminal of the switching element 17 are connected to each other, and the connection point thereof is connected to the winding 6 to flow the current iv. The emitter terminal of the switching element 18 and the collector terminal of the switching element 19 are connected to each other, and the connection point thereof is connected to the winding 7 to flow the current iw. The collector terminals of the switching elements 14, 16, 18 are connected to each other and connected to the high-potential DC line 33. Further, the emitter terminals of the switching elements 15, 17, 19 are connected to each other and connected to the low-potential DC line 34.

[0035] Thus, the controller 4 performs the on and off of the switching elements 14, 15, 16, 17, 18, 19 at appropriate timings in accordance with the generated gate signals 32, controls the currents iu, iv, iw flowing to the windings 5, 6, 7, and achieves the rotation control of the motor 2. The gate signals 32 are in the form of PWM (Pulse Width Modulation) signals in a manner that becomes a sinusoidal signal in which the currents iu, iv, iw have a 120-degree phase difference.

[0036] ​The voltage sensor 36 is connected to the high potential DC line 33 and the low potential DC line 34, and detects the potential difference therebetween. The potential difference between the high potential DC line 33 and the low potential DC line 34 is normally a high voltage of, for example, 100 V or more, so the voltage sensor 36 generates a DC voltage sensor signal 37 (Vdc) that has been converted to a low voltage that the controller 4 can detect, and inputs it to the controller 4.

[0037] The smoothing capacitor 38 included in the inverter 1 is connected between the high potential DC line 33 and the low potential DC line 34. The smoothing capacitor 38 has the function of suppressing the pulsation of the DC voltage that occurs due to the switching action of the switching elements 14 to 19.

[0038] In the battery 3, the terminal on the high potential side of the battery 3 is connected to the high potential DC line 33, and the terminal on the low potential side of the battery 3 is connected to the low potential DC line 34. Thus, it functions as a direct current power source that supplies power to the inverter 1 and the motor 2.

[0039] The torque command 39 (T*) given from, for example, the vehicle's higher-level controller such as an ECU (Electronic Control Unit) is input to the controller 4. The controller 4 performs torque control of the motor 2 based on this torque command 39 (T*).

[0040] Furthermore, although Figure 1 The stator and rotor temperatures of the motor 2 are input to the controller 4. The stator and rotor temperatures are desirably detected by temperature sensors, but it is also possible to configure so as to use temperatures that are inferred by a known method.

[0041] Figure 2 A processing block diagram for explaining the processing of the controller 4.

[0042] Figure 2 In the controller 4, the processing block includes a current command calculation section 40, a three-phase / two-phase conversion section 41, a current control section 42, a two-phase / three-phase conversion section 43, a speed calculation section 44, a PWM gate control signal generation section (control signal generation section) 45, a magnetic pole position correction section 46, a deviation calculator 47, a deviation calculator 48, and an adder 49. The three-phase voltage command value generation section is formed by the current command calculation section 40, the three-phase / two-phase conversion section 41, the current control section 42, the two-phase / three-phase conversion section 43, the deviation calculator 47, and the deviation calculator 48. Normally, the three-phase voltage command values are generated based on the magnetic pole position signal 9 (θ) detected by the magnetic pole position sensor 8, but in the present application, a corrected magnetic pole position signal 50 (θ') that has been corrected by the method described later is used instead of the magnetic pole position signal 9 (θ).

[0043] The controller 4 performs rotational control by flowing three-phase currents iu, iv, iw to the motor 2, and the inside of the controller 4 uses a method of so-called current vector control, that is, processing is performed in a coordinate system that is converted from a three-phase fixed coordinate to a two-phase rotational coordinate represented by a d-axis and a q-axis.

[0044] The current command operation section 40 calculates a d-axis current command value id* and a q-axis current command value iq*. The torque command 39 (T*), the rotational angular velocity ω, and the DC voltage sensor signal 37 (Vdc) are input to the current command operation section 40, and the d-axis current command value id* and the q-axis current command value iq* are calculated from these inputs. The rotational angular velocity ω is calculated in the speed operation section 44 from the corrected magnetic pole position signal 50 (θ').

[0045] In a synchronous motor having magnetic saliency such as an IPMSM, the torque T is represented as in the following expression (1) and depends on the current.

[0046] T = Pp - {Φ + (Ld - Lq)id} - iq (1)

[0047] In expression (1), id is the d-axis current, iq is the q-axis current, Pp is the number of pole pairs, Ld is the d-axis inductance, Lq is the q-axis inductance, and Φ is the magnet flux.

[0048] The corrected magnetic pole position signal 50 (θ') is input to the three-phase two-phase conversion section 41 and the two-phase three-phase conversion section 43, and is used for conversion between the two-phase coordinate of the d-axis and the q-axis and the three-phase coordinate.

[0049] The three-phase two-phase conversion section 41 performs coordinate conversion to the d-axis and the q-axis with respect to the current sensor signals 11 (iu), 12 (iv), 13 (iw) based on information of the corrected magnetic pole position signal 50 (θ'), and outputs a d-axis detected current id and a q-axis detected current iq.

[0050] The deviation calculator 47 calculates a deviation of the d-axis current command value id* output from the current command operation section 40 from the d-axis detected current id output from the three-phase two-phase conversion section 41, and outputs the d-axis current deviation Δid to the current control section 42. The deviation calculator 48 calculates a deviation of the q-axis current command value iq* output from the current command operation section 40 from the q-axis detected current iq output from the three-phase two-phase conversion section 41, and outputs the q-axis current deviation Δiq to the current control section 42.

[0051] The current control section 42 performs feedback control in a manner in which the d-axis differential current Δid representing a deviation of a command value as a target value from a measured value as an output value and the q-axis current deviation Δiq become zero, and calculates and outputs a d-axis voltage command Vd* and a q-axis voltage command Vq* as voltage commands in order to update the output value. The feedback control in the current control section 42 is performed by, for example, PI control. The d-axis voltage command value Vd* and the q-axis voltage command value Vq* output from the current control section 42 are input to the two-phase / three-phase conversion section 43, and three-phase voltage command values Vu*, Vv*, Vw* are calculated and output in accordance with the corrected magnetic pole position signal 50 (θ').

[0052] The PWM gate control signal generation section (control signal generation section) 45 generates six gate signals 32 as PWM signals by comparison of the three-phase voltage command values Vu*, Vv*, Vw* with a carrier not shown, and outputs them to the gate drive circuit 35.

[0053] The magnetic pole position correction section 46 is a portion that becomes a feature of the present application. In the magnetic pole position correction section 46, an offset error Δθ caused by an installation error of the magnetic pole position sensor 8 is calculated. The calculated offset error Δθ is added to the magnetic pole position signal 9 (θ) detected by the magnetic pole position sensor 8, and is used as a corrected magnetic pole position signal 50 (θ') for generation of the three-phase voltage command values and the like.

[0054] Figure 3 The specific procedure for calculating the offset error.

[0055] First, it is determined whether or not there is a request for angle error correction from a higher-level controller. For example, as one example of execution, the angle error correction is executed before shipment of a product after the magnetic pole position sensor 8 is assembled to the motor 2. In addition, it can be executed at the time of replacement of the inverter 1 or the motor 2, it can be executed in a case where a decrease in output or torque of the motor 2 is detected, and the like. The higher-level controller mounted in a vehicle, for example, outputs a request for angle error correction when it is determined that the angle error correction of the present application can be executed in accordance with information such as the rotational speed of the motor 2.

[0056] In a case where there is a request for angle error correction from the higher-level controller, three-phase short-circuiting is started. The three-phase short-circuiting is a state in which the switching elements 14, 16, 18 constituting the upper arm of the inverter 1 are simultaneously set to be on and the switching elements 15, 17, 19 constituting the lower arm are simultaneously set to be off, or a state in which the switching elements 14, 16, 18 constituting the upper arm are simultaneously set to be off and the switching elements 15, 17, 19 constituting the lower arm are simultaneously set to be on.

[0057] After the start of the three-phase short-circuiting, the passage of a prescribed threshold time is waited for. The threshold time is set to be, for example, 0.1 seconds. The reason for this will be described later. Figure 4 The reason for this will be described. Figure 4The diagram shows the current waveforms of phases U through W when a three-phase short circuit begins at a certain moment during the rotation of motor 2. For example... Figure 4 As shown, in the transient state immediately following a three-phase short circuit, the current waveform deviates. In this state, the d-axis and q-axis currents oscillate, making it impossible to perform magnetic pole position error correction with high precision. Therefore, magnetic pole position correction calculations must be performed after the deviations in each phase current converge and transition to a steady state.

[0058] The threshold time until the transition to a steady state can be calculated based on the characteristics of motor 2. The time constant in a motor with d-axis inductance Ld, q-axis inductance Lq, and stator resistance R can be expressed as 2Ld·Lq / {R·(Ld+Lq)}.

[0059] Furthermore, in this embodiment, the determination of when the state has reached a steady state is made by comparing it with a threshold time. However, other methods can also be used to make the determination, such as when the amplitude of each phase current has fallen within a specified range or when the oscillation of the d-axis current id and the q-axis current iq has fallen below a specified threshold.

[0060] Figure 3 In the process, when the three-phase short-circuit time has exceeded the threshold, a magnetic pole position correction calculation is performed. Details of the magnetic pole position correction calculation will be provided in [link to relevant documentation]. Figure 5 This will be described later. After the magnetic pole position correction calculation is completed, the three-phase short circuit ends, and the system returns to the normal PWM mode.

[0061] Figure 5 This is a block diagram illustrating the configuration of the magnetic pole position correction unit 46 in this embodiment. Figure 5 The block diagram, for Figure 3 The order of magnetic pole position correction operations will be explained.

[0062] picture Figure 3 As explained earlier, magnetic pole position correction calculations are performed during a three-phase short circuit. First, the dq transformation is performed on the currents detected during the three-phase short circuit to calculate the actual current phase βact. The actual current phase βact during a three-phase short circuit is expressed by equation (2). The current phase is referenced to the q-axis at 0 degrees and is positive in the counterclockwise direction.

[0063] βact = tan -1 (-id / iq)···(2)

[0064] Furthermore, the ideal current phase calculation unit 55 calculates the ideal current phase βideal based on the rotational speed (angular velocity ω) of the motor 2 and the stator temperature Ts. The ideal current phase βideal under three-phase short circuit is expressed by equation (3). Here, ω is the electric angular velocity [rad / s], Lq is the q-axis inductance [H], and R is the stator resistance [Ω].

[0065] βideal = tan -1 (-ω · Lq / R) · · · (3)

[0066] Here, the so-called ideal current phase βideal is the current phase in the case where there is no error in the magnetic pole position. The q-axis inductance Lq varies depending on the q-axis current, and the stator resistance R varies depending on the stator temperature. Among these, if the rotational speed is equal to or higher than a predetermined rotational speed (for example, 1000 rpm), the q-axis current hardly varies, so it can be regarded as being substantially fixed. However, the stator resistance R depends on the temperature, and as the temperature rises, the resistance value also rises. Therefore, in the present application, a correction operation is performed using the stator temperature Ts.

[0067] Figure 6 FIG. 3 is a graph showing the relationship between the rotational speed and the ideal current phase. The current phase of the vertical axis is expressed by the value when the q-axis is taken as the 0-degree reference. As shown in FIG. 3, in a region where the rotational speed is relatively low, the deviation caused by the temperature is large, and the error in the case where the temperature correction is not performed is large. Figure 6

[0068] As described above, after the ideal current phase βideal that takes into account the influence of the stator temperature Ts is calculated, the difference from the actual current phase βact is found. This difference corresponds to the offset error Δθ of the magnetic pole position sensor 8.

[0069] As described above, the power conversion device of the present embodiment is characterized by including: an inverter that converts a direct-current voltage into an alternating-current voltage to drive a synchronous motor; and a magnetic pole position correction section 46 that corrects the error of the magnetic pole position sensor 8 of the motor 2 and the rotor magnetic pole position. The magnetic pole position correction section 46 includes: an actual current phase operation section that operates the current phase βact from the current when the three-phase lines are short-circuited in the rotation of the motor 2; and an ideal current phase calculation section that calculates the ideal current phase βideal from the rotational speed of the rotor and the temperature of the stator. Furthermore, the magnetic pole position is corrected from the difference Δθ of the outputs of the actual current phase operation section and the ideal current phase calculation section.

[0070] Thus, the magnetic pole position can be found with high accuracy even in the low-speed region. In other words, by the present application, the offset error can be corrected in a simple manner, so the tolerance management at the time of assembly of the magnetic pole position sensor can be simplified. Furthermore, in the magnetic pole position error correction in the present embodiment, the current phase is directly found from the current value on the rotational coordinates, so the correction can be performed in a simpler configuration than the method of calculating the ideal current using the three-phase current on the fixed coordinates. Furthermore, since the offset angle is found using the three-phase short-circuit current, the error caused by the currentless time due to the switching does not affect, and the correction can be performed with high accuracy.

[0071] ​Further, in the present embodiment, the stator temperature Ts is used in the calculation of the ideal current phase, whereby the shift correction is made with higher accuracy using fewer parameters. However, the use of other parameters is not excluded. For example, the ideal current phase can be calculated taking into account not only the stator temperature but also the influence of the rotor temperature.

[0072] Next, a modification of the ideal current phase calculation section in Figure 5 will be described. Figure 7 A modification of the ideal current phase calculation section in -1 is an example in which the stator resistance R is calculated using a one-dimensional map according to the stator temperature Ts. The stator resistance R calculated using the one-dimensional map, the q-axis inductance taken as a fixed value, and the rotor speed (rotational speed ω) are used to calculate βideal = tan Figure 8 In a modification of the ideal current phase calculation section in

[0073] Embodiment 2

[0074] In the present embodiment, in addition to the configuration of the magnetic pole position correction section 46 shown in Figure 5 , a noise removal section is provided. The other aspects are the same as in Embodiment 1.

[0075] Ideally, the three-phase short-circuit current is supposed to be a three-phase symmetrical sinusoidal current as shown in the steady state of Figure 4 . However, in reality, it becomes a current in which noise is superimposed due to the deviation of the impedance of each phase, the spatial harmonics of the induced voltage, the error of the current sensor, and the like. Further, in a case where the steady state is not perfect, the dq-axis currents also oscillate. Therefore, in the present embodiment, by providing the noise removal section, the influence of these can be removed, and more accurate angle correction can be achieved. The noise removal section removes noise by a first-order lag filter or an averaging process.

[0076] Figure 9 A block diagram of a case where a noise removal section 56 is provided at the rear stage of the deviation calculation section which calculates the deviation of the actual current phase βact from the ideal current phase βideal. Alternatively, it can be configured such that noise removal sections 57, 58 are provided at the rear stages of the actual phase calculation section and the ideal current phase calculation section, respectively, as in Figure 10

[0077] Explanation of Symbols

[0078] ​1…inverter, 2…motor, 3…battery, 4…controller, 5-7…three-phase winding, 8…magnetic pole position sensor, 9…magnetic pole position signal, 10…current sensor, 11…U-phase current sensor signal, 12…V-phase current sensor signal, 13…W-phase current sensor signal, 14-19…switching element, 20-25…return diode, 26-31…gate drive signal, 32…gate signal, 33…high potential DC line, 34…low potential DC line, 35…gate drive circuit, 36…voltage sensor, 37…DC voltage sensor signal, 38…smoothing capacitor, 39…torque command, 40…current command operation section, 41…three-phase / two-phase conversion section, 42…current control section, 43…two-phase / three-phase conversion section, 44…speed operation section, 45…PWM gate control signal generation section (control signal generation section), 46…magnetic pole position correction section, 47…deviation calculator, 48…deviation calculator, 49…adder, 50…corrected magnetic pole position signal (θ'), 55…ideal current phase calculation section, 56-58…denoising section, 65…motor drive device (motor control device).

Claims

1. A power conversion device, characterized by, Possessing: an inverter that converts a direct-current voltage into an alternating-current voltage to drive a synchronous motor; and a magnetic pole position correction section that corrects an error of a rotation angle sensor and a rotor magnetic pole position of the synchronous motor, the magnetic pole position correction section possesses: an actual current phase operation section that dq-transforms each phase current at the time of a three-phase line-to-line short circuit in the rotation of the synchronous motor, and operates a current phase from a d-axis current and a q-axis current obtained after the transformation; and an ideal current phase calculation section that calculates an ideal current phase from a rotational speed of the rotor and a temperature of a stator, the power conversion device corrects the magnetic pole position from a difference between outputs of the actual current phase operation section and the ideal current phase calculation section.

2. The power conversion device according to claim 1, characterized in that the magnetic pole position correction section starts operating after three-phase currents are converted into a steady state after a three-phase line-to-line short circuit.

3. The power conversion device according to claim 1, characterized in that a noise removal section is provided for a difference between outputs of the actual current phase operation section and the ideal current phase calculation section, or the calculated actual phase and ideal phase.

4. The power conversion device according to claim 1, characterized in that the ideal current phase calculation section calculates an ideal current phase from a rotational speed of the rotor, a temperature of the stator, and a temperature of the rotor of the synchronous motor.

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