Inverter control device, electric vehicle system

Abstract

The inverter control device of the present invention suppresses the deviation between the detection result of the output current of the inverter and the actual current. The current detection unit (7) detects the three-phase AC current output from the inverter (3) or input to the inverter (3). The inverter control device (1) controls the inverter (3) based on the current detection value of the detection result of the three-phase AC current given by the current detection unit (7) and the specified current command value. The inverter control device (1) corrects the current detection value to correct the detection error of the three-phase AC current caused by the delay time of the filter (72) which is a filtering element of the current detection unit (7).

Description

Technical Field

[0001] The present invention relates to an inverter control device and an electric vehicle system using the same. Background Art

[0002] The rotating electrical machines (motors) used in electric vehicles, such as electric vehicles (EVs) and hybrid electric vehicles (HEVs), require high power and high torque response. Therefore, a three-phase inverter is commonly used to drive a permanent magnet rotating electrical machine (PM motor) using rare earth sintered magnets that retain high energy. The three-phase inverter converts the DC voltage generated by a DC power supply into a line-to-line voltage (AC voltage) of a desired voltage and frequency using PWM (pulse width modulation) control, thereby enabling variable speed drive of the motor.

[0003] The three-phase inverter installed in an electric vehicle usually detects the three-phase current output from the three-phase inverter and controls the three-phase current in a manner consistent with the current command based on the torque command, which is determined by the driver's pedal force. The three-phase current is detected using a current detection element and an A / D converter. The current detection element is located in the main circuit of the inverter, and the A / D converter converts the analog signal detected by the current detection element into a digital signal. As with conventional circuits, the current detection element is equipped with a filter for removing noise accompanying the signal line. This denoising filter can remove noise, but on the other hand, the three-phase current passing through the denoising filter has a delay element, so there is a deviation between the detected current and the actual current flowing into the inverter.

[0004] Furthermore, in recent years, motors with low inductance and short electrical time constants, such as low-voltage, high-current motors for automotive applications, have increased in popularity. Compared to conventional motors, the difference between the detected current and the actual current in such motors has increased.

[0005] In conventional inverters, this discrepancy between the detected current and the actual current results in a mismatch between the torque command and the actual output torque, resulting in a failure to provide the acceleration performance requested by the driver. Furthermore, the accuracy of the fail-safe functions designed to protect the inverter and motor decreases, potentially damaging the inverter or motor in the worst-case scenario.

[0006] To avoid such a discrepancy between the detected current and the actual current flowing to the inverter, a technique is known, for example, from Patent Document 1. Patent Document 1 describes a technique in which the current slope is detected via a filter with a small time constant, and multiple filters with different time constants are switched based on the detection result.

[0007] Prior art literature

[0008] Patent Literature

[0009] Patent Document 1: Japanese Patent No. 6050841 Summary of the Invention

[0010] Problems to be solved by the invention

[0011] The technique in Patent Document 1 requires multiple filters with different time constants, making it difficult to use when the substrate mounting area is limited. Furthermore, filters with short time constants cannot adequately remove noise, causing the slope of the detected current to deviate from the true value. Consequently, filter switching cannot be performed properly, potentially worsening the discrepancy between the detected current and the actual current. In particular, in motors with low inductance, as described above, current ripple increases, increasing the likelihood of a worsening discrepancy between the detected current and the actual current.

[0012] Technical means to solve the problem

[0013] The inverter control device of the present invention controls the inverter based on a current detection value of the AC current detection result given by a current detection unit that detects the AC current output from or input to the inverter and a specified current command value, and includes a correction unit that corrects either the current detection value or the current command value to correct the detection error of the AC current caused by the delay time of the filter element of the current detection unit.

[0014] The electric vehicle system of the present invention includes the inverter control device, the inverter controlled by the inverter control device, and a three-phase synchronous motor driven by the inverter, and travels using the rotational driving force of the three-phase synchronous motor.

[0015] Effects of the Invention

[0016] According to the present invention, it is possible to suppress a discrepancy between a detection result of an output current of an inverter and an actual current. BRIEF DESCRIPTION OF THE DRAWINGS

[0017] Figure 1 It is a block diagram showing the configuration of a motor drive device including an inverter control device according to a first embodiment of the present invention.

[0018] Figure 2 This is a functional block diagram of the inverter control device according to the first embodiment of the present invention.

[0019] Figure 3 This diagram explains the difference between the current command value and the actual current according to the modulation rate.

[0020] Figure 4 This diagram shows the relationship between the triangular wave signal, the actual current, and the current detection value.

[0021] Figure 5 This diagram shows the carrier harmonic voltage ripple that varies with the modulation rate.

[0022] Figure 6 This is a functional block diagram of a current detection value correction unit according to the first embodiment of the present invention.

[0023] Figure 7 This is a diagram showing an example of the relationship between the current command value and the current detection value when the modulation rate is changed.

[0024] Figure 8 This is a diagram showing an example of actual current and current detection values when the motor torque is reversed in sign while the motor speed is fixed, during power running and regeneration.

[0025] Figure 9 This diagram illustrates how the direction of divergence between the true value and the detected value changes depending on the detection position of the current ripple.

[0026] Figure 10 Graph showing an example of the relationship between the current command value and the current detection value when the modulation rate and the power supply voltage are changed.

[0027] Figure 11 This is a diagram showing an example of the relationship between the current command value and the current detection value when the modulation rate is changed.

[0028] Figure 12 Graph showing an example of the relationship between the current command value and the current detection value when the modulation rate and the switching frequency are changed.

[0029] Figure 13 This is a diagram showing an example of the relationship between the current command value and the current detection value when the filter delay time constant is changed.

[0030] Figure 14 This is a functional block diagram of an inverter control device according to a second embodiment of the present invention.

[0031] Figure 15 This is a functional block diagram of a current command value correction unit according to a second embodiment of the present invention.

[0032] Figure 16 This is a functional block diagram of a current detection value correction unit according to a third embodiment of the present invention.

[0033] Figure 17 This is a functional block diagram of a current detection value correction unit according to a fourth embodiment of the present invention.

[0034] Figure 18 This is a configuration diagram of an electric vehicle system according to a fifth embodiment of the present invention. DETAILED DESCRIPTION

[0035] (First embodiment)

[0036] Hereinafter, a first embodiment of the present invention will be described with reference to the drawings.

[0037] Figure 1 This is a block diagram showing the configuration of a motor drive device 6 including an inverter control device according to a first embodiment of the present invention. The motor drive device 6 includes an inverter control device 1 , a motor 2 , an inverter 3 , a high-voltage battery 5 , and a current detection unit 7 .

[0038] The inverter control device 1 outputs a gate signal to the inverter 3 based on a torque command T* corresponding to a target torque required of the vehicle, thereby controlling the inverter 3. The details of the inverter control device 1 will be described later.

[0039] The inverter 3 is connected to the motor 2 and the high-voltage battery 5 , and includes an inverter circuit 31 , a pulse width modulation signal output unit 32 , and a smoothing capacitor 33 .

[0040] The inverter circuit 31 includes upper arm switching elements Sup, Svp, and Swp, and lower arm switching elements Sun, Svn, and Swn. When the motor 2 is in the traction mode, these switching elements convert the DC power supplied from the high-voltage battery 5 into AC power and output it to the motor 2. Furthermore, when the motor 2 is in the regenerative mode, the AC power generated by the motor 2 is converted into DC power and output to the high-voltage battery 5. This allows the inverter 3 to convert between DC and AC power.

[0041] The pulse-width modulation signal output unit 32 outputs a pulse-width modulation signal (PWM signal) to each switching element of the inverter circuit 31 based on a gate signal from the inverter control device 1. Each switching element switches at a predetermined timing based on the PWM signal input from the pulse-width modulation signal output unit 32, thereby performing conversion between direct current and alternating current in the inverter circuit 31.

[0042] The smoothing capacitor 33 smoothes the DC power supplied from the high-voltage battery 5 to the inverter circuit 31 or the DC power output from the inverter circuit 31 to the high-voltage battery 5 .

[0043] The high-voltage battery 5 serves as a DC voltage source for the motor drive device 6. The power supply voltage E of the high-voltage battery 5, a DC voltage, is converted by the inverter 3 into a three-phase AC voltage in a pulsed form with a variable voltage and frequency, which is then applied to the motor 2 as a line voltage. Furthermore, the DC voltage E of the high-voltage battery 5 fluctuates significantly depending on the state of charge of the high-voltage battery 5.

[0044] Motor 2 is a synchronous motor that is rotationally driven by the supply of line voltage from inverter 3. A rotation position sensor 21 is mounted on motor 2 so that the phase of the three-phase AC voltage is controlled by inverter control device 1 according to the phase of the induced voltage of motor 2. Rotation position detector 22 calculates the rotation position θ of the rotor in motor 2 based on the output signal of rotation position sensor 21. Here, rotation position sensor 21 can be, for example, a rotary transformer composed of an iron core and windings. Alternatively, rotation position sensor 21 can be constructed using a magnetoresistive element such as a GMR sensor or a Hall element. In addition, instead of providing rotation position detector 22 in motor drive device 6, rotation position θ can be estimated based on the three-phase current or three-phase voltage of motor 2.

[0045] The current detection unit 7 detects U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw as three-phase AC currents flowing through the motor 2. The current detection unit 7 includes a current detection element 71, a filter 72, and an analog / digital (AD) converter 73.

[0046] The current detection element 71 is composed of a Hall element, etc., and detects the three-phase AC currents Iu, Iv, and Iw respectively, and outputs voltages corresponding to these current values. Figure 1 While the example shown here shows a current detection unit 7 having three current detection elements 71 corresponding to each phase of the three-phase AC current, the number of current detection elements 71 may be two, and the current value of the remaining phase may be calculated based on the fact that the sum of the three-phase currents is zero. Furthermore, the pulsed DC bus current flowing into the inverter 3 may be detected as the voltage across the shunt resistor Rsh inserted between the smoothing capacitor 33 and the inverter 3 (DC current detection value Idc), and the three-phase AC current may be calculated based on this detection result. For example, the DC current detection value Idc may be acquired at an appropriate time depending on the state of each switching element of the inverter 3, thereby reproducing the three-phase AC current using the DC current detection value Idc. Alternatively, the three-phase AC currents Iu, Iv, and Iw may be detected using any other method.

[0047] The filter 72 is configured using resistors and capacitors to remove noise from the voltage signal outputted by the current detection element 71. The time constant τ of the filter 72 is derived from the resistance component Rf and the capacitance component Cf of the filter 72 using the following equation (1).

[0048] τ=Rf×Cf···(1)

[0049] The analog-to-digital (AD) converter 73 receives the voltage signal input from the current detection element 71 via the filter 72 as analog data and converts it into digital data at a predetermined sampling rate. The digital data thus obtained is output from the AD converter 73 to the inverter control device 1. Thus, the inverter control device 1 can obtain the detected values of the three-phase AC current in the form of digital data.

[0050] Next, use Figure 2 , details of the inverter control device 1 will be described. Figure 2 This is a functional block diagram of an inverter control device 1 according to a first embodiment of the present invention. The inverter control device 1 of this embodiment includes a current command generator 11, a three-phase / dq current converter 12, a current controller 13, a dq / three-phase voltage converter 14, a gate signal generator 15, a speed calculator 16, a switching frequency generator 17, a triangular wave generator 18, a modulation factor calculator 19, and a current detection value corrector 20. The inverter control device 1 drives the inverter circuit 31 of the inverter 3 based on the d-axis current command Id* and the q-axis current command Iq* corresponding to the power supply voltage E and the torque command T*. The inverter control device 1 is comprised of, for example, a microcomputer, and these functional blocks can be implemented by executing a predetermined program in the microcomputer. Alternatively, some or all of these functional blocks can be implemented using hardware circuits such as logic ICs or FPGAs.

[0051] The current command generating unit 11 determines a d-axis current command Id* and a q-axis current command Iq* based on the torque command T* and the power supply voltage E using a motor torque relational expression or a map.

[0052] The three-phase / dq current conversion unit 12 calculates a d-axis current detection value Id1 and a q-axis current detection value Iq1 by performing dq conversion on the three-phase AC current detection values based on the digital data of the U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw output from the current detection unit 7 and the rotational position θ output from the rotational position detector 22. The d-axis current detection value Id1 and the q-axis current detection value Iq1 calculated by the three-phase / dq current conversion unit 12 are hereinafter referred to as "first d-axis current detection value Id1" and "first q-axis current detection value Iq1," respectively.

[0053] The current detection value correction unit 20 corrects the first d-axis current detection value Id1 and the first q-axis current detection value Iq1 calculated by the three-phase / dq current conversion unit 12 to correct the detection errors of the U-phase AC current Iu, the V-phase AC current Iv, and the W-phase AC current Iw caused by the delay time of the filter 72 included in the current detection unit 7. It then outputs the d-axis current detection value Id2 and the q-axis current detection value Iq2 corresponding to these correction results. The details of the current detection value correction unit 20 will be described later. The d-axis current detection value Id2 and the q-axis current detection value Iq2 calculated by the current detection value correction unit 20 by correcting the first d-axis current detection value Id1 and the first q-axis current detection value Iq1, respectively, will be referred to as the "second d-axis current detection value Id2" and the "second q-axis current detection value Iq2," respectively.

[0054] The current control unit 13 calculates the d-axis voltage command Vd* and the q-axis voltage command Vq* so that the d-axis current command Id* and the q-axis current command Iq* calculated by the current command generation unit 11 are consistent with the second d-axis current detection value Id2 and the second q-axis current detection value Iq2 calculated by the current detection value correction unit 20, respectively.

[0055] The dq / three-phase voltage conversion unit 14 calculates the three-phase voltage command values, namely the U-phase voltage command value Vu*, the V-phase voltage command value Vv*, and the W-phase voltage command value Vw*, by performing UVW transformation on the d-axis voltage command Vd* and the q-axis voltage command Vq* based on the d-axis voltage command Vd* and the q-axis voltage command Vq* output from the current control unit 13 and the rotation position θ output from the rotation position detector 22.

[0056] The gate signal generator 15 generates pulsed voltages for each of the U, V, and W phases based on the comparison results between the three-phase voltage command values (U-phase voltage command value Vu*, V-phase voltage command value Vv*, and W-phase voltage command value Vw*) output from the dq / three-phase voltage converter 14 and the triangular wave signal Tr output from the triangular wave generator 18. The gate signal generator 15 then generates gate signals for the switching elements of each phase of the inverter 3 based on the generated pulsed voltages. At this time, the gate signals Gup, Gvp, and Gwp for the upper arm of each phase are logically inverted to generate gate signals Gun, Gvn, and Gwn for the lower arm. The gate signals generated by the gate signal generator 15 are output from the inverter control device 1 to the pulse width modulation signal output unit 32 of the inverter 3, where they are converted into PWM signals. This controls the on / off state of each switching element in the inverter circuit 31, thereby adjusting the output voltage of the inverter 3.

[0057] The speed calculation unit 16 calculates the electrical angle frequency ωr corresponding to the rotation speed (rotation speed) of the motor 2 based on the temporal change in the rotation position θ.

[0058] The switching frequency generator 17 outputs a switching frequency fc based on the torque command T* and the electrical angle frequency ωr. Alternatively, a predetermined fixed switching frequency fc may be output.

[0059] The triangular wave generating unit 18 outputs a triangular wave signal Tr according to the switching frequency fc.

[0060] The modulation factor calculation unit 19 calculates the modulation factor mod of the inverter 3 based on the d-axis current command Id* and the q-axis current command Iq* calculated by the current command generation unit 11 and the power supply voltage E. To calculate the modulation factor mod, the modulation factor calculation unit 19 first calculates the d-axis voltage Vd and the q-axis voltage Vq according to the following equation (2). Equation (2) is the voltage equation for the motor 2.

[0061] Vd=R×Id-ωr×Lq×Iq

[0062] Vq=R×Iq+ωr×Ld×Id+ωr×Ke···(2)

[0063] In formula (2), Id and Iq represent the d-axis current and q-axis current, respectively. In this embodiment, formula (2) is calculated as Id = Id* and Iq = Iq*. In addition, ωr represents the electrical angle frequency of motor 2, which is calculated by speed calculation unit 16 as described above. Furthermore, Ld and Lq represent the d-axis inductance and q-axis inductance, respectively, Ke represents the induced voltage constant, and R represents the winding resistance. These values are predetermined based on the structure of motor 2.

[0064] Furthermore, the induced voltage constant Ke has temperature dependence. Therefore, the induced voltage constant Ke in equation (2) can be corrected to account for this temperature dependence using equation (3). Equation (3) shows the relationship between the temperature of the rotor of motor 2 and the induced voltage. According to equation (3), when the rotor temperature changes from the normal temperature T_nomi, the temperature of the rotor magnets changes, and the induced voltage changes linearly accordingly.

[0065] Ke=Ke_nomi+(T-Tnomi)×K···(3)

[0066] In equation (3), Ke_nomi represents the induced voltage constant at normal temperature, T_nomi represents the normal rotor temperature, and K represents the temperature dependence slope of the induced voltage. These values are predetermined based on the structure of motor 2. Furthermore, T represents the rotor temperature, which is obtained using a temperature sensor (not shown).

[0067] The modulation factor calculation unit 19 calculates the modulation factor mod using the following equation (4) from the d-axis voltage Vd and the q-axis voltage Vq derived using equation (2).

[0068] mod=2√(Vd 2 +Vq 2 ) / E···(4)

[0069] Furthermore, the voltage utilization factor defined by the following equation (5) may be used instead of the modulation factor mod. The same calculation can be performed regardless of which one is used, so the following description will be based on the case of using the modulation factor mod.

[0070] Voltage utilization factor = (effective value of line voltage) / E···(5)

[0071] Next, before describing in detail the current detection value correction unit 20 that is a feature of this embodiment, a phenomenon that has been the focus of the present invention will be described below.

[0072] Figure 3 This diagram explains the difference between the current command value and the actual current according to the modulation rate. Figure 3 2 shows an example of the relationship between the rotation speed of the motor 2, the current command value, the actual current, and the modulation factor when the current command value and the power supply voltage are fixed.

[0073] like Figure 3 As shown, as the speed of motor 2 increases and the modulation rate increases, the actual current flowing from inverter 3 to motor 2 deviates from the current command value of inverter 3, with the maximum deviation being approximately 0.8%. While this error is small in isolation, factors that influence the torque accuracy of motor 2 include rotor temperature (magnet temperature) T and winding resistance R. Therefore, to improve torque accuracy, inverter control device 1 must obtain a current detection value that minimizes the error caused by the deviation from the actual current. Furthermore, the current detection value is used to protect inverter 3 and motor 2. Therefore, from the perspective of preventing damage to inverter 3 and motor 2, obtaining a current detection value that is consistent with the actual current is also important.

[0074] Next, refer to Figures 4 to 6 Two reasons why a difference occurs between the current detection value acquired by the current detection unit 7 at the timing corresponding to the triangular wave signal Tr and the actual current, which corresponds to the modulation factor mod calculated by the above-mentioned equation (4), will be described.

[0075] First, the first reason is that the current detection unit 7 has a filter 72 as an RC filter for noise removal. Figure 1 As shown, the current detection unit 7 for detecting three-phase AC current is provided with a filter 72 composed of resistors and capacitors. The inverter control device 1 obtains the current detection value through the filter 72, which causes a delay in the current detection value relative to the actual current, causing a deviation from the true value.

[0076] Figure 4 Graph showing the relationship between the triangular wave signal Tr, the actual current, and the current detection value. Figure 4 In the figure, the U-phase AC current Iu among the three-phase AC currents is taken as an example, and the triangular wave signal Tr output from the triangular wave generating unit 18 is indicated by symbol 41, the actual value (actual current) of the U-phase AC current Iu is indicated by symbol 42, and the value (current detection value) of the U-phase AC current Iu in the current detecting unit 7 after passing through the filter 72 is indicated by symbol 43. In addition, the digital data value obtained by sampling and holding the current detection value 43 by the analog-to-digital (AD) converter 73 is indicated by symbol 44.

[0077] like Figure 4 As shown, the analog-to-digital (AD) converter 73 samples and holds the detected value of the U-phase AC current Iu at, for example, the peak side of the triangular wave signal Tr, i.e., the moment when the signal transitions from rising to falling, and at the valley side of the triangular wave signal Tr, i.e., the moment when the signal transitions from falling to rising, to obtain the detected value. The detected value is then output to the inverter control device 1. Due to limitations on the processing load of the inverter control device 1, the detected value of the U-phase AC current Iu may be obtained only at one of the peak side and the valley side of the triangular wave signal Tr.

[0078] right Figure 4 Comparing the actual current 42 with the current detection value 43 reveals that the current detection value 43 is slightly higher than the actual current 42 at the peak and valley points of the triangular wave signal Tr. Therefore, the digital data value 44 obtained based on the current detection value 43 is also higher than the actual current 42 at the sampling time. Thus, since the current detection unit 7 obtains the current detection value 43 via the filter 72, a delay occurs in the current detection value 43 relative to the actual current 42, resulting in a divergence.

[0079] The second reason is that the carrier harmonic voltage ripple that changes according to the modulation rate mod affects the current detection value.

[0080] Figure 5 This is a diagram showing the carrier harmonic voltage ripple that changes according to the modulation rate mod. As mentioned above, the current detection value of the three-phase AC current output from the current detection unit 7 to the inverter control device 1 is detected on the peak side and valley side of the triangular wave signal Tr. Therefore, the frequency component that mainly affects the current detection value is the component that is twice the switching frequency fc of the triangular wave signal Tr. According to Figure 5 It is known that the double component of the switching frequency fc (2fc+ / -f1) reaches its maximum value when the modulation rate mod is around 0.6. Furthermore, f1 is the fundamental frequency of the motor 2, which is given by the following equation (6).

[0081] f1=(motor speed) / 60×(number of motor poles) / 2···(6)

[0082] As described above, the current detection value obtained by current detection unit 7 deviates from the actual current by a factor corresponding to the modulation rate mod due to the "noise removal RC filter" and the "carrier harmonic voltage ripple that varies with the modulation rate." Therefore, in the present invention, the current detection value correction unit 20 in inverter control device 1 corrects the current detection value obtained from current detection unit 7 based on the RC filter's time constant, the switching frequency fc, and the power supply voltage E, eliminating this deviation from the actual current.

[0083] Next, the current detection value correction unit 20 will be described in detail below. Figure 6 This is a functional block diagram of the current detection value correction unit 20 according to the first embodiment of the present invention. The current detection value correction unit 20 of this embodiment is composed of functional blocks: a correction current calculation unit (traction) 201, a correction current calculation unit (regeneration) 202, a traction / regeneration determination unit 203, a switching unit 204, a current absolute value calculation unit 205, a correction gain calculation unit 206, a d-axis correction unit 207, and a q-axis correction unit 208.

[0084] The correction current calculation unit (traction) 201 and the correction current calculation unit (regeneration) 202 calculate the correction current ΔIcmp_p for traction and the correction current ΔIcmp_g for regeneration, respectively, based on the modulation factor mod, the power supply voltage E, the switching frequency fc, the filter delay time constant τ, and the current amplitude |I|. The following describes the relationship between these correction currents and the modulation factor mod, the power supply voltage E, the current amplitude |I|, the switching frequency fc, and the filter delay time constant τ, followed by a detailed description of the correction current calculation method.

[0085] First, the relationship between the correction current and the modulation rate mod will be described. Figure 7 : is a diagram showing an example of the relationship between the current command value and the current detection value when the modulation rate mod is changed. Figure 7 In the example, the power supply voltage E, the switching frequency fc, and the filter delay time constant τ are set to fixed values.

[0086] like Figure 7 As shown in FIG. 1 , the current detection value varies depending on the modulation rate mod, causing a deviation from the current command value. Therefore, in order to compensate for this deviation, the correction current must be changed according to the modulation rate mod.

[0087] In addition, if Figure 7 As shown in the figure, the deviation between the current command value and the current detection value is different in the traction and regeneration. That is, the current detection value is different relative to the current command value in the traction and regeneration. Figure 8 and Figure 9 The reasons are explained.

[0088] Figure 8 This is a diagram showing an example of actual current and current detection values when the motor torque is reversed while the motor speed is fixed, during traction and regeneration. Figure 8 As shown on the left side of , during traction, the actual current including the current ripple is detected when the current ripple is falling. Therefore, the current detection value with filter delay is larger than the current detection value without filter delay. On the other hand, Figure 8 As shown on the right side of the graph, during regeneration, the actual current including the current ripple is detected when the current ripple is rising. Therefore, the current detection value with filter delay is smaller than the current detection value without filter delay.

[0089] Figure 9 This figure illustrates the change in the direction of deviation between the true value and the detected value caused by the detection position of the current ripple. Figure 9 As shown on the left side of the figure, when the current ripple is detected at the time of the drop, a current higher than the true value is detected due to the delay of the filter. Figure 9 As shown on the right side of FIG, during regeneration when detection is performed at the rising edge of the current ripple, a current lower than the true value is detected due to the delay of the filter.

[0090] Therefore, something like Figure 7 As explained in the previous section, the detected current value relative to the commanded current value varies depending on the traction and regeneration conditions. Therefore, the correction current must also be adjusted accordingly.

[0091] Next, the relationship between the correction current and the power supply voltage E will be described. Figure 10 : is a diagram showing an example of the relationship between the current command value and the current detection value when the modulation rate mod and the power supply voltage E are changed. Figure 10 Here, the current amplitude |I|, the switching frequency fc, and the filter delay time constant τ are set to fixed values.

[0092] like Figure 10 As shown, the difference between the current command value and the current detection value changes in proportion to the power supply voltage E. This is because as the power supply voltage E increases, the current ripple increases. Therefore, to compensate for this difference, the correction current must be changed in proportion to the power supply voltage E.

[0093] Next, the relationship between the correction current and the current amplitude |I| will be described. Figure 11 : is a diagram showing an example of the relationship between the current command value and the current detection value when the modulation rate mod is changed. Figure 11In this case, the switching frequency fc and the filter delay time constant τ are set to fixed values.

[0094] like Figure 11 As shown in the figure, when the current command value is increased, the current detection value increases, and the difference between the current command value and the current detection value increases. Therefore, in order to compensate for this difference, the correction current must be changed according to the current amplitude |I|.

[0095] Next, the relationship between the correction current and the switching frequency fc will be described. Figure 12 : is a diagram showing an example of the relationship between the current command value and the current detection value when the modulation rate mod and the switching frequency fc are changed. Figure 12 Here, the power supply voltage E, the current amplitude |I|, and the filter delay time constant τ are all set to fixed values.

[0096] like Figure 12 As shown, the deviation between the current command value and the current detection value decreases as the switching frequency fc increases. This is because as the switching frequency fc increases, the current flowing to motor 2 changes slightly with a fine cycle, thereby reducing current ripple. Therefore, to compensate for this deviation, the correction current must be changed inversely proportional to the switching frequency fc.

[0097] Finally, the relationship between the correction current and the filter delay time constant τ is described. Figure 13 This is a diagram showing an example of the relationship between the current command value and the current detection value when the filter delay time constant τ is changed. Figure 13 Here, the power supply voltage E, the current amplitude |I|, and the switching frequency fc are set to fixed values.

[0098] like Figure 13 As shown in the figure, the longer the filter delay time constant τ increases, the greater the deviation between the current command value and the current detection value. The reason is that when the filter delay time constant τ becomes longer, Figure 9 The detection position of the current ripple shown is shifted. Therefore, to compensate for this deviation, the correction current must be changed according to the filter delay time constant τ.

[0099] By organizing the relationships described above, the traction correction current ΔIcmp_p and the regeneration correction current ΔIcmp_g are expressed as follows: (7) and (8), respectively. Specifically, the correction current calculation unit (traction) 201 calculates the traction correction current ΔIcmp_p using equation (7), and the correction current calculation unit (regeneration) 202 calculates the regeneration correction current ΔIcmp_g using equation (8).

[0100] ΔIcmp_p=E / fc×τ×f_p(mod)···(7)

[0101] ΔIcmp_g=E / fc×τ×f_g(mod)···(8)

[0102] In equations (7) and (8), E represents the power supply voltage, fc represents the switching frequency, and τ represents the filter delay time constant. Furthermore, f_p(mod) in equation (7) represents a coefficient determined by a correction table for traction that depends on the modulation rate, and f_g(mod) in equation (8) represents a coefficient determined by a correction table for regeneration that depends on the modulation rate. Furthermore, the values of these correction tables can be derived in advance through simulation or experimentation under the conditions where the power supply voltage E, switching frequency fc, and filter delay time constant τ are fixed. Furthermore, linear interpolation can be performed using a spectrum of multiple conditions obtained by varying these parameters, thereby determining f_p(mod) and f_g(mod). This also fully demonstrates the effects of the present invention.

[0103] Back to Figure 6 As described above, the running / regeneration determination unit 203 determines whether the operation of the inverter 3 is in the running region or the regeneration region based on the electrical angle frequency ωr and the torque command T*.

[0104] Specifically, if the product of the electrical angle frequency ωr and the torque command T* is zero or positive, the pull / regeneration determination unit 203 determines that it is a pull region and outputs "1". If it is negative, it determines that it is a regeneration region and outputs "0".

[0105] Switching unit 204 selects the output of either correction current calculation unit (traction) 201 or correction current calculation unit (regeneration) 202 based on the output of traction / regeneration determination unit 203, and outputs the selected output to correction gain calculation unit 206. Specifically, when the output of traction / regeneration determination unit 203 is "1," correction current ΔIcmp_p for traction, which is the output of correction current calculation unit (traction) 201, is selected as correction current ΔIcmp and output to correction gain calculation unit 206. Furthermore, when the output of traction / regeneration determination unit 203 is "0," correction current ΔIcmp_g for regeneration, which is the output of correction current calculation unit (regeneration) 202, is selected as correction current ΔIcmp and output to correction gain calculation unit 206.

[0106] The current absolute value calculation unit 205 calculates the current amplitude |I|, that is, the absolute value of the three-phase AC current, for use in correcting the current calculation and the correction gain calculation, according to the following equation (9).

[0107] |I|=√(Id 2 +Iq 2 )···(9)

[0108] The correction gain calculation unit 206 calculates a correction gain Kcmp for correcting the current detection value according to the following equation (10).

[0109] Kcmp=(|I|+ΔIcmp) / |I|···(10)

[0110] The d-axis correction unit 207 calculates the second d-axis current detection value Id2 by multiplying the first d-axis current detection value Id1 by the correction gain Kcmp calculated by the correction gain calculation unit 206 using the following equation (11).

[0111] Id2=Kcmp×Id1···(11)

[0112] Like the d-axis correction unit 207 , the q-axis correction unit 208 uses the following equation (12) to obtain the second q-axis current detection value Iq2 by multiplying the first q-axis current detection value Iq1 by the correction gain Kcmp calculated by the correction gain calculation unit 206 .

[0113] Iq2=Kcmp×Iq1···(12)

[0114] As described above, the current detection value correction unit 20 corrects the first d-axis current detection value Id1 and the first q-axis current detection value Iq1 based on the modulation factor mod, the power supply voltage E, the switching frequency fc, the filter delay time constant τ, and the current amplitude |I|, respectively, to calculate the second d-axis current detection value Id2 and the second q-axis current detection value Iq2. This effectively reduces any discrepancies between the current detection value detected by the current detection unit 7 via the filter 72 and the actual current flowing between the motor 2 and the inverter 3.

[0115] This embodiment reduces the discrepancy between the true and detected current values, which can be exacerbated by the modulation rate and the RC filter's delay time constant, without requiring additional sensors. This ensures that the motor torque command matches the actual output torque command, enabling highly accurate motor torque output. Furthermore, by detecting the true value of the motor's three-phase current, it prevents the current from exceeding the detection thresholds required to protect the inverter and motor, thereby preventing damage to the inverter and motor.

[0116] According to the first embodiment of the present invention described above, the following effects are achieved.

[0117] (1) The current detection unit 7 detects the three-phase AC current output from or input to the inverter 3. The inverter control device 1 controls the inverter 3 based on the current detection values based on the detection results of the three-phase AC current provided by the current detection unit 7, namely the first d-axis current detection value Id1 and the first q-axis current detection value Iq1, and the specified current command values, namely the d-axis current command Id* and the q-axis current command Iq*. The inverter control device 1 includes a current detection value correction unit 20. The current detection value correction unit 20 corrects the first d-axis current detection value Id1 and the first q-axis current detection value Iq1 to correct the detection error of the three-phase AC current caused by the delay time τ of the filter 72, which is a filtering element of the current detection unit 7. Therefore, it is possible to suppress the deviation between the detection result of the output current of the inverter 3 and the actual current.

[0118] (2) The current detection value correction unit 20 corrects the first d-axis current detection value Id1 and the first q-axis current detection value Iq1 based on the modulation factor mod or the voltage utilization rate of the inverter 3. Therefore, the current detection values can be corrected to suppress the divergence between the current command value, which varies based on the modulation factor mod or the voltage utilization rate, and the actual current.

[0119] (3) The current detection value correction unit 20 corrects the first d-axis current detection value Id1 and the first q-axis current detection value Iq1 according to equations (7) to (12) described above, based on the switching frequency fc of the inverter 3, the DC voltage applied to the inverter 3, i.e., the power supply voltage E, the delay time τ of the filter 72, and the AC current amplitude |I|. Therefore, the current detection values can be corrected to suppress any divergence between the current command value, which varies depending on the aforementioned parameters, and the actual current.

[0120] (4) The current detection value correction unit 20 calculates a correction gain Kcmp using the correction gain calculation unit 206 based on the modulation factor mod or voltage utilization factor of the inverter 3, the switching frequency fc of the inverter 3, the DC voltage applied to the inverter 3, i.e., the power supply voltage E, the delay time τ of the filter 72, and the amplitude |I| of the AC current. Specifically, the correction current calculation unit (traction) 201 and the correction current calculation unit (regeneration) 202 calculate the traction correction current ΔIcmp_p and the regeneration correction current ΔIcmp_g, respectively, according to equations (7) and (8). Subsequently, the correction gain calculation unit 206 uses these calculation results and the current amplitude |I| calculated by the current absolute value calculation unit 205 according to equation (9) to calculate the correction gain Kcmp according to equation (10). Furthermore, in d-axis correction unit 207 and q-axis correction unit 208, the first d-axis current detection value Id1 and the first q-axis current detection value Iq1 are multiplied by correction gain Kcmp using equations (11) and (12), respectively, thereby correcting the first d-axis current detection value Id1 and the first q-axis current detection value Iq1. Therefore, the current detection values can be corrected while reliably suppressing any divergence between the current command value and the actual current.

[0121] (Second embodiment)

[0122] Next, a second embodiment of the present invention will be described. In this embodiment, an example in which the current command value is corrected instead of the current detection value will be described.

[0123] Figure 14 1A is a functional block diagram of an inverter control device 1A according to a second embodiment of the present invention. Figure 2 The inverter control device 1 of the first embodiment shown differs in that it includes a current command value correction unit 20A in place of the current detection value correction unit 20. Furthermore, like the inverter control device 1, the inverter control device 1A of this embodiment is also configured, for example, by a microcomputer, and each functional block can be implemented by executing a predetermined program in the microcomputer. Alternatively, some or all of the functional blocks can be implemented using hardware circuits such as logic ICs or FPGAs.

[0124] The current command value correction unit 20A corrects the d-axis current command Id1* and q-axis current command Iq1* output from the current command generation unit 11 to correct detection errors in the U-phase AC current Iu, V-phase AC current Iv, and W-phase AC current Iw caused by the delay time of the filter 72 included in the current detection unit 7. The current command value correction unit 20A then outputs the d-axis current command Id2* and q-axis current command Iq2* corresponding to these correction results to the current control unit 13. Hereinafter, the pre-corrected d-axis current command Id1* and q-axis current command Iq1* output from the current command generation unit 11 will be referred to as the "first d-axis current command Id1*" and the "first q-axis current command Iq1*," respectively. The d-axis current command Id2* and the q-axis current command Iq2* calculated by correcting the first d-axis current command Id1* and the first q-axis current command Iq1* by the current command value correction unit 20A are referred to as “second d-axis current command Id2*” and “second q-axis current command Iq2*”, respectively.

[0125] Figure 15 This is a functional block diagram of the current command value correction unit 20A according to the second embodiment of the present invention. Figure 6 The current detection value correction unit 20 of the first embodiment shown is different in that it includes a correction gain calculation unit 206A, a d-axis correction unit 207A, and a q-axis correction unit 208A instead of the correction gain calculation unit 206, the d-axis correction unit 207, and the q-axis correction unit 208, respectively.

[0126] Correction gain calculation unit 206A calculates correction gain Kcmp* for correcting the current command value according to the following equation (13).

[0127] Kcmp*=(|I|-ΔIcmp) / |I|···(13)

[0128] The d-axis correction unit 207A obtains the second d-axis current command Id2* by multiplying the first d-axis current command Id1* by the correction gain Kcmp* calculated by the correction gain calculation unit 206A using the following equation (14).

[0129] Id2*=Kcmp*×Id1*···(14)

[0130] Like the d-axis correction unit 207A, the q-axis correction unit 208A uses the following equation (15) to obtain the second q-axis current command Iq2* by multiplying the first q-axis current command Iq1* by the correction gain Kcmp* calculated in the correction gain calculation unit 206A.

[0131] Iq2*=Kcmp*×Iq1*···(15)

[0132] As described above, the current command value correction unit 20A corrects the first d-axis current command Id1* and the first q-axis current command Iq1*, respectively, based on the modulation factor mod, the power supply voltage E, the switching frequency fc, the filter delay time constant τ, and the current amplitude |I|, thereby calculating the second d-axis current command Id2* and the second q-axis current command Iq2*. This effectively reduces any discrepancies between the current detection value detected by the current detection unit 7 via the filter 72 and the actual current flowing between the motor 2 and the inverter 3, as in the first embodiment.

[0133] (Third embodiment)

[0134] Next, a third embodiment of the present invention will be described. In this embodiment, a different functional configuration is used to realize the Figure 2 An example of the current detection value correction unit 20 will be described.

[0135] picture Figure 7 As explained in [1], the direction of the divergence between the current command value (which varies according to the modulation factor mod) and the detected current value differs between traction and regeneration, so the correction current must also be adjusted accordingly. However, the difference in the correction current between traction and regeneration is simply the direction, or sign, of the current command value. Therefore, in this embodiment, the functional configuration of the detected current value correction unit 20 is simplified to account for this.

[0136] Figure 16 This is a functional block diagram of the current detection value correction unit 20 according to the third embodiment of the present invention. Figure 6 The first embodiment shown is different in that the correction current calculation unit (regeneration) 202 is not provided, and a gain change unit 209 is provided instead of the switch unit 204 .

[0137] Gain changing unit 209 multiplies the traction correction current ΔIcmp_p output from correction current calculation unit (traction) 201 by a gain of either 1 or -1, based on the output of traction / regeneration determination unit 203. Specifically, when the output of traction / regeneration determination unit 203 is "1," traction correction current ΔIcmp_p is multiplied by 1 and output to correction gain calculation unit 206 as correction current ΔIcmp. Furthermore, when the output of traction / regeneration determination unit 203 is "0," traction correction current ΔIcmp_p is multiplied by -1, inverting its sign, and output to correction gain calculation unit 206 as correction current ΔIcmp. Thus, the sign of correction current ΔIcmp is inverted depending on whether inverter 3 is operating in the traction range or the regeneration range.

[0138] According to the third embodiment of the present invention described above, the functional configuration of the current detection value correction unit 20 can be simplified while achieving the same operational effects as those described in the first embodiment.

[0139] Furthermore, in the third embodiment of the present invention described above, the correction current calculation unit (pulling) 201 can be omitted instead of the correction current calculation unit (regeneration) 202. In this case, the sign of the gain multiplied by the gain changing unit 109 can be reversed. In other words, the current detection value correction unit 20 of this embodiment can be implemented by including either the correction current calculation unit (pulling) 201 or the correction current calculation unit (regeneration) 202.

[0140] Furthermore, the same calculation method as that of the third embodiment of the present invention described above can be applied to the current command value correction unit 20A described in the second embodiment. In this manner, the same operational effects as those described in the second embodiment can be achieved.

[0141] (Fourth embodiment)

[0142] Next, a fourth embodiment of the present invention will be described. In this embodiment, a different functional configuration is used to realize the present invention. Figure 2 An example of the current detection value correction unit 20 will be described.

[0143] Figure 17 This is a functional block diagram of the current detection value correction unit 20 according to the fourth embodiment of the present invention. Figure 6 The first embodiment shown is different in that the correction gain calculation unit 206 is not provided, and the correction current ΔIcmp is input to the d-axis correction unit 207 and the q-axis correction unit 208 instead of the correction gain Kcmp.

[0144] In the present embodiment, the d-axis correction unit 207 calculates the second d-axis current detection value Id2 by integrating the correction current ΔIcmp with respect to the first d-axis current detection value Id1 using the following equation (16).

[0145] Id2=Id1+ΔIcmp···(16)

[0146] Similar to the d-axis correction unit 207 , the q-axis correction unit 208 uses the following equation (17) to calculate the second q-axis current detection value Iq2 by integrating the correction current ΔIcmp with respect to the first q-axis current detection value Iq1 .

[0147] Iq2=Iq1+ΔIcmp···(17)

[0148] According to the fourth embodiment of the present invention described above, the functional configuration of the current detection value correction unit 20 can be simplified while achieving the same operational effects as those described in the first embodiment.

[0149] Furthermore, the same calculation method as that of the fourth embodiment of the present invention described above can be applied to the current command value correction unit 20A described in the second embodiment and the current detection value correction unit 20 described in the third embodiment. This also achieves the same operational effects as those described in the second and third embodiments.

[0150] (Fifth embodiment)

[0151] Next, use Figure 18 , an embodiment in which the inverter control device of the present invention is applied to an electric vehicle system is described.

[0152] Figure 18 FIG. 5 is a structural diagram of an electric vehicle system according to a fifth embodiment of the present invention. Figure 18 As shown, the electric vehicle system of this embodiment includes a powertrain that uses a motor 2 as a motor generator.

[0153] exist Figure 18 In the electric vehicle system, a front wheel axle 801 is rotatably journaled at the front of a vehicle body 800, and front wheels 802 and 803 are provided at both ends of the front wheel axle 801. A rear wheel axle 804 is rotatably journaled at the rear of the vehicle body 800, and rear wheels 805 and 806 are provided at both ends of the rear wheel axle 804.

[0154] A differential 811, serving as a power distribution mechanism, is provided in the center of the front wheel axle 801. It distributes the rotational driving force transmitted from the engine 810 via the transmission 812 to the left and right front wheel axles 801. The engine 810 and the motor 2 are mechanically coupled together via a belt 830 by a pulley 810a provided on the crankshaft of the engine 810 and a pulley 820a provided on the rotating shaft of the motor 2.

[0155] Thus, the rotational driving force of the motor 2 can be transmitted to the engine 810, and the rotational driving force of the engine 810 can be transmitted to the motor 2. In the motor 2, under the control of the inverter control device 1, the three-phase AC power output from the inverter 3 is supplied to the stator coils of the stator, causing the rotor to rotate, generating a rotational driving force corresponding to the three-phase AC power.

[0156] That is, the motor 2 operates as an electric motor driven by the inverter 3 according to the control of the inverter control device 1, and on the other hand, it operates as a generator that receives the rotational driving force of the engine 810 and rotates the rotor, thereby inducing an electromotive force in the stator coil of the stator to generate three-phase alternating current.

[0157] The inverter 3 is a power conversion device that converts the direct current supplied from the high-voltage battery 5, which serves as a high-voltage (42V or 300V) power source, into three-phase alternating current. It controls the three-phase alternating current flowing to the stator coil of the motor 2 according to the operating instruction value, corresponding to the magnetic pole position of the rotor.

[0158] The three-phase AC power generated by the motor 2 is converted to DC power by the inverter 3 to charge the high-voltage battery 5. The high-voltage battery 5 is electrically connected to the low-voltage battery 823 via a DC-DC converter 824. The low-voltage battery 823 constitutes the low-voltage (14V) power supply of the electric vehicle system and is used to power the starter 825 for initial starting (cold start) of the engine 810, the radio, lights, etc.

[0159] When the vehicle is stopped, such as at a traffic light (idle-stop mode), engine 810 is stopped. When restarting engine 810 (hot start), inverter 3 drives motor 2 to restart engine 810. Furthermore, in idle-stop mode, if the high-voltage battery 5 is insufficiently charged or engine 810 is not fully warmed up, engine 810 continues to drive without stopping. Furthermore, in idle-stop mode, auxiliary equipment such as the air conditioner compressor, which is driven by engine 810, must be powered. In these cases, motor 2 is driven to drive the auxiliary equipment.

[0160] Even in acceleration mode or high-load operation mode, motor 2 is driven to assist the driving of engine 810. Conversely, in charging mode, which requires charging of high-voltage battery 5, engine 810 generates electricity in motor 2 to charge high-voltage battery 5. In other words, a regenerative mode is implemented for braking or decelerating the vehicle.

[0161] In the electric vehicle system of this embodiment using the inverter control device 1, the divergence between the true current value and the detected current value, which is exacerbated by the modulation rate and the delay time constant of the RC filter, can be reduced without adding additional sensors. This ensures that the command value is consistent with the actual torque, enabling the output of highly accurate motor torque. Consequently, the driver's desired torque can be achieved with an inexpensive configuration. Furthermore, the true value of the motor's three-phase current can be detected, preventing it from exceeding the detection threshold used to protect the inverter and motor, thereby preventing damage to the inverter and motor and, consequently, preventing the electric vehicle from stalling.

[0162] Furthermore, the electric vehicle system of this embodiment is described for a hybrid vehicle, but similar effects can be obtained in the case of a plug-in hybrid vehicle, an electric vehicle, and the like.

[0163] In addition, in the above embodiments, a separate inverter control device is described, but as long as it has the above functions, the present invention can also be applied to an inverter device in which an inverter control device and an inverter are integrated, or a motor drive system in which an inverter device and a motor are integrated.

[0164] The present invention is not limited to the above-described embodiment, and various modifications can be made without departing from the spirit of the present invention.

[0165] Explanation of symbols

[0166] 1.1A…Inverter control unit, 2…Motor, 3…Inverter, 5…High-voltage battery, 6…Motor drive unit, 7…Current detection unit, 11…Current command generation unit, 12…Three-phase / dq current conversion unit, 13…Current control unit, 14…dq / three-phase voltage conversion unit, 15…Gate signal generation unit, 16…Speed calculation unit, 17…Switching frequency generation unit, 18…Triangle wave generation unit, 19…Modulation rate calculation unit, 20…Current detection value correction unit, 20A…Current command value correction unit, 21…Rotation position sensor, 22…Rotation position Detector, 31…inverter circuit, 32…pulse width modulation signal output unit, 33…smoothing capacitor, 71…current detection element, 72…filter, 73…analog / digital (AD) converter, 201…corrected current calculation unit (traction), 202…corrected current calculation unit (regeneration), 203…traction / regeneration determination unit, 204…switching unit, 205…current absolute value calculation unit, 206, 206A…corrected gain calculation unit, 207, 207A…d-axis correction unit, 208, 208A…q-axis correction unit, 209…gain changing unit.

Claims

1. An inverter control device that controls an inverter based on a current detection value obtained by a current detection unit and a predetermined current command value, wherein the current detection unit detects the AC current output from or input to the inverter, the inverter control device comprising: A correction unit is provided for correcting one of the current detection value and the current command value to correct a detection error of the AC current caused by a delay time of a filter element included in the current detection unit. The correction unit corrects either the current detection value or the current command value according to a modulation factor or a voltage utilization factor of the inverter.

2. The inverter control device according to claim 1, characterized in that: The correction unit corrects one of the current detection value and the current command value based on a switching frequency of the inverter, a DC voltage applied to the inverter, a delay time of the filter element, and an amplitude of the AC current.

3. The inverter control device according to claim 1, wherein: The correction unit calculates a correction gain based on the modulation rate or voltage utilization rate of the inverter, the switching frequency of the inverter, the DC voltage applied to the inverter, the delay time of the filter element, and the amplitude of the AC current, and multiplies the current detection value or the current command value by the correction gain to correct either the current detection value or the current command value.

4. The inverter control device according to claim 3, characterized in that: The correction unit calculates a correction current value based on the modulation rate or the voltage utilization rate, the switching frequency, the DC voltage, and the delay time, and calculates the correction gain based on the correction current value and the amplitude of the AC current.

5. The inverter control device according to claim 1, characterized in that: The correction unit calculates a correction current value based on the modulation rate or voltage utilization rate of the inverter, the switching frequency of the inverter, the DC voltage applied to the inverter, and the delay time of the filter element, and accumulates the correction current value with the current detection value or the current command value to correct one of the current detection value or the current command value.

6. The inverter control device according to claim 4 or 5, characterized in that: The correction unit inverts the sign of the correction current value according to whether the inverter is operating in a pull region or a regeneration region.

7. An electric vehicle system, characterized in that: have: The inverter control device according to any one of claims 1 to 6; the inverter controlled by the inverter control device; and a three-phase synchronous motor driven by the inverter, The electric vehicle system travels using the rotational driving force of the three-phase synchronous motor.

Images