High-accuracy, fast voltage and temperature sensor circuit

Abstract

This disclosure relates to high-accuracy, fast voltage and temperature sensor circuits. A temperature sensing circuit includes: a current generation circuit that generates an initial current (Iptat) proportional to an absolute temperature; and a voltage generation circuit configured to mirror Iptat using an adjustable current source to generate a scaled current, and supplying the scaled current to a first terminal of a resistor to generate a reference voltage at the first terminal. A second terminal of the resistor has a voltage complementary to the absolute temperature (Vctat) applied to the second terminal. An analog-to-digital converter (ADC) has a reference input that receives the reference voltage and a data input that receives Vctat or an external source voltage. The ADC generates an output code indicating the ratio between: a) Vctat or the external source voltage, and b) the reference voltage. Digital circuitry determines the temperature readout from the output code and calibrates the reference voltage and temperature readout based on the output code.

Description

[0001] CROSS-REFERENCE TO RELATED APPLICATIONS

[0002] This application claims priority to U.S. Provisional Patent Application No. 63 / 273,651, filed October 29, 2021, and U.S. Non-Provisional Patent Application No. 17 / 965,282, filed October 13, 2022, the entire contents of which are incorporated herein by reference to the fullest extent permitted by law. Technical Field

[0003] This application relates to the field of temperature and voltage sensing circuits, and more specifically, to temperature and voltage sensing circuits utilizing an adjustable current source in the generation of a temperature-independent reference voltage, which is used to generate a highly accurate digital representation of the temperature value of an integrated circuit chip in which the temperature sensing circuit is housed. Background Technology

[0004] System-on-a-Chip (SoC) devices are used in mobile devices such as smartphones and tablets, as well as in a variety of embedded systems. Some current SoCs are capable of temperature-sensing task scheduling and self-calibration regarding temperature to help reduce power consumption. Temperature and voltage sensors are also used in image sensing applications to adjust the voltage and temperature offset of the image sensor during use. To enable this functionality, such SoCs include on-chip temperature sensors integrated with other components of the SoC.

[0005] A voltage proportional to the absolute temperature Vptat can be generated as the difference between the base-emitter junction voltages of two bipolar junction transistors biased at different current densities. Mathematically, this can be expressed as: Vptat = ΔVbe = Vbe1 - Vbe2. This voltage, proportional to the absolute temperature Vptat, is relatively error-free because the errors in Vbe1 and Vbe2 caused by the transistors' lack of ideal performance cancel each other out.

[0006] The relationship between Vptat and temperature can be mathematically expressed as Vptat = kT / q ln(p), where T is the temperature in Kelvin, k is the Boltzmann constant, q is the electron charge, and p is the ratio of the current density of the bipolar junction transistor used to generate Vptat. An analog-to-digital converter (ADC) digitizes Vptat relative to a reference voltage Vref and outputs a ratio μ as a result, which can be calculated as μ = Vptat / Vref. This ratio can be appropriately scaled to generate a digital temperature readout in desired units, for example: Temperature (°C) = A*μ + B, where A and B are constants.

[0007] To achieve temperature independence, the reference voltage Vref is typically generated as the sum of a voltage proportional to the absolute temperature Vptat and a voltage complementary to the absolute temperature Vctat, such as... Figure 1A As shown, under ideal conditions, it will produce a truly temperature-independent reference voltage. A voltage complementary to the absolute temperature Vctat is generated as the base-emitter junction voltage Vbe of the bipolar junction transistor.

[0008] Challenges arise when it is desired to update the generated digital temperature readout at high frequencies (e.g., every 10 μs) using known techniques. To sample Vref and ΔVbe at high frequencies, the sampling capacitors would need to be charged and discharged very rapidly. However, this is not feasible for conventional temperature sensors because common bipolar junction transistors (BJTs) typically used in thermal sensor applications (e.g., parasitic substrate PNP BJTs available in standard CMOS processes) cannot be biased with higher currents (greater than 2–3 μA per BJT) and are not suitable for use as thermal sensors. Increasing the number of BJTs (e.g., multiple BJTs connected in parallel) to increase the current they can be biased with eventually leads to transistor capacitance dominating and / or resulting in area / power losses, and this approach has only been feasible so far, with the desired operating speed likely remaining unattainable.

[0009] By using bipolar junction transistors biased at different multiples of the same current to obtain ΔVbe, one can attempt to increase the speed. Figure 1B The image shows a sample temperature sensor 1 utilizing this technology, in which it can be observed that the temperature sensor 1 includes a PNP bipolar junction transistor QP2, whose emitter receives current I, whose collector is grounded, and whose base is connected to the base of another PNP bipolar junction transistor QP1. The emitter of bipolar junction transistor QP1 receives a scaled current pI (e.g., current I scaled by a scaling factor p), its collector is grounded, and its base is connected to the base of bipolar junction transistor QP2. The voltage ΔVbe is the difference between the base-emitter voltages of QP1 and QP2. The slowest node in temperature sensor 1 is the emitter of QP2. If current I is insufficient to meet the desired speed, and it is desired to scale the current by a factor of 100, resulting in temperature sensor 1 being scaled by a factor of 100 (for the same accuracy), then it is desired to scale bipolar junction transistors QP1 and QP2 by a factor of 100 (e.g., by using 100 bipolar junction transistors QP1 and QP2 connected in parallel), such that the combined current increases from pI to (p+1)*100. However, as explained, using this design, the transistor's capacitance eventually becomes dominant and / or leads to losses in power consumption and area. Therefore, this approach is only feasible so far, and the desired operating speed may still remain unattainable.

[0010] Another approach to increasing speed is to obtain ΔVbe by using bipolar junction transistors biased at different multiples of the same current. Figure 1C An example of this temperature sensor 1' is shown, in which it can be observed that the temperature sensor 1' includes a PNP bipolar junction transistor QP1, whose emitter is connected to node N1, whose collector is grounded, and whose base is connected to its collector to generate a voltage complementary to the absolute temperature Vctat of its emitter-base junction terminals. The temperature sensor 1' further includes a first adjustable resistor Rptat1 connected between node N1 and the drain of a p-channel transistor MP1. The source of p-channel transistor MP1 is connected to a power supply voltage VDD, and the gate of p-channel transistor MP1 is connected to the gate of p-channel transistor MP2. The source of p-channel transistor MP2 is also connected to the power supply voltage VDD, and its drain is connected to a second adjustable resistor Rptat2. The second adjustable resistor Rptat2 is connected between the drain of MP2 and node N2, and a resistor R1 is connected between node N2 and the emitter of PNP bipolar junction transistor QP2. The collector of PNP transistor QP2 is connected to ground, and its base is connected to the base of PNP transistor QP1 and ground. Therefore, note that both PNP transistors QP1 and QP2 are diode-coupled. Amplifier 2 has a non-inverting terminal connected to node N2 and an inverting terminal connected to node N1, as well as an output terminal connected to the gates of p-channel transistors MP1 and MP2.

[0011] In operation, amplifier 2 drives the gates of transistors MP1 and MP2 by changing their gate voltages and final currents, forcing the voltage at the inverting input terminal of amplifier 2 to equal the voltage at the non-inverting input terminal. This causes the base-emitter voltage Vbe1 of PNP transistor QP1 (which is complementary to the absolute temperature Vctat) to appear at node N2. Since resistor R1 is between voltages Vbe1 and Vbe2 (the base-emitter voltage of transistor QP2), the voltage across resistor R1 is Vbe1 - Vbe2, which can be called ΔVbe. The resulting current Iptat flowing through resistor R1 is:

[0012]

[0013] The current Iptat is proportional to the absolute temperature (ignoring the temperature change in the resistivity of R1) and also flows into PNP transistors QP1 and QP2.

[0014] Since the voltage at node N1 is Vctat (the base-emitter voltage Vbe of transistor QP1), by applying an appropriate PTAT voltage above Vctat at node N1 and appropriately scaling resistors R1 and Rptat1, Rptat2, a reference voltage Vref can be obtained at the drain of transistor MP1. The reference voltage Vref can therefore be expressed as:

[0015]

[0016] Please note that ΔVbe is the voltage proportional to the absolute temperature Vptat (the temperature coefficients of R1 and Rptat1 / Rptat2 are canceled out in the Vref expression):

[0017] Vref=Vctat+Vptat

[0018] The accuracy of this temperature readout depends primarily on the temperature independence of the reference voltage Vref. However, due to the lack of ideal transistor performance, errors are introduced. Mathematically, the resulting real-world Vbe can be expressed as: Vbe = Vbe0 - λT + C(T), where Vbe0 is the value of Vbe at 0°K, λ is the decay slope of Vbe0 with temperature, and C(T) is a nonlinear quantity.

[0019] The slope λ is process-dependent, thus introducing inaccuracies into Vbe, and consequently into the generated reference voltage Vref. Figure 1D The sample range of Vbe values ​​generated by different values ​​of slope λ can be seen, and it can be noted that the inaccuracy of Vbe introduced by slope λ is linear.

[0020] use Figure 1C The design allows Iptat to be adjusted to target specific needs. Figure 1DThe Vbe range shown is used to compensate for Vref in an attempt to keep Vref as temperature-independent as possible, so that the aforementioned μ, and therefore the temperature readout, is as accurate as possible. This adjustment of Iptat can be performed by adjusting the resistance values ​​of the adjustable resistors Rptat1 / Rptat2 and R1. However, the accuracy of this adjustment is limited by the resistance value of the smallest of the adjustable resistors Rptat1 / Rptat2 and R1. High-rate switching (e.g., switching every 10µs) requires very high currents in both branches, resulting in very small resistance values. For example, to adequately adjust Iptat and make the temperature calculation accurate to within 1°C, the resistance value of the smallest of the adjustable resistors Rptat1 / Rptat2 and R1 can be as low as 5Ω; therefore, the associated switch needs to have a resistance smaller than this value, which leads to high area consumption. Furthermore, the challenge lies in the fact that the design has two variables—the resistance of R1 and the resistances of Rptat1 / Rptat2 (which are matched to each other)—and these resistances cannot be effectively changed independently, as one resistance will affect the other.

[0021] Further development is needed to overcome these accuracy limitations and to facilitate calibration. Summary of the Invention

[0022] This document discloses a temperature sensing circuit comprising: a current generation circuit configured to generate an initial current proportional to an absolute temperature; and a voltage generation circuit configured to mirror the initial current proportional to the absolute temperature using an adjustable current source to generate a scaled current, and supplying the scaled current to a first terminal of a resistor to generate an internal reference voltage at the first terminal, wherein a second terminal of the resistor has a voltage complementary to the absolute temperature applied to the second terminal. An analog-to-digital converter (ADC) has a reference input configured to receive the internal reference voltage, and a data input configured to selectively receive either a voltage complementary to the absolute temperature or an external source voltage. The ADC is configured to generate an output code indicating a ratio between: a) the voltage complementary to the absolute temperature or the external source voltage, and b) the internal reference voltage. Digital circuitry is configured to determine a temperature readout from the output code and calibrate the internal reference voltage and the temperature readout based on the output code.

[0023] The digital circuit calibrates the internal reference voltage by passing a known reference voltage as an external source voltage to the analog-to-digital converter, and adjusts an adjustable current source to modify the magnitude of the scaling current, thereby modifying the internal reference voltage according to the output code, until the internal reference voltage is equal to the known reference voltage or equal to a known percentage of the known reference voltage.

[0024] After the internal reference voltage is calibrated, the digital circuit calibrates the temperature readout determination by comparing the temperature readout based on the output code with the known temperature, and adjusts the constant used to determine the temperature readout until the temperature readout matches the known temperature.

[0025] The output code is calculated as μ = Vctat / Vref, where Vctat is the voltage complementary to the absolute temperature, and Vref is the internal reference voltage.

[0026] The temperature readout is defined as T = A × (1 - μ) - B, where T is the temperature, A and B are constants, where A is an adjusted constant, and μ is the output code.

[0027] The digital circuitry is further configured to determine the voltage value of an external or internal power supply voltage or any other voltage by passing the external or internal power supply voltage or any other voltage as an external source voltage to the analog-to-digital converter and determining the voltage value based on an internal reference voltage and an output code.

[0028] The current generation circuit may include: a first PNP transistor having an emitter coupled to a first node, a collector coupled to ground, and a base coupled to the collector of the first PNP transistor; a second PNP transistor having an emitter coupled to a second node via a first resistor, a collector coupled to ground, and a base coupled to the base of the first PNP transistor; a first p-channel transistor having a source coupled to a power supply voltage, a drain coupled to the first node, and a gate; and a second p-channel transistor having a source coupled to a power supply voltage, a drain coupled to the second node, and a gate coupled to the gate of the first p-channel transistor. Equivalence may be implemented between the drain currents of the first p-channel transistor and the second p-channel transistor. The first resistor may be coupled between the second node and the emitter of the second PNP transistor.

[0029] Equality can be achieved by an operational amplifier between the drain currents of the first p-channel transistor and the second p-channel transistor, the operational amplifier having a non-inverting terminal coupled to the second node, an inverting terminal coupled to the first node, and an output coupled to the gate of the first p-channel transistor and the second p-channel transistor.

[0030] The voltage generation circuit may include: an adjustable current source coupled between the power supply voltage and the third node; a resistor coupled between the third node and the fourth node; and a diode coupled to the PNP transistor, which generates a voltage complementary to the absolute temperature at the fourth node.

[0031] The adjustable current source can be an adjustable transistor device having a source coupled to a power supply voltage, a drain coupled to a third node, and a gate coupled to the gates of a first p-channel transistor and a second p-channel transistor.

[0032] The first switch may be coupled to selectively apply a voltage complementary to absolute temperature or an external source voltage to the data input of the analog-to-digital converter. The input circuit may include: a filter; a second switch that selectively applies an external supply voltage to the filter; a third switch that selectively provides a known reference voltage to the filter; and a fourth switch that selectively provides the output from the filter to the data input of the analog-to-digital converter. Attached Figure Description

[0033] Figure 1A It is a graph showing how a temperature-independent reference voltage (Vref) is generated in a temperature sensor by summing a voltage proportional to the absolute temperature (Vptat) and a voltage complementary to the absolute temperature (Vctat).

[0034] Figure 1B This is a schematic diagram of a first-in-the-art bandgap voltage generator.

[0035] Figure 1C This is a schematic diagram of a second prior art bandgap voltage generator.

[0036] Figure 1D It is a graph showing how the slope of the base-emitter junction voltage across the temperature of the bipolar junction transistor used to generate Vctat (also known as Vbe) depends on the process (which varies between transistors due to process variations).

[0037] Figure 2 This is a schematic diagram of the voltage and temperature sensors described in this article.

[0038] Figure 3A This shows the process when performing voltage (Vref) calibration. Figure 2 Voltage and temperature sensors.

[0039] Figure 3B This shows the process when performing temperature measurement calibration. Figure 2 Voltage and temperature sensors.

[0040] Figure 3C This shows when temperature sensing is performed. Figure 2 Voltage and temperature sensors.

[0041] Figure 3D This illustrates when performing power supply voltage sensing. Figure 2 Voltage and temperature sensors Detailed Implementation

[0042] The following disclosure enables those skilled in the art to make and use the subject matter disclosed herein. The general principles described herein can be applied to embodiments and applications other than those detailed above without departing from the spirit and scope of this disclosure. This disclosure is not intended to be limited to the embodiments shown, but is consistent with the widest scope of the principles and features disclosed or suggested herein. Note that in the following description, any described resistor or resistor is a discrete or integrated device unless otherwise stated, and is not simply an electrical lead between two points. Therefore, any described resistor or resistor coupled or connected between two points has a greater resistance than a lead or trace between those two points, and such a resistor or resistor should not be construed as a lead or trace. In other words, the resistor described herein is not a lead or trace.

[0043] Now refer to Figure 2 Temperature sensor 5 is described. This temperature sensor 5 is configured to be incorporated and integrated within a single integrated circuit chip, and operates to report the temperature of a region of that single integrated circuit chip.

[0044] The temperature sensor 5 includes the following circuitry: a current generation circuit 10 that generates a current proportional to the absolute temperature Iptat; a reference voltage generation circuit 15 that generates a reference voltage Vref from Iptat and outputs a voltage Vbe complementary to the absolute temperature; an analog-to-digital converter (ADC) 20 that receives the reference voltages Vref and Vbe from the voltage generation circuit 15 or receives an input voltage from the input circuit 16; and a digital circuit 25 that reads the output of the ADC 20 and controls the input circuit 16 and the voltage generation circuit 15 to achieve calibration and sensing.

[0045] First, the structure of each circuit will be described, and then the structure of temperature sensor 5 will be described.

[0046] More specifically, the current generation circuit 10 includes a first PNP transistor QP1, whose emitter is connected to node N1, whose collector is grounded, and whose base is connected to the base of a second PNP transistor QP2. A resistor R1 is connected between node N2 and the emitter of transistor QP2, and the collector of transistor QP2 is grounded. The source of the first p-channel transistor MP1 is connected to the power supply voltage VDD, its drain is connected to node N1, and its gate is connected to the gate of the second p-channel transistor MP2. The source of transistor MP2 is connected to VDD, and its drain is connected to node N2. The non-inverting terminal of operational amplifier 11 is connected to node N2, its inverting terminal is connected to node N1, and its output is connected to the gates of transistors MP1 and MP2.

[0047] Note that, compared to the prior art, the current generation circuit 10 may not have a resistor connected between node N1 and the drain of transistor MP1, and may not have a resistor connected between node N2 and the drain of transistor MP2—the scaling of Iptat is performed in circuit 15 to generate Vptat and thus generate Vref (which is used in voltage and temperature sensing and calibration).

[0048] The voltage generation circuit 15 includes an adjustable current source (shown as an adjustable third p-channel transistor MP3, representing multiple such transistors connected in parallel) connected between the power supply voltage VDD and node N3, with its gate connected to the gates of transistors MP1 and MP2. Transistors MP1, MP2, and MP3 are matched to improve the Iptat current mirror. In the case where the adjustable current source (also referred to as the current DAC) is multiple third p-channel transistors MP3 connected in parallel, the source of transistor MP3 is connected to VDD, its drain is connected to node N3, and its gate is connected to the gates of transistors MP1 and MP2 and the output of operational amplifier 11. Resistor R2 is connected between nodes N3 and N4. The emitter of the third PNP transistor QP3 is connected to node N4, its collector is connected to ground, and it is diode-coupled, so its base is connected to its collector.

[0049] Analog-to-digital converter (ADC) 20 derives its reference voltage from node N3 (with or without voltage buffer), its input is connected to node N5, and its output bit ADCOUT is connected to digital circuitry 25. Switch S1 selectively connects node N4 to node N5. Through switches S1 to S4, Vbe, Vsupply, and Vref_Ext (after filtering) can be provided through different channels of the ADC and as different inputs (without loss of generality), or through a single channel using multiplexing (multiplexing techniques) outside the ADC.

[0050] Input circuit 16 includes filter circuit 17, which has a first input selectively connected by switch S2 to receive a supply voltage for testing Vsupply, a second input selectively connected by switch S3 to receive an external reference voltage Vref_ext, and an output selectively connected by switch S4 to node N5. Filter circuit 17 can be used to remove noise from the voltage signals Vsupply and Vref_ext, and also appropriately scale these voltages to bring them within the operating range of ADC 20.

[0051] Digital circuit 25 includes a sequencer and a thermal sensor controller 26, which primarily receives a clock signal Fclk used by digital circuit 25 and receives data from data formatter 27. Data formatter 27 has inputs that receive the output ADCOUT of ADC 20, a first value A, and a second value B, and outputs that provide DATAREADY and DATAOUT signals. DATA_READY indicates that data formatter 27 is ready to output another data word, and DATA_OUT is the data word being output (e.g., a digital representation of voltage or temperature sensing / calibration data).

[0052] In operation, operational amplifier 11 drives the gates of transistors MP1 and MP2 to force the voltage at the inverting input of amplifier 11 to be equal to the voltage at the non-inverting input of amplifier 11, and thus forces the gate-to-drain voltages of transistors MP1 and MP2 to be equal. This results in the base-emitter voltage Vbe1 of PNP transistor QP1 (which is a voltage complementary to the absolute temperature Vctat) appearing at the drain of transistor MP2. Those skilled in the art will understand that operational amplifier 11 can have a finite offset voltage, which can manifest as an error between its input terminals. Therefore, it may be necessary to chop operational amplifier 11 to average the inaccuracies generated by these offset voltages. Since resistor R1 is between voltages Vbe1 and Vbe2 (the base-emitter voltage of transistor QP2), the voltage across resistor R1 is Vbe1 - Vbe2, which can be referred to as ΔVbe. The resulting current Iptat0 applied through resistor R1 is proportional to the absolute temperature and flows into PNP transistors QP2 and QP1.

[0053] The current Iptat0 can be expressed as:

[0054]

[0055] The voltage at node N1 is Vctat (the base-emitter voltage Vbe of transistor QP1). Since the adjustable current source MP3 is connected to transistors MP1 and MP2 as a current mirror, the PTAT current Iptat (which is a multiple of Iptat0, the multiple depending on how many parallel-connected transistors MP3 are activated) originates from the drain of the adjustable current source MP3 and flows through resistor R2, thereby generating a reference voltage Vref at node N3. The reference voltage Vref can be expressed as:

[0056]

[0057] like Figure 3AAs shown, temperature sensor 5 first operates in the voltage calibration phase. During the calibration phase, switches S3 and S4 are closed while switches S1 and S2 are open, causing ADC 20 to receive Vref from I0 at its input as its reference voltage and Vref_ext. The value of Vref_ext is selected within the expected input range of the ADC. The goal is to increase the scaling factor of the conducting transistor MP3 so that the resulting reference voltage Vref from block 15 equals the bandgap reference voltage. The digital word ADCOUT output by ADC 20 represents the ratio between Vref_ext and Vref. Based on this digital word ADCOUT, digital circuitry 25 adjusts the adjustable current source MP3 (i.e., activates more parallel-connected transistors MP3 to increase the amplitude of Iptat, or activates fewer parallel-connected transistors MP3 to decrease the amplitude of Iptat, and thus causes a corresponding change in Vref). Specifically, when ADCOUT digitally represents a value greater than the expected ratio Vref_ext / Vref (e.g., 0.5), the adjustable current source MP3 is adjusted to increase the magnitude of Iptat, thereby increasing Vref and decreasing the Vref_Ext / Vref ratio. When ADCOUT digitally represents a value less than the expected ratio Vref_ext / Vref, the adjustable current source MP3 is adjusted to decrease the magnitude of Iptat, thereby decreasing Vref and increasing the Vref_Ext / Vref ratio. Temperature sensor 5 continues to operate during this voltage calibration, meaning that ADC 20 again periodically samples its input to periodically generate new ADCOUT values, and one adjustment of the adjustable current source MP3 is performed by digital circuitry 25 for each new ADCOUT value.

[0058] This adjustment is performed until the ratio Vref_ext / Vref, expressed digitally by ADCOUT, reaches its maximum possible accuracy. This completes the calibration of the reference voltage Vref (correcting the Vbe range; correcting the Vbe slope and scaling error or scaling ratio error between the current generation circuit 10 and the voltage generation circuit 15), and the calibrated version of the internal reference Vref can be referred to as Vref_Cal. The number of branches of the resulting MP3 (or voltage calibration code VCALCODE) can be stored in a one-time programmable (OTP) memory or any other memory, so that it can be used to reproduce Vref_Cal whenever the silicon die in which the temperature sensor 5 is incorporated is needed.

[0059] Temperature sensor 5 then operates during the temperature calibration phase, such as... Figure 3BAs shown. During the temperature calibration phase, switch S1 is closed while the other switches are open. Therefore, ADC 20 receives the calibrated reference voltages Vref_Cal and Vbe at its input, and the digital word ADCOUT output by ADC 20 represents the ratio between Vbe and Vref_Cal. Remembering that Vbe = Vctat and calculating μ = Vctat / Vref_Cal here, the temperature can be calculated by digital circuit 25 as:

[0060] T = A × (1 - μ) - B

[0061] In this implementation, A and B are constants, and the initial, standard, pre-known values ​​are used for both A and B. To perform temperature calibration, digital circuitry 25 compares the calculated temperature T with an input specifying an actual known temperature, and adjusts constant A based on a mathematical comparison between the calculated temperature T and the known die temperature (in thermal equilibrium with the surrounding area) undergoing temperature calibration. Specifically, A is adjusted so that the calculated temperature T matches the die temperature accurately measured externally. B is set according to the unit used to report the temperature (e.g., Celsius, Kelvin) and also corrects for any offset errors.

[0062] The temperature calibration can be performed in a chamber with a known temperature. For example, a chip with an integrated temperature sensor 5 can be placed in a chamber maintained at 27°C (and left there for a sufficient time to achieve thermal equilibrium with its surroundings), which is a known temperature, and the temperature calibration can be performed.

[0063] According to a non-limiting example, if the calculated temperature T is less than the known temperature, A is increased and the calculated temperature T is recalculated. If the calculated temperature T is greater than the known temperature, A is decreased and the calculated temperature T is recalculated. After each recalculation, the comparison between the calculated temperature T and the known temperature is performed again, and the constant A is adjusted again, and this continues until the calculated temperature T matches the known temperature to a sufficient accuracy of A. This completes the calibration of the temperature calculation and sensing, and the calibrated version of A can be called A_Calcode and can be stored in permanent memory for accurate temperature readout of the silicon die in which the temperature sensor 5 is incorporated.

[0064] Temperature sensing in normal operating mode can now be performed after Vref calibration and temperature calibration to find the chip's current temperature. Now refer to... Figure 3C Describes temperature sensing in this mode, where switch S1 is closed and the other switches are open.

[0065] ADC 20 receives Vref_Cal as its reference voltage and Vbe at its input, and the digital word ADCOUT output by ADC 20 represents the ratio between Vbe and Vref_Cal. Remembering that Vbe = Vctat and μ = Vctat / Vref_Cal, the temperature can be calculated by digital circuit 25 as:

[0066] T = A_calcode × (1-μ) - B

[0067] Therefore, temperature sensor 5 works normally here, and temperature sensing can continue for the expected time.

[0068] In some applications, it may be desirable for the temperature sensor 5 to have the additional capability of testing voltage so that the current absolute value of that voltage can be determined. In the example shown, the supply voltage Vsupply is the sensed voltage.

[0069] In this voltage sensing mode, switches S2 and S4 are closed, while switches S1 and S3 are open. As a result, Vsupply is provided as input to ADC 20, and similarly, ADCOUT represents the ratio between Vsupply and Vref_cal. Since vref_cal is a known value from ADCOUT, the value of Vsupply can be calculated.

[0070] The design of temperature sensor 5 offers numerous advantages. For example, the output impedance of voltage generation circuit 15 can be scaled as needed to suit ADC 20 and the desired sampling rate without compromising accuracy. Unlike prior art designs, this scaling does not involve a corresponding 8-fold scaling of the parallel current consumed by the bipolar junction transistor or the area. Furthermore, since calibration is performed in the current domain, there is no issue of dealing with small calibration resistor sizes. As understood, in addition to scaling the Iptat current generated by the adjustable current source MP3, the sizes of resistor R2 and transistor QP3 are also scaled to produce the correct reference voltage Vref and CTAT voltage Vbe. Therefore, overall, temperature sensor 5 allows for a significantly higher temperature conversion rate than prior art designs.

[0071] As another advantage, due to the scalability of the circuit, the buffering of the reference voltage Vref used in the ADC can be eliminated, and in fact, Vref buffering is not performed in the temperature sensor 5, resulting in area and power savings without loss of accuracy.

[0072] Finally, it is obvious that modifications and variations may be made to the content described and shown herein without departing from the scope of this disclosure as defined by the appended claims.

[0073] Although the invention has been described with respect to a limited number of embodiments, those skilled in the art to which this invention pertains will understand that other embodiments can be conceived without departing from the scope of the invention disclosed herein. Therefore, the scope of this disclosure will be limited only by the appended claims.

Claims

1. A temperature sensing circuit, comprising: A current generating circuit configured to generate an initial current proportional to the absolute temperature; A voltage generation circuit is configured to use an adjustable current source to mirror the initial current proportional to the absolute temperature to generate a scaled current, and to supply the scaled current to a first terminal of a resistor to generate an internal reference voltage at the first terminal, wherein a second terminal of the resistor has a voltage complementary to the absolute temperature applied to the second terminal. An analog-to-digital converter having a reference input configured to receive the internal reference voltage and a data input configured to selectively receive either the voltage complementary to an absolute temperature or an external source voltage, wherein the analog-to-digital converter is configured to generate an output code indicating a ratio between: a) the voltage complementary to an absolute temperature or the external source voltage, and b) the internal reference voltage; as well as A digital circuit configured to determine a temperature readout from the output code and to calibrate the internal reference voltage and the temperature readout based on the output code.

2. The temperature sensing circuit of claim 1, wherein the digital circuit calibrates the internal reference voltage by means of: The known reference voltage is passed to the analog-to-digital converter as the external source voltage; and The adjustable current source is adjusted to modify the magnitude of the scaling current, and the internal reference voltage is modified according to the output code until the internal reference voltage is equal to the known reference voltage or equal to a known percentage of the known reference voltage.

3. The temperature sensing circuit of claim 2, wherein the digital circuit calibrates the temperature readout determination after calibration of the internal reference voltage by: The temperature readout based on the output code will be compared with a known temperature; and Adjust the constant used to determine the temperature readout until the temperature readout matches the known temperature.

4. The temperature sensing circuit according to claim 3, wherein the output code is calculated as: μ = Vctat / Vref Where Vctat is the voltage complementary to the absolute temperature, and Vref is the internal reference voltage.

5. The temperature sensing circuit according to claim 4, wherein the temperature readout is determined as: T = A × (1 - μ) - B Where T is the temperature, A and B are constants, where A is the adjusted constant, and μ is the output code.

6. The temperature sensing circuit of claim 3, wherein the digital circuit is further configured to determine the voltage value by passing an external power supply voltage or an internal power supply voltage or any other voltage as the external source voltage to the analog-to-digital converter, wherein the voltage value is determined based on the internal reference voltage and the output code.

7. The temperature sensing circuit according to claim 3, wherein the current generating circuit comprises: A first PNP transistor has an emitter coupled to a first node, a collector coupled to ground, and a base coupled to the collector of the first PNP transistor. The second PNP transistor has an emitter coupled to a second node via a first resistor, a collector coupled to ground, and a base coupled to the base of the first PNP transistor. A first p-channel transistor has a source coupled to a power supply voltage, a drain coupled to the first node, and a gate. as well as The second p-channel transistor has a source coupled to the power supply voltage, a drain coupled to the second node, and a gate coupled to the gate of the first p-channel transistor. The equality is implemented between the drain currents of the first p-channel transistor and the second p-channel transistor; and The first resistor is coupled between the second node and the emitter of the second PNP transistor.

8. The temperature sensing circuit of claim 7, wherein the equivalence is implemented by an operational amplifier between the drain currents of the first p-channel transistor and the second p-channel transistor, the operational amplifier having a non-inverting terminal coupled to the second node, an inverting terminal coupled to the first node, and an output coupled to the gates of the first p-channel transistor and the second p-channel transistor.

9. The temperature sensing circuit according to claim 7, wherein the voltage generation circuit comprises: The adjustable current source is coupled between the power supply voltage and the third node; The resistor coupled between the third node and the fourth node; as well as A diode-coupled PNP transistor generates the voltage that is complementary to the absolute temperature at the fourth node.

10. The temperature sensing circuit of claim 9, wherein the adjustable current source comprises an adjustable transistor device having a source coupled to the power supply voltage, a drain coupled to the third node, and a gate coupled to the gates of the first p-channel transistor and the second p-channel transistor.

11. The temperature sensing circuit of claim 7, further comprising a first switch coupled to selectively apply the voltage complementary to the absolute temperature or the external source voltage to the data input of the analog-to-digital converter; further comprising an input circuit including: filter; A second switch selectively applies an external power supply voltage to the filter; A third switch selectively supplies the known reference voltage to the filter; as well as A fourth switch selectively provides the output from the filter to the data input of the analog-to-digital converter.

12. A temperature sensing method, comprising: Use an adjustable current source to generate scaled current; An internal reference voltage is generated from the scaling current, which is proportional to the absolute temperature. The internal reference voltage and the voltage complementary to the absolute temperature or the external source voltage are digitized to generate an output code representing the ratio between: a) the voltage complementary to the absolute temperature or the external source voltage, and b) the internal reference voltage; The temperature readout is determined from the output code; as well as The internal reference voltage and temperature readout are determined based on the output code.

13. The method of claim 12, wherein the internal reference voltage is calibrated by using a known reference voltage as the external source voltage, and the method further comprises adjusting the adjustable current source to modify the magnitude of the scaling current, thereby modifying the internal reference voltage according to the output code until the internal reference voltage is equal to the known reference voltage.

14. The method of claim 13, wherein after the internal reference voltage is calibrated, the temperature readout is determined to be calibrated by: The temperature read from the output code is compared with a known temperature; and Adjust the constant used to determine the temperature readout until the temperature readout matches the known temperature.

15. The method of claim 14, further comprising: The voltage value of the external power supply voltage is determined by using the external power supply voltage as the external source voltage; as well as The voltage value is determined based on the internal reference voltage and the output code.

16. The method of claim 15, wherein the output code is calculated as: μ = Vctat / Vref Where Vctat is the voltage complementary to the absolute temperature, and Vref is the internal reference voltage.

17. The method of claim 16, wherein the temperature readout is determined as: T = A × (1 - μ) - B Where T is the temperature, A and B are constants, where A is the adjusted constant, and μ is the output code.

18. A temperature sensing circuit, comprising: Voltage generation circuit, the voltage generation circuit being configured to: Generate a voltage that is complementary to the absolute temperature; as well as An adjustable current proportional to the absolute temperature is supplied to a first terminal of a resistor to generate an internal reference voltage at the first terminal, wherein a second terminal of the resistor receives the voltage complementary to the absolute temperature. An analog-to-digital converter having a reference input configured to receive the internal reference voltage and a data input configured to selectively receive either the voltage complementary to an absolute temperature or an external source voltage, wherein the analog-to-digital converter is configured to generate an output code indicating a ratio between: a) the voltage complementary to an absolute temperature or the external source voltage, and b) the internal reference voltage; as well as A digital circuit configured to determine a temperature readout from the output code and to calibrate the internal reference voltage and the temperature readout based on the output code.

19. The temperature sensing circuit of claim 18, wherein the digital circuit calibrates the internal reference voltage by: The known reference voltage is passed to the analog-to-digital converter as the external source voltage; and The magnitude of the current, which is proportional to the absolute temperature, is modified, and then the internal reference voltage is modified according to the output code until the internal reference voltage is equal to the known reference voltage.

20. The temperature sensing circuit of claim 19, wherein the digital circuit determines the temperature readout by calibrating the internal reference voltage after calibration by: The temperature read from the output code is compared with a known temperature; and Adjust the constant used to determine the temperature readout until the temperature readout matches the known temperature.

21. The temperature sensing circuit according to claim 20, wherein the output code is calculated as: μ = Vctat / Vref Where Vctat is the voltage complementary to the absolute temperature, and Vref is the internal reference voltage.

22. The temperature sensing circuit of claim 21, wherein the temperature readout is determined as: T = A × (1 - μ) - B Where T is the temperature, A and B are constants, where A is the adjusted constant, and μ is the output code.

23. The temperature sensing circuit of claim 20, wherein the digital circuit is further configured to determine the voltage value of an external power supply voltage by: passing the external power supply voltage as the external source voltage to the analog-to-digital converter, and determining the voltage value based on the internal reference voltage and the output code.

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