A switching conversion system with multi-mode constant current control

Abstract

The present invention discloses a switching conversion system with multi-mode constant current control, which is characterized in that it includes a bandgap reference circuit unit, a resistance measurement circuit unit, a switch drive circuit, an over-temperature protection circuit unit, a switch delay compensation circuit, a constant current error amplification circuit, a constant current operation circuit, a zero-crossing detection circuit, a current sampling comparison circuit and an ARM main control unit; the bandgap reference circuit unit includes a start-up circuit, a bandgap reference core circuit and an under-voltage latch circuit; the present invention realizes a stable current output of 1.5 A for the primary-side feedback constant current drive circuit without an auxiliary winding under an input voltage of 85-220 Vac and a maximum load of 12 V. The linear regulation rate of the circuit is lower than ±2%, the load regulation rate is lower than ±1.5%, and the conversion efficiency in different working environments is higher than 85%. By detecting the magnitude of the secondary current and adjusting the drive voltage V DRI according to the detected secondary current, a fast turn-off synchronous rectification drive circuit in multiple modes is realized.

Description

Technical Field

[0001] The present invention relates to the technical field of constant current control, and in particular to a switching conversion system with multi-mode constant current control. Background Art

[0002] In the current environment where electronic technology is becoming increasingly mature, switching power supply management ICs have a large number of applications in many fields and have become a major research hotspot. As a commonly used component of switching power supplies, flyback converters are often seen because of their relatively simple structure, which not only has high output efficiency and low manufacturing cost, but also greatly reduces the interaction between input and output due to the presence of transformers. Especially for mobile and portable devices with increasing applications, in order to achieve high power conversion efficiency, almost each of them is equipped with at least one adapter. In this case, flyback converters have obtained great development.

[0003] Flyback converters are mainly divided into two categories: secondary side feedback and primary side regulation (PSR). Comparing the two, although the former can obtain relatively accurate voltage and current outputs, its constant current and constant voltage control effects are poor under high temperature conditions. Constant current control also has extensive applications in LED driver power supplies. Compared with constant voltage control, its output current is stable and not affected by input voltage fluctuations and environmental interference.

[0004] Therefore, it is necessary to provide a switching conversion system with multi-mode constant current control to solve the above technical problems. Summary of the Invention

[0005] The purpose of this part is to outline some aspects of the embodiments of the present invention and briefly introduce some preferred embodiments. Some simplifications or omissions may be made in this part, as well as in the abstract and title of the present application, to avoid obscuring the purpose of this part, the abstract, and the title, but such simplifications or omissions shall not be used to limit the scope of the present invention.

[0006] To solve the above technical problems, the present invention provides the following technical solution. A switching conversion system with multi-mode constant current control, characterized in that it includes a bandgap reference circuit unit, a resistance measurement circuit unit, a switch drive circuit, an over-temperature protection circuit unit, a switch delay compensation circuit, a constant current error amplification circuit, a constant current operation circuit, a zero-crossing detection circuit, a current sampling comparison circuit, and an ARM main control unit; the bandgap reference circuit unit includes a startup circuit, a bandgap reference core circuit, and an under-voltage latch circuit; the purpose of the startup circuit is to ensure that the bandgap reference can break away from the "degeneracy point" during startup. When the bandgap reference starts up normally, the triode turns off and the startup circuit will automatically become invalid; when the system supply voltage is low, the under-voltage latch circuit will cause errors in the comparator and some logic circuits. Only when the input voltage meets the requirements for normal operation of the circuit, will it give an enable signal to control the normal operation of each module circuit of the control system circuit; the resistance measurement circuit unit includes a matrix switch, a four-wire resistance test circuit, a resistance measurement circuit, a filter amplification circuit, and a 24-bit ADC sampling circuit; the input end of the matrix switch is connected to the output of the current monitoring circuit in the constant current output circuit unit; the ARM main control unit stores computer programs, PID control, PWM constant current control, and PFM constant current control.

[0007] As a preferred embodiment of the switching conversion system with multi-mode constant current control according to the present invention, the input end of the switch drive circuit is connected to the output end of the ARM main control unit, and the output end of the switch drive circuit is connected to the input end of the matrix switch.

[0008] As a preferred embodiment of the switching conversion system with multi-mode constant current control according to the present invention, the constant current operation circuit includes an output current estimation Vio generation circuit.

[0009] As a preferred embodiment of the switching conversion system with multi-mode constant current control according to the present invention, the over-temperature protection circuit unit can further improve the reliability of the OTP module by setting hysteresis, and avoid system instability caused by continuous temperature fluctuations at the threshold point.

[0010] As a preferred embodiment of the switching conversion system with multi-mode constant current control according to the present invention, the resistance measurement circuit unit further includes an external metering interface connected to the matrix switch, which is used to connect external metering devices for constant current output and resistance measurement accuracy metering.

[0011] As a preferred embodiment of the switching conversion system with multi-mode constant current control according to the present invention, the switch delay compensation circuit is used to superimpose with the primary current on the CS sampling resistor, and the superimposed current acts on the sampling resistor as the positive input of the comparator. Its purpose is to enable the comparator to make a flip in advance before the primary peak current arrives to compensate for the total turn-off delay Δt inside the circuit.

[0012] As a preferred embodiment of the switching conversion system with multi-mode constant current control according to the present invention, the zero-crossing detection circuit correspondingly outputs a voltage monitoring and protection circuit, which can take protection measures in time when short circuit or open circuit occurs at the output end, enhancing the safety and reliability of the system.

[0013] As a preferred embodiment of the switching conversion system with multi-mode constant current control according to the present invention, the current sampling and comparison circuit includes a leading edge blanking (LEB) circuit, a switching delay compensation circuit and a current sampling and comparison circuit.

[0014] Advantages of the present invention: The system is a constant current modulation implementation scheme integrating median sampling, and at the same time adopts a structure of cascading two power transistors to replace the auxiliary winding to complete the sampling of the end point of the demagnetization time, further reducing the system energy consumption. The control circuit part integrates the input cable compensation technology, improving the constant current accuracy under different input line voltages. In addition, the circuit also integrates protection circuits such as over-temperature protection and primary side short circuit protection, effectively improving the reliability and safety of the system. The reference voltage is dynamically adjusted by the self-regulating loop to generate the LED drive voltage, realizing the minimum saturation voltage drop of the constant current circuit; and the self-regulating loop realizes the precise dynamic adjustment of the reference voltage through current charging and discharging, without additional filter capacitors, and can realize single-chip integration, reducing the system cost and complexity. BRIEF DESCRIPTION OF THE DRAWINGS

[0015] In order to more clearly illustrate the technical solutions of the embodiments of the present invention, the drawings required for the description of the embodiments will be briefly introduced below. Obviously, the drawings in the following description are only some embodiments of the present invention, and those of ordinary skill in the art can obtain other drawings without creative efforts based on these drawings.

[0016] Among them:

[0017] Figure 1 Schematic diagram of the ordinary PWM control mode of the switching conversion system with multi-mode constant current control provided by an embodiment of the present invention;

[0018] Figure 2 Schematic diagram of the PWM regulation mode with input voltage detection of the switching conversion system with multi-mode constant current control provided by an embodiment of the present invention;

[0019] Figure 3 Schematic circuit diagram of the PFM constant current control mode of the switching conversion system with multi-mode constant current control provided by an embodiment of the present invention;

[0020] Figure 4 General structure diagram of the bandgap reference of the switching conversion system with multi-mode constant current control provided by an embodiment of the present invention;

[0021] Figure 5 The bandgap reference circuit diagram of the switching conversion system with multi-mode constant current control according to an embodiment provided by the present invention;

[0022] Figure 6 The simulation curve diagram of the temperature coefficient of the bandgap reference of the switching conversion system with multi-mode constant current control according to an embodiment provided by the present invention;

[0023] Figure 7 The undervoltage latch circuit diagram of the switching conversion system with multi-mode constant current control according to an embodiment provided by the present invention;

[0024] Figure 8 The overtemperature protection circuit diagram of the switching conversion system with multi-mode constant current control according to an embodiment provided by the present invention. Detailed implementation manners

[0025] In order to make the above objects, features, and advantages of the present invention more obvious and understandable, the following will describe the detailed implementation manners of the present invention with reference to the accompanying drawings of the specification.

[0026] Many specific details are set forth in the following description in order to fully understand the present invention. However, the present invention may also be implemented in other ways different from those described herein. Those skilled in the art can make similar extensions without departing from the connotation of the present invention. Therefore, the present invention is not limited by the specific embodiments disclosed below.

[0027] Embodiment 1

[0028] Refer to Figure 1-8, the first embodiment of the present invention, a switching conversion system with multi-mode constant current control, characterized in that it includes a bandgap reference circuit unit, a resistance measurement circuit unit, a switch drive circuit, an over-temperature protection circuit unit, a switch delay compensation circuit, a constant current error amplification circuit, a constant current operation circuit, a zero-crossing detection circuit, a current sampling comparison circuit, and an ARM main control unit; the bandgap reference circuit unit includes a start-up circuit, a bandgap reference core circuit, and an under-voltage latch circuit; the purpose of the start-up circuit is to ensure that the bandgap reference can break away from the "degenerate point" during startup. When the bandgap reference starts up normally, the triode turns off, and the start-up circuit will automatically become invalid; when the system supply voltage is low, the under-voltage latch circuit will cause errors in comparators and some logic circuits. Only when the input voltage meets the requirements for normal circuit operation, will it give an enable signal to control the normal operation of each module circuit of the control system circuit; the resistance measurement circuit unit includes a matrix switch, a four-wire resistance test circuit, a resistance measurement circuit, a filter amplification circuit, and a 24-bit ADC sampling circuit; the input end of the matrix switch is connected to the output of the current monitoring circuit in the constant current output circuit unit; the ARM main control unit stores computer programs, PID control, PWM constant current control, and PFM constant current control. Specifically, the principle is the same as that of the traditional control method. The constant current control method with primary side feedback is divided into two modes: pulse width modulation PWM and pulse frequency modulation PFM. The following will detail these two control methods. PWM constant current control. This control method calculates the value of the primary side peak current through known condition data, then compares this value with the primary side current sampling value through a comparator to generate a proportional time signal. Finally, the signal is transmitted to the PWM controller to generate a signal for controlling the power switch, achieving constant current control.

[0029]

[0030]

[0031] In the above control process, the demagnetization time T of the secondary side is obtained by detecting n - 1 cycles d and the switching period T s , and the value of Vpref is calculated in n - 1 cycles. Then it is used to control the nth cycle. Then, there is a delay time in this method, so the control delay will definitely affect the accuracy of the output current. As Figure 1 described.

[0032] As Figure 2 shown, it is the principle circuit of the PWM control method with input voltage detection. The duty cycle signal for controlling the PWM controller can be obtained through known conditions:

[0033]

[0034] In both cases of comparison, there is a time delay, and the output results will have errors. Moreover, when the input conditions change, as can be seen from the previous formula, it is necessary to measure this to meet the known conditions of the formula, which undoubtedly makes the circuit structure more cumbersome and may even reduce the system efficiency.

[0035] PFM constant current control, calculates the output voltage V0 through the information of the auxiliary winding sampling terminal

[0036] Obtains the demagnetization time, and then processes the input information through the OSC module to obtain the system period T s . After comparing it with the primary side sampling information, a signal for controlling the switching tube is generated through an RS flip-flop. The I0 can be obtained from the following formula:

[0037]

[0038] L p Is the inductance value of the primary side of the transformer. The constant current idea of this constant current modulation method is: when the value of I p，pk is known, to ensure that the output current I0 remains constant, the value of T s ·V o must be constant.

[0039] In the PFM constant current control mode, due to the instability of the primary side inductance value L P of the transformer itself, the measurement difficulty is relatively large, and the results obtained will have errors, so a compensation circuit is needed for error compensation. As Figure 3 shown.

[0040] By comparing the above two control methods, it can be seen that in the PFM control scheme, the deviation of the primary inductance directly affects the output accuracy, and a compensation circuit is needed to compensate it. In the PWM control scheme, in the current mode, the duty cycle of the switch is controlled by the change of the peak current threshold: there is an inverse proportional relationship between the output voltage and the threshold. The switching converter in the PWM mode has the advantages of high response and high output accuracy. In this mode, the efficiency is relatively high during overload, but when the load decreases, the duty cycle will gradually decrease; when the load is light, as the frequency increases, the power loss of the power switch tube increases, thereby reducing the light load efficiency.

[0041] The PWM control method can achieve constant current and has the characteristics of high full-load efficiency and low noise. Constant current control is a method of controlling the opening time of the output pulse by comparing and amplifying the current sampling signal and the error amplification signal. The constant current control mode can obtain the voltage across the switching tube from the resistor and can achieve current protection. The current feedback method has the advantages of high voltage regulation rate, fast dynamic response, and stable output.

[0042] PID control is also known as proportional-integral-derivative control. Its principle is simple, its working performance is stable, and it is easy to control, making it the most commonly used controller in engineering practice. The flyback circuit can be divided into two operating states, namely CCM and DCM. When the circuit is in the DCM operating mode, problems are not likely to occur during the stable operating state of the circuit. Therefore, relatively simple PI regulation can be used to solve these problems and obtain the desired control effect. PI control is carried out based on the operating error of the circuit. The required control quantity is calculated according to this error, so as to make the system achieve a stable effect. Among them, the larger the integral regulation output and the proportional regulation output, the larger the corresponding error quantity will be, and vice versa. Compared with the two, the effect of integral regulation is better.

[0043] The overall structure of PI regulation. In this figure, one end of the operational amplifier is the error signal, and the input V c (t) is the regulated value. The relationship between the error quantity and the regulated quantity is as follows:

[0044]

[0045] The above formula contains many arithmetic terms and is relatively complex to calculate. It is necessary to simplify the formula, that is, sample the integral term with data to make it discrete. After simplification, we get:

[0046]

[0047] where T is the sampling period

[0048]

[0049]

[0050] If we use these two formulas to present the relationship of PI regulation, we need a lot of cumbersome calculations. Therefore, we combine the two formulas to get:

[0051] V c (n) = V c (n - 1) + K p (e(n) - e(n - 1)) + K iT e(n)

[0052] where K P 、K i 、K iT The values are as follows:

[0053]

[0054]

[0055] K iT = K iInternal structure of the PI module drawn according to the PI increment expression.

[0056] Embodiment 2

[0057] Refer to Figure 1-8 , the second embodiment of the present invention. In the design of analog circuits, various stable voltage references and current references are often essential. Voltage references are generally used for power supply of each digital - analog module in the circuit and the comparison reference potential of each comparator. Current references are used for current biasing of various amplifiers and current mirrors, etc. An absolute temperature - independent voltage reference is usually very difficult to directly implement through a circuit. However, it can be assumed that there are two voltages V1 and V2 with opposite temperature coefficients, where V1 is positively correlated with temperature and V2 is negatively correlated with temperature. Then, by selecting appropriate weighting parameters α1 and α2 and adding them together, finally the temperature coefficients of the two just cancel each other out. Then the finally superimposed V ref = α1V1 + α2V2 is the required voltage reference with zero temperature coefficient. For this reason, circuit designers have proposed the concept of a bandgap voltage reference (BGR). As Figure 4 shown in the general structure of the bandgap voltage reference, it is obtained by superimposing a positive - temperature - coefficient voltage and a negative - temperature - coefficient voltage, so that the voltage remains basically constant over a large temperature change range.

[0058] (1) Negative - temperature - coefficient voltage

[0059] The base - emitter voltage of a bipolar transistor, or more popularly, the forward voltage of a pn diode, has a negative temperature coefficient. For the forward - conduction stage of a bipolar device, its collector current can be written as:

[0060] I C = I S exp(V BE / VT) (1.1)

[0061] where

[0062] V T = kT / q (1.2)

[0063] The saturation current I s is proportional to u is the mobility of minority carriers, n i is the intrinsic carrier concentration of silicon. Among them, the relationship between the minority - carrier concentration u and temperature can be expressed as u ∝ u0T m , m ≈ - 3 / 2, and at the same time, the relationship between n i and temperature can be expressed as n i2 ∝T 3 exp[-E g / (kT)], where E g ≈1.12eV, is the bandgap energy of silicon.

[0064] Furthermore, the expression for the saturation current I S can be obtained as follows:

[0065]

[0066] where b is the proportionality coefficient, and at the same time, Equation (1.1) can be rewritten as

[0067] V BE = V T exp(I C / I S ) (1.4)

[0068] Assuming that the value of I C remains unchanged, the temperature coefficient of the base-emitter voltage of the bipolar transistor can be expressed as:

[0069]

[0070] According to Equation (1.3), we have

[0071]

[0072] Finally, by combining Equations (1.5) and (1.6), the temperature coefficient of VBE at a given temperature T can be obtained as

[0073]

[0074] It can be seen from this that it is related to the magnitude of its own value of V BE itself. When V BE ≈750mV and T = 300°K,

[0075] Therefore, the V BE obtained here is a voltage with a negative temperature coefficient.

[0076] (2) Positive temperature coefficient voltage

[0077] When two bipolar transistors operate at different current densities, the difference in their base-emitter voltages, ΔV BE is a voltage value with a positive temperature coefficient. Assuming that two identical transistors (Is1 = Is2) are biased with collector currents of I0 and nI0 respectively, it can be calculated that, neglecting their base currents:

[0078] ΔV BE=V BE1 - VBE2 =V T ln n (1.8)

[0079] Its difference ΔV BE has a temperature coefficient of:

[0080]

[0081] It can be seen that the voltage ΔV here BE exhibits a positive temperature coefficient. And this temperature coefficient is independent of the temperature itself or the collector current.

[0082] (3) Bandgap reference circuit

[0083] According to the above description, by superimposing the obtained positive and negative temperature coefficient voltages, a well-behaved zero-temperature coefficient voltage reference V REF =α1V BE +α2(V T lnn). At room temperature Let α1 = 1. According to it can be calculated that α2lnn≈17.2. At this time, the zero-temperature coefficient voltage reference is:

[0084] V REF =V BE +17.2V T ≈1.25V (1.10)

[0085] As Figure 5 shown is the bandgap reference circuit of the present invention, which mainly includes two parts: a start-up circuit and a bandgap reference core circuit.

[0086] MPi and MNi represent PMOS transistors and NMOS transistors respectively, and Qi represents a PNP type bipolar transistor. In the figure, Q1:Q2:Q3 = 4:1:1. The purpose of the start-up circuit is to ensure that the bandgap reference can break away from the "degeneracy point" during start-up. When the bandgap reference starts up normally, the MN8 transistor is turned off and the start-up circuit will automatically become ineffective. At the same time, a cascaded common-source common-gate (cadcode) current mirror structure is adopted in the design of the current mirror, aiming to improve the current mirror accuracy of the two paths of Q1 and Q2, reduce the influence of the VCC input voltage and temperature fluctuations on the current mirror, and further improve the power supply rejection ratio (PSRR) of the bandgap reference.

[0087] As Figure 5 shown, after adopting the cascode current mirror structure, the voltages at points x and y are strictly the same. Therefore, it can be deduced that the PTAT (positive temperature coefficient) current at point x is:

[0088]

[0089] This positive temperature coefficient current is copied by the current mirror branches of MP21 and MP32, and flows through R2 and transistor Q3 simultaneously, finally obtaining the bandgap reference voltage V BGR :

[0090]

[0091] At room temperature, from Equation (1.10), it can be known that to obtain a bandgap voltage reference with zero temperature coefficient,

[0092]

[0093] Finally, according to Equation (1.13), appropriate resistance values of R1 and R2 are selected. Through the simulation software, the simulation temperature range is set from -50°C to 125°C. The final simulation results are as Figure 6 shown. In the given temperature range, the maximum output of the bandgap reference is 1.195V, the minimum output is 1.191V, its maximum temperature offset value is 4mV, and the temperature coefficient is 0.057mV / °K, meeting the design specifications.

[0094] For the under-voltage lockout (UVLO) circuit, during the initial power-on stage or power-off stage of the system, the supply voltage of the system may not meet the operating requirements of the internal circuits of the system. For example, when the supply voltage of the system is low, it may cause errors in comparators and some logic circuits. Therefore, it is necessary to design an under-voltage lockout circuit that will give an enable signal and control the normal operation of each module circuit of the control system circuit only when the input voltage meets the normal operating requirements of the circuit. When the voltage is low, the enable signal is invalid, the circuits of each module of the system are turned off, and the system enters the standby state.

[0095] As Figure 7 shown, the under-voltage lockout circuit designed in this paper, D2 and D3 are Zener diodes with nominal voltages of 5.6V and 7.3V respectively. The analysis of its working principle is as follows. When the system is powered on and the VDD voltage rises continuously, at first, the VDD voltage is less than the sum of the reverse voltages of Zener diode D2 and Zener diode D3, and transistor HN1 is turned off. At this time, the potential at point x is equal to the VDD voltage. After passing through the buffer circuit, the output of UVLO is equal to the VDD voltage. As the VDD voltage continues to rise, when the VDD voltage is greater than the reverse voltages of the two Zener diodes, transistor HN1 gradually conducts. When HN1 is fully conducting, the voltage at point x is set to zero. At this time, the output voltage of UVLO is zero, and the UVLO module fails, and each working module of the circuit operates normally. The voltage at this time is called UVLO off .

[0096] Over-temperature protection circuit; Since the process parameters of CMOS devices are closely related to the operating temperature, if the operating temperature of the chip is too high, it may cause abnormal operation of the entire system. In addition, if a fault occurs inside the chip, such as a device breakdown causing a short circuit, the most direct manifestation of the chip in this case is a sharp rise in the junction temperature. Without protection, it is very likely to cause the chip to burn out. Therefore, based on the above two points, it is necessary to design a temperature detection and protection circuit, namely the OTP circuit, inside the system. First, the PTAT current in the bandgap reference (BGR) circuit is replicated through a current mirror. This current then flows through resistors R1 and R2 to generate a positive temperature coefficient voltage V b1 , and from Equation (1.11), the value of this voltage can be derived as:

[0097] V b1 = V T ln4·k (1.30)

[0098] where k is the ratio of the resistance at point x in Figure 7 to the sum of the resistances of R1 and R2 here (assuming that the N2 transistor is turned off at this time). Since the same type of resistors are used in the design of the BGR circuit and the OTP circuit, k is a quantity independent of the temperature coefficient. And the V T here has a positive temperature coefficient, so V b1 also has a positive temperature coefficient.

[0099] At room temperature, by setting an appropriate resistance value of R1 such that V b1 is less than the turn-on voltage of the triode Q1, Q1 is turned off, and its collector voltage V C1 is set to a high potential. After passing through the inverter, the final OTP output signal is at a low potential, and the system operates normally. When the chip operates abnormally and the internal temperature continues to rise, then the V b1 voltage also continues to rise. Eventually, it reaches the turn-on voltage of the Q1 transistor, and the Q1 transistor conducts, directly pulling the V C1 potential to zero. After passing through the inverter, the final OTP output is high-level effective, and the system is immediately turned off to protect the chip. Figure 8 as shown

[0100] At room temperature, the OTP output signal is low, so the gate voltage of N2 is high. At this time, N2 conducts and shorts the resistor R2. The PTAT current copied by the current mirror is only applied to R1. Therefore, the turn-on threshold T on of the OTP circuit can be set by adjusting the resistance value of R1. When the system temperature is too high and the OTP output is high, the gate voltage of N2 is low and N2 turns off. The resistor R2 is connected in series in the circuit. At this time, the PTAT current flows through both R1 and R2, which means that when the system temperature drops and is about to recover, the PTAT current needs to drop to a lower value before Q1 can turn off again. That is, when the system recovers, it needs to drop to a lower temperature for the OTP output to flip to low. This temperature is called the turn-off threshold T off . Here, the value of T can be set by adjusting the resistance value of R2 off . By setting the hysteresis, the over-temperature protection circuit can further improve the reliability of the OTP module and avoid system instability caused by continuous temperature fluctuations at the threshold point.

[0101] Conversion Efficiency Test and Comparison

[0102] The conversion efficiency shown by the system under different input voltages or different load conditions is different. The power supply drive system generally has a higher conversion efficiency under the condition of half load. In the case of light load, the conversion efficiency will decrease relatively. Therefore, it is necessary to test the working efficiency of the system under different input and load conditions respectively. The input end of the system is connected to a dedicated adjustable regulated power supply. The input voltage of the system can be directly set through the regulated power supply, and the input power of the system can be recorded. The output end of the system is connected to a test electronic load. The electronic load can be set in the constant current mode to maintain the output current while changing the load voltage at the output end, and the output power consumption of the system can be directly displayed. Finally, the read data is processed to calculate the conversion efficiency of the system under the current working conditions. Set the input voltage range to 85V - 265V and the load setting range to 25% - 100%. Test the conversion efficiency of the 1.5A constant current driver under different conditions. The test data is shown in Table 1.

[0103] Input voltage 25% load efficiency 50% load efficiency 75% load efficiency Full load efficiency Average efficiency 85V 86.57％ 86.74％ 86.23％ 85.78％ 86.33％ 130V 86.96％ 87.65％ 86.45％ 85.98％ 86.76％ 190V 87.01％ 87.37％ 87.10％ 86.86％ 87.09％ 220V 86.88％ 87.23％ 86.76％ 86.92％ 86.95％ 265V 86.72％ 87.12％ 87.02％ 87.11％ 86.99％

[0104] Table 1 Test Results of Conversion Efficiency of Constant Current Drive System Supporting CCM / QR Mode

[0105] As can be seen from Table 1, under various input conditions, the average output efficiency of the system reaches over 86%, meeting the requirements of the average efficiency index. The lowest efficiency occurs when the input voltage is 85V and the output is at full load, which is 85.78%. The lowest full-load efficiency is still higher than 85%, meeting the requirements of the full-load efficiency index. At the same time, the conversion efficiency of the system under all conditions is higher than 85%, meeting the national six-level energy consumption requirements. First, by eliminating auxiliary windings, optocouplers and other related components, the overall efficiency of the system is improved. At the same time, the integration of the quasi-resonant technology also significantly reduces the operating losses of the system. Table 2 lists the measured conversion efficiency results of a traditional 12V / 1.5A flyback constant-current driver with an auxiliary winding, which is used as a comparison for this design to further highlight the optimization of this design in terms of the working efficiency index.

[0106] Input voltage 25% load efficiency 50% load efficiency 75% load efficiency Full load efficiency Average efficiency 85V 86.10％ 86.23％ 85.91％ 85.32％ 85.89％ 130V 86.26％ 86.48％ 85.89％ 85.46％ 86.10％ 190V 86.51％ 86.61％ 86.83％ 86.72％ 86.67％ 220V 86.75％ 86.74％ 86.79％ 86.89％ 86.79％ 265V 86.52％ 86.63％ 86.80％ 87.01％ 86.74％

[0107] Table 2 Test Results of the Conversion Efficiency of a Traditional Constant-Current Drive System with an Auxiliary Winding

[0108] Comparing Table 1 and Table 2, it can be seen that after eliminating the auxiliary winding and adopting the valley conduction technology, the efficiency of the constant-current driver has been significantly improved under different input voltages and different load conditions. From the average efficiency column, the average efficiency of the constant-current drive system with an auxiliary winding is increased by about 0.3% compared with the traditional one. It can be further concluded that after eliminating the auxiliary winding, since the number of components in the whole system decreases, the power consumption also decreases, and finally the overall efficiency is improved. At the same time, since the valley conduction technology reduces the conduction loss of the power transistor, the efficiency is also improved.

[0109] In summary, a system-level simulation platform for the constant-current drive chip is built. By selecting appropriate parameters of the peripheral components, the startup process of the chip, the protection module, and the constant-current output accuracy under normal working conditions are mainly analyzed, verifying that the designed circuit functions normally, and various parameters such as the line regulation rate and load regulation rate of the system meet the design specifications. By building a physical test platform for the constant-current drive system, the conversion efficiency of the constant-current drive system under different working conditions is tested, and finally it is verified that the average efficiency and full-load efficiency of the system meet the specified indicators.

[0110] It should be understood that in the development process of any actual implementation, such as in any engineering or design project, a large number of specific implementation decisions can be made. Such development efforts may be complex and time-consuming, but for those ordinary technical personnel who benefit from this disclosure, without excessive experimentation, the development efforts will be a routine task of design, manufacturing, and production.

[0111] It should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and not to limit them. Although the present invention has been described in detail with reference to the preferred embodiments, those of ordinary skill in the art should understand that the technical solutions of the present invention can be modified or equivalently replaced without departing from the spirit and scope of the technical solutions of the present invention, and they should all be covered within the scope of the claims of the present invention.

Claims

1. A switching conversion system with multi-mode constant current control, characterized in that: It includes a bandgap reference circuit unit, a resistance measurement circuit unit, a switch drive circuit, an over-temperature protection circuit unit, a switch delay compensation circuit, a constant current error amplifier circuit, a constant current operation circuit, a zero-crossing detection circuit, a current sampling comparison circuit and an ARM main control unit; the bandgap reference circuit unit includes a startup circuit, a bandgap reference core circuit and an undervoltage lockout circuit; the purpose of the startup circuit is to ensure that the bandgap reference can be separated from the "degenerate point" when starting. When the bandgap reference is started normally, the transistor is turned off and the startup circuit will automatically fail; the undervoltage lockout circuit will give an enable signal only when the input voltage meets the normal working requirements of the circuit, and the circuits of each module of the control system circuit work normally; The resistance measurement circuit unit includes a matrix switch, a four-wire resistance test circuit, a resistance measurement circuit, a filter amplifier circuit and a 24-bit ADC sampling circuit; the input end of the matrix switch is connected to the output of the current monitoring circuit in the constant current output circuit unit; The ARM main control unit stores computer programs, PID control, PWM constant current control and PFM constant current control; The constant current operation circuit includes an output current estimation Vio generation circuit; The switch delay compensation circuit is used to superimpose the primary current on the CS sampling resistor, and the superimposed current then acts on the sampling resistor as the positive input of the comparator. Its purpose is to make the comparator flip in advance before the primary peak current arrives to compensate for the total shutdown delay Δt inside the circuit.

2. The switch conversion system with multi-mode constant current control according to claim 1, characterized in that: The input end of the switch driving circuit is connected to the output end of the ARM main control unit, and the output end of the switch driving circuit is connected to the input end of the matrix switch.

3. The switch conversion system with multi-mode constant current control according to claim 1, characterized in that: The over-temperature protection circuit unit further improves the reliability of the OTP module by setting hysteresis, avoiding system instability caused by continuous temperature fluctuations at the threshold point.

4. The switch conversion system with multi-mode constant current control according to claim 1, characterized in that: The resistance measurement circuit unit also includes an external metering interface connected to the matrix switch, which is used to connect an external metering device to perform constant current output and resistance measurement accuracy metering.

5. The switch conversion system with multi-mode constant current control according to claim 1, characterized in that: The output voltage monitoring protection circuit corresponding to the zero-crossing detection circuit can take protective measures in time when a short circuit or open circuit occurs at the output end, thereby enhancing the safety and reliability of the system.

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