Sensorless Active Disturbance Rejection Control Method for Permanent Magnet Synchronous Motor Based on Nonlinear Flux Observation

Abstract

The present invention provides a sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux linkage observation, which solves the problem of poor control effect of the permanent magnet synchronous motor, including: step 1, measuring the induced current of the permanent magnet synchronous motor to obtain a rotor position estimation value and a speed estimation value; step 2, using the speed estimation value and a given speed command value, obtaining a q-axis current command value through a speed loop auto-disturbance rejection controller; step 3, transforming the induced current to obtain a d-axis current and a q-axis current, and then obtaining a d-axis voltage command value and a q-axis voltage command value; step 4, converting the d-axis voltage command value and the q-axis voltage command value into an axis voltage command value and an axis voltage command value, and inputting them into an SVPWM modulator to control the permanent magnet synchronous motor. The present invention uses a nonlinear flux linkage observer as a position observer, uses a rotor position estimation value and a speed estimation value to calculate the rotor position and speed estimation, and realizes sensorless control of the permanent magnet synchronous motor.

Description

Technical Field

[0001] The invention relates to the field of motor control, and in particular to a sensorless self-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation. Background Art

[0002] With the rapid development of aerospace, permanent magnet synchronous motors with high power density and low energy consumption have become a hot research topic for aircraft-borne pump loads. Permanent magnet synchronous motors are synchronous motors that use permanent magnet excitation to generate a synchronous rotating magnetic field. Their application in aircraft-borne pump loads has significant advantages, mainly due to their high efficiency, high power factor, low noise, low vibration and precise speed regulation performance.

[0003] The working principle of a permanent magnet synchronous motor is as follows: There is a permanent magnet on the rotor. When current passes through the stator of the motor, a rotating magnetic field is formed on the stator. Since the magnetic field of the permanent magnet is constant, when the rotating magnetic field and the magnetic field of the permanent magnet interact, a torque is generated, which causes the motor to rotate. Existing aircraft-mounted pump loads all use mechanical position / speed sensors to obtain the position or speed signal of the motor.

[0004] Permanent magnet synchronous motors are widely used due to their high efficiency, high performance and high reliability. Traditional control methods require mechanical position / speed sensors to accurately obtain rotor position information. However, aircraft-borne pump loads work in high temperature, high pressure and high electromagnetic interference environments, which makes mechanical position / speed sensors prone to failure, thereby causing traction system failures, resulting in large torque shocks and endangering aviation safety. In order to fundamentally eliminate this safety hazard, researchers have proposed position sensorless drive technology. Position sensorless drive technology is an advanced motor control technology that aims to reduce dependence on traditional mechanical sensors. It mainly estimates rotor position and speed through voltage and current measurements and the application of specific algorithms. It has the advantages of strong anti-interference ability, high integration and long service life.

[0005] In the control of permanent magnet synchronous motors, the commonly used flux observer plays an important role in the motor control system. The flux observer observes the flux state inside the motor in real time and converts it into a voltage signal or other processable signal form to achieve closed-loop control of the permanent magnet synchronous motor. In practice, there are a large number of disturbances in the permanent magnet synchronous motor control system, and these disturbances can be divided into periodic disturbances and non-periodic disturbances according to different types. Among them, the typical representative of non-periodic disturbances is the change of load torque. Due to the requirements of the working environment, the load will change frequently, resulting in fluctuations in the motor speed; due to the influence of factors such as the inverter dead time, the 5th harmonic and 7th harmonic are converted into the 6th harmonic as a periodic disturbance after coordinate transformation, which will affect the current loop control performance, cause magnetic field orientation error, and affect the control effect. Compared with the conventional flux observer, the nonlinear flux observer has the characteristics of being applicable to a wide speed range because its state variables are independent of the speed, and it has a simple structure and is easy to implement. Summary of the invention

[0006] The purpose of the present invention is to provide a sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor based on nonlinear flux observation, so as to solve the problem of poor control effect of the permanent magnet synchronous motor. The nonlinear flux observer is used as a position observer, and the rotor position and speed estimation are calculated using the rotor position estimation value and the speed estimation value to realize the sensorless control of the permanent magnet synchronous motor. The speed loop, d-axis and q-axis current auto-disturbance rejection controller are used to improve the response current waveform of the permanent magnet synchronous motor, improve the rotor position estimation performance, and use the space vector pulse width modulation (Space Vector Pulse Width Modulation, SVPWM) device to optimize the position sensorless control performance of the permanent magnet synchronous motor based on the nonlinear flux observer.

[0007] The technical solution for achieving the purpose of the present invention is as follows:

[0008] The present invention provides a sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation, comprising the following steps:

[0009] Step 1: Measure the induced current of the permanent magnet synchronous motor and obtain the estimated value of the rotor position according to the induced current and speed estimate ;

[0010] Step 2: Using the estimated speed value and the given speed command value , the q-axis current command value is obtained through the speed loop anti-disturbance controller ;

[0011] Step 3: transform the induced current to obtain the d-axis current i d and q-axis current iq , according to the d-axis current i d , the d-axis voltage command value is obtained through the d-axis current anti-disturbance controller , according to the q-axis current i q , the q-axis voltage command value is obtained through the q-axis current anti-disturbance controller ;

[0012] Step 4: The d-axis voltage command value obtained in step 3 is and the q-axis voltage command value Convert to α-axis voltage command value and β-axis voltage command value , the α-axis voltage command value and β-axis voltage command value The input SVPWM modulator controls the permanent magnet synchronous motor.

[0013] In one embodiment of the present invention, the step 1 comprises:

[0014] Step 11: measure the induced current of the permanent magnet synchronous motor in the ABC coordinate system, and perform Clarke transformation on the induced current to obtain the α-axis current i α and the β-axis current i β ;

[0015] Step 12: Use the α-axis current i obtained in step 11 α and the β-axis current i β , the α-axis voltage command value of the previous cycle and the β-axis voltage command value of the previous cycle As input to the nonlinear flux observer model, obtain the rotor position estimate The sine and cosine components of

[0016] Step 13: The estimated rotor position obtained in step 12 After normalizing the sine and cosine components, the rotor position estimate is calculated through the phase-locked loop and speed estimate .

[0017] In one embodiment of the present invention, the step 11 is specifically as follows:

[0018] The three-phase induced current i of the permanent magnet synchronous motor in the ABC coordinate system is a 、i b 、i c Take samples;

[0019] The three-phase induced current i a 、i b 、i cPerform Clarke transformation to the αβ coordinate system to obtain the α-axis current i α and the β-axis current i β .

[0020] In one embodiment of the present invention, the nonlinear flux observer model is expressed as:

[0021] x ^ ˙ = y + c 2 or ( x ^ ) [ ψ m 2 − ‖ or ( x ^ ) ‖ 2 ] ， c ＞ 0 or ( x ^ ) = x ^ − L [ i α i β ] y = − R s [ i α i β ] + [ u α ， 1 * u β , 1 * ] Formula (1)

[0022] In the formula, represents the state variables of the nonlinear flux observer, represents the derivative of the state variable of the nonlinear flux observer, represents the intermediate variable of magnetic flux observation, represents the observer gain, represents a vector function, represents the permanent magnet flux, express The norm of ; represents the stator inductance of the permanent magnet synchronous motor, Represents the stator resistance of the permanent magnet synchronous motor;

[0023] The state variables of the nonlinear flux observer are calculated using formula (1): Then, according to formula (2)

[0024] [ cos i ^ r sin i ^ r ] = 1 ψ m ( x ^ − L [ i α i β ] ) Formula (2)

[0025] The rotor position estimate is calculated The sine and cosine components of [ cos i ^ r sin i ^ r ] .

[0026] In one embodiment of the present invention, in step 2, the speed loop anti-disturbance control is expressed as:

[0027] Formula (3)

[0028] In the formula, represents the intermediate variable of the speed loop, represents the estimated value of the total disturbance of the speed loop obtained by using the speed loop expansion state observer, represents the gain of the speed loop anti-disturbance controller, represents the bandwidth of the speed loop anti-disturbance controller, represents the speed estimation value obtained by using the speed loop expansion state observer;

[0029] The speed loop expansion state observer is expressed as:

[0030] Formula (4)

[0031] In the formula, represents the speed observation error, represents the q-axis current command value calculated in the previous cycle, β s1 represents the first gain of the speed loop expansion state observer, β s2 represents the second gain of the speed loop expansion state observer; the speed estimate obtained in step 1 , calculated using the speed loop expansion state observer as well as .

[0032] In one embodiment of the present invention, in step three, the d-axis current auto-disturbance rejection controller is expressed as:

[0033] Formula (5)

[0034] In the formula, represents the intermediate variable corresponding to the d-axis, represents the estimated value of the total disturbance on the d-axis obtained using the d-axis extended state observer, represents the calculated d-axis disturbance, Represents the gain of the current signal; represents the bandwidth of the d-axis current ADRC, represents the d-axis current command value, represents the estimated value of the d-axis current obtained using the d-axis extended state observer;

[0035] The d-axis expansion state observer is expressed as:

[0036] Formula (6)

[0037] In the formula, represents the d-axis current observation error, Indicates the d-axis voltage command value calculated in the previous cycle, represents the first gain of the d-axis extended state observer, represents the second gain of the d-axis extended state observer; through the d-axis current , calculated using the d-axis extended state observer as well as .

[0038] In one embodiment of the present invention, in step 3, the q-axis current auto-disturbance rejection controller is expressed as:

[0039] Formula (7)

[0040] In the formula, represents the intermediate variable corresponding to the q-axis, represents the estimated value of the total disturbance on the q-axis obtained by using the q-axis extended state observer, represents the calculated q-axis disturbance, Represents the gain of the current signal; represents the bandwidth of the q-axis current ADRC, Indicates the command value of the q-axis current, represents the estimated value of q-axis current obtained by using the q-axis extended state observer;

[0041] The q-axis extended state observer is expressed as:

[0042] Formula (8)

[0043] In the formula, represents the q-axis current observation error, Indicates the q-axis voltage command value calculated in the previous cycle, represents the first gain of the q-axis extended state observer, represents the second gain of the q-axis extended state observer; through the q-axis current , calculated using the q-axis extended state observer as well as .

[0044] Compared with the prior art, the present invention has the following beneficial effects:

[0045] The present invention uses a nonlinear flux observer as a position observer and uses the rotor position estimate The rotor position and speed estimation values ​​are used to calculate the rotor position and speed estimation to realize sensorless control of the permanent magnet synchronous motor. The response current waveform of the permanent magnet synchronous motor is improved by the anti-disturbance controller to improve the rotor position estimation performance. The SVPWM modulator is used to optimize the position sensorless control performance of the permanent magnet synchronous motor based on the nonlinear flux observer. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] FIG1 is a flow chart of a sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation provided by the present invention;

[0047] FIG2 is a block diagram of the active disturbance rejection control principle of a permanent magnet synchronous motor based on a nonlinear flux observer provided by the present invention;

[0048] FIG3 is a structural block diagram of a nonlinear flux observer provided by the present invention;

[0049] FIG4 is a structural diagram of a speed loop anti-disturbance control controller provided by the present invention;

[0050] FIG5 is a structural diagram of a d-axis current auto-disturbance rejection controller provided by the present invention;

[0051] FIG6 is a structural diagram of a q-axis current auto-disturbance rejection controller provided by the present invention;

[0052] FIG7 is a disturbance calculation block diagram of the current loop auto-disturbance rejection controller provided by the present invention;

[0053] FIG8 is a loading curve of load torque provided by the present invention;

[0054] FIG9 is a speed waveform diagram when both the speed loop and the current loop use PI control;

[0055] FIG10 is a speed waveform diagram of the speed loop and the current loop after the auto-disturbance rejection controller proposed by the present invention is adopted;

[0056] Figure 11 The A-phase current waveform when the speed loop and current loop adopt PI control;

[0057] Figure 12 This is a schematic diagram of harmonic order when the speed loop and current loop adopt PI control;

[0058] Figure 13 The A-phase current waveform after the speed loop and the current loop adopt the auto-disturbance rejection controller proposed by the present invention;

[0059] Figure 14 It is a schematic diagram of the harmonic order after the speed loop and the current loop adopt the auto-disturbance rejection controller proposed by the present invention. DETAILED DESCRIPTION

[0060] The present invention is described in detail below in conjunction with the various embodiments shown in the accompanying drawings, but it should be noted that these embodiments are not limitations of the present invention, and any equivalent transformations or substitutions in functions, methods, or structures made by ordinary technicians in the field based on these embodiments are all within the scope of protection of the present invention.

[0061] Please refer to Figure 1. An embodiment of the present invention provides a sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor based on nonlinear flux observation. The embodiment of the present invention is based on a sensorless auto-disturbance rejection control strategy for a permanent magnet synchronous motor based on a nonlinear flux observer. The principle block diagram of the auto-disturbance rejection control for a permanent magnet synchronous motor based on a nonlinear flux observer is shown in Figure 2. The sensorless auto-disturbance rejection control strategy for a permanent magnet synchronous motor obtains a rotor position estimate and a speed estimate based on a nonlinear flux observer. The difference between the speed estimate and the given speed of the permanent magnet synchronous motor is passed through an active disturbance rejection controller (ADRC) of a speed loop to obtain a command value of the axis current in the q direction. The sensorless auto-disturbance rejection control strategy for a permanent magnet synchronous motor adopts i d=0 control, the difference between the command current and the feedback current of the dq axis passes through the active disturbance rejection control (ADRC) of the current loop to obtain the dq axis command voltage, and the permanent magnet synchronous motor is controlled by space vector pulse width modulation (SVPWM), thereby realizing the position sensorless closed-loop structure of the permanent magnet synchronous motor, and the nonlinear flux observer structure frame is shown in Figure 3. Considering the existence of disturbances, the speed loop anti-disturbance controller, the d-axis current anti-disturbance controller, and the q-axis current anti-disturbance controller of this embodiment realize the permanent magnet synchronous motor position sensorless anti-disturbance control strategy based on the nonlinear flux observer. Among them, the speed loop ADRC regards the load torque and electromagnetic torque as the total disturbance, takes the motor speed as the input and the q-axis current as the output. The structure of the speed loop ADRC is shown in Figure 4. The current loop ADRC regards the parts except the d-axis voltage command value and the q-axis voltage command value as the total disturbance. The disturbance of the motor speed is compensated by calculating the total disturbance and combining the second-order extended state observer. The structure of the d-axis current ADRC is shown in Figure 5, the structure of the q-axis current ADRC is shown in Figure 6, and the current loop disturbance calculation principle of the current loop ADRC is shown in Figure 7.

[0062] The sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation according to an embodiment of the present invention comprises the following steps 1 to 4:

[0063] Step 1: Measure the induced current of the permanent magnet synchronous motor and obtain the estimated value of the rotor position based on the induced current and speed estimate .

[0064] The above step 1 includes the following steps 11, 12 and 13.

[0065] Step 11: Measure the induced current of the permanent magnet synchronous motor in the ABC coordinate system, and perform Clarke transformation on the induced current to obtain the α-axis current i α and the β-axis current i β .

[0066] Specifically, step 11 is as follows: in the ABC coordinate system, the three-phase induced current i of the permanent magnet synchronous motor is a 、i b 、i c Sampling; the three-phase induced current i a 、i b 、i c Perform Clarke transformation to the αβ coordinate system to obtain the α-axis current i α and the β-axis current i β .

[0067] Step 12: In the αβ coordinate system, use the α-axis current i obtained in step 11 α and the β-axis current i β , the α-axis voltage command value of the previous cycle and the β-axis voltage command value of the previous cycle As input to the nonlinear flux observer model, obtain the rotor position estimate The sine and cosine components of .

[0068] Specifically, step 12 is as follows: In the αβ coordinate system, the α-axis current i obtained in step 11 is α and the β-axis current i β , the α-axis voltage command value of the previous cycle and the β-axis voltage command value of the previous cycle As input to the nonlinear flux observer model, the actual rotor position estimate is obtained using the nonlinear flux observer model The sine and cosine components of . Among them:

[0069] The mathematical model of permanent magnet synchronous motor is expressed as:

[0070] { L d dt [ i α i β ] = − R s [ i α i β ] + oh ψ m [ s i n i r − c o s i r ] + [ u α u β ] T e = 3 P 4 ψ m ( i α c o s i r − i β s i n i r )

[0071] Where, L represents the stator inductance of the permanent magnet synchronous motor, represents the derivation, represents the stator resistance of the permanent magnet synchronous motor, represents the α-axis current, represents the β-axis current, Indicates the motor speed, represents the permanent magnet flux, represents the rotor position angle, represents the α-axis voltage command value, represents the β-axis voltage command value, represents the electromagnetic torque, Indicates the number of motor poles;

[0072] Among them, the nonlinear flux observer model is expressed as:

[0073] x ^ ˙ = y + c 2 or ( x ^ ) [ ψ m 2 − ‖ or ( x ^ ) ‖ 2 ] ， c ＞ 0 or ( x ^ ) = x ^ − L [ i α i β ] y = − R s [ i α i β ] + [ u α ， 1 * u β , 1 * ] Formula (1)

[0074] In formula (1), represents the state variables of the nonlinear flux observer, represents the derivative of the state variable, represents the intermediate variable of magnetic flux observation, represents the observer gain, represents a vector function, represents the permanent magnet flux, express The norm of represents the stator inductance of the permanent magnet synchronous motor, The embodiment of the present invention combines the above expression to construct a position observation model of a nonlinear flux observer.

[0075] The nonlinear flux observer calculates the rotor position estimate After normalizing the sine and cosine components, the estimated rotor position is calculated through the phase-locked loop and estimated speed ; The state variables of the nonlinear flux observer are calculated using formula (1): Then, according to formula (2):

[0076] [ cos i ^ r sin i ^ r ] = 1 ψ m ( x ^ − L [ i α i β ] ) Formula (2)

[0077] Calculate the estimated rotor position The sine and cosine components of [ cos i ^ r sin i ^ r ] It should be noted that the formula (2) in the embodiment of the present invention is obtained by Transformation 1 ψ m ( x − L [ i α i β ] ) = [ cos i r sin i r ] , the expressions of state variable x and intermediate variable y: x = L [ i α i β ] + ψ m [ c o s i s i n i ] , y = − R s [ i α i β ] + [ u α ， 1 * u β ， 1 * ] , the relationship between the state variable x and the intermediate variable y is ; Define the expression of vector function η(x) as: or ( x ) = x − L [ i α i β ] , the norm expression of the vector function η(x) is: .in, represents the intermediate variable, represents the stator resistance of the permanent magnet synchronous motor, Indicates the motor speed, represents the permanent magnet flux, Indicates the parameters, represents the stator inductance of the permanent magnet synchronous motor, Represents the rotor position angle. Therefore, the expression of the sine and cosine components including the estimated rotor position calculated by the observer in the embodiment of the present invention is formula (2).

[0078] Step 13: Contain the estimated value of the rotor position After normalizing the sine and cosine components, the speed estimation value is obtained through phase-locked loop calculation .

[0079] Step 2: Use the estimated speed and the given speed command value , the q-axis current command value is obtained through the speed loop anti-disturbance controller .

[0080] Specifically, step 2 includes: the estimated speed value output in step 1 and the given speed command value Input speed loop ADRC, speed estimation and the given speed command value The difference is passed through the speed loop anti-disturbance controller to obtain the q-axis current command value Among them: the speed loop anti-disturbance controller is expressed as:

[0081] Formula (3)

[0082] In formula (3), represents the intermediate variable of the speed loop, represents the estimated value of the total disturbance of the speed loop obtained by using the speed loop expansion state observer, represents the gain of the speed loop anti-disturbance controller, represents the bandwidth of the speed loop anti-disturbance controller, represents the speed estimation value obtained by using the speed loop expansion state observer;

[0083] The speed loop expansion state observer is expressed as:

[0084] Formula (4)

[0085] In formula (4), represents the speed observation error, represents the q-axis current command value calculated in the previous cycle, β s1 represents the first gain of the speed loop expansion state observer, β s2 represents the second gain of the speed loop expansion state observer. The speed estimate obtained in step 1 , calculated using the speed loop expansion state observer as well as .

[0086] Step 3: Transform the induced current to obtain the d-axis current i d and q-axis current i q , according to the d-axis current i d , the d-axis voltage command value is obtained through the d-axis current anti-disturbance controller , according to the q-axis current i q , the q-axis voltage command value is obtained through the q-axis current anti-disturbance controller .

[0087] Specifically, in step 3 of the embodiment of the present invention, the α-axis current i α and the β-axis current i β The d-axis current i is obtained by Park transformation to the dq coordinate system d and q-axis current i q , for the d-axis current i d The d-axis current ADRC is used to perform proportional and integral regulation (PI) to obtain the d-axis voltage command value. ; For q-axis current i q The q-axis current self-disturbance rejection controller is used to perform proportional and integral regulation to obtain the q-axis voltage command value. .

[0088] More specifically, in step three of the embodiment of the present invention, the d-axis current auto-disturbance rejection controller is expressed as:

[0089] Formula (5)

[0090] In formula (5), represents the intermediate variable corresponding to the d-axis, represents the estimated value of the total disturbance on the d-axis obtained using the d-axis extended state observer, represents the calculated d-axis disturbance, Represents the gain of the current signal; represents the bandwidth of the d-axis current ADRC, represents the d-axis current command value, represents the estimated value of the d-axis current obtained using the d-axis extended state observer;

[0091] The d-axis expansion state observer in the embodiment of the present invention is expressed as:

[0092] Formula (6)

[0093] In formula (6), represents the d-axis current observation error, Indicates the d-axis voltage command value calculated in the previous cycle, represents the first gain of the d-axis extended state observer, represents the second gain of the d-axis extended state observer; through the d-axis current , calculated using the d-axis extended state observer as well as .

[0094] More specifically, in step three of the embodiment of the present invention, the q-axis current auto-disturbance rejection controller is expressed as:

[0095] Formula (7)

[0096] In formula (7), represents the intermediate variable corresponding to the q-axis, represents the estimated value of the total disturbance on the q-axis obtained by using the q-axis extended state observer, represents the calculated q-axis disturbance, Represents the gain of the current signal; represents the bandwidth of the q-axis current ADRC, Indicates the command value of the q-axis current, represents the estimated value of the q-axis current obtained using the q-axis extended state observer.

[0097] The q-axis expansion state observer in the embodiment of the present invention is expressed as:

[0098] Formula (8)

[0099] In formula (8), represents the q-axis current observation error, Indicates the q-axis voltage command value calculated in the previous cycle, represents the first gain of the q-axis extended state observer, represents the second gain of the q-axis extended state observer; through the q-axis current , calculated using the q-axis extended state observer as well as .

[0100] It should be noted that the speed loop expansion state observer in step 2 of the embodiment of the present invention and the d-axis expansion state observer and the q-axis expansion state observer in step 3 can use a second-order expansion state observer, which is established through a single-input single-output system, and the establishment process specifically includes:

[0101] (1) Establish a first-order single-input single-output system, the expression is:

[0102]

[0103] In the formula, represents the state variable derivative, represents the dynamic response of the system, represents the external disturbance, b represents the gain of the input signal, Represents input.

[0104] (2) Establish a second-order extended state observer, expressed as:

[0105]

[0106] In the formula, e represents the observation error, z 1 represents the estimated state variable, Represents a variable, represents the derivative of the estimated state variable, z 2 represents the estimated disturbance, β 1 represents the first gain of the second-order extended state observer, represents the derivative of the estimated disturbance, β 2 represents the second gain of the second-order extended state observer.

[0107] Step 4: The d-axis voltage command value obtained in step 3 and q-axis voltage command value Convert to α-axis voltage command value and β-axis voltage command value , the α-axis voltage command value and β-axis voltage command value The input SVPWM modulator controls the permanent magnet synchronous motor.

[0108] More specifically, step 4 of the embodiment of the present invention is as follows: and q-axis voltage command value The α-axis voltage command value is obtained by inverse Park transformation and β-axis voltage command value ; α-axis voltage command value and β-axis voltage command value Input SVPWM modulator; SVPWM modulator outputs control signal to control the operation of permanent magnet synchronous motor.

[0109] In an embodiment of the present invention, when the permanent magnet synchronous motor is running in a high-speed area and the inverter dead time is small, the position sensorless drive system is relatively less affected by the inverter dead time and can be ignored. Since the load of the permanent magnet synchronous motor changes frequently, the motor speed fluctuates greatly, which makes it difficult to maintain the output torque of the motor stable. In addition, there is a current sampling error, which causes an error in the estimated rotor position signal, resulting in a magnetic field orientation error, which in turn reduces the position sensorless control performance. In order to improve the sensorless control performance of permanent magnet synchronous motors as aircraft-borne pump loads, an embodiment of the present invention has developed a permanent magnet synchronous motor self-disturbance control method based on a nonlinear flux observer for sensorless control of permanent magnet synchronous motors to improve the control performance of the motor after disturbances.

[0110] The principle block diagram of the permanent magnet synchronous motor self-disturbance rejection control based on the nonlinear flux observer of the embodiment of the present invention is shown in Figure 2. The estimated speed and the estimated rotor position are solved by the nonlinear flux observer to form a position sensorless control closed loop structure; the anti-disturbance performance of the permanent magnet synchronous motor in the presence of disturbances is enhanced by constructing an anti-disturbance rejection controller. Figure 4 is the speed loop self-disturbance rejection controller structure, Figure 5 is the d-axis current self-disturbance rejection controller structure, Figure 6 is the q-axis current self-disturbance rejection controller structure, Figure 7 is the disturbance calculation principle block diagram of the current loop self-disturbance rejection controller, and Figure 8 is the loading curve of the load torque of the load start under loading conditions. The initial load is 1.5Nm, and 0.5Nm is loaded every 0.2s after 1s. The load reaches the maximum at 1.6s, with a maximum of 3.5Nm. FIG9 is a speed waveform diagram when both the speed loop and the current loop use PI control. The speed waveform is 8000rpm, and the speed reaches the rated speed at 0.67s. After reaching the rated speed, the estimated speed fluctuation increases significantly. When the load changes suddenly, the speed drops severely, and the time required to recover to 8000rpm is long. FIG10 is a speed waveform diagram after the speed loop and the current loop are replaced with the proposed auto-disturbance rejection controller. The speed waveform is 8000rpm, and the speed reaches the rated speed at 0.5s. After reaching the steady state, the estimated speed fluctuation is significantly smaller than FIG9. When the load changes suddenly, the speed fluctuation is small, and the time required to recover to 8000rpm is short. By comparing FIG9 and FIG10, it can be seen that after using the sensorless auto-disturbance rejection control method of the embodiment of the present invention, the time required for the actual speed to reach the rated speed is significantly reduced, and the speed fluctuation is smaller when the load changes suddenly. Figure 11 and Figure 12 The following are the A-phase current waveform and harmonic order diagrams when both the speed loop and the current loop use PI control. Figure 11 and Figure 12 It can be seen that the A-phase current is at the maximum limit value most of the time, and the current tracking performance is poor. After the speed loop and the current loop are replaced by the anti-disturbance controller, the waveform and harmonic number of the A-phase current are as follows: Figure 13 and Figure 14 As shown, compared Figure 11 and Figure 12 It can be seen that after adding the ADRC, the current waveform is significantly reduced and the tracking performance is significantly improved.

[0111] In summary, the embodiments of the present invention realize the permanent magnet synchronous motor active disturbance rejection control based on the nonlinear flux observer; the embodiments of the present invention reduce the motor speed fluctuation caused by load fluctuation through the speed loop active disturbance rejection controller (ADRC); the embodiments of the present invention reduce the 5th harmonic or 7th harmonic caused by factors such as the inverter dead time through the current loop active disturbance rejection controller (ADRC), and reduce the content of the 6th harmonic converted after coordinate transformation in the response current, thereby enhancing the sinusoidality of the response current, reducing the influence of periodic disturbances on the current loop in the positionless control, and enhancing the robustness of the control system; the embodiments of the present invention improve the position estimation accuracy by improving the position sensorless control performance of the permanent magnet synchronous motor based on the nonlinear flux observer.

[0112] The series of detailed descriptions listed above are only specific descriptions of feasible implementation methods of the present invention. They are not intended to limit the scope of protection of the present invention. Any equivalent implementation methods or changes that do not deviate from the technical spirit of the present invention should be included in the scope of protection of the present invention.

[0113] It will be apparent to those skilled in the art that the invention is not limited to the details of the exemplary embodiments described above and that the invention can be implemented in other specific forms without departing from the spirit or essential features of the invention. Therefore, the embodiments should be considered exemplary and non-limiting in all respects, and the scope of the invention is defined by the appended claims rather than the foregoing description, and it is intended that all variations falling within the meaning and scope of the equivalent elements of the claims be included in the invention. Any reference numeral in a claim should not be considered as limiting the claim to which it relates.

[0114] In addition, it should be understood that although the present specification is described according to implementation modes, not every implementation mode contains only one independent technical solution. This description of the specification is only for the sake of clarity. Those skilled in the art should regard the specification as a whole. The technical solutions in each embodiment may also be appropriately combined to form other implementation modes that can be understood by those skilled in the art.

Claims

1. A sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor based on nonlinear flux observation, characterized in that: The following steps are involved: Step 1: Measure the induced current of the permanent magnet synchronous motor and obtain the estimated value of the rotor position according to the induced current and the estimated speed ω m ; Step 2: Using the estimated speed value ω m and the given speed command value The q-axis current command value is obtained through the speed loop anti-disturbance controller Step 3: transform the induced current to obtain the d-axis current i d and q-axis current i q , according to the d-axis current i d , the d-axis voltage command value is obtained through the d-axis current anti-disturbance controller According to the q-axis current i q , the q-axis voltage command value is obtained through the q-axis current anti-disturbance controller Step 4: The d-axis voltage command value obtained in step 3 is and the q-axis voltage command value Converted to α-axis voltage command value by inverse Park transformation The d-axis voltage command value obtained in step 3 is and the q-axis voltage command value Converted to β-axis voltage command value by inverse Park transformation The α-axis voltage command value and β-axis voltage command value Input SVPWM modulator to control permanent magnet synchronous motor; Wherein, the step 1 comprises: Step 11: measure the induced current of the permanent magnet synchronous motor in the ABC coordinate system, and perform Clarke transformation on the induced current to obtain the α-axis current i α and the β-axis current i β ; Step 12: Use the α-axis current i obtained in step 11 α and the β-axis current i β , the α-axis voltage command value of the previous cycle and the β-axis voltage command value of the previous cycle As input to the nonlinear flux observer model, obtain the rotor position estimate The sine and cosine components of ; Step 13: The estimated rotor position obtained in step 12 After normalizing the sine and cosine components, the rotor position estimate is calculated through the phase-locked loop and the estimated speed ω m ; The nonlinear flux observer model is expressed as: In the formula, represents the state variables of the nonlinear flux observer, represents the derivative of the state variable of the nonlinear flux observer, y represents the intermediate variable of flux observation, γ represents the observer gain, represents the vector function, ψ m represents the permanent magnet flux, express norm, L represents the stator inductance of the permanent magnet synchronous motor, R s Represents the stator resistance of the permanent magnet synchronous motor; The state variables of the nonlinear flux observer are calculated using formula (1): Then, according to formula (2) The rotor position estimate is calculated The sine and cosine components of 2. The sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation according to claim 1 is characterized in that: The step 11 is specifically as follows: The three-phase induced current i of the permanent magnet synchronous motor in the ABC coordinate system is a 、i b 、i c Take samples; The three-phase induced current i a 、i b 、i c Perform Clarke transformation to the αβ coordinate system to obtain the α-axis current i α and the β-axis current i β .

3. The sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation according to claim 1 is characterized in that: In step 2, the speed loop anti-disturbance control is expressed as: In the formula, C s0 represents the intermediate variable of the speed loop, represents the estimated value of the total disturbance of the speed loop obtained by using the speed loop expansion state observer, b s represents the gain of the speed loop anti-disturbance controller, K p represents the bandwidth of the speed loop anti-disturbance controller, represents the speed estimation value obtained by using the speed loop expansion state observer; The speed loop expansion state observer is expressed as: In formula (4), e s represents the speed observation error, represents the q-axis current command value calculated in the previous cycle, β s1 represents the first gain of the speed loop expansion state observer, β s2 represents the second gain of the speed loop expansion state observer; the speed estimation value ω obtained in step 1 m , calculated using the speed loop expansion state observer as well as 4. The sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation according to claim 1 is characterized in that: In step 3, the d-axis current ADRC is expressed as: In the formula, C d0 represents the intermediate variable corresponding to the d-axis, represents the d-axis total disturbance estimate obtained using the d-axis extended state observer, d d represents the calculated d-axis disturbance, b represents the gain of the current signal; ω d represents the bandwidth of the d-axis current ADRC, represents the d-axis current command value, represents the estimated value of the d-axis current obtained using the d-axis extended state observer; The d-axis expansion state observer is expressed as: In the formula, e d represents the d-axis current observation error, represents the d-axis voltage command value calculated in the previous cycle, β d1 represents the first gain of the d-axis extended state observer, β d2 represents the second gain of the d-axis extended state observer; through the d-axis current i d , calculated using the d-axis extended state observer as well as 5. The sensorless auto-disturbance rejection control method for a permanent magnet synchronous motor with nonlinear flux observation according to claim 1 is characterized in that: In step 3, the q-axis current ADRC is expressed as: In the formula, C q0 represents the intermediate variable corresponding to the q-axis, represents the estimated value of the total disturbance on the q-axis obtained by using the q-axis extended state observer, d q represents the calculated q-axis disturbance, b represents the gain of the current signal; ω q represents the bandwidth of the q-axis current ADRC, Indicates the command value of the q-axis current, represents the estimated value of q-axis current obtained by using the q-axis extended state observer; The q-axis extended state observer is expressed as: In the formula, e q represents the q-axis current observation error, represents the q-axis voltage command value calculated in the previous cycle, β q1 represents the first gain of the q-axis extended state observer, β q2 represents the second gain of the q-axis extended state observer; through the q-axis current i q , calculated using the q-axis extended state observer as well as

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