Half-wave rectification LLC converter and control IC

Abstract

The invention provides a half-wave rectification LLC converter and a control IC capable of preventing an increase in circulating current when an output current increases or when an output voltage is low and variable. The half-wave rectification LLC converter and the control IC have: a first series circuit in which a main switch and a resonance switch are connected in series; a second series circuit in which the primary winding and the resonance capacitor are connected in series; a transformer having a primary winding and a secondary winding; and a control unit that turns on and off the main switch and the resonance switch. The control unit has a first drive signal for turning on / off the main switch, a second drive signal for turning on / off the resonant switch, and a third drive signal for turning on / off the resonant switch after the second drive signal. Furthermore, the control unit turns on the second drive signal during a period of 1 / 2 of the resonance period from a time that is 1 / 2 of the previous resonance period than the timing at which the excitation current of the transformer becomes zero.

Description

Technical Field

[0001] The invention relates to a half-wave rectification LLC converter and a control IC. Background Art

[0002] In the past, a half-wave rectifier LLC converter was proposed as a power supply circuit for an isolated DC-DC converter. Patent document 1 discloses an operation method for a power converter that controls a main switch and a resonant switch. The operation method disclosed in patent document 1 proposes a control that turns the resonant switch on and off twice during the off period of the main switch. Regarding the first on-off of the resonant switch, after the main switch is turned off, the resonant switch is turned on during the period when the current flows in the body diode of the resonant switch, and the resonant current generated by the resonant capacitor and the leakage inductance is used to transmit power to the secondary side. Regarding the second on-off of the resonant switch, it is performed immediately before the main switch to achieve ZVS (zero volt switching) of the main switch, thereby achieving a high-efficiency, low-noise power supply.

[0003] Patent Document 1: U.S. Patent No. 10892687

[0004] However, in the control method disclosed in Patent Document 1, as in complementary control, immediately after the main switch is turned off, when current flows in the body diode connected in parallel with the resonant switch, the resonant switch is turned on. Only the resonant current that flows when the resonant switch is turned on is sent as power to the secondary side. Therefore, when the output current increases, the on-time of the main switch increases and the amplitude of the excitation current increases, the operating frequency decreases. In this case, since the time for sending power to the secondary side remains unchanged, the peak value of the current flowing on the secondary side increases by more than a proportional relationship with respect to the increase in the output current. Therefore, the effective value of the current flowing on the secondary side also increases by more than a proportional relationship with respect to the increase in the output current, which increases the loss of the resistance component of the secondary side circuit and reduces the efficiency. For example, when the secondary side diode is set to a synchronous rectifier composed of a MOSFET, the problem becomes apparent. Summary of the invention

[0005] The present invention has been made in view of the problems in the prior art. Another object of the present invention is to provide a half-wave rectification LLC converter and a control IC that can prevent an increase in circulating current associated with an increase in output current.

[0006] The half-wave rectification LLC converter and control IC of the present invention are characterized in that they have: a first series circuit, in which a main switch and a resonant switch are connected in series at both ends of a DC power supply; a second series circuit, in which a primary winding and a resonant capacitor are connected in series at both ends of the resonant switch; a transformer, which has a primary winding and a secondary winding electromagnetically coupled to the primary winding with opposite polarity; a control unit, which turns on and off the main switch and the resonant switch; a rectification and smoothing circuit, which rectifies and smoothes the voltage of the secondary winding; and an output voltage detection unit, which detects the output voltage of the rectification and smoothing circuit, and the control unit has a first drive signal, a second drive signal and a third drive signal. The first drive signal turns the main switch on and off, the second drive signal turns the resonant switch on and off during a period when the main switch is off, and the third drive signal turns the resonant switch on and off after the second drive signal. The second drive signal is generated as follows: after the main switch is turned on and off by the first drive signal, there is a flyback period in which energy stored in the transformer is released from the secondary winding, the resonant switch is turned on by the second drive signal during the flyback period to flow the resonant current, and after the resonant current ends, the resonant switch is turned off by the second drive signal during an excitation current charging period in which the resonant capacitor is charged by the excitation current.

[0007] According to the present invention, it is possible to provide a half-wave rectification LLC converter and a control IC capable of preventing an increase in circulating current associated with an increase in output current. BRIEF DESCRIPTION OF THE DRAWINGS

[0008] Figure 1 1 is a diagram showing a configuration of a half-wave rectification LLC converter operated by a control IC according to the present embodiment.

[0009] Figure 2 FIG. 1 is a diagram showing an operation waveform of a half-wave rectification LLC converter as a comparative example.

[0010] Figure 3 It is a diagram showing operation waveforms of the half-wave rectification LLC converter and the control IC according to the first embodiment.

[0011] Figure 4 1 and 2 show operation waveforms when the on timing of the resonant switch is shifted backward in the half-wave rectification LLC converter and the control IC according to the first embodiment.

[0012] Figure 5 It is a diagram showing operation waveforms of the half-wave rectification LLC converter and the control IC according to the first embodiment.

[0013] Figure 6 This is a block diagram showing a control unit according to the first embodiment.

[0014] Figure 7It is a diagram showing the configuration of the excitation current detection means according to the first embodiment.

[0015] Figure 8 It is a diagram showing the configuration of the enable signal generating unit according to the first embodiment.

[0016] Fig. 9A It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the first embodiment.

[0017] Fig. 9B It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the first embodiment.

[0018] Fig. 9C It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the first embodiment.

[0019] Fig.9D It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the first embodiment.

[0020] Fig.10 This is a block diagram of a control unit according to the second embodiment.

[0021] Fig.11 It is a diagram showing the configuration of an enable signal generating unit according to the second embodiment.

[0022] Fig.12 It is a diagram showing an operation waveform of the enable signal generating unit according to the second embodiment.

[0023] Fig.13 It is a diagram showing the characteristics of the enable signal generating unit according to the second embodiment.

[0024] Fig.14A It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the second embodiment.

[0025] Fig. 14B It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the second embodiment.

[0026] Fig. 14C It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the second embodiment.

[0027] Fig.14D It is a diagram showing the operation waveform of the half-wave rectification LLC converter according to the second embodiment.

[0028] Fig.15 It is a diagram showing the change of the operating frequency with respect to the output current in the first embodiment and the second embodiment.

[0029] Description of symbols

[0030] Vin: DC power supply; QM: main switch; QR: resonant switch; T: transformer; Np: primary winding; Ns: secondary winding; Na: auxiliary winding; Lm: excitation inductance; Llk: leakage inductance; Cr, Cv: resonant capacitor; DM, DR: body diode; Ds: diode; Ca, Co: output capacitor; Ra1, Ra2, Rs: resistor; VgsM, VgsR: drive signal; 10: half-wave LLC converter; 100a, 100b: control unit; 111: output voltage detection unit; 1 12: zero-crossing detection unit; 113: excitation current detection unit; 114, COMosc: comparator; 115: SR latch; 121: P1 generation unit; 122: P2 generation unit; 130a, 130b: enable signal generation unit; 131: subtraction circuit; 141: P0 timing generation unit; 142: disable signal generation unit; 143: P2 delay unit; 144: P1 threshold generation unit; 151: 1st drive unit; 152: 2nd drive unit; 200: output voltage detection unit. DETAILED DESCRIPTION

[0031] The following describes in detail a half-wave LLC converter 10 and a control IC according to several embodiments of the present invention with reference to the drawings. The same or corresponding parts in the drawings of the drive circuits of the embodiments are denoted by the same reference numerals and their description is omitted.

[0032] (Structure of Half-wave LLC Converter 10)

[0033] A half-wave rectification LLC converter (hereinafter, referred to as a half-wave LLC converter 10 ) is one of power supply circuit topologies of an isolation type DC-DC converter.

[0034] Figure 1 is a diagram showing the structure of a half-wave LLC converter 10. The half-wave LLC converter 10 includes a first parallel circuit in which a main switch QM and a body diode DM are connected in parallel at one end of a DC power source Vin. In addition, the half-wave LLC converter 10 includes a second parallel circuit in which a resonant switch QR and a body diode DR are connected in parallel at the other end of the DC power source Vin. The first parallel circuit and the second parallel circuit are connected in series to form a half-bridge circuit.

[0035] In addition, the half-wave LLC converter 10 has a transformer T having a primary winding Np and a secondary winding Ns. In addition, the half-wave LLC converter 10 has a series resonant circuit connected to both ends of the resonant switch QR, the series resonant circuit being a resonant reactor Lr, a resonant capacitor Cr, and a primary winding Np connected in series.

[0036] In addition, the half-wave LLC converter 10 connects a diode Ds as a rectifying element and an output capacitor Co in series to the secondary winding Ns. In addition, the half-wave LLC converter 10 is an insulating type DC-DC converter in which the polarities of the primary winding Np and the secondary winding Ns of the transformer T are connected in such a manner that the voltage generated in the secondary winding Ns when the main switch QM is turned on reversely biases the diode Ds.

[0037] The resonant reactor Lr can utilize the leakage inductance Llk of the transformer T. In addition, the body diodes DM and DR are unnecessary when the main switch QM and the resonant switch QR are semiconductor elements that can allow current to flow in opposite directions even when they are in an off state.

[0038] For example, when MOSFETs are used for the main switch QM and the resonant switch QR, current can flow in the opposite direction through the body diode generated in the structure. In addition, it is known that even if the voltage between the gate and the source is below the threshold, the current flows in the opposite direction due to the voltage between the gate and the drain in GaN-FETs (gallium nitride). In addition, it is also possible to make it unnecessary by turning on the main switch QM and the resonant switch QR while the current flows in the diode.

[0039] First, the control timing of the resonance switch QR for preventing an increase in the circulating current when the output current increases, which is a disadvantage of conventional control, will be described.

[0040] Figure 2 An example of an operation waveform of control confirmed by simulation in a conventional half-wave rectification LLC converter is shown.

[0041] exist Figure 2 In the example shown, the output voltage is 20V and the rated output current is assumed to be 3A. Figure 2 In the example shown, the on-time of the main switch QM is controlled for the higher half of the rated output current, i.e., 1.5A to 3A, and the on-time of the main switch QM is fixed and frequency controlled for the lower half of the rated output current, i.e., below 1.5A. Figure 2 In the example shown, it is assumed that the burst mode is entered when the output current is less than 1 / 10 of the rated output current. Figure 2 The example shown is an operating waveform when the output current changes from 0.3 A to 3 A when the upper limit of the operating frequency is set to 150 kHz, the circuit constants are set so that the output current is 0.3 A which is 1 / 10 of the rated output current and the operating frequency does not become an audible frequency.

[0042] like Figure 2 As shown in the figure, since the frequency of the action waveform at 0.3A is low, its time width is twice that of the other action waveforms on the horizontal axis. Figure 2 As can be seen from the operation waveform shown, zero volt switching (hereinafter referred to as ZVS) can be achieved: in the full load region, when the switch is turned on, the voltage across the switch is made zero volt.

[0043] However, in the conventional control method, after the main switch is turned off, as in general complementary control, the resonant switch is turned on when current flows in the body diode connected in parallel with the resonant switch. And the power transmission to the secondary side is only the resonant current Ire flowing when the resonant switch QR is turned on.

[0044] In addition, general complementary control refers to a control in which the resonant switch QR is turned on with a dead time so as to completely compensate for the off period of the main switch QM. As a result, during the period when the main switch QM is turned on, the difference in the voltage Vr of the resonant capacitor Cr is applied from the DC power supply Vin to the primary winding Np of the transformer T, and the excitation current increases. In addition, during the period when the resonant switch QR is turned on by the complementary control, the voltage Vr in the opposite direction is applied to the primary winding Np of the transformer T, so the excitation current decreases.

[0045] Therefore, when the output current Io increases, the on-time of the main switch QM increases, and the amplitude ILmp of the excitation current ILm increases, the operating frequency decreases. This operating frequency decrease itself is not a problem, but when the resonance period is set to Tcrlk, the time Tcrlk / 2 for transmitting power to the secondary side remains unchanged, so the peak value of the current flowing on the secondary side increases more than the proportional relationship with respect to the increase in the output current.

[0046] Therefore, the effective value of the current flowing on the secondary side also increases proportionally to the increase in the output current, increasing the loss of the resistance component of the secondary side circuit and reducing the efficiency. For example, this becomes a problem when the secondary side diode is set as a synchronous rectifier composed of a MOSFET.

[0047] Furthermore, it can be seen that the current flowing through the resonance switch QR also increases by more than a proportional relationship with respect to the increase in the output current.

[0048] This is because when the increase in the output current is controlled by increasing the excitation current, the excitation current ILm after the resonant current Ire stops flowing is not output to the secondary side. As a result, the circulating excitation current ILmr that charges the resonant capacitor Cr flows from the source to the drain of the resonant switch QR or through the body diode of the resonant switch QR, charging the resonant capacitor Cr.

[0049] The increase in the circulating excitation current ILmr after the resonant current stops flowing causes various problems. The circulating excitation current ILmr is a circulating current that circulates on the primary side, which increases the current flowing through the resonant switch QR, the second body diode, the primary winding Np of the transformer T, and the resonant capacitor Cr, increasing the loss of each element and deteriorating the efficiency.

[0050] Therefore, if the on / off timing of the resonant switch QR after the main switch is turned off is performed with the timing of the conventional complementary control, the circulating current increases when the output current increases, resulting in deterioration of efficiency. This suggests that since the slope of the excitation current decrease becomes gentle even when the output voltage decreases, the circulating current increases, and therefore, in a power supply device that makes the output voltage variable, when the output voltage is set to a lower level, the efficiency deteriorates.

[0051] (First embodiment)

[0052] Figure 3 The operation waveform when the control of the present invention is applied is shown. Figure 3 In FIG. 1 , the drive signal VgsM is a drive signal of the main switch QM. In addition, the drive signal VgsM is equivalent to the first drive signal. In addition, the voltage VdsM is a voltage of the main switch QM. In addition, the current IdM is a current of the main switch QM. In addition, the excitation current ILm is a current of the excitation inductance Lm of the transformer T. In addition, the threshold signal Vth1 is a threshold value for determining the start timing of the period T4 described later.

[0053] In addition, the drive signal VgsR is a drive signal of the resonant switch QR. In addition, the P1 signal is the first drive signal in the drive signal VgsR, and the P2 signal is the second drive signal in the drive signal VgsR. In addition, the first drive signal VgsR is equivalent to the second drive signal, and the second drive signal VgsR is equivalent to the third drive signal.

[0054] In addition, the voltage VdsR is the voltage of the resonant switch QR. In addition, the current IdR is the current of the resonant switch QR. In addition, the voltage Vrm is the voltage of the diode Ds. In addition, the current If is the current of the diode Ds.

[0055] exist Figure 3 In the example shown, there is a period T1 in which, when the resonance switch QR is in the off state, the main switch QM is turned on to increase the current of the magnetizing inductance Lm and the leakage inductance Llk, thereby charging the resonance capacitor Cr.

[0056] In addition, Figure 3In the example shown, after the period T1 ends, there is a period T2 as follows: During the period T2, the resonant switch QR remains off, and the main switch QM is turned off. During the period T2, a part of the current of the excitation inductance Lm and the current of the leakage inductance Llk are charged into the resonant capacitor Cr, and the remaining part of the excitation inductance Lm is output to the secondary side.

[0057] In addition, Figure 3 In the example shown, after the period T2 ends, there is a period T3: During the period T3, the main switch QM and the resonant switch QR remain off, and the current of the excitation inductor Lm is output to the secondary side. This period is the same as the operation of a normal flyback converter, and the period T3 corresponds to the flyback period.

[0058] In addition, Figure 3 In the example shown, in the middle of the period T3, there is a period T4 in which the main switch QM is kept off and the resonant switch QR is turned on. The period T4 is a period in which the voltage of the resonant capacitor Cr is applied to the primary winding Np, the resonant current of the resonant capacitor Cr and the leakage inductance Llk flows, and the current of a half cycle of the resonant current stops flowing and ends.

[0059] In addition, Figure 3 In the example shown, after the period T4 ends, there is a period T5 as follows: During the period T5, the excitation current is charged to the resonance capacitor Cr, and when the resonance capacitor Cr is charged, the main switch QM remains off, the resonance switch QR is turned off, and the excitation current ends at zero. The period T5 corresponds to the excitation current charging period.

[0060] In addition, Figure 3 In the example shown, after the period T5 ends, there is the following period T6: During the period T6, a resonance action is performed between the capacitance component of the resonant capacitor Cv equivalently connected between the output of the half bridge and GND and the inductance component (Lm+Llk) of the primary winding Np.

[0061] In addition, Figure 3 In the example shown, in the middle of the period T6, there is a period T7. In the period T7, the main switch QM is kept off, the resonance switch QR is turned on, and the excitation current is increased with a current in the opposite direction to the period T1.

[0062] And, in Figure 3 In the example shown, after the period T7 ends, there is a period T8 . In the period T8 , the main switch QM is kept off, the resonance switch QR is turned off, and the voltage of the main switch QM is reduced.

[0063] The control method of the half-wave LLC converter 10 according to the first embodiment is a control method having a feature of controlling the timing of the period T4 so that the time of the period T5 is equal to or shorter than the resonance period Tcrlk.

[0064] exist Figure 3 The control method of this embodiment shown and Figure 2 The conventional control method shown is characterized in that a period T3 as a flyback period is newly provided. Therefore, as the load current increases, the period T5 becomes longer in the conventional example (comparative example), but the control method of the present invention has a feature that the period T3 as a flyback period becomes longer.

[0065] In the above existing control, refer to Figure 4 , which means that the circulating current increases due to the increase in load current. In contrast, in the control of this embodiment, the increase in circulating current disappears.

[0066] Figure 4 (a) is related to Figure 2 (f) Same timing. Figure 4 (b) to (f) are, with Figure 4 (a) The same circuit constants, through the P0 signal (refer to Figure 6 ) After the main switch QM is disconnected, the simulation-based action waveform is changed when the timing of turning on the resonant switch QR is changed by the P1 signal between the time t2 to t6 when the excitation current decreases:

[0067] Figure 4 (a) is when the operation is performed under the existing control, the period of period T5 becomes 1.2×Tcrlk. In addition, Figure 4 (b) is an operation waveform when the timing of the P1 signal is delayed and the period of the period T5 is set to Tcrlk.

[0068] also, Figure 4 (c) is an operation waveform when the timing of the P1 signal is further delayed and the duration of the period T5 is set to 0.7×Tcrlk. Figure 4 (d) is an operation waveform when the timing of the P1 signal is further delayed and the duration of the period T5 is set to 0.4×Tcrlk. Figure 4 (e) is an operation waveform when the timing of the P1 signal is further delayed and the duration of the period T5 is set to 0.1×Tcrlk.

[0069] exist Figure 4 In the example shown, it can be seen that the waveform of the current IdM of the main switch QM does not change, but the current IdR of the resonance switch QR and the current If of the secondary-side diode Ds change significantly.

[0070] According to the simulated data, the average current and effective current of the current IdR of the resonant switch QR are as follows.

[0071] exist Figure 4 In (a), the average current is 210mAave and the effective current is 1192Arms. Figure 4 In (b), the average current is 209mAave and the effective current is 1061Arms. Figure 4 In (c), the average current is 207mAave and the effective current is 846Arms. Figure 4 In (d), the average current is 206mAave and the effective current is 712mArms. Figure 4 In (e), the average current is 205mAave and the effective current is 657mArms.

[0072] Furthermore, the average current and effective current of the current If of the diode Ds are as follows.

[0073] exist Figure 4 In (a), the average current is 3.0Aave and the effective current is 8.00Arms. Figure 4 In (b), the average current is 3.0Aave and the effective current is 7.33Arms. Figure 4 In (c), the average current is 3.0Aave and the effective current is 6.12Arms. Figure 4 In (d), the average current is 3.0Aave and the effective current is 5.35Arms. Figure 4 In (e), the average current is 3.0Aave and the effective current is 4.80Arms.

[0074] Since the average current IdR of the resonant switch QR is equal to the average current of the main switch QM, it can be seen that Figure 4 As the input current decreases from (a) to (e), this means that the effective current flowing through the resonant switch QR and the transformer T decreases, and the loss caused by the on-resistance of the resonant switch QR and the resistance component of the transformer T decreases, thereby improving efficiency.

[0075] In addition, this means that the further the P1 signal is delayed, the smaller the period T5 is. In the period T5, the energy of the magnetizing inductance Lm accumulated in the period T1 cannot be output to the secondary side, and the resonance capacitor Cr is charged, so that the circulating current increases. That is, the effect is shown that the period T3 in which the energy of the magnetizing inductance Lm accumulated in the period T1 can be output to the secondary side is newly provided.

[0076] Furthermore, as the P1 signal is delayed further, the period T3 becomes longer, and the stored excitation energy can be directly output to the secondary side, and the period T5 becomes shorter, thereby reducing the circulating current.

[0077] Therefore, when the pulse width of the P1 signal is Tcrlk / 2 and the P1 off timing is set to the moment when the excitation current is zero, the loss is the smallest.

[0078] Therefore, the on width of the P1 signal is set to Tcrlk / 2, and the on timing is set to be Tcrlk / 2 before the timing when the excitation current is zero. However, since a margin is required when manufacturing products, when referring to Figure 4 When the waveform is of the form of a period T3, for example, Figure 4 In the embodiment with the period T3 as in (b), the period T5 may be made shorter than the resonance period Tcrlk.

[0079] Figure 5 These are operation waveforms simulated by a model when the control method of the present invention is applied. Figure 5 The example shown is similar to Figure 2 As can be seen from the example, the peak values ​​of the current IdR of the resonant switch QR and the current If of the secondary-side rectifying element, namely the diode Ds, become lower, and the effective current decreases.

[0080] (Structure of Half-wave LLC Converter 10)

[0081] The half-wave LLC converter 10 of this embodiment relates to a control method of the main switch QM and the resonant switch QR and a control IC for controlling the main switch QM and the resonant switch QR. In the first embodiment, the control IC corresponds to the control unit 100a. In addition, as described above, the details of the circuit of the half-wave LLC converter 10 are Figure 1 as shown in the description.

[0082] The key point of control in this embodiment is to shorten the period T5 as much as possible and extend the period T3 as much as possible. In addition, the disconnection timing of the resonant switch QR is preferably the period T5 that can achieve ZVS. Considering the deviation of the on timing and the on width, the period T5 is preferably less than Tcrlk with a margin. In addition, the period T4 is 1 / 2 of the resonant period. Therefore, when the moment when the excitation current is zero is set to tz, the on time of the P1 signal is set to tz-1.5×Tcrlk to tz-Tcrlk / 2, the on width is set to be greater than Tcrlk / 2, and the disconnection time is set to before tz.

[0083] Regarding the on timing of the P1 signal, the value of the threshold signal Vth1 and the pulse width of P1 may be set so that the P1 detects the excitation current and turns on at time t4 that matches the value of the predetermined threshold signal Vth1 and turns off at period T5 after Tcrlk / 2.

[0084] (Structure of Half-wave LLC Converter 10 and Control IC)

[0085] use Figure 1 and Figure 6 , a half-wave LLC converter 10 and a control IC according to the first embodiment will be described.

[0086] like Figure 1 As shown, the half-wave LLC converter 10 has a first series circuit in which a main switch QM and a resonant switch QR are connected in series at both ends of a DC power supply Vin, and has a second series circuit in which a primary winding Np and a resonant capacitor Cr are connected in series at both ends of the resonant switch QR.

[0087] The main switch QM and the resonant switch QR are formed of MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) and have body diodes DM and DR between the drain and the source, respectively.

[0088] In addition, MOSFET has an output capacitor Coss between the drain and the source. The capacitance of this output capacitor Coss appears in the resonant operation. For the convenience of explaining the operation, Figure 1 In FIG. 1 , a resonant capacitor Cv is connected to both ends of the resonant switch QR, but no additional capacitor other than the output capacitor Coss of the MOSFET is required.

[0089] In addition, the main switch QM and the resonant switch QR may also be switch elements of compound semiconductors such as GaN (gallium nitride) and SiC (silicon carbide). In addition, the main switch QM and the resonant switch QR may also be composed of IGBT (Insulated Gate Bipolar Transistor). In addition, when the element structure does not have a body diode, it can be replaced by adding body diodes DM and DR. In this specification, for the sake of convenience, the main switch QM and the resonant switch QR are set to MOSFET.

[0090] The half-wave LLC converter 10 includes a transformer T including a primary winding Np and a secondary winding Ns electromagnetically coupled to the primary winding Np with opposite polarity. The transformer T is configured such that the coupling coefficient between the primary winding Np and the secondary winding Ns is less than 1, so that the primary winding Np has an excitation inductance Lm and the primary winding Np has a leakage inductance Llk. In addition, a diode Ds and an output capacitor Co as a rectifying element are connected in series at both ends of the secondary winding Ns of the transformer T, and a first rectifying circuit for rectifying and smoothing the voltage on the secondary side is connected. In addition, the first rectifying circuit corresponds to a rectifying and smoothing circuit.

[0091] Furthermore, the transformer T includes an auxiliary winding Na, and a second rectifying circuit including a rectifying element Da and an output capacitor Ca connected in series is connected to both ends of the auxiliary winding Na.

[0092] Furthermore, the half-wave LLC converter 10 includes an output voltage detection unit 200 that detects the output voltage Vo, a winding voltage detection unit that detects the winding voltage, and an IdM detection unit that detects the current IdM of the main switch QM.

[0093] Here, if Figure 1 As shown in FIG. 1 , the current detection unit (IdM detection unit) of the main switch QM is connected to a series circuit of a capacitor Cs and a resistor Rs at both ends of the resonant capacitor Cr, and the capacitance of the capacitor Cs is set to be less than 1 / 10 of the capacitance of the resonant capacitor Cr. A method of connecting a detection resistor in series with the resonant capacitor as in the prior art may also be used, but by Figure 1 With such a configuration, the power consumed by the resistor Rs is reduced to less than 1 / 10. In addition, the current detection unit (IdM detection unit) can also detect the current of the resonant switch QR.

[0094] The control unit 100a turns the main switch QM on and off according to the output signals of the output voltage detection unit 200 and the IdM detection unit. In addition, the control unit 100a turns the resonance switch QR on and off twice during the off period of the main switch QM by using the first pulse signal P1 and the second pulse signal P2.

[0095] like Figure 6 As shown, the control unit 100a includes a current source Icc. In addition, the control unit 100a generates a Vfb signal indicating a feedback voltage by using the output signal of the output voltage detection unit 200. Furthermore, the control unit 100a includes a comparator 114 that compares the Isense signal with the Vfb signal.

[0096] The comparator 114 is connected so that when the value of the ISense signal exceeds the value of the Vfb signal, it becomes High (high level) and is output to the R terminal of the SR-FF 115 (SR flip-flop). In addition, the POS signal is input to the S terminal of the SR-FF 115, so that the following P0 signal is output: the P0 signal sets the output of the SR-FF 115 to High by the falling edge of the POS signal, and sets the output of the SR-FF 115 to Low (low level) by the output of the comparator 114.

[0097] In addition, the control unit 100a includes a first drive unit 151, which receives the P0 signal and generates a signal for driving the main switch QM of the half-wave LLC converter 10. The first drive unit 151 controls the on-width of the drive signal VgsM for the main switch QM, thereby controlling the output voltage to a constant voltage. In addition, the drive signal VgsM corresponds to the first drive signal.

[0098] Furthermore, the control unit 100a includes an output voltage detection unit 111 that receives the auxiliary winding voltage Vna, generates a VoD signal indicating a voltage proportional to the output voltage, and outputs the VoD signal to the P1 threshold generation unit 144. The VoD signal corresponds to the second voltage signal.

[0099] In addition, the control unit 100a has a zero-crossing detection unit 112, which receives the auxiliary winding voltage Vna and outputs a ZCD signal to the P2 delay unit 143. The ZCD signal is a pulse signal of zero voltage that detects that the voltage of the auxiliary winding Na changes from a negative voltage to a positive voltage. In addition, the ZCD signal is equivalent to the zero-crossing signal. In addition, the P2 delay unit 143 is equivalent to the second pulse delay unit.

[0100] Furthermore, the control unit 100a includes an excitation current detection unit 113 that receives the auxiliary winding voltage Vna, generates an LmC signal indicating a voltage proportional to the excitation current, and outputs the LmC signal to the P1 generation unit 121. The LmC signal corresponds to the first voltage signal.

[0101] The P1 threshold value generating unit 144 generates a threshold value signal Vth1 multiplied by a preset ratio based on the VoD signal outputted from the output voltage detecting unit 111, and outputs the threshold value to the P1 generating unit 121 so that the threshold value becomes a value indicated by the threshold value signal Vth1 described later. That is, when the value of the VoD signal increases or decreases, the value of the threshold value signal Vth1 also increases or decreases proportionally. In addition, the P1 threshold value generating unit 144 is equivalent to the first pulse threshold value generating unit.

[0102] The P1 generation unit 121 sets the signal P1 to High and sends a signal of a preset pulse width based on the LmC signal of the exciting current detection unit 113 and the threshold signal Vth1 of the P1 threshold generation unit 144. The P1 generation unit 121 corresponds to the first pulse generation unit.

[0103] Here, the preset ratio in the P1 threshold value generating unit 144 and the preset pulse width in the P1 generating unit 121 are determined as follows: In addition, the time when the exciting current becomes zero is referred to as time tz.

[0104] The timing at which the P1 signal generated by the P1 generation unit 121 becomes High is 0.5×Tcrlk to 1.5×Tcrlk before the time tz. In the present embodiment, Tcrlk represents a resonance period. The P1 signal corresponds to the first pulse signal.

[0105] The timing when the P1 signal goes Low is made earlier than time tz. The pulse width of the P1 signal is set to be greater than 0.5×Tcrlk. The threshold signal Vth1 of the P1 threshold generation unit 144 and the pulse width of the P1 generation unit 121 are determined so as to generate the P1 signal with such a pulse width.

[0106] By making the above settings, even if the output voltage changes, the timing when the P1 signal becomes High is earlier than the above-mentioned 0.5×Tcrlk to 1.5×Tcrlk, and the timing when the P1 signal becomes Low is earlier than the time tz when the excitation current becomes zero. In addition, when the output voltage does not change, the threshold signal Vth1 may be a threshold voltage predetermined in a manner to achieve the timing described above. The predetermined threshold voltage is equivalent to the third voltage signal.

[0107] Furthermore, the control unit 100 a includes an enable signal generating unit 130 a that receives the P1 signal, the POS signal, and the Vfb signal, outputs an Enb signal that is a pulse signal based on the voltage of the Vfb signal, and outputs it to the P2 delay unit 143 .

[0108] In addition, the control unit 100a includes a P2 delay unit 143. The P2 delay unit 143 receives the Enb signal and the ZCD signal, uses the ZCD signal when the Enb signal is High as a trigger, and outputs a P2Dy signal for turning on the resonant switch QR to the P2 generation unit 122 after a predetermined delay time.

[0109] Furthermore, the control unit 100a includes a P2 generation unit 122, which receives the P2Dy signal, outputs a P2 signal of a preset pulse width, and outputs it to the second driving unit 152. The P2 generation unit 122 corresponds to a second pulse generation unit, and the P2 signal corresponds to a second pulse signal.

[0110] Furthermore, the control unit 100a includes a P0 timing generation unit 141, which outputs a preset pulse signal, namely a P0S signal, according to the edge of the P2 signal. Furthermore, the control unit 100a includes a second driving unit 152, which receives the P1 signal and the P2 signal and outputs a driving signal VgsR for driving the resonant switch QR of the half-wave LLC converter 10.

[0111] Figure 7 The excitation current detection unit 113 is composed of an operational amplifier. Figure 7 The integration circuit shown in the figure integrates the auxiliary winding voltage Vna, and the exciting current detection unit 113 outputs an LmC signal representing a voltage proportional to the exciting current.

[0112] Figure 8 1 is a diagram showing the structure of an enable signal generating unit 130a of a control unit 100a according to a first embodiment of the present invention. The enable signal generating unit 130a includes: a reference voltage Vfbmin; and an amplifier OP-Amp, which outputs a voltage VT obtained by amplifying the voltage difference between a Vfb signal representing a feedback voltage and the reference voltage Vfbmin. In addition, the enable signal generating unit 130a includes a VRamp circuit. The VRamp circuit connects a reference current Iosc, a capacitor Cosc charged with the reference current Iosc, and a switch Sosc for resetting the capacitor Cosc in parallel. In the VRamp circuit, the switch Sosc is turned on by the P0S signal and turned off by the edge of the P1 signal, and the reference current Iosc charges Cosc to generate a ramp voltage VR.

[0113] In addition, the enable signal generating unit 130a has a comparator COMosc that connects the voltage VT to the -terminal (negative terminal) and the ramp voltage VR to the +terminal (positive terminal), and sends out a pulse signal, namely the Enb signal. When the ramp voltage VR exceeds the voltage VT, the pulse signal becomes High.

[0114] FIG. 9A to FIG. 9D 1 and 2 are operation waveforms of the half-wave LLC converter 10 to which the control of the control unit 100 a of the half-wave LLC converter 10 according to the first embodiment is applied.

[0115] FIG. 9A to FIG. 9D(a) to (e) are operation waveforms when the output current decreases as the output current moves from (a) to (e). According to the configuration of the output voltage detection unit 200 and the control unit 100a, the voltage of the Vfb signal decreases as the output current decreases.

[0116] Fig. 9A (a) shows a waveform when the output current is large, and is an operation waveform when the Vfb signal is higher than the reference voltage Vfbmin.

[0117] In period T1, at time t1, the P0S signal goes High, the SR-FF 115 is set, and the P0 signal goes High. When the drive signal VgsM goes High and the main switch QM is turned on, the current of the main switch QM increases, and the voltage of the Isense signal rises.

[0118] When the voltage of the Isense signal reaches the voltage of the Vfb signal at time t2, the comparator output becomes High, the SR-FF 115 is reset, and the P0 signal becomes Low. In addition, the drive signal VgsM becomes Low, and the main switch QM is turned off.

[0119] During the period T1, when the voltage of the resonant capacitor Cr is set to Vcr, the difference between the voltage of the DC power supply Vin and Vcr is applied to the leakage inductance Llk and the excitation inductance Lm. Therefore, the current Itran of the primary winding Np of the transformer T is expressed by the following formula (1), and increases with the passage of time. The excitation inductance Lm and the leakage inductance Llk both accumulate the current of Itran. In addition, a negative voltage proportional to the voltage difference between Vin and Vcr is generated for the auxiliary winding voltage Vna.

[0120] [Formula 1]

[0121] Itran={(Vin-Vcr) / (Lm+Llk)}×t...(1)

[0122] When the main switch QM is turned off at time t2, the auxiliary winding voltage Vna changes from negative to positive through zero, but since the Enb signal is Low, the output of the P2 delay unit 143 remains zero.

[0123] In the period T2, when QL is turned off at time t2, the energy accumulated in the magnetizing inductance Lm and the leakage inductance Llk generates a counter electromotive force.

[0124] Since the leakage inductance Llk is not coupled to the secondary winding Ns, it charges the clamp capacitor Cr together with a part of the energy of the excitation inductance Lm, and is not output from the secondary winding Ns to the first rectifier circuit. On the other hand, the excitation inductance Lm is an inductance coupled to the secondary winding Ns. Therefore, the back electromotive force is generated in the direction of turning on the diode Ds connected to the secondary winding Ns of the transformer T, and the remaining part of the energy of the excitation inductance Lm charged to the resonance capacitor Cr is output to the secondary side.

[0125] Then, when the leakage energy is fully charged into the resonance capacitor Cr at time t3, the energy of the magnetizing inductance Lm is fully output to the secondary side.

[0126] In conventional control, the resonance switch QR is turned on during the period T2 , but in the present embodiment, the resonance switch QR is basically not turned on during the period T2 , which is different from conventional control.

[0127] However, when the load current decreases and the voltage of the Vfb signal decreases, the period T1 becomes shorter, and as a result, when the maximum value of the LmC signal approaches the value of the threshold signal Vth1, the resonant switch QR is turned on during the period T2. However, such an operation is also included in the scope of this embodiment.

[0128] During the period T3, both the main switch QM and the resonant switch QR are turned off. At time t3, the energy stored in the leakage inductance Llk becomes zero, and the energy stored in the magnetizing inductance Lm is output from the secondary winding Ns to the first rectifier circuit. This period is the same as the operation of a normal flyback converter, and the period T3 corresponds to the flyback period.

[0129] During periods T2 and T3, since the energy stored in the excitation inductance Lm is output from the secondary winding Ns to the first rectifier circuit, a voltage proportional to the output voltage is generated with respect to the auxiliary winding voltage Vna, and the output of the excitation current detection unit 113 decreases in proportion to the excitation current.

[0130] In the period T4 , at time t4 , when the value of the LmC signal decreases and becomes equal to the threshold signal Vth1 , the P1 generating unit 121 turns on the resonance switch QR.

[0131] Since the energy of the exciting inductance Lm is output to the secondary side, when the resonant switch QR is turned on, a resonant current flows through the resonant capacitor Cr and the leakage inductance Llk.

[0132] Therefore, the P1 generation unit 121 has an on width of 1 / 2 or more of the resonance period Tcrlk of the resonance capacitor Cr and the leakage inductance Llk. In addition, the time t6 when the excitation current becomes zero is set to tz, the P1 threshold generation unit 144 sets the value of the threshold signal Vth1, and the P1 generation unit 121 sets the pulse width of the P1 signal so that the P1 signal is disconnected between (tz-Tcrlk) and the time tz.

[0133] During period T5, after 1 / 2Tcrlk has passed since the resonant switch QR was turned on, the resonant current is limited by the excitation current, and the excitation current charges the resonant capacitor Cr through the resonant switch QR. In addition, during period T5, the current flows from the source to the drain of the resonant switch QR. Period T5 is equivalent to the excitation current charging period.

[0134] Since the P1 signal is generated under the conditions described above, the P1 signal is turned off during the period T5. During this period, even if the resonance switch QR is turned off, the current path does not change because the current is commutated to the body diode DR after the switch is turned off.

[0135] When the excitation current becomes zero at time t6, the period T5 ends. During the period T6, the excitation current becomes zero at time t6, but the resonant switch QR is turned off, so the excitation current does not flow in the negative direction as in the general complementary control. Therefore, resonance occurs between the resonant capacitor Cv and the inductance of the primary winding Np of the transformer T, that is, the excitation inductance Lm and the leakage inductance Llk, and the main switch QM, the resonant switch QR, and the voltage of each winding vibrates at the resonance period Tcvlm.

[0136] In the enable signal generating unit 130a, the ramp voltage VR starts to rise when the P1 signal is turned off. Since the ramp voltage VR can detect the first zero crossing after the excitation current becomes zero, it may be configured to start to rise when the output of the excitation current detecting unit 113, that is, the LmC signal, becomes zero.

[0137] exist Fig. 9A In (a), since the voltage of the Vfb signal is higher than the voltage of the reference voltage Vosc, the voltage VT of the enable signal generating unit 130a is zero. In the period T5, when the P1 signal becomes Low, the enable signal, namely the Enb signal, becomes High.

[0138] Since the P2 delay unit 143 receives the ZCD signal after the Enb signal becomes High, and sends a signal with a preset pulse width, Fig. 9A In (a), a signal P2Dy having a preset pulse width is output based on the initial ZCD signal.

[0139] Regarding the preset pulse width, since the ZCD signal detects the zero crossing of the auxiliary winding voltage Vna oscillating with a period of Tcvlm from negative to positive, the pulse width of the P2Dy signal is preset to Tcvlm / 4. As a result, the time t7 of the edge in the pulse signal of the P2Dy signal becomes the minimum value of the voltage across the resonant switch QR.

[0140] In this example, it is explained that the zero-crossing detection unit 112 detects the zero-crossing of the auxiliary winding voltage Vna from negative to positive, and the P2Dy signal is Tcvlm / 4. The zero-crossing detection unit 112 detects the zero-crossing of the auxiliary winding voltage Vna from positive to negative, and even if the pulse width of the P2Dy signal is set to (3 / 4)×Tcvlm, the time t7 of the edge in the pulse signal of the P2Dy signal becomes the minimum value of the voltage across the resonant switch QR.

[0141] The period T7 is a period during which the resonance switch QR is turned on and the excitation current in the negative direction flows. The time t7 is an edge of the P2Dy signal. The P2 generation unit 122 generates a signal P2 of a preset pulse width using the edge of the P2Dy signal and outputs it to the second drive unit 152 .

[0142] The preset pulse width is set so that the negative current flows through the inductance of the primary winding Np of the transformer T during the period T7 to thereby store energy, and the resonance capacitor Cv is charged to a voltage equal to or higher than the voltage of the DC power supply Vin, thereby setting the main switch QM to zero volts.

[0143] The second turning-on timing of the resonance switch QR in the present embodiment is the same as the conventional non-complementary control, but the resonance current does not flow.

[0144] Therefore, when the voltage of the resonance capacitor Cr is set to the voltage Vr, the on width Ton2 of the P2 signal, which is a pulse for making the capacitance of the resonance capacitor Cv zero, is expressed by the following equation (2).

[0145] [Formula 2]

[0146]

[0147] The average value of the voltage Vr of the resonance capacitor Cr is N=Np / Ns times the turns ratio of the output voltage Vo. Therefore, when Vr=N·Vo is considered, it is expressed by the following equation (3).

[0148] [Formula 3]

[0149]

[0150] Therefore, when the on-width of Ton2 that enables ZVS when the maximum value of the input voltage is Vinmax is set to Ton2max, it is expressed by the following equation (4).

[0151] [Formula 4]

[0152]

[0153] Furthermore, the P2 generation unit 122 generates a P2 signal having a preset pulse width Ton2max using the edge of the P2Dy signal and outputs the signal to the drive unit 2 , thereby achieving ZVS even when the input voltage decreases.

[0154] The above is an example of setting, and the on-width given by the above formula when the input voltage is Vintyp less than Vinmax can also be pre-set to Ton2typ. In this case, there are the following advantages: when the input voltage is the maximum value, it is turned on at the minimum point of the voltage instead of turning on at ZVS, but the current in the negative direction can be reduced and higher efficiency can be achieved when the input voltage is below Vtyp.

[0155] In addition, since the pulse width of the optimal Ton2 is correlated with the output voltage and the input voltage, the P2 generation unit 122 uses a unit that detects the input voltage and the output voltage and corrects Ton2 using the input and output voltages, thereby achieving the optimal ZVS. In addition, even if the input voltage and the output voltage are changed, the P2 generation unit 122 can achieve the optimal ZVS and can constitute a high-efficiency power supply.

[0156] In period T8, the P0 timing generating unit 141 sends a P0S signal of a preset pulse width according to the edge of the P2 signal. The P0 signal becomes High by the edge of the P0S signal, and the main switch QM is turned on, thereby ending.

[0157] This is a period during which, when the resonance switch QR is turned off at time t8 , the resonance capacitor Cv is discharged by the negative current flowing through the inductance of the primary winding Np.

[0158] During this period T8, the inductance component of the primary winding Np and the resonant capacitor Cv resonate. When the resonant period is Tcvlm, during the period T8, the time DT from when the voltage of the main switch QM drops due to the voltage resonance and the resonant switch QR is turned off to when the resonant voltage reaches the minimum point is expressed by the following formula (5).

[0159] [Formula 5]

[0160]

[0161] The P0 timing generating unit 141 sends out a POS signal having a pulse width preset also based on the above-mentioned formula according to the edge of the P2 signal.

[0162] Furthermore, since Vac=N·Vo, the time DT is expressed by the following equation (6).

[0163] [Formula 6]

[0164]

[0165] The P0 timing generation unit 141 uses a unit that detects the input voltage and the output voltage and corrects the pulse width of the POS signal under the input and output conditions, thereby realizing a more preferable ZVS, and the input voltage changes, and a high-efficiency power supply can be formed. Therefore, the P0 timing generation unit 141 may also use a unit that detects the input voltage and the output voltage and corrects the pulse width of the POS signal under the input and output conditions.

[0166] Fig. 9B (b) is the output current ratio Fig. 9A Since the voltage of the Vfb signal is lower than that of (a), the current peak value, that is, IdMp, is controlled to a lower value compared to (a), and the period T3 becomes shorter, but the value of the Vfb signal becomes a value slightly higher than the value of the reference voltage Vfbmin, so the action after period T4 becomes the same as (a).

[0167] Fig. 9B (c) is the action waveform when the output current is further reduced, and is the action waveform when the voltage of the Vfb signal is equal to the value of the reference voltage Vfbmin. Fig. 9B Similarly, IdMp is controlled to a lower value, and the period T3 becomes shorter, but the value of the Vfb signal is equal to the value of the reference voltage Vfbmin, so the period after T4 becomes the same as Fig. 9A (a) The same action.

[0168] Fig. 9C (d) is the action waveform when the output current is further reduced, and Vfb becomes a voltage slightly lower than Vfbmin. Fig. 9C The IdMp of (d) is restricted to Fig. 9B (c) is approximately the same degree. And, Fig. 9C IdMp in (d) amplifies the small potential difference between the value of the Vfb signal and the value of the reference voltage Vfbmin, increases the voltage VT, and delays the Enb signal, thereby lengthening the period T6 and making it possible to control the output voltage to be constant.

[0169] Fig.9D (e) is the output current ratio Fig. 9CIn the operation waveform when (d) is further reduced, IdMp remains substantially the same, and the period T6 becomes longer, so that the output voltage is controlled to be constant.

[0170] As described above, the control unit 100a of the half-wave LLC converter 10 of the first embodiment includes the excitation current detection unit 113, which detects the excitation current of the transformer T to generate the timing of the time t4. The control IC has the following characteristics: based on the output of the excitation current detection unit 113, that is, the LmC signal, the timing of the period T4 is determined when the excitation current reaches the value of the predetermined threshold signal Vth1. In addition, Figure 7 This is a circuit configuration example of the exciting current detection means 113, and by integrating the winding voltage of the transformer T, a voltage proportional to the exciting current can be obtained.

[0171] In addition, as described above, the control unit 100a has: an excitation current detection unit 113, which detects the excitation current of the transformer T and outputs a first voltage signal proportional to the excitation current; and an output voltage detection unit 111, which outputs a second voltage signal proportional to the output voltage. In addition, the first voltage signal is equivalent to the LmC signal. In addition, the second voltage signal is equivalent to the VoD signal. The control unit 100a sets the on timing of the second drive signal based on the first voltage signal and the threshold value set based on the second voltage signal, thereby allowing a part of the excitation current to flow in the rectification and smoothing circuit. Therefore, when the output voltage becomes lower, the slope of the excitation current decreases becomes gentle, and when the output voltage increases, the slope of the excitation current decreases becomes steep, so the value of the threshold signal Vth1 is adjusted in proportion to the output voltage.

[0172] Furthermore, the control unit 100a is characterized in that the time when the voltage across the resonant switch QR reaches the minimum value is detected, and the start time of the period T7, i.e., the time when the resonant switch QR is turned on, is controlled to be the time when the voltage across the resonant switch QR reaches the minimum value. As a method for detecting the time when the voltage across the resonant switch QR reaches the minimum value, the control unit 100a has a zero-crossing detection unit 112, which detects the zero-crossing time of the voltage of the auxiliary winding Na and sends a zero-crossing signal (ZCD signal). In addition, the control unit 100a has a P2 generation unit 122, which sends a P2 signal for turning on the resonant switch QR after a predetermined delay time based on the ZCD signal.

[0173] Furthermore, the P2 generation unit 122 includes a P2 delay unit 143 that delays the P2Dy signal according to the signal of the enable signal generation unit 130a. Thus, the half-wave LLC converter 10 can extend the period T6 to reduce the frequency when the load current decreases.

[0174] As described above, the control unit 100a of the first embodiment includes a first output voltage control unit as a unit for controlling the output voltage, and the first output voltage control unit controls the on-width of the main switch QM. In addition, the control unit includes a second output voltage control unit as a unit for controlling the output voltage, and the second output voltage control unit controls the period T6 during which both the main switch QM and the resonant switch QR are turned off. Therefore, compared with the conventional control method, the circulating current is reduced, and ZVS can be achieved in a wide range of the output current, and a high-efficiency and low-noise power supply can be achieved.

[0175] (Second embodiment)

[0176] like FIG. 9A to FIG. 9D As shown in FIG. 1 , when observing the change in the operating frequency in the first embodiment, the switching point between the first output voltage control means for controlling the on-width of the main switch QM and the second output control means for controlling the period T6 becomes the highest frequency. For example, in order to miniaturize the power supply, it is necessary to increase the operating frequency when the output current is large to reduce the transformer T and capacitor as passive components constituting the power supply.

[0177] Hereinafter, as a second embodiment, a method of controlling the operating frequency by coordinating the first output control means and the second output control means in order to realize a more efficient and compact power supply device will be described.

[0178] Fig.10 1 is a block diagram showing a control unit 100 b for realizing control of the half-wave LLC converter 10 according to the second embodiment.

[0179] The control unit 100b of the second embodiment is different from the control unit 100a of the first embodiment in that it includes a disable signal generating unit 142. The following describes the differences from the control unit 100a of the first embodiment.

[0180] The control unit 100b includes a disable signal generating unit 142 that outputs a disable signal Dis to a P2 delay unit 143. The disable signal Dis becomes High by the POS signal and becomes Low after a preset delay time upon receiving the P1 signal.

[0181] Furthermore, the control unit 100 b includes an enable signal generating unit 130 b , which is reset by the POS signal and outputs an Enb signal which is a pulse signal based on the voltage of the Vfb signal, and outputs the Enb signal to the P2 delay unit 143 .

[0182] Furthermore, the control unit 100 b has a P2 delay unit 143 , which receives the Dis signal, the Enb signal, and the ZCD signal, uses the ZCD signal when the Dis signal is Low and the Enb signal is High as a trigger, and outputs a P2Dy signal to the P2 generation unit 122 .

[0183] Fig.11 1 is a diagram showing a configuration of an enable signal generating unit 130 b of a control unit 100 b according to the second embodiment.

[0184] The enable signal generating unit 130b includes: a reference voltage Vosc; a subtraction circuit, which outputs a voltage VT obtained by subtracting the voltage of the Vfb signal from the reference voltage Vosc; and a reference current Iosc. In addition, the enable signal generating unit 130b includes a VRamp circuit. In the VRamp circuit, a capacitor Cosc charged with the reference current Iosc and a switch Sosc for resetting the capacitor Cosc are connected in parallel, respectively. When the POS signal becomes L, Iosc charges Cosc to generate a ramp voltage VR. In addition, the enable signal generating unit 130b is composed of a comparator COMosc that connects the voltage VT to the negative terminal and connects the ramp voltage VR to the positive terminal, and sends out a pulse signal, namely, an Enb signal, which becomes High when the ramp voltage VR exceeds the voltage VT.

[0185] Fig.12 It is shown Fig.11 FIG. 1 is a diagram of an operation waveform of the enable signal generating unit 130 a. Fig.12 (a) to (d) show the operation waveforms when the output current increases as the output current moves to the right in the figure. In addition, according to the configuration of the output voltage detection unit 200 and the control unit 100b, the voltage of the Vfb signal rises as the output current increases.

[0186] The reference current Iosc is a preset current source, so the slope of the ramp voltage VR increases at a constant rate. The more the output current increases, the more the value of the Vfb signal rises. When the voltage difference with the reference voltage Vosc decreases, the voltage VT decreases, so the time until the Enb signal becomes High becomes shorter. When the value of the Vfb signal is higher than the value of the reference voltage Vosc, it becomes High at the time when the POS signal becomes Low.

[0187] Fig.13 It is shown Fig.11 The horizontal axis represents the value of the Vfb signal, and the vertical axis represents the time Enb_Delay from when the Sosc is turned off to when the Enb signal becomes High.

[0188] The reference current Iosc is a preset current source, and the slope of the ramp voltage VR increases at a constant rate. In addition, the voltage VT is the difference between the preset reference voltage Vosc and the voltage of the Vfb signal. Therefore, Fig.13 As shown in FIG. 1 , as a characteristic of the enable signal generating unit 130b of the second embodiment, the time Enb_Delay from when the switch Sosc is turned off to when the Enb signal becomes High is expressed in the following formula (7). As shown in formula (7), the time Enb_Delay is controlled to be negatively proportional to the Vfb signal with respect to the Vfb signal. Where a = -(Cosc / Iosc), b = Cosc×(Vosc / Iosc).

[0189] [Formula 7]

[0190] Enb_Delay=a×Vfb+b(a<0,B>0)…(7)

[0191] FIG. 14A to FIG. 14D 1 and 2 are operation waveforms of the half-wave LLC converter 10 to which the control unit 100 b of the half-wave LLC converter 10 according to the second embodiment is applied.

[0192] FIG. 14A to FIG. 14D (a) to (d) are operation waveforms when the output current decreases as the output current moves from (a) to (d). According to the configuration of the output voltage detection unit 200 and the control unit 100b, the voltage of the Vfb signal decreases as the output current decreases.

[0193] Fig.14A (a) shown is an operation waveform when the output current is large and the voltage of the Vfb signal is higher than the voltage of the reference voltage Vosc.

[0194] The action of period T1 and FIG. 9A to FIG. 9D The operation is the same as in the first embodiment shown.

[0195] When the main switch QM is turned off at time t2, the auxiliary winding voltage Vna changes from negative to zero, but Fig.14A In (a), since the Vfb voltage is higher than the reference voltage Vosc, the voltage VT of the enable signal generating unit 130b is 0. In addition, the enable signal Enb signal becomes High, but since the disable signal Dis is High, the output of the P2 delay unit 143 remains zero.

[0196] The actions during period T2 to period T6 are the same as FIG. 9A to FIG. 9D The operation is the same as in the first embodiment shown.

[0197] The disable signal generating unit 142 is set to become Low after a preset delay time has passed since the P1 signal is turned off. As for the preset delay time, since the P1 signal is turned off based on the time when the excitation current becomes zero, the delay time is preset so that the first zero crossing after the excitation current becomes zero can be detected. FIG. 14A to FIG. 14D In the example shown, the setting is made immediately after time t6 when the exciting current becomes zero.

[0198] exist Fig.14A In (a), since the voltage of the Vfb signal is higher than the voltage of the reference voltage Vosc, the voltage VT of the enable signal generating unit 130b is zero, and the enable signal, ie, the Enb signal, is already High.

[0199] The P2 delay unit 143 receives the ZCD signal after the Enb signal is High and the Dis signal becomes Low, and sends a signal with a preset pulse width. Fig.14A In (a) shown, the P2 delay unit 143 outputs a signal P2Dy having a preset pulse width according to the ZCD signal after the Dis signal becomes Low.

[0200] The ZCD signal output from the zero-crossing detection unit 112 detects the zero-crossing from negative to positive of the auxiliary winding voltage Vna oscillating at the resonance period Tcvlm. Therefore, the pulse width of the P2Dy signal is set to Tcvlm / 4 in advance, so that the time t7 of the edge in the pulse signal of the P2Dy signal becomes the minimum value of the voltage across the resonant switch QR.

[0201] In this example, the zero-crossing detection unit 112 detects the zero-crossing of the auxiliary winding voltage Vna from negative to positive. In addition, regarding the P2Dy signal, the zero-crossing detection unit 112 detects the zero-crossing of the Vna voltage from positive to negative, and even if the pulse width of the P2Dy signal is set to (3 / 4)×Tcvlm, the edge in the pulse signal of the P2Dy signal becomes the time t7. Therefore, the P2Dy signal becomes the minimum value of the voltage across the resonant switch QR.

[0202] The actions during period T7 and period T8 are the same as FIG. 9A to FIG. 9D The operation is the same as in the first embodiment shown.

[0203] Fig. 14B (b) is the output current ratio Fig.14A (a) is an operation waveform in which the voltage Vfb becomes a value lower than the voltage of the reference voltage Vosc when the voltage is small.

[0204] The voltage of the Vfb signal decreases, and accordingly, the current peak value of IdM, i.e., IdMp, decreases, and the period T1 becomes shorter. Since the voltage of the Vfb signal is smaller than the value of the reference voltage Vosc, the enable signal generation unit 130b outputs a voltage VT based on the difference between the values ​​of Vosc and the Vfb signal, and the Enb_Delay time becomes longer. As a result, the Enb signal becomes High at a time later than the time when the Dis signal becomes Low. Therefore, according to the ZCD signal after the Enb signal becomes High, the P2Dy signal is sent, and then the P2 turn-on signal is output. In (b), at the minimum value of the second voltage vibration, the resonant switch QR is turned on. Therefore, the periods T1 and T3 become shorter, but the period T6 becomes longer, and the operating frequency is reduced.

[0205] Fig. 14C and Fig.14D (c) and (d) are the output current ratios. Fig. 14B (b) is an operation waveform in which the voltage of the Vfb signal becomes a value further lower than the voltage of the reference voltage Vosc when the voltage further decreases.

[0206] As the voltage of the Vfb signal decreases, the current peak value of IdL further decreases accordingly, and the period T1 becomes shorter. The periods T1 and T3 become shorter, but the period T6 becomes longer, and the operation frequency further decreases.

[0207] When the operating frequencies of the half-wave LLC converter 10 of the first embodiment and the half-wave LLC converter 10 of the second embodiment are compared, when the output current is 3A, both are 106kHz and remain unchanged. When the output current is 2A, the operating frequency changes from 131kHz to 108kHz. In addition, when the output current is 1.5A, the operating frequency changes from 151kHz to 89kHz. In addition, when the output current is 1A, the operating frequency changes from 101kHz to 71kHz. And, when the output current is 0.5A, the operating frequency changes from 54kHz to 48kHz.

[0208] Fig.15 It is a diagram showing the operation frequency in the second embodiment. Fig.15 (a) in FIG. 1 is a variation of the operating frequency in the output current of the first embodiment. Fig.15 (b) in FIG. 1 is a variation of the operating frequency in the output current of the second embodiment.

[0209] Therefore, when compared with the control method of the first embodiment, the frequency variation can be suppressed, and a high-efficiency and low-noise power supply can be realized within the output variation range.

[0210] (Third embodiment)

[0211] In the first and second embodiments, a control method is described in which the resonant switch is turned on and off twice during the off period of the main switch. The characteristic is that the first on and off of the resonant switch is performed at a timing that reduces the circulating current and can provide a high-efficiency converter. The second on and off of the resonant switch is to achieve ZVS of the main switch and provide a high-efficiency and low-noise power supply. However, depending on the input voltage, output voltage, and turns ratio of the transformer, the voltage vibration during period T6 can be increased, and sometimes the second on and off of the resonant switch may not necessarily be performed. In the first and second embodiments, it is possible to change to a control method without the second on and off of the resonant switch.

[0212] (Other embodiments)

[0213] With reference to the accompanying drawings, the embodiments are described in detail, but the embodiments are not limited by the contents described in the above embodiments. In addition, the structural elements described above include structural elements and substantially the same structural elements that can be easily thought of by a person skilled in the art. Moreover, the structures described above can be appropriately combined. In addition, various omissions, substitutions or changes of the structure can be made within the scope of the gist of the embodiments.

[0214] The features of the half-wave LLC converter 10 and the control IC are described below.

[0215] The half-wave LLC converter 10 and the control IC of the first embodiment have a first series circuit in which a main switch QM and a resonant switch QR are connected in series at both ends of a DC power supply Vin. In addition, the half-wave LLC converter 10 and the control IC have a second series circuit in which a primary winding Np and a resonant capacitor Cr are connected in series at both ends of the resonant switch QR. In addition, the half-wave LLC converter 10 and the control IC have a transformer T having a primary winding Np and a secondary winding Ns electromagnetically coupled to the primary winding Np with opposite polarity. In addition, the half-wave LLC converter 10 and the control IC have a control unit 100a that turns on and off the main switch QM and the resonant switch QR. In addition, the half-wave LLC converter 10 and the control IC have: a rectifying and smoothing circuit that rectifies and smoothes the voltage of the secondary winding Ns; and an output voltage detection unit 200 that detects the output voltage of the rectifying and smoothing circuit. The control unit 100a includes a first drive signal for turning the main switch QM on and off, and a second drive signal for turning the resonant switch QR on and off during a period when the main switch QM is off. The control unit 100a is characterized in that the second drive signal is generated in such a manner that after the main switch is turned on and off by the first drive signal, there is a flyback period in which energy stored in the transformer is released from the secondary winding, the resonant switch is turned on by the second drive signal during the flyback period to flow a resonant current, and after the flow of the resonant current ends, the resonant switch is turned off by the second drive signal during an excitation current charging period in which the resonant capacitor is charged by the excitation current.

[0216] According to this configuration, the half-wave LLC converter 10 and the control IC can reduce circulating current and realize a high-efficiency power supply.

[0217] The half-wave LLC converter 10 and the control IC of the second embodiment have a first series circuit in which a main switch QM and a resonant switch QR are connected in series at both ends of a DC power supply Vin. In addition, the half-wave LLC converter 10 and the control IC have a second series circuit in which a primary winding Np and a resonant capacitor Cr are connected in series at both ends of the resonant switch QR. In addition, the half-wave LLC converter 10 and the control IC have a transformer T having a primary winding Np and a secondary winding Ns electromagnetically coupled to the primary winding Np with opposite polarity. In addition, the half-wave LLC converter 10 and the control IC have a control unit 100a that turns on and off the main switch QM and the resonant switch QR. In addition, the half-wave LLC converter 10 and the control IC have: a rectifying and smoothing circuit that rectify and smoothes the voltage of the secondary winding Ns; and an output voltage detection unit 200 that detects the output voltage of the rectifying and smoothing circuit. The control unit 100a has a first drive signal, a second drive signal, and a third drive signal, wherein the first drive signal turns the main switch QM on and off, the second drive signal turns the resonant switch QR on and off during the period when the main switch QM is off, and the third drive signal turns the resonant switch on and off after the second drive signal. In addition, the control unit 100a is characterized in that the second drive signal is generated in such a manner that after the main switch is turned on and off by the first drive signal, there is a flyback period in which energy accumulated in the transformer is released from the secondary winding, the resonant switch is turned on by the second drive signal during the flyback period to flow the resonant current, and after the resonant current ends, the resonant switch is turned off by the second drive signal during the excitation current charging period in which the resonant capacitor is charged by the excitation current.

[0218] According to this configuration, the half-wave LLC converter 10 and the control IC can reduce circulating current, thereby realizing a high-efficiency and low-noise power supply.

[0219] The half-wave LLC converter 10 and the control unit 100a of the control IC of the third embodiment may also include an excitation current detection unit 113, which detects the excitation current of the transformer T and outputs a first voltage signal proportional to the excitation current. In addition, the control unit 100a may also include an output voltage detection unit 111, which outputs a second voltage signal proportional to the output voltage. The control unit 100a may also generate the second drive signal in the following manner: the timing of turning on and off the second drive signal is set according to the first voltage signal and the threshold value set based on the second voltage signal, thereby having a flyback period in which the energy stored in the transformer is released from the secondary winding, turning on the resonant switch by the second drive signal during the flyback period to flow the resonant current, and turning off the resonant switch by the second drive signal during the excitation current charging period in which the resonant capacitor is charged by the excitation current after the resonant current ends.

[0220] According to this configuration, even if the output voltage of the half-wave LLC converter 10 and the control IC changes, the circulating current can be reduced, and a high-efficiency and low-noise power supply can be realized.

[0221] The half-wave LLC converter 10 and the control unit 100a of the control IC of the fourth embodiment may also include an excitation current detection unit 113, which detects the excitation current of the transformer T and outputs a first voltage signal proportional to the excitation current. In addition, the control unit 100a may also include a preset third voltage signal. The control unit 100a may also generate the second drive signal in the following manner: the timing of turning on and off the second drive signal is set according to the first voltage signal and the threshold value set based on the second voltage signal, thereby having a flyback period in which the energy stored in the transformer is released from the secondary winding, turning on the resonant switch by the second drive signal during the flyback period to flow the resonant current, and turning off the resonant switch by the second drive signal during the excitation current charging period in which the resonant capacitor is charged by the excitation current after the resonant current ends.

[0222] According to this configuration, the half-wave LLC converter 10 and the control IC can reduce the circulating current with a control unit that is simpler than the second embodiment, and can realize a high-efficiency and low-noise power supply.

[0223] The half-wave LLC converter 10 and the control unit 100a of the control IC of the fifth aspect may also include a zero-crossing detection unit 112 that outputs a zero-crossing signal indicating that a zero voltage is detected when the voltage of the auxiliary winding Na of the transformer T changes from a negative voltage to a positive voltage or from a positive voltage to a negative voltage. In addition, the control unit 100a may further include a second pulse generating unit that sends a second pulse signal for turning on the resonant switch QR after a predetermined delay time based on the zero-crossing signal.

[0224] According to this configuration, since the half-wave LLC converter 10 and the control IC are turned on at the minimum voltage of the resonant switch, a high-efficiency and low-noise operation can be achieved.

[0225] The half-wave LLC converter 10 and the control unit 100a of the control IC of the sixth embodiment may also include an enable signal generating unit 130a, which generates an enable signal based on the feedback voltage detected by the output voltage detecting unit 200. In addition, the control unit 100a may also include a second pulse delay unit, which is used to delay the third drive signal based on the zero-crossing signal and the enable signal. In addition, when the load current decreases, the second pulse delay unit may also delay the third drive signal to reduce the frequency.

[0226] According to this configuration, the half-wave LLC converter 10 and the control IC can extend the period and reduce the frequency when the load current decreases.

[0227] Regarding the control unit 100b of the half-wave LLC converter 10 and the control IC of the seventh embodiment, the control unit 100b may also have an enable signal generating unit 130b, which generates an enable signal based on the feedback voltage detected by the output voltage detecting unit 200. In addition, the control unit 100b may also have a disable signal generating unit 142, which generates a disable signal based on the second drive signal. In addition, the control unit 100b may also have a second pulse delay unit, which is used to delay the third drive signal based on the zero-crossing signal, the enable signal and the disable signal. In addition, the second pulse delay unit may also delay the third drive signal when the load current decreases to reduce the frequency.

[0228] According to this configuration, the half-wave LLC converter 10 and the control IC can suppress frequency fluctuations, and can realize a power supply with higher efficiency and lower noise within the output fluctuation range.

Claims

1. A half-wave rectifier LLC converter and a control IC, characterized in that: The half-wave rectifier LLC converter and control IC have: a first series circuit having a main switch and a resonant switch connected in series at both ends of a DC power supply; a second series circuit having a primary winding and a resonant capacitor connected in series at both ends of the resonant switch; a transformer having the primary winding and a secondary winding electromagnetically coupled to the primary winding with opposite polarity; a control unit that turns the main switch and the resonant switch on and off; A rectifying and smoothing circuit, which rectifies and smoothes the voltage of the secondary winding; as well as an output voltage detection unit, which detects the output voltage of the rectifying and smoothing circuit, The control unit has a first drive signal and a second drive signal, wherein the first drive signal turns the main switch on and off, and the second drive signal turns the resonant switch on and off while the main switch is off. The control unit generates the second drive signal in the following manner: after the main switch is turned on and off by the first drive signal, there is a flyback period in which energy stored in the transformer is released from the secondary winding, during which the resonant switch is turned on by the second drive signal to flow a resonant current, and after the flow of the resonant current ends, during an excitation current charging period in which the resonant capacitor is charged with energy stored in the transformer, the resonant switch is turned off by the second drive signal.

2. A half-wave rectifier LLC converter and a control IC, characterized in that: The half-wave rectifier LLC converter and control IC have: a first series circuit having a main switch and a resonant switch connected in series at both ends of a DC power supply; a second series circuit having a primary winding and a resonant capacitor connected in series at both ends of the resonant switch; a transformer having the primary winding and a secondary winding electromagnetically coupled to the primary winding with opposite polarity; a control unit that turns the main switch and the resonant switch on and off; a rectifying and smoothing circuit, which rectifies and smoothes the voltage of the secondary winding; and an output voltage detection unit, which detects the output voltage of the rectifying and smoothing circuit, The control unit includes a first drive signal, a second drive signal, and a third drive signal, wherein the first drive signal turns the main switch on and off, the second drive signal turns the resonant switch on and off while the main switch is off, and the third drive signal turns the resonant switch on and off after the second drive signal. The control unit generates the second drive signal in the following manner: after the main switch is turned on and off by the first drive signal, there is a flyback period in which energy stored in the transformer is released from the secondary winding, during which the resonant switch is turned on by the second drive signal to flow a resonant current, and after the flow of the resonant current ends, during an excitation current charging period in which the resonant capacitor is charged with energy stored in the transformer, the resonant switch is turned off by the second drive signal.

3. The half-wave rectifier LLC converter and control IC according to claim 1 or 2, characterized in that: The control unit includes: an excitation current detection unit that detects the excitation current of the transformer and outputs a first voltage signal proportional to the excitation current; and an output voltage detection unit that outputs a second voltage signal proportional to the output voltage. The control unit generates the second drive signal in the following manner: according to the first voltage signal and a threshold value set based on the second voltage signal, a turn-on timing of the second drive signal is set, an turn-on width of the second drive signal is set to be greater than half of the resonant period of the resonant current, and a turn-off timing of the second drive signal is set to be during the charging period of the excitation current.

4. The half-wave rectifier LLC converter and control IC according to claim 1 or 2, characterized in that: The control unit includes: an excitation current detection unit that detects the excitation current of the transformer and outputs a first voltage signal proportional to the excitation current; and a predetermined third voltage signal. The control unit generates the second drive signal in the following manner: based on the first voltage signal and the third voltage signal, the on timing of the second drive signal is set, the on width of the second drive signal is set to be greater than half of the resonant period of the resonant current, and the off timing of the second drive signal is set to be the charging period of the excitation current.

5. The half-wave rectified LLC converter and control IC according to claim 2, wherein: The control unit also has: a zero-crossing detection unit, which outputs a zero-crossing signal, wherein the zero-crossing signal indicates that a zero voltage is detected when the voltage of the auxiliary winding of the transformer changes from a negative voltage to a positive voltage, or from a positive voltage to a negative voltage; as well as The second pulse generating unit sends a second pulse signal for turning on the resonant switch after a predetermined delay time based on the zero-crossing signal.

6. The half-wave rectified LLC converter and control IC according to claim 5, wherein: The control unit also has: an enable signal generating unit, which generates an enable signal according to the feedback voltage detected by the output voltage detecting unit; and a second pulse delay unit, which is used to delay the third driving signal according to the zero-crossing signal and the enable signal, When the load current decreases, the second pulse delay unit delays the third driving signal to reduce the frequency.

7. The half-wave rectified LLC converter and control IC according to claim 5, wherein: The control unit also has: an enable signal generating unit, which generates an enable signal according to the feedback voltage detected by the output voltage detecting unit; as well as a disable signal generating unit, which generates a disable signal according to the second driving signal; as well as a second pulse delay unit, which is used to delay the third driving signal according to the zero-crossing signal, the enable signal and the disable signal, When the load current decreases, the second pulse delay unit delays the third driving signal to reduce the frequency.

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