Model predictive pulse mode control for multi-converter systems

Abstract

The invention relates to a method for controlling an electrical converter system (10) having at least two electrical converters (28a, 28b), comprising: determining a pulse pattern (A1, U1; A1, U2) of each electrical converter (28a, 28b) as a function of a reference converter voltage of the respective converter (28a, 28b); a2, U2), wherein each pulse pattern (A1, U1; a2, U2) comprises a switching time (A1; A2) for the respective electrical converter (28a, 28b); a2) and a switching transition (U1; u2); it is determined that the pulse pattern (A1, U1; a2, U2) and determining a measured sum current (isum) and at least one measured difference current (idiff) from the measurements in the electrical converter system (10); determining a sum current error (isum, err) and at least one difference current error (idiff, err), the sum current error (isum, err) being determined by subtracting the measured sum current (isum) from the sum current reference equation (I) and the difference current error (idiff, err) being determined by subtracting the corresponding measured difference current (idiff) from the difference current reference equation (II); determining a sum flux correction ([delta] [psi] sum) by multiplying the gain equation (III) by the sum current error (isum, err) and determining at least one difference flux correction ([delta] [psi] diff) by multiplying the gain (Ls) by the corresponding difference current error (idiff, err); mapping the sum flux correction ([delta] [psi] sum) and the at least one difference flux correction ([delta] [psi] diff) to a converter flux correction ([delta] [psi] 1, abc, [delta] [psi] 2, abc) of each electrical converter (28a, 28b); the pulse pattern (A1, U1; A1 ') of each electrical converter (28a, 28b) is modified by shifting the switching time. A2, U2) such that the converter flux correction ([delta] [psi] 1, abc, [delta] [psi] 2, abc) is compensated by the corrected pulse pattern (u1, abc, u2, abc) with the shifted switching time; and applying the modified pulse pattern (u1, abc, u2, abc) to the electrical converter (28a, 28b).

Description

Technical Field

[0001] The present invention relates to the field of high power converter control. In particular, the present invention relates to a method, a computer program, a computer readable medium and a controller for controlling an electric converter system. The present invention also relates to an electric converter system. Background Art

[0002] Traditionally, carrier-based pulse width modulation (CB-PWM) using linear control schemes (such as voltage-oriented control) has been used to solve the control problem. However, carrier-based pulse width modulation usually requires a high pulse number (i.e., the ratio between the device switching frequency and the fundamental component) to meet the harmonic requirements. For medium voltage systems, high pulse numbers are not an option due to high switching losses. In order to meet harmonic standards while also operating the converter at a low switching frequency, the use of OPP is a stopgap measure. OPP is an offline calculated switching sequence that produces the best harmonic distortion for a given number of pulses. However, in order to obtain low harmonic distortion, OPP abandons having one switching transition per phase during the modulation interval. Therefore, there is no fixed modulation interval and regularly spaced moments where the fundamental component can be conveniently sampled. This results in sampling both the fundamental component and the ripple, making the controller design more complicated. Typically, traditional linear controllers are simply tuned slowly using additional filters to not react to the ripple. Therefore, the controller is sluggish during transients. Moreover, using a linear controller with the OPP requires the OPP to have a smooth angle, thus creating an additional constraint in the optimization problem, which degrades the harmonic performance of the OPP.

[0003] To this end, the model predictive pulse mode control (MPP 3 C), see for example US 8766 570 B2 or EP 2469 692 A1. 3 MP C is a predictive control method that achieves fast closed-loop control of the OPP by manipulating the on / off instances of the OPP. 3 C solves the high bandwidth control problem by controlling the instantaneous flux of the electric machine around its (optimal) steady-state trajectory (i.e., fundamental component and ripple together). No filters or estimators are required to extract the fundamental component, and no constraints are imposed on the smoothness of the switching angles. This results in a controller with deadbeat-like performance during transients, while also having the excellent harmonic performance of OPP during steady-state operation.

[0004] As described in EP 2 891 241 B1, during large transients and disturbances, additional pulses can be inserted to shorten the settling time.

[0005] In EP 3 729 637 A1, in order to extend the method to high pulse numbers, a carrier-based MP 3C method to generate CB-PWM pattern online and then feed it to MP 3 C switch time correction mechanism.

[0006] In addition to controlling flux, MP 3 C is also suitable for controlling the current of a single converter system (see EP 3 320 609 A1) and for controlling the current of a static synchronous compensator based on a modular multilevel converter (see EP 3 142 236 A1).

[0007] WO 2019 / 137916 A1 describes a model predictive control based on an online determined pulse width modulation switching pattern.

[0008] SHEN JIE et al., “A high-performance 2x27 MVA machine test bench based on multilevel IGCT converters”, 2014 IEEE ENERGY CONVERSION CONGRESS AND EXPOSITION (ECCE), IEEE, September 14, 2014, pp. 3224 to 3233, describes a parallel connection concept for converters which also takes into account controlling the circulating current so that the output current and output mode are not affected (see Chapter IV).

[0009] US 10 516 328 B2 describes a model predictive control method for an electrical converter, which may include another control objective, such as controlling the circulating current.

[0010] EP 3 142 236 A1 describes a control method for an electrical converter in which an optimized pulse pattern is corrected by model predictive control based on a compensation flux error.

[0011] Quevedo Daniel E., "Predictive Control in Power Electronics and Drives: Basic Concepts, Theory, and Methods" in "Advanced and Intelligent Control in Power Electronics and Drives", December 31, 2014, Springer Link, pages 181 to 226 is a chapter of a book discussing model predictive control for power electronics. Summary of the invention

[0012] When working with a dual converter system or more generally a multi-converter system where each converter is controlled using an OPP, it may be necessary to account for the coupling between two or more converter line currents, either when calculating the reference current trajectory or when making decisions about the desired manipulation of switching instances of multiple OPPs.

[0013] The object of the present invention is to provide a fast and reliable model predictive control method for a multi-converter system.

[0014] This object is achieved by the subject-matter of the independent claims. Further exemplary embodiments are evident from the dependent claims and the following description.

[0015] A first aspect of the invention relates to a method for controlling an electric converter system having at least two electric converters. The at least two electric converters may be connected in series and / or in parallel. They may be connected to an electric grid and / or an electric machine. The electric converter may be a low voltage converter or a medium voltage converter adapted to handle voltages up to 1 kV or more. The method may be automatically performed by a controller of the electric converter system.

[0016] According to an embodiment of the present invention, the method comprises: determining a pulse mode for each electrical converter according to a reference converter voltage of the corresponding converter, wherein each pulse mode comprises a switching time and a switching transition of the corresponding electrical converter. An external control loop may provide a reference voltage for each electrical converter, which reference voltage may be determined, for example, based on a reference torque when connected to an electric machine and based on an active power reference and a reactive power reference when connected to a power grid. The pulse mode may comprise a sequence of switching times and switching transitions, the switching times being provided as switching angles, and the switching transitions being indicative of which level the output voltage should be switched to. The pulse modes of different electrical converters may be different. The pulse mode determined in this step may be regarded as a nominal pulse mode, which will be corrected to a corrected pulse mode later.

[0017] According to an embodiment of the invention, the method comprises: determining a sum current reference and at least one difference current reference generated by a pulse pattern. In particular, for N electrical converters, N-1 difference current references may be determined. According to the pulse pattern of each electrical converter, the ripple current reference may be determined according to the steady-state trajectory of the ripple current by direct calculation or according to a lookup table. The current reference may be a superposition of a fundamental current reference and a ripple current reference. The sum current reference may be the sum of these current references, and the difference current reference may be the difference of two such sum current references.

[0018] In particular, the effect of each of the nominal pulse patterns applied to the electrical converter system on the converter output current is estimated. A ripple current reference may be determined based on a dynamic system model. The system model may include a grid impedance estimate and may incorporate compensation for rotations of the voltage vectors of the electrical converters relative to each other due to the transformer configuration. Such phase shifts and rotations may be caused by one electrical converter connected to a star-connected winding of the transformer and another electrical converter connected to a delta-connected winding of the transformer. The ripple current reference is transformed into a sum current ripple reference and a difference current ripple reference. The sum current reference and the difference current reference may be obtained by superimposing the ripple current reference on the corresponding sum fundamental current reference and the difference fundamental current reference.

[0019] According to an embodiment of the invention, the method comprises: determining a measured sum current and at least one measured difference current from measurements in an electrical converter system. In the converter system, the output currents of the electrical converters may also be measured directly or estimated from other measurements. The measured sum current may be the sum of these currents. The measured difference current may be the difference of two such measured output currents.

[0020] According to an embodiment of the invention, the method comprises: determining a sum current error and at least one differential current error, wherein the sum current error is determined by subtracting a measured sum current from a sum current reference, and the differential current error is determined by subtracting a corresponding measured differential current from a differential current reference. An instantaneous sum current error and an instantaneous differential current error are determined by subtracting a sum current measurement and a differential current measurement from a current reference.

[0021] According to an embodiment of the invention, the method comprises determining a sum flux correction by multiplying a gain such as a sum model inductance by a sum current error and determining at least one differential flux correction by multiplying a gain such as a differential model inductance by a corresponding differential current error. The sum current error and the differential current error are converted into a sum flux correction and a differential flux correction. This step involves multiplying the two current errors by some gain terms, such as a sum inductance and a differential inductance. The method can determine the best and only set of gain terms (i.e., sum inductance and differential inductance) to be used based on the model parameters.

[0022] According to an embodiment of the invention, the method comprises: mapping the sum flux correction and at least one difference flux correction to a converter flux correction of each electrical converter. For example, in case of a three-phase system, the sum flux correction and the difference flux correction are mapped to a three-phase flux correction of each electrical converter. The mapping may take into account an appropriate rotation matrix to compensate for the configuration of the converter system. For example, an electrical converter connected to a delta-connected transformer on the secondary side will have its voltage vector rotated by 30 degrees on the primary side.

[0023] According to an embodiment of the invention, the method comprises: modifying the pulse pattern of each electrical converter by shifting the switching times such that a converter flux correction is compensated by the modified pulse pattern with the shifted switching times. The nominal pulse pattern of each electrical converter is modified by time shifting the switching moments and / or the switching times. The converter flux correction of each electrical converter, such as a three-phase flux correction, is provided to a model predictive control method, such as described in US 8 766 570 B2 or EP 2 469 692 A1, which are incorporated herein by reference.

[0024] This model prediction method can rely on a prediction model, a cost function, an optimization phase and a rolling horizon strategy. In general, a sequence of future control inputs (here, shifted switching times) is calculated over the prediction horizon so that the cost function is minimized by solving the optimization problem. Out of the sequence of control inputs, only the first element at the current time step can be applied to the electrical converter system. At the next time step, new measurements and / or estimates are obtained, and a modified sequence of future control inputs is calculated over the prediction horizon shifted by one time step. This results in a rolling horizon strategy.

[0025] The winding flux vector is controlled along the winding flux reference trajectory using a fast closed-loop control by correcting the switching times of the pulse pattern within the prediction time domain. The switching times may be shifted in time such that for each electrical converter, the difference between the flux generated by the corrected pulse pattern and the flux generated by the nominal pulse pattern at the end of the prediction time domain is equal to the converter flux correction.

[0026] According to an embodiment of the invention, the method comprises: applying a modified pulse pattern to the electrical converters. For each electrical converter, at least some of the switching transitions of the modified pulse pattern are applied to semiconductor switches of the electrical converter at shifted switching times.

[0027] The method can be applied to single-phase electrical converter systems, three-phase electrical converter systems and multi-phase electrical converter systems. The electrical converter can have a shared DC link or a separate DC link. The electrical converter can be a two-level converter, a three-level converter, and generally a multi-level converter. The electrical converter can include a controllable semiconductor switch, such as a transistor and a thyristor, which is configured to convert a first AC or DC voltage into a second AC or DC voltage, which can have two levels, three levels or more levels.

[0028] Through this method, MP 3Method C uses a fast closed-loop switching time correction mechanism to control multiple current errors of a multi-converter system using a common pulse pattern, such as described in US 8 766 570 B2 or EP 2 469 692 A1. By this method, a steady-state reference trajectory of the harmonic currents in the multi-converter system can be calculated and the measured current errors relative to the reference are converted in an optimal way into a unique flux correction for each individual electrical converter.

[0029] This method does not require a linear PI control loop or an anti-windup scheme. Contrary to the case of a linear controller, it also does not require filtering of the current measurements of the different electrical converters. Moreover, it does not separate the fundamental and harmonic components of the measured current.

[0030] According to an embodiment of the invention, a reference converter voltage of each electrical converter is determined based on a sum converter reference and at least one difference converter reference. The sum converter reference may be or may include a sum reference current and / or a sum reference voltage. The difference converter reference may be or may include a difference reference current and / or a difference reference voltage. The sum converter reference and at least one difference converter reference may be provided by an external control loop (e.g., a DC link controller, a voltage controller, and / or other active and reactive power reference generators).

[0031] According to an embodiment of the invention, the method further comprises: determining a positive phase sequence reference and a negative phase sequence reference for the sum converter reference and the at least one difference converter reference. In general, a set of three unbalanced phasors can be decomposed into symmetrical components, which are the positive phase sequence, the negative phase sequence and the zero phase sequence that are added to obtain the three unbalanced phasors. In the present case, the sum converter reference and the at least one difference converter reference can be decomposed in this way.

[0032] According to an embodiment of the invention, a reference converter voltage of each electrical converter is determined based on a positive phase sequence reference and a negative phase sequence reference. In this way, a pulse pattern for the electrical converter can be determined based on the positive phase sequence component and the negative phase sequence component of the sum and difference fundamental voltage reference for the electrical converter. The fundamental voltage reference can be calculated based on a steady-state system equation as a function of the PCC (point of common coupling) voltage and the positive phase sequence component and the negative phase sequence component of the sum and difference fundamental current reference that can be provided by an external control loop.

[0033] According to an embodiment of the present invention, the positive phase sequence reference includes a sum reference current, at least one difference reference current, a reference converter voltage and at least one difference reference converter voltage. The reference of the positive phase sequence dq current can be determined by an external control loop (e.g., a DC link controller, a voltage controller and / or other active and reactive power reference generators). The positive phase sequence and at least one difference reference converter voltage can be determined based on the positive phase sequence sum reference current and at least one positive phase sequence difference reference current.

[0034] According to an embodiment of the present invention, the negative phase sequence reference includes a reference current and a reference converter voltage. The reference of the negative phase sequence and current can be determined by an external control loop (e.g., a negative phase sequence dq current injector triggered by an asymmetric fault). The negative phase sequence reference converter voltage can be determined based on the negative phase sequence and reference current.

[0035] According to an embodiment of the invention, the pulse pattern is an optimized pulse pattern stored in a lookup table with respect to a modulation index. All pulse patterns can be calculated offline with respect to a specific optimization target (such as low harmonic distortion) and can be stored in a controller implementing the method. In particular, the switching angle (i.e., switching time) and the switching transition can be stored in a lookup table.

[0036] In this case, the modulation index of each electrical converter is determined from the respective converter voltage and / or possibly the low-pass filtered dc-link voltage, and the optimized pulse pattern is loaded from a look-up table based on the modulation index.

[0037] The phase of each electrical converter relative to the rotating frame (dq frame) may also be determined from the respective converter voltage and / or a possible low-pass filtered dc-link voltage.

[0038] Furthermore, the optimal steady-state converter line current trajectories for the model predictive control method may be stored in a lookup table together with the corresponding optimized pulse patterns.

[0039] In case of optimized pulse mode, the MP 3 C method to control the current of a converter system with more than one electrical converter. To this end, reference and current trajectories and reference difference current trajectories are calculated based on external control loops and the OPPs selected by these external control loops. The current errors measured relative to these quantities are converted into flux correction terms to be assigned to each electrical converter. MP 3 The C method uses these flux correction terms to adjust the switching times. In multi-converter systems, this extension ensures that there is no coupling between the electrical converters since the current reference and / or switching time corrections are properly accounted for. When compared to traditional linear control schemes, MP 3 C's fast transient response while preserving the spectrum of the optimized pulse mode.

[0040] According to an embodiment of the present invention, the ripple current trajectory is stored in a lookup table for each pulse mode. In the lookup table, a current ripple trajectory can be stored, which contains high-order current harmonics for the pulse mode. The steady-state trajectory of each electrical converter can be composed of a fundamental frequency component and high-order harmonics. The steady-state trajectory is the trajectory of the system state under steady-state conditions given the applied pulse mode. The high-order harmonics generated by the optimized pulse mode can be pre-calculated offline and can be stored in a lookup table together with the corresponding optimized pulse mode. Then, the high-order harmonics can be added to the fundamental component.

[0041] According to an embodiment of the invention, a sum current reference and at least one difference current reference are determined from a ripple current trajectory of the electrical converter, and in particular from a steady-state trajectory of the electrical converter. At a given time, i.e. at a converter reference angle of the electrical converter, values ​​from the steady-state trajectory are determined, and from these values, a sum current reference and at least one difference current reference are calculated. The converter reference angle is calculated as the sum of the phase of the electrical converter and the PCC voltage angle. The phase of the electrical converter can be determined by the sum reference converter voltage and the difference reference converter voltage.

[0042] According to an embodiment of the present invention, a pulse pattern is determined online during operation of an electrical converter system. For example, the pulse pattern is determined using carrier-based pulse width modulation or space vector modulation. The switching times of the pulse pattern and the sequence of switching transitions can be determined online by simulating pulse width modulation using mathematical methods. The carrier-based pulse pattern can be generated online for each electrical converter. For example, WO 2019 / 137916 A1, incorporated by reference, describes online calculation of switching commands based on pulse width modulation.

[0043] According to an embodiment of the invention, different transformer configurations between electrical converters are compensated by rotating at least one converter flux correction. Such rotation can be done by rotating the corresponding quantities in the complex plane or by using a rotation matrix in the state space. The method can also compensate for phase shifts when mapping voltage references to individual electrical converters. Rotation of the quantities of the converters relative to each other in the state space can be caused by one electrical converter connected to a star-connected winding of the transformer and another electrical converter connected to a delta-connected winding of the transformer.

[0044] According to an embodiment of the invention, the converter flux correction of each electric converter is scaled using the DC link voltage of the electric converter. Each electric converter may comprise an individual DC link. In order to take into account the effect of different DC link voltages, the converter flux correction is scaled.

[0045] According to an embodiment of the present invention, at least two electrical converters are connected in parallel. For example, at least two electrical converters are connected to the secondary winding of a transformer, which is connected to the power grid via a primary winding. As another example, at least two electrical converters are connected to the windings of an electric machine. The electric machine may be a generator, such as a generator of a wind turbine. The method relates to a multi-converter system connected to a power grid or an electric machine. In general, the method can be applied to a power grid or to an electric machine in the presence of another management controller. The method can also be applied to a power grid and an electric machine simultaneously.

[0046] According to an embodiment of the invention, at least two electrical converters are connected in series. The input of one electrical converter can be connected to the output of another electrical converter. In this case, the differential current can be zero, but other quantities (e.g., differential voltage and / or differential flux correction) can be different from zero.

[0047] Other aspects of the invention relate to a computer program for controlling an electric converter system, which, when executed by a processor, is adapted to carry out the steps of the method as described herein, and to a computer readable medium in which such a computer program is stored. The method may be executed as software in one or more processors.

[0048] The computer readable medium may be a hard disk, a USB (Universal Serial Bus) storage device, a RAM (Random Access Memory), a ROM (Read Only Memory), an EPROM (Erasable Programmable Read Only Memory) or a FLASH memory. The computer readable medium may also be a data communication network that allows downloading program code, such as the Internet. In general, the computer readable medium may be a non-transitory medium or a transient medium.

[0049] Another aspect of the invention relates to a controller for controlling an electric converter system. The controller is adapted to perform a method as described herein. The controller may include one or more processors in which the method is executed as software. The method may also be implemented at least partially or completely in hardware such as a DSP or FPGA.

[0050] Another aspect of the invention relates to an electric converter system controlled as described herein.

[0051] According to an embodiment of the invention, the electrical converter comprises at least two electrical converters connected in parallel and / or in series; and a controller as described herein.

[0052] In general, it should be understood that features of the method as described above and below may be features of the computer program, computer readable medium, controller and electrical converter as described above and below, and vice versa.

[0053] These and other aspects of the invention will be apparent from and elucidated with reference to the embodiments described hereinafter. BRIEF DESCRIPTION OF THE DRAWINGS

[0054] The subject matter of the invention will be explained in more detail in the following text with reference to exemplary embodiments illustrated in the drawings.

[0055] Figure 1 A circuit diagram of a converter system according to an embodiment of the invention is schematically shown.

[0056] Figure 2 A circuit diagram of an electrical converter is schematically shown.

[0057] Figure 3 The equivalent circuit diagram of the transformer is shown.

[0058] Figure 4 A block diagram of a controller according to an embodiment of the present invention is shown and a method according to an embodiment of the present invention is illustrated.

[0059] Figure 5 , Figure 6 , Figure 7 and Figure 8 Shows Figure 4 Block diagram of the box.

[0060] Fig. 9 and Fig.10 Shows Figure 8 Block diagram of the box.

[0061] Fig.11 and Fig.12 An equivalent circuit diagram of a converter system according to an embodiment of the present invention is shown.

[0062] Fig.13 An equivalent circuit diagram of an electric machine is shown.

[0063] The reference numerals used in the drawings and their meanings are listed in summary form in the reference numeral list. In principle, identical components have the same reference numerals in the figures. DETAILED DESCRIPTION

[0064] Converter system

[0065] Figure 1 Schematically shown is a converter system 10, which may be a wind energy conversion system, such as a gearless wind energy conversion system. The converter system 10 comprises a rotating electric machine 12, which is connected to a transformer 16 via two or more converter branches 14a, 14b.

[0066] The rotating electric machine 12 may be a generator, for example a synchronous generator or a motor. In the following, it is assumed that the rotating electric machine 12 is a synchronous generator without loss of generality. The rotating electric machine 12 may be connected to a wind turbine 18 .

[0067] The rotating electric machine 12 may have two or more independent winding systems connected to one of the converter branches 14a, 14b. The transformer 16 may have two or more secondary windings 22a, 22b powered by the converter branches 14a, 14b. The secondary winding 22a may be star-connected. The secondary winding 22b may be delta-connected.

[0068] Each converter branch 14a, 14b can be composed of a rectifier 24, a DC link 26 and an inverter 28. Both the rectifier 24 and the inverter 28 can be two-level converters or three-level converters as depicted here, or they can have even more voltage levels. The rectifier 24 and the inverter 28 can be back-to-back converters and / or 3LNPC (three-level neutral point clamped) converters. The machine-side rectifier 24 converts the AC voltage of the corresponding winding system into a DC voltage, which is supplied to the DC link 26 and the grid-side inverter 28. The inverter 28 converts the DC voltage into an AC output voltage, which is supplied to the corresponding secondary windings 22a, 22b of the transformer 16.

[0069] The transformer 16 further comprises a primary winding 30, which is connected to a grid 32. The transformer 16 may be a wye / delta-wye transformer.

[0070] The electrical converter system 10 and the rectifier 24 and the inverter 28 are controlled by a controller 34. The controller 34 may include a processor and a computer-readable medium. The computer-readable medium may store a computer program for executing the control method as described herein. Alternatively or additionally, the controller 34 may include a circuit system configured to execute the control method, for example, an FPGA, an ASIC, and / or a DSP. The controller 34 may be configured to modulate and / or switch the rectifier 24 and the inverter 28. In particular, the rectifier 24 and / or the inverter 28 may be switched using an OPP.

[0071] The control method that can be implemented by the controller 34 will be described below for the inverter 28 or more generally for the electrical converter 28 connected to the transformer 16. Figure 11 and 2. These components of the converter system 10 are emphasized in FIG. 1. However, more generally, the control method can also be applied to a rectifier 24 as an electrical converter connected to a winding system of an electric machine 12. In both cases, the AC outputs of the electrical converters 28 (or 24) are connected in parallel. In the case of two converters 28 (or 24), the converter system 10 can be represented as a dual converter system.

[0072] Even more generally, the control method may also be applied to more than two electrical converters 28 (or 24) connected in parallel or connected in series via their outputs.

[0073] Figure 2 An example of a three-level three-phase NPC (Neutral Point Clamped) electrical converter 24, 28 is shown which may be controlled by a method as described herein. The electrical converter 24, 28 comprises an NPC half-bridge 36 for each phase.

[0074] Each electrical converter 24, 28 has a total dc link voltage v dc,x The electrical converter is denoted by x∈{1,2} below. Back to Figure 1 , the secondary winding 22a connected to the first converter 28a (x=1) is connected in a Y configuration, while the secondary winding 22b connected to the second converter 28b (x=2) is connected in a delta configuration. The primary side 30 of the transformer connected to the grid 32 (more precisely, at the point of common coupling) is connected in a Y configuration. This overall configuration implies that the modulation signal of the converter 28b is phase-shifted by 30° with respect to the modulation signal of the converter 28a.

[0075] Assume that the voltages of the top dc link capacitor and the bottom dc link capacitor are equal, thereby evenly dividing the total dc link voltage, i.e., Each phase leg of the NPC converter 28a, 28b can synthesize three voltage levels relative to the neutral point N, which can be described as:

[0076]

[0077] where p∈{a,b,c} is a particular phase and u(x,p)∈{-1,0,1} is the so-called switch position of phase p. x，abc =[v x，a v x，b v x，c ] T and u x，abc =[u x，a u x，b u x，c ] T As the three-phase (converter output) voltage and the three-phase switch position, respectively, it can be obtained:

[0078]

[0079] By v pcc，abc =[v pcc，a v pcc，b v pcc，c ] T The voltage at the point of common coupling (with respect to the star point of the primary winding 30 ) is referred to as the PCC voltage for short.

[0080] like Figure 3 As shown, the grid 32 is modeled as a grid with a rigid pure sinusoidal voltage source (ie, an infinite bus) v g，abc =[v g，a v g， b v g，c ] T The series inductor L g and resistor R g Grid parameters and grid voltage sources are unknown quantities, however, the former can potentially be obtained in a slower time scale via a grid impedance estimation scheme.

[0081] Reference Frame

[0082] The standard three-phase abc frame is indicated as ξ abc =[ξ a ξ b ξ c ] T Note that the abc quantities are always indicated by their subscripts. The stationary (orthogonal) reference frame (also called the αβ reference frame) is represented by ξ αβ =[ξ α ξ β ] T indicates that, and

[0083] ξ αβ =Kξ abc ,

[0084] The Clarke transformation matrix is ​​defined as:

[0085]

[0086] Likewise, the inverse transform is defined as:

[0087] ξ abc =K -1 ξ αβ ,

[0088] The (pseudo) inverse of K is as follows:

[0089]

[0090] Whenever the subscript is removed from a quantity, assuming the αβ reference frame, ξ = ξ αβ .

[0091] The (synchronous) rotating (orthogonal) reference frame (also called dq reference frame) is given by ξ dq =[ξ d ξ q ] T indicates that, and

[0092] ξ dq =P(φ pcc )ξ abc ,

[0093] The Park transformation matrix is ​​defined as:

[0094]

[0095] The d-axis of the rotating reference frame is aligned with (i.e., synchronized with) the PCC voltage. Therefore, the angle between the d-axis and the α-axis is given by φ pcc =ω1t, where ω1 is the fundamental (angular) frequency of the grid. Typically, φ is obtained from a synchronous reference frame phase-locked loop (PLL) pcc . The inverse transform is as follows:

[0096] ξ abc =P -1 (ξ pcc )ξ dq ,

[0097] Where P(φ pcc ) has the following (pseudo) inverse:

[0098]

[0099] use

[0100]

[0101] As a clockwise rotation matrix, we get:

[0102] ξ dq =R(φ pcc )ξ αβ .

[0103] Mathematical Model of Dual Converter System

[0104] Figure 3 The transformer equivalent circuit of the dual converter system 10 in the αβ reference frame is presented. By v1=[v 1，α v 1，β ] T and v2=[v 2，α v 2，β ]T The output voltages of converters 28a, 28b are described as:

[0105]

[0106] in

[0107]

[0108] The 30° (clockwise) rotation is imposed due to the delta configuration of the converter 28b. The transformer 16 is comprised of its primary leakage inductance and secondary leakage inductance (L p and L s ) and the corresponding resistance (R p and R s The line currents of converters 28a, 28b are represented by i1 and i2, respectively.

[0109] according to Figure 3 , a dynamic model based only on known quantities can be derived:

[0110]

[0111] Where L t =2L p +L s and R t =2R p +R s , and the state space matrix is ​​defined as:

[0112]

[0113] and

[0114] If the grid voltage and grid impedance are known, the dynamic characteristics in (1) will also be calculated using v g Replace v pcc and use For replacement (R t , L t ) holds true afterwards. This dynamic characteristic based on grid estimation will also be referred to as equation (1) in the rest of the text.

[0115] The currents of the two converter branches 14a, 14b are coupled through all state space matrices in (1). Using two separate current controllers to control this dynamic system without accurately accounting for the coupling may lead to adverse effects such as unequal loading of the converter branches 14a, 14b and increased oscillatory behavior during transients. 1,abc ) changes not only i1 but also i2, so this holds.

[0116] Another representation is used by conversion to a sum and difference model. Taking ξ1 and ξ2 as the quantities of the converters 28a, 28b (x=1, 2), respectively, the sum model is defined as:

[0117] ξ sum =ξ1+ξ2, (2)

[0118] The difference model is defined as:

[0119] ξ diff =ξ1-ξ2. (3)

[0120] It is obvious that and Applying (2) and (3) to the differential equation of (1) yields:

[0121]

[0122] where i sum =i1+i2,i diff =i1-i2, v sum =v1+v2 and v diff =v1-v2 are the sum current, difference current, sum voltage and difference voltage respectively. The sum current is equal to the grid current, i sum =i g Furthermore, in the dynamic characteristics above, whenever we use For replacement (R t , L t ) can be used when g Replace v pcc Equation (4) can also be described in dq frame as:

[0123]

[0124] Control Purpose

[0125] The control objectives for the dual converter system 10 are as follows:

[0126] 1. When connected to a symmetrical grid without faults, a deterministic voltage spectrum and current spectrum are generated according to two pre-calculated OPPs during steady-state operation,

[0127] 2. Follow any current reference, which may for example involve positive and negative phase sequence active and reactive power delivery,

[0128] 3. Operate in a safe and reliable manner by maintaining a set of system variables within certain ranges, and

[0129] 4. Enable fast current response during dynamic events such as grid faults.

[0130] The harmonic requirements in the first objective must be met using the offline calculated OPP. During steady-state operation, the controller 34 should only modify the OPP to remove unmodeled disturbances. Under ideal conditions, the controller 34 should not modify the OPP at all, thereby keeping the good harmonic performance of the OPP intact. This objective can be achieved by accounting for the coupling between the two converter branches 14a, 14b as described below.

[0131] Harmonic current trajectory

[0132] Assuming the converter current reference ( and ) is provided by some external control loop to meet power delivery and safe operation requirements. The ripple (i.e., higher order harmonics) components ( and ) as a reference. The control method then superimposes the reference of the ripple on the fundamental component as follows:

[0133]

[0134] This is allowed due to the linear system model.

[0135] Will and The steady-state trajectory of the ripple is represented as . The ripple trajectory is derived based on a dynamic model that may include an estimated grid model. It is assumed that the grid impedance estimate can be used to calculate and Otherwise, set R g =0 and L g = 0. Based on the nominal pulse mode in (1) and The ripple current can be calculated as:

[0136]

[0137] in and are the fundamental components of the pulse mode of converters 28a, 28b (x=1, 2), respectively. g has only the fundamental component, which for a linear system implies that it only affects the fundamental component of the current, so it is not in (6). Due to the periodicity of the trajectory caused by the periodic input to the linear system, it can be calculated by integration according to (6) for t∈[0,T1], The steady-state trajectory and By knowing the following boundary conditions:

[0138]

[0139] Current Error and Converter Flux Correction

[0140] Based on the integration of (4) and ignoring the resistance, it is possible to predict T in the time domain p At the end, the sum and difference currents are predicted as follows:

[0141]

[0142] By respectively p The applied voltage correction Δv sum and Δv diff Adding this to the sum and difference voltages gives:

[0143]

[0144] It follows that:

[0145]

[0146] According to equation (7), it can be directly concluded that for the current (Δi sum , Δi diff ) can be modified by adjusting the sum flux and the difference flux (Δψ sum , Δψ diff ) is realized by a unique set of total corrections. sum , Δψ diff ) can be mapped to each electrical converter 28a, 28b (x=1, 2) as follows:

[0147]

[0148] These equations present a novel method for obtaining the desired individual converter flux corrections of the dual converter system 10 by multiplying the two current errors with a gain term. The set of gains defined above is derived from the model parameters.

[0149] In the final step, T p The total correction of the converter flux of each converter on T needs to be mapped to a pulse pattern correction on the same time domain. By implementing a quadratic procedure or a deadbeat variant, this mapping step can be performed for each converter 28a, 28b individually. In the case of the deadbeat variant, the time domain T is extended. p Until at least two switching transitions in each phase are taken into account. Both mapping methods are described in US Pat. No. 8,766,570 B2 or EP 2,469,692 A1, which are incorporated herein by reference.

[0150] Although the converter flux is corrected, the converter current is still the quantity that is directly controlled; in other words, there is no need to estimate the (virtual) converter flux.

[0151] In summary, it is shown that any current correction can be mapped into sum and difference flux corrections via (7). The converter flux correction can be obtained using (8). The MPS of US 8 766 570 B2 or EP 2 469 692 A1 can be used, for example. 3 Method C obtains a time correction of the pulse pattern for each converter 28a, 28b based on the assigned individual converter flux corrections.

[0152] Current-Based Model Predictive Control Algorithm for Dual Converter Systems

[0153] Figure 4 A high-level control diagram of a model predictive control method for the converter system 10 is shown. The blocks in this diagram and the following diagrams may represent modules of the controller 34 or software functionality of the control method.

[0154] The variables with subscripts ps and ns refer to the positive phase sequence component and the negative phase sequence component, respectively. ps,dq =R(φ pcc )ξ ps is transformed into a (synchronous) rotating reference frame. The negative phase sequence is transformed into a rotating reference frame rotating in the relative direction, such as ξ ns,dq =R(-φ pcc )ξ ns For quantities of interest, such as the PCC voltage, ξ can be obtained in the αβ reference frame via a positive phase sequence estimator and a negative phase sequence estimator. ps and ns During an asymmetric operating mode (eg, an asymmetric grid fault) a negative phase sequence exists.

[0155] Block 40 represents a synchronous reference frame phase locked loop (PLL) and phase sequence estimator which receives the synchronous reference frame phase locked loop (PLL) and phase sequence estimator at a common coupling point v pcc The three-phase voltage measured at the dq reference frame is used to determine the positive phase sequence component and the negative phase sequence component v of the voltage. pcc,ps,dq and v pcc,ns,dq and its angle φ pcc .

[0156] These quantities are passed to the fundamental component responsible for determining the positive phase sequence reference and The positive phase sequence reference generation block 42 is passed to the fundamental component responsible for determining the negative phase sequence reference and The negative phase sequence reference generation block 44.

[0157] All of these quantities are passed to the internal current-based model predictive controller 46, which also receives the angle φ between the d-axis and the α-axis. pcc , the angle φ pcc Provided by block 40; and the measured and current and the difference current i sum ,i diff The controller 46 determines the three-phase switch positions u of the converters 28a, 28b. 1,abc and u 2,abc .

[0158] The blocks 42, 44, 46 are explained in detail below. For convenience, the time dependency is discarded.

[0159] Normal phase sequence reference generation

[0160] Figure 5 and Figure 6 The control diagram for the positive phase sequence reference generation block 42 is shown in more detail.

[0161] Positive phase sequence dq current and The references are calculated by some external control loops (such as DC link controllers, voltage controllers and other active and reactive power reference generators). Given these quantities, the positive phase sequence reference generation block 42 determines the positive phase sequence sum converter (reference) voltage and the positive phase sequence difference converter (reference) voltage and In contrast to the ripple reference trajectory derivation, this block is based on steady-state equations with respect to the PCC voltage and ignores the estimated grid impedance model.

[0162] Citing (5), using the steady-state equation (i.e., ), for and converter voltage, it can be calculated and

[0163]

[0164] And for the difference converter voltage:

[0165]

[0166] variable and is passed to the internal current controller 46 .

[0167] Negative phase sequence reference generation

[0168] Figure 7The control diagram for the positive phase sequence reference generation block 44 is shown in more detail.

[0169] Negative phase sequence current The reference is determined by some external control loop (e.g., a negative phase sequence triggered by an asymmetric fault q current injector). Given these quantities, Figure 7 The feedforward controller shown determines the negative phase sequence converter voltage reference Figure 7 The controller portion shown is similar to Figure 5 The controller portion of the illustrated positive phase sequence reference generation block 42 is similar except that the variables are processed using a negative phase sequence.

[0170] From (5), it can be shown that the differential equation describing the converter in the dq reference frame with a negative phase sequence is as follows:

[0171]

[0172] Note that since the reference frame with the negative phase sequence is rotating clockwise, (9) is similar to (5), but with −ω1 replacing ω1.

[0173] The negative phase sequence reference is generated based on the steady-state version of (9):

[0174]

[0175] For summing the converter voltage references. If required, a difference reference can be added to each axis.

[0176] These variables and is passed to the internal current controller.

[0177] Internal current-based model predictive controller

[0178] Figure 8 The internal current-based model predictive controller 46 is shown in greater detail.

[0179] Once the converter voltage is determined and and fundamental current and references, they are passed to the (fast) internal current-based MP 3 C controller 46.

[0180] The internal controller 46 includes a modulation signal calculator 48, two mode loaders 50, a current reference determination block 52 and a mode controller 54. The converter reference angles φ1 and φ2 are calculated as the phases γ1 and γ2 and the PCC voltage angle φ pcc The sum.

[0181] The pulse mode loader 50 explained below requires a modulation index or modulation signal determined by the modulation index and modulation signal calculator 48. In a first step, the sum voltage and the difference voltage are mapped to the converter voltage as follows:

[0182]

[0183] The converter voltages are then scaled using their respective dc link voltages to calculate the modulation signal,

[0184]

[0185] Thereafter, the modulation index and phase (relative to the dq frame) are calculated as:

[0186]

[0187] The modulation signal in abc for each converter is calculated as:

[0188]

[0189] It can also be supplemented with injection of a common mode component.

[0190] Given a modulation index m x , for a single phase, each mode loader 50 supplies an array A of switching angles x =[α x,1 α x,2 …α x,d ] and the array U of switch transitions x =[u x,1 u x,2 …u x,d ], which together form the optimized pulse pattern (A x , U x ). Array and / or optimized pulse pattern (A x , U x ) are stored in a lookup table in the controller 34 and / or the pattern loader 50. The contents of the lookup table may be calculated off-line.

[0191] The optimized pulse pattern (A) determined by the pattern loader 50 x , U x ) is provided to the current reference determination block 52 and the mode controller block 54.

[0192] In some cases, it is preferred to use a carrier-based pulse width modulation method, such as the MP described in WO 2019 / 137916A1. 3C method. This method does not utilize OPP, but as an alternative, it proposes to calculate the modulation signal in abc to obtain the switching sequence. For example, the application case may be a low voltage fault ride-through. In this case, the mode loader 50 may generate a carrier-based pulse width modulation sequence, for example, as described in WO 2019 / 137916A1. Modulation signal can be used to determine the switching sequence over several modulation cycles. Then, the array A is filled with the switching angles and transitions of the PWM signal. x and U x .

[0193] Fig. 9 The current reference determination block 52 is shown in more detail. Once the appropriate pulse mode is selected, the sum current reference and the difference current reference are determined. and

[0194] As explained above, a ripple current reference may be determined and superimposed on the fundamental current reference, which is performed in block 52. The offline calculated ripple component is read out from the harmonic lookup tables 56. Each harmonic lookup table 56 contains the effect of the converter voltage on two converter branch currents. Using the superposition property of linear systems, the effect of the converter voltage in (6) can be stored separately and then added in another step to calculate the line current and ripple / harmonic reference. and ) reads the converter reference angles φ1 and φ2 and Then the sum current and the difference current and The ripples of are calculated as the sum and difference of the converter branch currents, respectively. In the final step, the ripple current reference is added to the fundamental component to calculate the current controller 54 which is used as a reference. and

[0195] In case of carrier based approach, reading the ripple reference can be skipped and only the fundamental component reference is used.

[0196] Fig.10 The pulse mode controller 54 is shown in more detail. The pulse mode controller 54 converts the nominal pulse mode (A) provided by the mode loader 50 into a pulse mode. x , U x ) is corrected to the corrected pulse mode u 1,abc and u 2,abc , then, the modified pulse pattern u 1,abc and u 2,abcis applied to the converter system 10 .

[0197] The pulse mode controller 54 acts as a high bandwidth current controller that regulates the current i sum and the differential current i diff and their (steady-state) reference and The sum current error and difference current error are defined as:

[0198]

[0199] As explained above, the required and differential flux corrections are as follows:

[0200]

[0201] Δψ diff =L s i diff,err .

[0202] Then, for each converter 28a, 28b, the sum flux correction and the difference flux correction are mapped to three-phase flux corrections as follows:

[0203]

[0204] It is important to note that due to u 2,abc The switching times will be corrected, so the flux correction must be multiplied by D- 1 to take into account the impact of the incremental configuration of converter 28b. The converter fluxes are then scaled using their respective DC link voltages as follows:

[0205]

[0206] Then, in block 58, the time correction required for each phase p to achieve the correction is obtained, wherein the transition time / switching angle A is shifted x To compensate for the flux correction Δψ x,abc This can be done using MP as explained in US 8 766 570 B2 or EP 2 469 692 A1 3 C method to complete.

[0207] The method may apply a rolling horizon, ie at each sampling instant the correction to the switching instant is calculated, but only the first instant of the optimal control input is applied to the converter 28a, 28b and the process is repeated at subsequent sampling instants.

[0208] In summary, the mode controller 54 implements the following steps:

[0209] 1. Calculate the current error as follows:

[0210]

[0211] 2. Convert the current error into sum flux correction and difference flux correction as follows:

[0212]

[0213] Δψ diff =L s i diff，err .

[0214] 3. Map the sum flux correction and the difference flux correction to the converter flux correction as follows:

[0215]

[0216] 4. Scale the flux correction of each converter using its respective DC link voltage as follows:

[0217]

[0218] 5. A time correction is calculated in block 58, for example according to the method described in US Pat. No. 8,766,570 B2 or EP 2,469,692 A1.

[0219] Fig.11 An equivalent circuit of a converter system 10 is shown in which more than two electrical converters 28 are connected in parallel to a transformer 16. The transformer may have N secondary windings 22 and one primary winding 30, each secondary winding 22 being connected to an electrical converter 28. The method as described above can be generalized to N converters 28 connected in parallel.

[0220] Fig.12 An equivalent circuit of a converter system 10 in which at least two electrical converters are connected in series to a transformer 16 is shown. The method can be generalized to a converter system 10 in which N converters are operated in series. As in the parallel case, this is achieved by defining and N-1 difference quantities for the converter quantity. Note that since each converter has the same current, the difference current is zero in the case of a series connection. However, other differences may not necessarily be zero (e.g., flux corrections assigned to the converters).

[0221] Fig.13 An equivalent circuit of a converter system 10 in which two electrical converters 28 are connected to an electric machine 12 is shown. The method can be generalized to such a converter system 10 by simply replacing the above linear system model with the linear system model of the winding system of the electric machine 12. Note that Figure 3 , Ls is replaced by Ls -M s , L p Replaced by M s , and R p =0.

[0222] Although the invention has been described and illustrated in detail in the drawings and the foregoing description, such description and illustration should be considered illustrative or exemplary rather than restrictive; the invention is not limited to the disclosed embodiments. Other variations to the disclosed embodiments can be understood and implemented by a person skilled in the art and practicing the claimed invention upon study of the drawings, the disclosure and the appended claims. In the claims, the word "comprising" does not exclude other elements or steps, and the indefinite article "a" or "an" does not exclude a plurality. A single processor or controller or other unit may perform the functions of several items described in the claims. The fact that certain measures are recited in mutually different dependent claims does not indicate that a combination of these measures cannot be used to advantage. Any reference signs in the claims should not be interpreted as limiting the scope.

Claims

1. A method for controlling an electric converter system (10) having at least two electric converters (28a, 28b), the method comprising: According to the reference converter voltage of each converter (28a, 28b) determining a pulse pattern (A1, U1; A2, U2) of a respective electric converter (28a, 28b), wherein each pulse pattern (A1, U1; A2, U2) comprises a switching time (A1; A2) and a switching transition (U1; U2) for the respective electric converter (28a, 28b); Determine the current reference generated by the pulse mode (A1, U1; A2, U2) and at least one differential current reference and determining the measured and current (i sum ) and at least one measured differential current (i diff ); wherein according to the pulse mode (A1, U1; A2, U2) of each electrical converter (28a, 28b), the ripple current reference is based on the steady-state ripple current trajectory is determined, and the current reference is a superposition of a fundamental current reference and the ripple current reference of the electrical converter (28a, 28b), wherein the sum current reference is the sum of the current references of the electrical converters, and a difference current reference is the difference of the two current references of the electrical converters, wherein the measured output current of the electrical converter (28a, 28b) is measured or estimated based on other measurements, wherein the measured and current (i sum ) is the sum of the measured output currents, and the measured difference current (i diff ) is the difference between the two measured output currents; Determine and current error (i sum,err ) and at least one differential current error (i diff,err ), where the sum of the current errors (i sum,err ) by taking the current reference from the Subtract the measured current (i sum ) is determined, and the differential current error (i diff,err ) by taking the difference current reference from The measured difference current (i diff ) is determined by By adding the gain Multiply the sum of the current errors (i sum,err ) to determine and flux correction (Δψ sum ) and by setting the gain (L s ) multiplied by the corresponding differential current error (i diff,err ) to determine at least one differential flux correction (Δψ diff ); The sum flux correction (Δψ sum ) and the at least one differential flux correction (Δψ diff ) is mapped to the converter flux correction (Δψ) of each electrical converter (28a, 28b) 1,abc , Δψ 2,abc ); The pulse pattern (A1, U1; A2, U2) of each electrical converter (28a, 28b) is modified by shifting the switching time so that the converter flux is modified (Δψ 1,abc , Δψ 2,abc ) with shifted switching time and the modified pulse mode (u 1,abc ,u 2,abc )compensate; as well as Applying the modified pulse pattern (u) to the electrical converter (28a, 28b) 1,abc ,u 2,abc ).

2. The method according to claim 1, wherein the reference converter voltage of each electrical converter (28a, 28b) According to and converter reference and at least one differential converter reference is determined, wherein the sum converter reference is a sum reference current and / or a sum reference voltage, and the difference converter reference is a difference reference current and / or a difference reference voltage.

3. The method according to any one of claims 1 and 2, further comprising: Determine the reference for and converter and at least one differential converter reference Normal phase sequence reference and / or negative phase sequence reference wherein the reference converter voltage of each electrical converter (28a, 28b) is determined according to the positive phase sequence reference and the negative phase sequence reference.

4. The method according to claim 3, The positive phase sequence reference includes and reference current At least one differential reference current and the reference converter voltage and at least one differential reference converter voltage The negative phase sequence reference includes and reference current and reference converter voltage 5. The method according to any one of the preceding claims, wherein the pulse pattern (A1, U1; A2, U2) is an optimized pulse pattern, and the optimized pulse pattern is stored in a lookup table (56) relative to the modulation index (m1, m2); wherein the modulation index (m1, m2) and / or phase of each electrical converter (28a, 28b) relative to the rotating frame (γ1, γ2) is determined according to the corresponding converter voltage and / or dc link voltage.

6. The method according to any one of the preceding claims, The steady-state ripple current trajectory Stored in a look-up table (56) for each pulse pattern (A1, U1; A2, U2).

7. The method according to claims 1 to 4, wherein during operation of the electrical converter system (10), the pulse pattern (A1, U1; A2, U2) is determined online; The pulse pattern (A1, U1; A2, U2) is determined using carrier-based pulse width modulation.

8. The method according to any one of the preceding claims, wherein the converter flux correction of each electrical converter (28a, 28b) uses the DC link voltage (v dc,1 , v dc,2 ) and is scaled.

9. The method according to any one of the preceding claims, wherein the at least two electrical converters (28) are connected in parallel; The at least two electrical converters (28) are connected to the secondary winding (22) of a transformer (16), the transformer (16) being connected to a power grid (32) via a primary winding (30) or the at least two electrical converters (28) are connected to the windings of an electric machine (12).

10. The method according to any one of the preceding claims, The at least two electrical converters (28) are connected in series.

11. A computer program for controlling an electric converter system (10), said computer program being adapted to perform the steps of the method according to any of the preceding claims when executed by a processor.

12. A computer-readable medium having stored therein the computer program according to claim 11.

13. A controller (34) for controlling an electric converter system (10), the controller (34) being adapted to perform the method according to any one of claims 1 to 10.

14. An electrical converter system (10), comprising: at least two electrical converters (28a, 28b) connected in parallel and / or in series; A controller (34) according to claim 13.

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