Method and system for calculating core loss of full-speed-domain multiphase ferrocobalt motor

Abstract

The invention discloses a core loss calculation method and system for a full-speed-domain multiphase ferrocobalt motor, the multiphase ferrocobalt motor is decoupled to a fundamental wave sub-plane and a harmonic wave sub-plane through coordinate transformation, the core loss generated by low-frequency harmonic current is reduced by using a maximum six-vector SVPWM control strategy, and the control performance is improved; the magnetic density is obtained through the sideband harmonic current under the static coordinate system, the magnetic density is substituted into a traditional iron core loss separation formula, fitting is carried out again, and the iron core loss separation formula is obtained and used for formulating motor modulation strategies under different working conditions, so that the iron core loss generated by the high-frequency harmonic current is restrained, and the efficiency of a motor control system is improved.

Description

Technical Field

[0001] The invention belongs to the field of multi-phase motor drive control, and in particular relates to a method for calculating the core loss of a multi-phase cobalt-iron alloy motor. Background Art

[0002] Multiphase permanent magnet motors have the characteristics of small torque pulsation, excellent fault tolerance, and high control flexibility. They have received widespread attention in all-electric aircraft and ship electric propulsion systems. Compared with traditional silicon steel alloy materials, cobalt-iron alloy materials have higher saturation flux, so they can achieve efficient and lightweight motor design while having better electromagnetic properties. However, cobalt-iron alloy materials will produce greater core losses. The main reason is that the magnetic flux amplitude of cobalt-iron alloy materials is much larger than that of traditional silicon steel alloy materials, and the loss is positively correlated with the magnetic flux amplitude, and cobalt-iron alloy materials have a higher loss coefficient than traditional silicon steel alloy materials. Core losses are mainly caused by fundamental current and high-frequency harmonic current, so it is of great significance to analyze the changes in core losses caused by high-frequency harmonic currents in the full speed range. Summary of the invention

[0003] In view of this, the present invention provides a method for calculating the core loss of a multi-phase cobalt-iron alloy motor in a full-speed range.

[0004] The present invention achieves the above technical objectives through the following technical means.

[0005] A method for calculating the core loss of a multiphase cobalt-iron alloy motor in the full speed range:

[0006] The multiphase cobalt-iron alloy motor is decoupled into the fundamental sub-plane and the harmonic sub-plane through coordinate transformation, and the maximum six-vector SVPWM control strategy is used to select the six large vectors closest to the reference voltage vector in the fundamental sub-plane, so that the projection of the six large vectors in the harmonic sub-plane is zero, thereby reducing the core loss caused by the low-frequency harmonic current;

[0007] The sideband harmonic voltages at one to five times the switching frequency in the stationary coordinate system are determined by the phase voltage expression. The sideband harmonic voltages in the stationary coordinate system are converted into the sideband harmonic voltages in the rotating coordinate system through coordinate transformation, and then the sideband harmonic currents in the rotating coordinate system are determined. The sideband harmonic currents in the rotating coordinate system are converted into the sideband harmonic currents in the stationary coordinate system through coordinate transformation. The magnetic flux density is obtained from the sideband harmonic currents in the stationary coordinate system, and is substituted into the traditional core loss separation formula, and refitted to obtain the core loss separation formula, which is used to formulate motor modulation strategies under different working conditions, thereby suppressing the core loss caused by high-frequency harmonic currents.

[0008] Furthermore, the projection of the six large vectors in the harmonic sub-plane is made zero. Specifically, the six large vectors closest to the reference voltage vector in the fundamental sub-plane are synthesized, the vector action time is calculated, the switching sequence of the maximum six-vector SVPWM is determined, and a centralization process is performed.

[0009] Furthermore, the phase voltage expression is obtained based on the modulation wave expression. In the maximum six-vector SVPWM control strategy, the zero vector is evenly divided to determine the modulation wave expression. Through Fourier analysis, the modulation wave expression is rewritten and simplified to determine the phase voltage expression.

[0010] Furthermore, the core loss separation formula is obtained by refitting, which is:

[0011] P Fe = x h fi α +χ c f 2 i 2 +χ e f 1.5 i 1.5

[0012] Where P Fe is the core loss, f is the frequency, i is the sideband harmonic current in the stationary coordinate system, χ h , χ c , χ e are the hysteresis loss coefficient, eddy current loss coefficient and additional loss coefficient obtained by re-fitting, and α is the hysteresis loss index.

[0013] Furthermore, the coordinate transformation is performed using a Clark transformation matrix and a Park transformation matrix of the multiphase cobalt-iron alloy motor, and the Clark transformation matrix and the Park transformation matrix are obtained according to the vector space decoupling principle.

[0014] A core loss calculation system for a full-speed multiphase cobalt-iron alloy motor, comprising:

[0015] The maximum six-vector SVPWM control module burns the maximum six-vector control strategy program into the digital signal processor through the code debugger. The digital signal processor amplifies the gate signal through the drive circuit and outputs it to the switching device to achieve the maximum six-vector control;

[0016] The core loss calculation module uses the least squares method to fit the data in MATLAB, and confirms the core loss coefficient by minimizing the sum of squares of errors, and then calculates the core loss.

[0017] An electronic device comprising a memory and a processor;

[0018] The memory is used to store computer programs;

[0019] The processor is used to execute the computer program and implement the above-mentioned method for calculating the core loss of the full-speed multi-phase cobalt-iron alloy motor when executing the computer program.

[0020] A storage medium stores a computer program, which, when executed by a processor, enables the processor to execute the above-mentioned method for calculating the core loss of a full-speed multi-phase cobalt-iron alloy motor.

[0021] The beneficial effects of the present invention are:

[0022] 1) The present invention proposes a control strategy based on maximum six-vector SVPWM, which effectively suppresses low-frequency harmonic currents and improves control performance.

[0023] 2) The present invention obtains high-frequency harmonic currents of one to five times the switching frequency, which have a greater impact on core loss, and is helpful for analyzing the changing pattern of high-frequency harmonics in the full speed domain.

[0024] 3) The present invention refits the core loss separation formula, which is beneficial to calculate the core loss generated by high-frequency harmonic current under different working conditions.

[0025] 4) By comparing the core loss generated by high-frequency current in the full speed range, the present invention can adjust the modulation strategy of the motor according to different working conditions to suppress the core loss and improve the efficiency of the motor control system. BRIEF DESCRIPTION OF THE DRAWINGS

[0026] Figure 1 A winding connection topology diagram of the three-neutral-point multi-phase permanent magnet motor of the present invention;

[0027] Figure 2 This is the maximum six-vector SVPWM control block diagram of the present invention;

[0028] FIG3( a ) is a schematic diagram of the fundamental wave sub-plane vector of the present invention;

[0029] FIG3( b ) is a schematic diagram of the z1-z2 sub-plane vector of the present invention;

[0030] FIG3( c ) is a schematic diagram of the z3-z4 sub-plane vector of the present invention;

[0031] Figure 4 This is a graph of the core loss amplitude at full speed obtained from the experiment described in the present invention;

[0032] FIG5(a) is a schematic diagram of theoretical values ​​of sideband harmonic currents at one times the switching frequency in a stationary coordinate system according to the present invention;

[0033] FIG5( b ) is a schematic diagram of experimental values ​​of sideband harmonic currents at one times the switching frequency in the stationary coordinate system of the present invention;

[0034] FIG5(c) is a schematic diagram of the theoretical value of the sideband harmonic current at twice the switching frequency in the stationary coordinate system of the present invention;

[0035] FIG5(d) is a schematic diagram of experimental values ​​of sideband harmonic current at twice the switching frequency in the stationary coordinate system of the present invention;

[0036] FIG5(e) is a schematic diagram of theoretical values ​​of sideband harmonic currents at three times the switching frequency in a stationary coordinate system according to the present invention;

[0037] FIG5( f ) is a schematic diagram of experimental values ​​of sideband harmonic current at three times the switching frequency in the stationary coordinate system of the present invention;

[0038] FIG5(g) is a schematic diagram of the theoretical value of the sideband harmonic current at four times the switching frequency in the stationary coordinate system of the present invention;

[0039] FIG5(h) is a schematic diagram of experimental values ​​of sideband harmonic current at four times the switching frequency in the stationary coordinate system of the present invention;

[0040] FIG5(i) is a schematic diagram of theoretical values ​​of sideband harmonic currents at five times the switching frequency in a stationary coordinate system according to the present invention;

[0041] FIG5(j) is a schematic diagram of experimental values ​​of sideband harmonic current at five times the switching frequency in the stationary coordinate system of the present invention;

[0042] Figure 6 This is a comparison chart of core losses caused by different high-frequency harmonic currents in the full speed range of the present invention. DETAILED DESCRIPTION

[0043] In order to make the purpose, technical solution and advantages of the present invention more clearly understood, the present invention is further described in detail below in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present invention and are not used to limit the present invention.

[0044] The method for calculating the core loss of a full-speed multiphase cobalt-iron alloy motor described in this embodiment includes three parts: maximum six-vector SVPWM control strategy, derivation of high-frequency harmonic current expression, and re-fitting of core loss. Figure 1-6 Provide specific instructions.

[0045] Step 1) The multiphase permanent magnet motor is a system with strong coupling, nonlinearity, and multiple variables, so it is necessary to control the motor through coordinate transformation. Figure 1The nine-phase permanent magnet motor control system shown in the figure obtains the Clark transformation matrix and Park transformation matrix of the multi-phase permanent magnet motor according to the vector space decoupling principle:

[0046]

[0047] Where θ is the rotor position angle, T αβ is the transformation matrix from the stationary coordinate system to the α-β coordinate system, T dq It is the transformation matrix from the α-β coordinate system to the rotating coordinate system, d represents the d-axis component of the current, voltage and flux, q represents the q-axis component of the current, voltage and flux, z1 represents the z1-axis component of the current, voltage and flux, z2 represents the z2-axis component of the current, voltage and flux, z3 represents the z3-axis component of the current, voltage and flux, z4 represents the z4-axis component of the current, voltage and flux, o1, o2 and o3 are zero-sequence components.

[0048] Step 2) Based on the Clark transformation matrix and Park transformation matrix of step 1), the nine-phase permanent magnet motor is decoupled into the fundamental wave sub-plane, the z1-z2 sub-plane and the z3-z4 sub-plane through coordinate transformation, see respectively Figure 3(a) , 3(b) , 3(c), where the z1-z2 subplane and the z3-z4 subplane do not participate in electromechanical energy conversion.

[0049] The maximum six-vector SVPWM suppresses low-frequency harmonic currents and reduces the core loss caused by low-frequency harmonic currents by selecting the six large vectors closest to the reference voltage vector in the fundamental sub-plane so that their projections in the two harmonic sub-planes (z1-z2 sub-plane and z3-z4 sub-plane) are zero, and has the maximum bus voltage utilization.

[0050] Taking the first sector (i.e., I in FIG. 3 (a)) as an example, the voltage vectors v70, v57, v71, v56, v58, and v69 are selected for synthesis, the vector action time is calculated, the switching sequence of the maximum six-vector SVPWM is determined, and a centralization process is performed to achieve zero projection of v70, v57, v71, v56, v58, and v69 in the z1-z2 subplane and the z3-z4 subplane. The vector action time is calculated in the following way:

[0051]

[0052] Where V α is the α-axis component of the voltage vector, V β is the β-axis component of the voltage vector, V z1 is the z1-axis component of the voltage vector, V z2 is the z2-axis component voltage vector, V z3 is the z3 axis component of the voltage vector, Vz4 is the z4-axis component of the voltage vector, T s is the action time of a switching cycle, U dc is the DC bus voltage, are the α-axis components of the six large vectors, are the β-axis components of the six large vectors, are the z1 axis components of the six large vectors, are the z2-axis components of the six large vectors, are the z3 axis components of the six large vectors, is the z4 axis component of the six large vectors, and T1…T6 is the action time of the six large vectors.

[0053] The control block diagram of the maximum six-vector SVPWM is as follows Figure 2 As shown in the figure, n is the feedback speed, n * is a given speed, i q is the q-axis feedback current, i d is the d-axis feedback current, is the given current of d axis, i z1 is the feedback current of z1 axis, The given current for the z1 axis, i z2 is the feedback current of the z2 axis, is the given current of the z2 axis, i z3 is the feedback current of the z3 axis, The given current of the z3 axis, i z4 is the feedback current of the z4 axis, is the given current of z4 axis, θ e is the rotor position angle, i s is the phase current.

[0054] The maximum six-vector SVPWM control strategy suppresses low-frequency harmonic currents and reduces the core loss caused by low-frequency harmonic currents; however, the core loss caused by high-frequency harmonic currents accounts for a large proportion of the total core loss, so the calculation of the core loss caused by high-frequency harmonic currents is achieved in the following way.

[0055] Step 3) Since the zero vector is evenly divided in the maximum six-vector SVPWM control strategy, the expression of the modulation wave is:

[0056]

[0057] Where a is the modulation range, and 0≤a≤1, ω o is the fundamental angular frequency.

[0058] Through Fourier analysis, the expression of the modulated wave (Formula (1)) is rewritten as follows:

[0059]

[0060] In the formula, n=6k+3, k=0, 1, 2,…

[0061] Formula (2) can be further simplified as:

[0062] y(t)≈(sin(ω o t)+μsin(3ω o t))(3)

[0063] In the formula, the intermediate quantity Intermediate

[0064] According to the modulation wave expression shown in formula (3), the A phase voltage expression of the nine-phase permanent magnet motor is:

[0065]

[0066] In the formula, ω c is the carrier angular velocity, u a is the A phase voltage, the intermediate quantity A0=2M(sin(ω o t)+μsin(3ω o t)),

[0067]

[0068] Because the sideband current harmonics at one to five times the switching frequency have a greater impact on the core loss, where ω c ±2ω o , 2ω c ±ω o 、3ω c ±2ω o , 4ω c ±ω o , 5ω c ±2ω o The high-frequency harmonic current at the frequency has the most significant impact on the core loss. Therefore, based on the A-phase voltage expression, the sideband harmonic voltage at one times the switching frequency to five times the switching frequency in the stationary coordinate system is further determined.

[0069] The sideband harmonic voltage at one-time switching frequency in the stationary coordinate system can be expressed as:

[0070] ε1≈β 1-2 cos((ω c ±2ω o )t)(5)

[0071] In the formula, the intermediate quantity J0 is the zero-order Bessel function, J1 is the first-order Bessel function, J2 is the second-order Bessel function, and J4 is the fourth-order Bessel function.

[0072] The sideband harmonic voltage at twice the switching frequency in the stationary coordinate system can be expressed as:

[0073] ε2≈β 2-1 sin(2ω c ±ω o )(6)

[0074] In the formula, the intermediate quantity

[0075] The sideband harmonic voltage at three times the switching frequency in the stationary coordinate system can be expressed as:

[0076] ε3≈β 3-2 cos((3ω c ±2ω o )t)(7)

[0077] In the formula, the intermediate quantity

[0078] The sideband harmonic voltage at four times the switching frequency in the stationary coordinate system can be expressed as:

[0079] ε4≈β 4-1 sin(4ω c ±ω o ) (8)

[0080] In the formula, the intermediate quantity

[0081] The sideband harmonic voltage at five times the switching frequency in the stationary coordinate system can be expressed as:

[0082] ε5≈β 5-2 cos((5ω c ±2ω o )t) (9)

[0083] In the formula, the intermediate quantity

[0084] Step 4) For further in-depth analysis, based on the Clark transformation matrix and the Park transformation matrix in step 1), the sideband harmonic voltage in the stationary coordinate system is converted into the sideband harmonic voltage in the rotating coordinate system through coordinate transformation.

[0085] The sideband harmonic voltage at one-time switching frequency in the rotating coordinate system is expressed as:

[0086]

[0087] Where: φ0 represents the initial torque angle;

[0088] The sideband harmonic voltage at twice the switching frequency in the rotating coordinate system is expressed as:

[0089]

[0090] The sideband harmonic voltage at three times the switching frequency in the rotating coordinate system is expressed as:

[0091]

[0092] The sideband harmonic voltage at four times the switching frequency in the rotating coordinate system is expressed as:

[0093]

[0094] The sideband harmonic voltage at five times the switching frequency in the rotating coordinate system is expressed as:

[0095]

[0096] Therefore, the sideband harmonic current expression in the rotating coordinate system can be further obtained:

[0097]

[0098]

[0099] Step 5) The sideband harmonic current expression in the rotating coordinate system is converted to the stationary coordinate system through coordinate change to obtain the sideband harmonic current expression in the stationary coordinate system:

[0100]

[0101] In the formula, the intermediate quantity L d is the d-axis inductance, L q is the q-axis inductance.

[0102] Step 6) The traditional core loss separation formula is a function of magnetic flux density, which is not conducive to calculating the core loss in the full speed range; the traditional core loss separation formula is:

[0103] P Fe =k h f α +k c f 2 B 2 +k e f 1.5 B 1.5 (35)

[0104] Where P Fe is the core loss, k h , kc , k e They are the coefficients of hysteresis loss, eddy current loss and additional loss respectively, α is the hysteresis loss index, f is the frequency, and B is the magnetic flux density.

[0105] Since the current is positively correlated with the magnetic flux density, the sideband harmonic current expression in the stationary coordinate system obtained in step 5) is substituted into the following formula:

[0106] F=Ni=HL (36)

[0107] Calculate the magnetomotive force F, and then substitute it into the following formula to calculate the magnetic flux density B:

[0108] B=μH=μF / L=μNi / L(37)

[0109] Substitute the calculated magnetic flux density B into the traditional core loss separation formula and refit it to obtain a new core loss separation formula:

[0110] P Fe = x h fi α +χ c f 2 i 2 +χ e f 1.5 i 1.5 (38)

[0111] Where i is the sideband harmonic current in the stationary coordinate system, N is the number of turns of the coil, H is the magnetic field intensity, L is the magnetic path length, μ is the vacuum permeability, and χ h , χ c , χ e These are the coefficients of hysteresis loss, eddy current loss and additional loss of the re-fitted core loss separation formula.

[0112] The core loss separation formula obtained by refitting will change with the modulation range, which is beneficial for the analysis of high-frequency core loss.

[0113] The core loss amplitude of the nine-phase cobalt-iron alloy motor in the full speed range obtained by experiment is as follows: Figure 4 As shown, it can be seen that with the increase of speed, the increase of core loss is more obvious.

[0114] The comparison between the theoretical and experimental values ​​of high-frequency current harmonics in the full speed domain is shown in Figure 5(a), (b), (c), (d), (e), (f), (g), (h), (i), and (j). The variation pattern of the experimental data is basically consistent with the theoretical derivation, indicating that the sideband harmonic current expression in the stationary coordinate system obtained in step 5) is correct.

[0115] Comparison of core losses caused by high-frequency harmonic currents at different frequencies Figure 6 As shown, as the modulation range changes, the dominant core loss changes accordingly.

[0116] The core loss calculation system of the full-speed multi-phase cobalt-iron alloy motor provided by the embodiment of the present invention includes:

[0117] The maximum six-vector SVPWM control module burns the maximum six-vector control strategy program into the DSP (Digital Signal Processing) through CCS (Code Composer Studio, code debugger). The DSP amplifies the gate signal through the drive circuit and outputs it to the switch device to achieve the maximum six-vector control;

[0118] The core loss calculation module uses the least squares method to fit the data in MATLAB, and confirms the core loss coefficient by minimizing the sum of squares of errors, and then calculates the core loss.

[0119] The phase current of the nine-phase permanent magnet motor is recorded by an oscilloscope and input into the ANSYS software. After simulation by the finite element simulation model and Fourier analysis, the core loss amplitude at each switching frequency is obtained and compared with the core loss value obtained by the new core loss separation formula.

[0120] It should be noted that the embodiments of the present invention can be implemented by hardware, software, or a combination of software and hardware. The hardware portion can be implemented using dedicated logic; the software portion can be stored in a memory and executed by an appropriate instruction execution system, such as a microprocessor or dedicated design hardware. A person of ordinary skill in the art will appreciate that the above-mentioned devices and methods can be implemented using computer executable instructions and / or contained in a processor control code, such as a carrier medium such as a disk, CD or DVD-ROM, a programmable memory such as a read-only memory (firmware), or a data carrier such as an optical or electronic signal carrier. Such code is provided on the carrier medium. The device and its modules of the present invention can be implemented by hardware circuits such as very large-scale integrated circuits or gate arrays, semiconductors such as logic chips, transistors, etc., or programmable hardware devices such as field programmable gate arrays, programmable logic devices, etc., can also be implemented by software executed by various types of processors, and can also be implemented by a combination of the above-mentioned hardware circuits and software, such as firmware.

[0121] The above embodiments are only used to illustrate the design ideas and features of the present invention, and their purpose is to enable those skilled in the art to understand the content of the present invention and implement it accordingly, and to obtain a method for calculating the core loss of a multi-phase cobalt-iron alloy motor. Therefore, any equivalent changes or modifications made based on the principles and design ideas disclosed by the present invention are within the scope of protection of the present invention.

Claims

1. A method for calculating the core loss of a multiphase cobalt-iron alloy motor in the full speed range, characterized by: The multiphase cobalt-iron alloy motor is decoupled into the fundamental sub-plane and the harmonic sub-plane through coordinate transformation, and the maximum six-vector SVPWM control strategy is used to select the six large vectors closest to the reference voltage vector in the fundamental sub-plane, so that the projection of the six large vectors in the harmonic sub-plane is zero, thereby reducing the core loss caused by the low-frequency harmonic current; The sideband harmonic voltages at one to five times the switching frequency in the stationary coordinate system are determined by the phase voltage expression. The sideband harmonic voltages in the stationary coordinate system are converted into the sideband harmonic voltages in the rotating coordinate system through coordinate transformation, and then the sideband harmonic currents in the rotating coordinate system are determined. The sideband harmonic currents in the rotating coordinate system are converted into the sideband harmonic currents in the stationary coordinate system through coordinate transformation. The magnetic flux density is obtained from the sideband harmonic currents in the stationary coordinate system, and is substituted into the traditional core loss separation formula, and refitted to obtain the core loss separation formula, which is used to formulate motor modulation strategies under different working conditions, thereby suppressing the core loss caused by high-frequency harmonic currents.

2. The method for calculating the core loss of a full-speed multiphase cobalt-iron alloy motor according to claim 1, characterized in that: The projection of the six large vectors in the harmonic sub-plane is made zero. Specifically, the six large vectors closest to the reference voltage vector in the fundamental sub-plane are synthesized, the vector action time is calculated, the switching sequence of the maximum six-vector SVPWM is determined, and a centralization process is performed.

3. The method for calculating the core loss of a full-speed multiphase cobalt-iron alloy motor according to claim 1, characterized in that: The phase voltage expression is obtained based on the modulation wave expression. In the maximum six-vector SVPWM control strategy, the zero vector is evenly divided, thereby determining the modulation wave expression. Through Fourier analysis, the modulation wave expression is rewritten and simplified to determine the phase voltage expression.

4. The method for calculating the core loss of a full-speed multiphase cobalt-iron alloy motor according to claim 1, characterized in that: The core loss separation formula is obtained by refitting, which is: P Fe =x h fi α +x c f 2 I 2 +x e f 1.5 I 1.5 Where P Fe is the core loss, f is the frequency, i is the sideband harmonic current in the stationary coordinate system, χ h , χ c , χ e are the hysteresis loss coefficient, eddy current loss coefficient and additional loss coefficient obtained by re-fitting, and α is the hysteresis loss index.

5. The method for calculating the core loss of a full-speed multiphase cobalt-iron alloy motor according to claim 1, characterized in that: The coordinate transformation is performed using the Clark transformation matrix and the Park transformation matrix of the multi-phase cobalt-iron alloy motor, and the Clark transformation matrix and the Park transformation matrix are obtained according to the vector space decoupling principle.

6. A system for implementing the method for calculating the core loss of a full-speed multiphase cobalt-iron alloy motor according to any one of claims 1 to 5, characterized in that: include: The maximum six-vector SVPWM control module burns the maximum six-vector control strategy program into the digital signal processor through the code debugger. The digital signal processor amplifies the gate signal through the drive circuit and outputs it to the switching device to achieve the maximum six-vector control; The core loss calculation module uses the least squares method to fit the data in MATLAB, and confirms the core loss coefficient by minimizing the sum of squares of errors, and then calculates the core loss.

7. An electronic device, characterized in that: including memory and processor; The memory is used to store computer programs; The processor is used to execute the computer program and implement the core loss calculation method of the full-speed multi-phase cobalt-iron alloy motor as described in any one of claims 1-5 when executing the computer program.

8. A storage medium, characterized in that: The storage medium stores a computer program, and when the computer program is executed by the processor, the processor executes the method for calculating the core loss of a full-speed multi-phase cobalt-iron alloy motor according to any one of claims 1 to 5.

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