Wireless charging system based on focus state estimation and dynamic fusion compensation

Abstract

This invention provides a wireless charging system based on focus state estimation and dynamic fusion compensation, relating to the field of wireless charging technology. The system consists of two main parts: a ground power supply unit and an on-board charging unit. The two are connected and work collaboratively via a bidirectional wireless communication link and a wireless energy transmission channel based on magnetic field coupling. The ground power supply unit receives DC power input from a DC bus converted from AC mains power and transmits energy to the on-board charging unit via magnetic field coupling; simultaneously, it receives communication signals from the on-board charging unit. The on-board charging unit receives energy from the ground power supply unit via magnetic field coupling, converts it into electrical energy, and supplies power to the vehicle's electrical load. Simultaneously, an observer-based control method is employed to estimate and dynamically compensate for the total disturbance to stabilize the output voltage of the system in real time and transmit control commands to the ground power supply unit.

Description

Technical Field

[0001] This invention relates to the field of wireless charging technology, and in particular to a wireless charging system based on focus state estimation and dynamic fusion compensation. Background Technology

[0002] With the rapid iteration of the new energy vehicle industry, electric vehicles have become a core vehicle for replacing traditional fuel vehicles and reducing carbon emissions in the transportation sector. However, the traditional wired charging mode suffers from problems such as cumbersome plugging and unplugging, interface wear and tear, safety hazards in inclement weather, and low maintenance efficiency in public settings, which hinder the further popularization of electric vehicles. Against this backdrop, wireless charging technology for electric vehicles, with its advantages of no physical contact, convenient operation, and strong environmental adaptability, has become an important development direction in the field of new energy vehicle charging. Through contactless power transmission, it effectively solves many problems of wired charging and lays the foundation for the coordinated implementation of autonomous driving, intelligent parking, and automatic charging.

[0003] In wireless charging systems for electric vehicles, energy is transferred through loosely coupled energy transfer coils. In practical applications, variations in vehicle parking position or chassis height can alter the relative positions of the transmitting and receiving coils, causing fluctuations in their mutual inductance. This change in mutual inductance directly alters the equivalent circuit parameters of the resonant network in the wireless charging system, causing a shift in the circuit's resonant frequency and severely impacting the energy transfer efficiency and control performance. Traditional control methods often lack sufficient response speed or control precision when faced with such drastic parameter changes, making it difficult to quickly restore the system to a stable operating state. This leads to output voltage oscillations, affecting charging efficiency and system reliability. Therefore, a control strategy capable of rapidly compensating for mutual inductance changes and maintaining stable and efficient system operation is urgently needed. Summary of the Invention

[0004] The technical problem to be solved by the present invention is to address the shortcomings of the prior art by providing a wireless charging control system for electric vehicles based on focus state estimation and dynamic fusion compensation, and designing an improved active disturbance rejection controller to control the H-bridge inverter in the wireless charging system for electric vehicles to suppress the impact of changes in the mutual inductance of the power transfer coil on the system stability.

[0005] To solve the above-mentioned technical problems, the technical solution adopted by the present invention is: a wireless charging system based on focus state estimation and dynamic fusion compensation, which consists of two main parts: a ground power supply device and an on-board charging device. The two are connected and work together through a two-way wireless communication link and a wireless energy transmission channel based on magnetic field coupling.

[0006] The ground power supply device is connected to a DC bus converted from AC mains power to receive DC power input, converts DC power into high-frequency AC power, and after resonance optimization, transmits energy to the on-board charging device through magnetic field coupling; at the same time, it receives communication signals from the on-board charging device through a two-way wireless communication circuit.

[0007] The on-board charging device receives energy from the ground power supply device through magnetic field coupling and converts it into electrical energy. It then rectifies the electrical energy to output DC power, regulates the voltage of the DC power, and supplies power to the vehicle's electrical load. At the same time, it adopts an observer-based control method to estimate and dynamically compensate for the total disturbance of the system in real time to stabilize the output voltage and transmits control commands to the ground power supply device.

[0008] Furthermore, the ground power supply device includes an H-bridge inverter A, a primary-side LCC resonant network B, a primary-side transmitting coil C, a primary-side execution controller K, and a ground-controlled communication module J;

[0009] The H-bridge inverter A is connected to the DC bus converted from AC mains power to receive DC power input and output high-frequency AC power.

[0010] The primary-side LCC resonant network B receives the high-frequency AC power output from the H-bridge inverter A and outputs power optimized by resonance.

[0011] The primary-side transmitting coil C receives the resonant optimized electrical energy output by the primary-side LCC resonant network B, and simultaneously transmits energy to the on-board charging device through magnetic field coupling.

[0012] The primary-side execution controller K receives the communication signal transmitted by the ground-controlled communication module J and sends control commands to the H-bridge inverter A;

[0013] The ground-controlled communication module J is connected to the primary-side execution controller K and is used to transmit communication signals to the primary-side execution controller K, while simultaneously receiving communication signals from the on-board charging device through a bidirectional wireless communication circuit.

[0014] Furthermore, the on-board charging device includes a secondary receiving coil D, a secondary series compensation and rectification network E, a BUCK topology DC / DC converter F, an electric vehicle battery load G, a bilateral disturbance suppression control device H, and an on-board main control communication module I;

[0015] The secondary receiving coil D receives energy from the primary transmitting coil C of the ground power supply device through magnetic field coupling and outputs electrical energy.

[0016] The secondary-side series compensation and rectification network E receives the electrical energy output from the secondary-side receiving coil D and outputs rectified DC electrical energy.

[0017] The BUCK topology DC / DC converter F receives DC power from the secondary-side series compensation and rectifier network E, and outputs DC power after voltage regulation; at the same time, it receives control signals from the bilateral disturbance suppression control device H.

[0018] The electric vehicle battery load G receives DC power regulated by the output voltage of the BUCK topology DC / DC converter F, supplies power to the vehicle's electrical loads, and provides output voltage information to the bilateral disturbance suppression control device H.

[0019] The bilateral disturbance suppression control device H estimates and compensates for disturbances caused by the mutual inductance changes due to the offset of the primary side transmitting coil C and the secondary side receiving coil D in real time, sends control signals to the BUCK topology DC / DC converter F, compensates for the output voltage error caused by the disturbance, calculates the control quantity u, and sends control signals to the primary side execution controller K through the vehicle main control communication module I and the ground controlled communication module J.

[0020] The vehicle-mounted main control communication module I connects to the bilateral disturbance suppression control device H to receive control signals, and establishes a two-way wireless communication link with the ground-controlled communication module J of the ground power supply device to exchange communication status information and control signals.

[0021] Furthermore, the H-bridge inverter A is provided with port A1, port A2 and port A3. Port A1 is configured to connect to the DC bus converted from AC mains power to receive DC power input. Port A2 receives control commands from the primary side execution controller K. Port A3 is configured to output high-frequency AC power.

[0022] The primary-side LCC resonant network B has ports B1 and B2. Port B1 is connected to port A3 of the H-bridge inverter A to receive high-frequency AC power, and port B2 is used to output the power after resonance optimization.

[0023] The primary-side transmitting coil C has a port C1, which is connected to port B2 of the primary-side LCC resonant network B to receive the resonant optimized electrical energy. At the same time, the primary-side transmitting coil C transmits energy to the on-board charging device through magnetic field coupling.

[0024] The primary-side execution controller K has ports K1 and K2. Port K1 is configured to send control commands to the H-bridge inverter A, and port K2 is used to receive communication signals transmitted by the ground-controlled communication module J.

[0025] The ground-controlled communication module J is equipped with port J1, which is connected to port K2 of the primary-side execution controller K to transmit communication signals and receive communication signals from the on-board charging device through a bidirectional wireless communication circuit.

[0026] Furthermore, the bilateral disturbance suppression control device H consists of a tracking differentiator H1, a dynamic fusion compensator H2, and a focus state estimator H3;

[0027] The focus state estimator H3 receives the output voltage information provided by the electric vehicle battery load G, and outputs the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate to the dynamic fusion compensator H2.

[0028] The tracking differentiator H1 receives a voltage reference signal from an external signal source, smooths the input voltage reference signal into a continuously changing target tracking signal with the current electric vehicle battery load G output voltage as the initial value and the voltage reference signal as the final value, and simultaneously generates its differential signal.

[0029] The dynamic fusion compensator H2 receives the target tracking signal output by the tracking differentiator H1. and differential signal The system outputs estimated values ​​of the electric vehicle battery load output voltage and the electric vehicle battery load output voltage change rate from the focus state estimator H3, and outputs control signals to drive the BUCK topology DC / DC converter F, and outputs control signals to the on-board main control communication module I.

[0030] Furthermore, the secondary receiving coil D is provided with a port D1, which receives energy from the primary transmitting coil C of the ground power supply device through magnetic field coupling, and the port D1 is configured to output electrical energy;

[0031] The secondary-side series compensation and rectification network E has ports E1 and E2. Port E1 is connected to port D1 of the secondary-side receiving coil D to receive electrical energy, and port E2 is used to output rectified DC power.

[0032] The BUCK topology DC / DC converter F has ports F1, F2 and F3. Port F1 is connected to port E2 of the secondary-side series compensation and rectification network module E to receive DC power. Port F2 is used to output DC power after voltage regulation. Port F3 is used to receive the control signal output by the bilateral disturbance suppression control device H.

[0033] The electric vehicle battery load G has ports G1, G2 and G3. Port G1 is connected to port F2 of the BUCK topology DC / DC converter F to receive DC power after voltage regulation. Port G3 is used to supply power to the vehicle's electrical load. Port G2 is used to provide output voltage information to the bilateral disturbance suppression control device H.

[0034] The tracking differentiator H1 has ports H1-1 and H1-2. Port H1-1 is configured to receive a voltage reference signal from an external signal source, and port H1-2 is connected to the dynamic fusion compensator H2 to output a smoothed signal.

[0035] The focus state estimator H3 has ports H3-1 and H3-2. Port H3-2 is configured to receive the output voltage information of the electric vehicle battery load port G2. Port H3-1 is used to output the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate to port H2-2 of the dynamic fusion compensator H2.

[0036] The dynamic fusion compensator H2 has ports H2-1, H2-2, H2-3 and H2-4. Port H2-1 is connected to port H1-2 of the tracking differentiator H1 to receive signals. Port H2-2 is connected to port H3-1 of the focus state estimator H3 to receive the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate. Port H2-3 is configured to output control signals to drive the BUCK topology DC / DC converter F. Port H2-4 is configured to output control signals to the vehicle main control communication module I.

[0037] The vehicle-mounted main control communication module I is provided with port I1, which is connected to port H2-4 of the dynamic fusion compensator H2 to receive control signals, and establishes a two-way wireless communication link with the ground controlled communication module J of the ground power supply device to exchange communication status information and control commands.

[0038] Furthermore, the bilateral disturbance suppression control device H compensates for the output voltage error caused by disturbances, and the specific method for calculating the control quantity u is as follows:

[0039] Step 1: The focus state estimator H3 samples the actual output voltage of the electric vehicle battery load G in real time. ;

[0040] Step 2: The focus state estimator H3 combines the currently sampled output voltage. It performs state observation and uses its internally constructed ultra-smooth kernel function to calculate the estimated value of the electric vehicle battery load output voltage. and the estimated value of the rate of change of output voltage of electric vehicle battery load and will and Output to dynamic fusion compensator H2;

[0041] Step 3: The tracking differentiator H1 receives the voltage reference signal provided by the external signal source. ;

[0042] Step 4: Tracking differentiator H1 receives voltage reference signal Furthermore, a discrete iterative algorithm based on the fastest control synthesis function is used to perform transient process arrangement calculations on the voltage reference signal that changes abruptly. The process involves creating a smooth trajectory curve from the current output voltage value of the electric vehicle battery load G to the target value, while simultaneously generating a target tracking signal for this trajectory. and its differential signal And transmit it to the dynamic fusion compensator H2;

[0043] Step 5: The dynamic fusion compensator H2 receives the target tracking signal from the tracking differentiator H1. and its differential signal and the output from the focus state estimator H3 and Calculate the tracking error of the output voltage. Tracking error of output voltage change rate ,in ;

[0044] Step 6: The dynamic fusion compensator H2 tracks the output voltage based on the calculated tracking error. Tracking error of output voltage change rate The initial control quantity is calculated by applying a nonlinear proportional element. ;

[0045] Step 7: Dynamically fuse compensator H2 to the initial control quantity Dynamic fusion compensation calculations are performed to obtain the final control quantity u, which is then output to the BUCK topology DC / DC converter F to adjust its switching action.

[0046] Step 8: The dynamic fusion compensator H2 simultaneously outputs the final control quantity u to the vehicle main control communication module I, and sends the control command to the ground power supply device through the wireless communication link;

[0047] Step 9: Return to step 4 and continue with output voltage sampling and adjustment.

[0048] Furthermore, the specific method by which the primary-side execution controller K sends control commands to the H-bridge inverter A is as follows:

[0049] Step S1: The primary-side execution controller K receives the control quantity u transmitted from the ground-controlled communication module J;

[0050] Step S2: The primary-side execution controller K generates a reference square wave signal with a duty cycle of 50% as the drive signal for the switching transistors Q1 and Q4 of the H-bridge inverter A.

[0051] Step S3: The primary-side execution controller K uses the control quantity u as a relative delay. This relative delay is used to control the square wave signals of the switching transistors Q2 and Q3 of the H-bridge inverter A relative to the square wave signals of the switching transistors Q1 and Q4 of the H-bridge inverter A, and the dimensionless ratio of the signal's working cycle. This relative delay is applied to the reference square wave signal generated in step S2, thereby generating another drive signal to control the switching transistors Q2 and Q3 of the H-bridge inverter A. The phase difference between the two signals constitutes the regulated phase shift angle. The phase shift angle is used to control the effective value of the high-frequency AC power output by the H-bridge inverter A by adjusting the phase difference between the two drive signals of the H-bridge inverter, thereby regulating the energy transmitted to the vehicle side to stabilize the output voltage.

[0052] Step S4: The primary-side execution controller K sends the phase shift angles corresponding to the two generated drive signals to the H-bridge inverter A;

[0053] Step S5: The primary side execution controller K continuously monitors the communication port of the ground controlled communication module J to determine whether a new control quantity u has been received. If a new control quantity is received, it immediately returns to step S3 and updates the relative delay to adjust the phase shift angle. If no new signal is received, it maintains the output of the current drive signal.

[0054] Furthermore, the focal state estimator H3 is structurally and functionally reconstructed based on the nonlinear extended state observer in traditional active disturbance rejection control, removing the input channel of the control quantity u and the channel specifically used to output the total disturbance estimate. The port is used, and the nonlinear function fal is replaced with the ultra-smooth kernel function usk to calculate the estimated value of the electric vehicle battery load output voltage. and the estimated value of the rate of change of output voltage of electric vehicle battery load The specific method is as follows:

[0055] Obtain the actual output voltage of the electric vehicle battery load G. ;

[0056] Based on actual output voltage and Calculate the state observation error ;

[0057] Update the state observations based on the following dynamic equation:

[0058] ;

[0059] ;

[0060] in, For observer bandwidth parameters, for The first derivative, for The first derivative, This is a super-smoothing kernel function used to map the state observation error e.

[0061] Output updated and ;

[0062] The super-smoothing kernel function It is configured to map the observation error e using the following functional relationship:

[0063] when hour,

[0064] ;

[0065] when hour,

[0066] ;

[0067] Where e is the state observation error, To meet The nonlinear parameters, The width parameter of the linear interval is greater than 0.

[0068] Furthermore, the dynamic fusion compensator H2 is structurally and functionally reconstructed based on the nonlinear state error feedback controller in traditional active disturbance rejection control, and the initial control quantity is calculated. and the initial control quantity The specific method for performing dynamic fusion compensation calculations to obtain the final control quantity u is as follows:

[0069] Receive target tracking signal from tracking differentiator H1 and its differential and the estimated output voltage of the electric vehicle battery load from the focus state estimator H3. And the estimated value of the rate of change of output voltage of electric vehicle battery load. ;

[0070] Set the actual output voltage of the input in the nonlinear extended state observer Equal to 0 simplifies the state equation of the nonlinear extended state observer. Simultaneously, it sets the control quantity u to the initial control quantity. compositional relationship Substituting the simplified state equations and using the bandwidth method to simplify the observer gain parameters, while linearizing the nonlinear function fal, we obtain the modified state equations; then, we compare the modified state equations with the composition relation. Perform a Laplace transform to obtain the Laplace transform of the control variable u. To the initial control quantity Laplace transform transfer function To achieve the initial control quantity Perform dynamic fusion compensation;

[0071] ;

[0072] in, For controller bandwidth, and , This is the bandwidth ratio factor. This is the estimated control gain value; It is a complex frequency.

[0073] The beneficial effects of adopting the above technical solution are as follows: The wireless charging system based on focal state estimation and dynamic fusion compensation provided by this invention can directly and without conversion set the duty cycle of the vehicle-side BUCK converter and the relative delay of the primary-side H-bridge inverter (thus controlling the phase shift angle) simultaneously with a unified control quantity u, achieving coordinated and direct control of the primary and secondary side energy. The structure is simple and the response is direct. Simultaneously, for traditional active disturbance rejection control, by embedding the disturbance estimation function from the observer into the controller, a focal state estimator focused on state estimation and a dynamic fusion compensator responsible for disturbance compensation and control are formed. This architecture can achieve accurate active disturbance rejection without precise modeling of disturbances such as mutual inductance. With its control theory core, it achieves extremely fast adjustment speed to changes in mutual inductance while ensuring no overshoot. Attached Figure Description

[0074] Figure 1 A schematic diagram of the structure of a wireless charging system based on focus state estimation and dynamic fusion compensation provided in an embodiment of the present invention;

[0075] Figure 2 This is a schematic diagram of the working process of the bilateral disturbance suppression control device provided in an embodiment of the present invention;

[0076] Figure 3 A schematic diagram of the workflow of the primary-edge execution controller provided in an embodiment of the present invention;

[0077] Figure 4 The structural connection diagram of the wireless charging system based on focus state estimation and dynamic fusion compensation provided in the embodiment of the present invention;

[0078] Figure 5The diagrams provided in this embodiment of the invention are performance verification diagrams of the bilateral disturbance suppression control device and the conventional controller. (a) is a diagram of coil mutual inductance disturbance variation, (b) is a diagram of the output voltage of the electric vehicle battery load G under the action of the bilateral disturbance suppression control device, (c) is a diagram of the output voltage of the electric vehicle battery load G under the action of the conventional proportional-integral-derivative controller, and (d) is a diagram of the output voltage of the electric vehicle battery load G under the action of the conventional active disturbance rejection controller. Detailed Implementation

[0079] The specific embodiments of the present invention will be described in further detail below with reference to the accompanying drawings and examples. The following examples are for illustrative purposes only and are not intended to limit the scope of the invention.

[0080] In this embodiment, a wireless charging system based on focus state estimation and dynamic fusion compensation, such as... Figure 1 , Figure 4 As shown, it consists of two main parts: a ground power supply device and an on-board charging device. The two are connected and work together through a two-way wireless communication link and a wireless energy transmission channel based on magnetic field coupling.

[0081] The ground power supply device is connected to a DC bus converted from AC mains power to receive DC power input, converts DC power into high-frequency AC power, and after resonance optimization, transmits energy to the on-board charging device through magnetic field coupling; at the same time, it receives communication signals from the on-board charging device through a two-way wireless communication circuit.

[0082] The on-board charging device receives energy from the ground power supply device through magnetic field coupling and converts it into electrical energy. It then rectifies the electrical energy to output DC power, regulates the voltage of the DC power, and supplies power to the vehicle's electrical load. At the same time, it adopts an observer-based control method to estimate and dynamically compensate for the total disturbance of the system in real time to stabilize the output voltage and transmits control commands to the ground power supply device.

[0083] The ground power supply device includes an H-bridge inverter A, a primary-side LCC resonant network B, a primary-side transmitting coil C, a primary-side execution controller K, and a ground-controlled communication module J;

[0084] The H-bridge inverter A is connected to the DC bus converted from AC mains power to receive DC power input and output high-frequency AC power.

[0085] The primary-side LCC resonant network B receives the high-frequency AC power output from the H-bridge inverter A and outputs power optimized by resonance.

[0086] The primary-side transmitting coil C receives the resonant optimized electrical energy output by the primary-side LCC resonant network B, and simultaneously transmits energy to the on-board charging device through magnetic field coupling.

[0087] The primary-side execution controller K receives the communication signal transmitted by the ground-controlled communication module J and sends control commands to the H-bridge inverter A;

[0088] The ground-controlled communication module J is connected to the primary-side execution controller K and is used to transmit communication signals to the primary-side execution controller K, while simultaneously receiving communication signals from the on-board charging device through a bidirectional wireless communication circuit.

[0089] In this embodiment, the H-bridge inverter A is provided with port A1, port A2 and port A3. Port A1 is configured to connect to the DC bus converted from AC mains power to receive DC power input. Port A2 receives control commands from the primary side execution controller K. Port A3 is configured to output high-frequency AC power.

[0090] The primary-side LCC resonant network B has ports B1 and B2. Port B1 is connected to port A3 of the H-bridge inverter A to receive high-frequency AC power, and port B2 is used to output the power after resonance optimization.

[0091] The primary-side transmitting coil C has a port C1, which is connected to port B2 of the primary-side LCC resonant network B to receive the resonant optimized electrical energy. At the same time, the primary-side transmitting coil C transmits energy to the on-board charging device through magnetic field coupling.

[0092] The primary-side execution controller K has ports K1 and K2. Port K1 is configured to send control commands to the H-bridge inverter A, and port K2 is used to receive communication signals transmitted by the ground-controlled communication module J.

[0093] The ground-controlled communication module J is equipped with port J1, which is connected to port K2 of the primary-side execution controller K to transmit communication signals and receive communication signals from the on-board charging device through a bidirectional wireless communication circuit.

[0094] The on-board charging device includes a secondary receiving coil D, a secondary series compensation and rectification network E, a BUCK topology DC / DC converter F, an electric vehicle battery load G, a bilateral disturbance suppression control device H, and an on-board main control communication module I.

[0095] The secondary receiving coil D receives energy from the primary transmitting coil C of the ground power supply device through magnetic field coupling and outputs electrical energy.

[0096] The secondary-side series compensation and rectification network E receives the electrical energy output from the secondary-side receiving coil D and outputs rectified DC electrical energy.

[0097] The BUCK topology DC / DC converter F receives DC power from the secondary-side series compensation and rectifier network E, and outputs DC power after voltage regulation; at the same time, it receives control signals from the bilateral disturbance suppression control device H.

[0098] The electric vehicle battery load G receives DC power regulated by the output voltage of the BUCK topology DC / DC converter F, supplies power to the vehicle's electrical loads, and provides output voltage information to the bilateral disturbance suppression control device H.

[0099] The bilateral disturbance suppression control device H estimates and compensates for disturbances caused by the mutual inductance changes due to the offset of the primary transmitting coil C and the secondary receiving coil D in real time, sends control signals to the BUCK topology DC / DC converter F, compensates for the output voltage error caused by the disturbance, quickly stabilizes the output voltage of the electric vehicle wireless charging system, calculates the control quantity, and sends control signals to the primary execution controller K through the on-board main control communication module I and the ground controlled communication module J.

[0100] The vehicle-mounted main control communication module I connects to the bilateral disturbance suppression control device H to receive control signals, and establishes a two-way wireless communication link with the ground-controlled communication module J of the ground power supply device to exchange communication status information and control signals.

[0101] In this embodiment, the secondary receiving coil D is provided with a port D1, which receives energy from the primary transmitting coil C of the ground power supply device through magnetic field coupling. The port D1 is configured to output electrical energy.

[0102] The secondary-side series compensation and rectification network E has ports E1 and E2. Port E1 is connected to port D1 of the secondary-side receiving coil D to receive electrical energy, and port E2 is used to output the rectified DC power.

[0103] The BUCK topology DC / DC converter F has ports F1, F2 and F3. Port F1 is connected to port E2 of the secondary series compensation and rectification network E to receive DC power. Port F2 is used to output DC power after voltage regulation. Port F3 is used to receive the control signal output by the bilateral disturbance suppression control device H.

[0104] The electric vehicle battery load G has ports G1, G2 and G3. Port G1 is connected to port F2 of the BUCK topology DC / DC converter F to receive voltage-regulated DC power. Port G3 is used to supply power to the vehicle's electrical load. Port G2 is used to provide output voltage information to the bilateral disturbance suppression control device H.

[0105] In this embodiment, the bilateral disturbance suppression control device H consists of a tracking differentiator H1, a dynamic fusion compensator H2, and a focal state estimator H3.

[0106] The focus state estimator H3 receives the output voltage information provided by the electric vehicle battery load G, and outputs the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate to the dynamic fusion compensator H2.

[0107] The tracking differentiator H1 receives a voltage reference signal from an external signal source, smooths the input voltage reference signal into a continuously changing target tracking signal with the current electric vehicle battery load G output voltage as the initial value and the voltage reference signal as the final value, and simultaneously generates its differential signal.

[0108] The dynamic fusion compensator H2 receives the target tracking signal output by the tracking differentiator H1. and differential signal The system outputs the estimated values ​​of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate from the focus state estimator H3, outputs control signals to drive the BUCK topology DC / DC converter F, and outputs control signals to the vehicle main control communication module I.

[0109] In this embodiment, the tracking differentiator H1 is provided with port H1-1 and port H1-2. Port H1-1 is configured to receive a voltage reference signal from an external signal source, and port H1-2 is connected to the dynamic fusion compensator H2 to output a smoothed signal.

[0110] The focus state estimator H3 has ports H3-1 and H3-2. Port H3-2 is configured to receive the output voltage information of the electric vehicle battery load port G2. Port H3-1 is used to output the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate to port H2-2 of the dynamic fusion compensator H2.

[0111] The dynamic fusion compensator H2 has ports H2-1, H2-2, H2-3 and H2-4. Port H2-1 is connected to port H1-2 of the tracking differentiator H1 to receive signals. Port H2-2 is connected to port H3-1 of the focus state estimator H3 to receive the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate. Port H2-3 is configured to output control signals to drive the BUCK topology DC / DC converter F. Port H2-4 is configured to output control signals to the vehicle main control communication module I.

[0112] The vehicle-mounted main control communication module I is equipped with port I1, which is connected to port H2-4 of the dynamic fusion compensator H2 to receive control signals and establish a two-way wireless communication link with the ground controlled communication module J of the ground power supply device to exchange communication status information and control commands.

[0113] In this embodiment, the hardware configuration of the wireless charging system based on focus state estimation and dynamic fusion compensation is as follows: the system design power is 8W and the target output voltage is 20V.

[0114] In the ground power supply unit, the H-bridge inverter A adopts a full-bridge topology. Its switching transistors can be Infineon's CoolMOS™ or IR series IGBTs. The switching frequency is set to 85kHz, and the input voltage of port A1, which is connected to the DC bus, is 200V.

[0115] The parameters of the primary-side LCC resonant network B have been optimized. The resonant inductance L1 is 48.59 μH, which can use a ferrite core or an amorphous core. The series resonant capacitor Cp is 77 nF, which can be a thin-film capacitor. The parallel capacitor Cf is 72.153 nF. These parameters are calculated according to the resonant frequency formula to ensure that the system operates efficiently near the resonant point of 85 kHz.

[0116] The primary-side transmitting coil C is wound with Litz wire and has a circular or rectangular shape. When the coil is circular, its diameter is 800mm; when the coil is rectangular, its length and width are 600mm×400mm and its inductance is 94.127μH. Ferrite sheets are used for magnetic shielding and magnetic conduction.

[0117] In the on-board charging device, the secondary receiving coil D corresponds to the primary coil C in terms of specifications, with an inductance of 94.127μH. The series compensation capacitor Cs of the secondary series compensation and rectifier network E is 37.247nF, and the rectifier bridge can be a silicon carbide Schottky diode.

[0118] The switching transistors for the BUCK topology DC / DC converter F can be Infineon's OptiMOS™, with a switching frequency of 85kHz, an output filter inductor of 1000μH, and an output filter capacitor of 1000μF.

[0119] The primary-side execution controller K in the ground power supply unit and the bilateral disturbance suppression control device H in the vehicle-mounted charging unit are the control cores of the wireless charging system, both implemented by digital signal processors. The functions of the bilateral disturbance suppression control device H are programmed into the TI TMS320F2837x or TMS320F2833x series DSP on the vehicle side. The primary-side execution controller K can be implemented using Microchip's dsPIC33EP series MCU or TI's C2000 series DS. The sampling period for both the primary-side execution controller K and the bilateral disturbance suppression control device H is set to 0.05882μs.

[0120] When the primary and secondary coils of the ground power supply unit and the on-board charging unit are properly aligned, the mutual inductance M is approximately 24.295 μH. To verify the anti-offset performance of the control method, the simulated mutual inductance variation range in this embodiment is 15 μH to 32 μH.

[0121] The vehicle-mounted main control communication module I includes a first microprocessor unit, a first wireless transceiver chip, a first antenna assembly, and a first interface circuit. The first microprocessor unit is configured to execute communication protocol processing, the first wireless transceiver chip is responsible for signal modulation and demodulation, the first antenna assembly realizes the transmission and reception of electromagnetic waves, and the first interface circuit is connected to the bilateral disturbance suppression control device H through port I1.

[0122] The ground-controlled communication module J includes a second microprocessor unit, a second wireless transceiver chip, a second antenna assembly, and a second interface circuit. The second microprocessor unit is configured to execute communication protocol processing, the second wireless transceiver chip is responsible for signal modulation and demodulation, the second antenna assembly transmits and receives electromagnetic waves, and the second interface circuit is connected to the primary-side execution controller K via port J1. The hardware selection for the vehicle-mounted main control communication module I and the ground-controlled communication module J must be compatible with the system's operating frequency and power consumption requirements. The first and second microprocessor units can be embedded MCUs, and the first and second wireless transceiver chips support wireless communication standards for specific frequency bands.

[0123] In this embodiment, the bilateral disturbance suppression control device H compensates for the output voltage error caused by disturbances, calculates the control quantity u, and provides a specific method for stabilizing the output voltage of the electric vehicle wireless charging system, such as... Figure 2 As shown, it includes the following steps:

[0124] Step 1: The focus state estimator H3 samples the actual output voltage of the electric vehicle battery load G in real time. ;

[0125] Step 2: The focus state estimator H3 combines the currently sampled output voltage. It performs state observation and uses its internally constructed ultra-smooth kernel function to calculate the estimated value of the electric vehicle battery load output voltage. and the estimated value of the rate of change of output voltage of electric vehicle battery load and will and Output to dynamic fusion compensator H2;

[0126] Step 3: The tracking differentiator H1 receives the voltage reference signal provided by the external signal source. ;

[0127] Step 4: Tracking differentiator H1 receives voltage reference signal Furthermore, a discrete iterative algorithm based on the fastest control synthesis function is used to perform transient process arrangement calculations on the voltage reference signal that changes abruptly. The process involves creating a smooth trajectory curve from the current output voltage value of the electric vehicle battery load G to the target value, while simultaneously generating a target tracking signal for this trajectory. and its differential signal And transmit it to the dynamic fusion compensator H2;

[0128] Step 5: The dynamic fusion compensator H2 receives the target tracking signal from port H1-2 of the tracking differentiator H1 through port H2-1. and its differential signal The output from port H3-1 of the focus state estimator via port H2-2 and Calculate the tracking error of the output voltage. Tracking error of output voltage change rate ,in ;

[0129] Step 6: The dynamic fusion compensator H2 tracks the output voltage based on the calculated tracking error. Tracking error of output voltage change rate The initial control quantity is calculated by applying a nonlinear proportional element. ;

[0130] Step 7: Dynamically fuse compensator H2 to the initial control quantity Dynamic fusion compensation calculation is performed to obtain the final control quantity u, and it is output to the controlled port F3 of the BUCK topology DC / DC converter F through port H2-3 to adjust its switching action;

[0131] Step 8: The dynamic fusion compensator H2 simultaneously outputs the final control quantity u through port H2-4 to port I1 of the vehicle main control communication module I, and sends the control command to the ground power supply device through the wireless communication link.

[0132] Step 9: Return to step 4 and continue with output voltage sampling and adjustment.

[0133] In this embodiment, the specific method for the primary-side execution controller K to issue control commands to the H-bridge inverter A is as follows: Figure 3 As shown, it includes the following steps:

[0134] Step S1: The primary side execution controller K receives the control quantity u transmitted from port J1 of the ground controlled communication module J through its communication port K2.

[0135] Step S2: The primary-side execution controller K generates a reference square wave signal with a duty cycle of 50% as the drive signal for the switching transistors Q1 and Q4 of the H-bridge inverter A.

[0136] Step S3: The primary-side execution controller K uses the control quantity u as a relative delay. This relative delay is used to control the square wave signals of the switching transistors Q2 and Q3 of the H-bridge inverter A relative to the square wave signals of the switching transistors Q1 and Q4 of the H-bridge inverter A, and the dimensionless ratio of the signal's working cycle. This relative delay is applied to the reference square wave signal generated in step S2, thereby generating another drive signal to control the switching transistors Q2 and Q3 of the H-bridge inverter A. The phase difference between the two signals constitutes the regulated phase shift angle. The phase shift angle is used to control the effective value of the high-frequency AC power output by the H-bridge inverter A by adjusting the phase difference between the two drive signals of the H-bridge inverter A, thereby precisely regulating the energy transmitted to the vehicle side to stabilize the system output voltage.

[0137] Step S4: The primary-side execution controller K sends the phase shift angles corresponding to the two generated drive signals to the controlled port A2 of the H-bridge inverter A through its output port K1.

[0138] Step S5: The primary-side execution controller K continuously monitors its communication port J1 to determine whether a new control quantity u has been received. If a new control quantity is received, it immediately returns to step S3 and updates the relative delay to adjust the phase shift angle. If no new signal is received, it maintains the output of the current drive signal.

[0139] In constructing the dynamic fusion compensator H2 and the focal state estimator H3 of the bilateral disturbance suppression control device H, this invention restructures the traditional nonlinear state error feedback controller (NLSEF) and extended state observer (NLESO) in terms of structure and function, and then uses the improved controller and the improved observer as the dynamic fusion compensator H2 and the focal state estimator H3, respectively.

[0140] In traditional Active Disturbance Rejection Control (ADRC), NLESO is the core of ADRC. For a second-order control system, extending it to a third-order control system involves the following process:

[0141] Define the expansion state:

[0142] ;

[0143] in, To control the output voltage of the system, for The first derivative, i.e., the rate of change of voltage. It is the sum of disturbances, that is, the sum of internal disturbances and external disturbances. Let y be the external disturbance, t be time, and y be the control system output. yes y The first derivative; for a general second-order system, only the first derivative is needed. , That works, but in ADRC, the sum of disturbances is extended to a third state, which makes it easier to calculate and estimate disturbances and realize active disturbance rejection control.

[0144] The control system equations can then be written in the following state-space form:

[0145] ;

[0146] in, yes The first derivative, yes The first derivative, yes The first derivative, The total disturbance value is the sum of the system's internal and external disturbances; it is unknown but bounded. b is the control gain, the control quantity. It is a value between 0 and 1, and is used to control the relative delay of the square wave control signals Q2 and Q3 of the H-bridge inverter A and the duty cycle of the BUCK topology DC / DC converter F.

[0147] The corresponding state equation for the third-order nonlinear extended state observer (NLESO) (more general in discrete form) is:

[0148] ;

[0149] Where e is the deviation between the observed state value and the actual output value of the control system. , , These are the estimated values ​​for the output voltage of the electric vehicle battery load, the estimated value for the rate of change of the output voltage of the electric vehicle battery load, and the estimated value for the total disturbance, respectively. , , They are , , The first derivative, , , The observer gain coefficient determines the convergence speed of the observer; It is an estimated value of the control gain b; It is a key nonlinear function. , , These are all power-order parameters of the nonlinear function fal, used to adjust the nonlinear characteristics of the function fal. , , The smaller the value of , the greater the gain of the function in the small error range where the error e is close to zero. correspond , Taking the largest value means that the estimate of the actual output y of the system is relatively stable and robust; correspond , A smaller value means that the observer is more sensitive to the estimation of the rate of change of the output; Corresponding total disturbance estimate , The smallest value results in the observer having the highest sensitivity to the estimation of the "total disturbance";

[0150] nonlinear functions The expression is:

[0151] ;

[0152] Wherein, parameter α satisfies The gain α determines the nonlinear shape. A smaller α results in a larger gain with small errors, effectively reducing steady-state error; a smaller gain with large errors enhances smoothness and robustness. δ is the width of the linear interval, used to avoid high-frequency flutter near zero.

[0153] Typical values ​​of the power parameter of the nonlinear function fal: , , It is a common configuration, but can be adjusted as needed.

[0154] For a nonlinear state error feedback controller (NLSEF), it utilizes the estimate from a nonlinear extended state observer (NLESO). , The target tracking signal given by the tracking differentiator H1 and its differential signal Calculate the initial control quantity ;

[0155] The equations established by the nonlinear state error feedback controller (NLSEF) are as follows:

[0156] ;

[0157] in, , It is the gain of a traditional nonlinear ADRC controller (equivalent to the P and D gains in PID). , It is a non-linear parameter of the fal function, usually To enhance the smoothness of the differential terms;

[0158] Finally, the total perturbation estimate is obtained using the Nonlinear Extended State Observer (NLESO). For the initial control quantity Dynamic compensation is performed to obtain the control variable:

[0159] ;

[0160] When traditional Active Disturbance Rejection Control (ADRC) is applied to wireless charging systems for electric vehicles, its inherent control structure, observer design, and parameter tuning methods have several shortcomings, limiting the performance ceiling of the wireless charging system in dealing with complex operating conditions such as mutual inductance fluctuations and load changes. Firstly, in the core Nonlinear State Error Feedback Unit (NLSEF), traditional designs place disturbance compensation after feedback control, forming a series compensation structure. In this structure, the total disturbance estimated by the Nonlinear Extended State Observer (NLESO) needs to pass through a separate subtractor before it can be applied to the final control variable. This results in a certain lag in disturbance compensation, failing to achieve deep coordination with the error feedback, and affecting the dynamic response speed and compensation accuracy of the wireless charging system to rapid disturbances such as sudden changes in mutual inductance.

[0161] Traditional nonlinear extended state observers (NLESO) have relatively complex structures. Their dynamic equations typically include two input channels: the controlled output y and the control quantity u, and require a dedicated [device / system]. This is used to estimate the total disturbance. While this complete third-order structure provides comprehensive state and disturbance information, it also increases computational complexity. More importantly, the dynamic relationship between the observer and the controller lacks a clear, theory-guided coordination mechanism. Their bandwidth parameters often need to be tuned independently, which can lead to improper parameter coordination. If the controller bandwidth is much higher than the observer bandwidth, the control action may be based on inaccurate observation information, easily causing overshoot or oscillation.

[0162] In terms of parameter tuning, traditional Active Disturbance Rejection Control (ADRC) faces the problem of numerous parameters and strong coupling. It requires simultaneous adjustment of the gains of multiple nonlinear functions of NLESO (such as...). , , The process of determining the control gain of NLSEF is highly dependent on personal experience and repeated trial and error, lacking a systematic simplification method. It is not only cumbersome but also difficult to guarantee optimal robustness under different operating conditions, reducing the engineering usability and portability of the control method.

[0163] Furthermore, the design of traditional Active Disturbance Rejection Control (ADRC) still relies to some extent on the mathematical model of the controlled object, and its controller parameters are usually optimized for a specific controlled object model or compensation topology. When the topology of the control system changes, such as switching from an LCC / S resonant network to an SS resonant network, or when the type of the front-end DC / DC converter is different, the parameters of the traditional controller may no longer be applicable and need to be readjusted, which limits its generalization ability under different hardware configurations.

[0164] Finally, the nonlinear function used in traditional NLSEF The function itself suffers from insufficient smoothness and limited approximation accuracy. The derivative discontinuities at the error zero and segmentation points easily induce high-frequency chattering in the control quantity, leading to additional harmonics in the power switching devices and affecting the electromagnetic compatibility (EMC) and acoustic performance of the control system. Furthermore, its simple mathematical form makes it difficult to perfectly balance response speed and control smoothness over a wide error range, hindering further improvements in the overall control quality of electric vehicle wireless charging systems.

[0165] To address the aforementioned problems, this invention structurally and analytically reconstructs the traditional nonlinear function fal, proposing a novel ultra-smooth kernel function (USK), the mathematical expression of which is as follows:

[0166] ;

[0167] The design of this ultrasmoothing kernel function incorporates the following key innovations:

[0168] Smooth approximation of high-order polynomials over small error intervals:

[0169] exist Within the interval, the traditional simple linear or low-power form is abandoned. Instead, a combined polynomial consisting of a linear principal term, a cubic term, and a quintic term based on Taylor expansion theorem is adopted. .

[0170] Design Principle: This structure is obtained by Taylor expansion of ideal nonlinear characteristics or by solving under specific smoothing constraints (such as requiring C² continuity at ±δ). Linear principal terms. The core gain characteristics of the traditional function with small errors are retained; while the added odd-order higher-order terms... and Its coefficients are precisely calculated to ensure that the function is accurate at the piecewise points. The function achieves a continuous connection between the function and the external function in terms of higher-order derivatives (at least first and second order), thus mathematically guaranteeing the overall smoothness of the function.

[0171] Smooth transition of rational fractions over large error intervals:

[0172] exist The interval, the function consists of two terms: the first term It inherits the ability of traditional functions to "suppress" large errors, ensuring the rapid response and stability of the control system.

[0173] The second item is a correction item: The design is a rational fractional function, and its function is:

[0174] Smoothing Compensation: When |e| is slightly greater than δ, this term smooths the first term, making it more consistent with the internal higher-order polynomials. A high-order smooth connection is achieved, completely eliminating the "inflection point".

[0175] Asymptotic convergence: As |e| increases, the value of this term decays rapidly, and the behavior of the entire function asymptotically approaches that of the function. This does not affect the dynamic performance under large errors.

[0176] In this embodiment, the focused state estimator H3 (FSE) is obtained as follows: a simplified design of the traditional nonlinear extended state observer (NLESO) is implemented to optimize its observation focus and improve efficiency. The main improvement lies in removing the input channel for the control variable u and the dedicated output for estimating the total disturbance from the traditional structure. The port is then used, and the fal function is replaced with the usk function. The dynamics of the focus state estimator H3 are described by the following set of nonlinear differential equations:

[0177] ;

[0178] After this simplification, the focus state estimator H3 no longer directly outputs the total disturbance estimate. Instead, it focuses its observational capabilities on the system's key state variables, such as the estimated output voltage of the electric vehicle battery load. and the estimated value of the rate of change of output voltage of electric vehicle battery load .

[0179] The method for obtaining the Dynamic Fusion Compensator (DFC) H2 of this invention is based on the reconstruction of the nonlinear state error feedback controller structure in traditional Active Disturbance Rejection Control (ADRC). In traditional ADRC, the nonlinear extended state observer (NLESO) independently estimates and outputs the total disturbance value estimate. This value is then used for explicit disturbance compensation in the control law. To improve performance and simplify the structure, this invention uses the total disturbance value estimate... By incorporating disturbance compensation into the control law of NLSEF in advance, the disturbance compensation can take effect earlier within the controller, thereby enhancing the dynamic response of the system.

[0180] To perform frequency domain analysis and parameter tuning of this improved disturbance compensation circuit, it is necessary to establish a starting point from the initial control input. The transfer relationship to the final control quantity u is determined. As a mathematical derivation method, during the frequency domain transformation of the control system model, the input y=0 of the nonlinear extended state observer (NLESO) is set. This setting transforms the NLESO into a single-input form that depends only on the control quantity u, thereby converting the dynamic compensation relationship of the system into an analyzable transfer function Gc(s). This transfer function Gc(s) effectively characterizes the dynamic characteristics of disturbance compensation and is used to replace the explicit disturbance compensation term in the traditional control law. This replacement not only simplifies the controller structure but also improves the dynamic compensation performance through frequency domain optimization.

[0181] Specifically, the input of the nonlinear extended state observer (NLESO) is first set to y=0, transforming it into an observer with only the control variable u as input. The state equation is:

[0182] ;

[0183] Composition relationship of control quantities Substituting into the above equation and eliminating the final control quantity u, we obtain a value based on the initial control quantity. A closed-loop system model with the observer state as the variable and the input as the input.

[0184] The observer gain parameter is obtained by using the bandwidth parameterization method. Configured to match observer bandwidth The relevant form; performing a Laplace transform on the state equation y=0 to convert it from the time domain to the complex frequency domain;

[0185] To perform frequency domain analysis and parameter tuning, the nonlinear function is... Equivalent linearization is performed, and a bandwidth parameterization method is introduced. Specifically, the observer gain parameter is... Configured to match observer bandwidth The relevant form, namely:

[0186] ;

[0187] Meanwhile, under the assumption of equivalent linearization, the nonlinear function is... It is approximately a proportional relationship, that is Based on this approximation, a Laplace transform is performed on the state equation y=0 to convert it from the time domain to the complex frequency domain.

[0188] In the complex frequency domain, extract the total disturbance estimate. For the output signal channel, the Laplace transform of the control quantity u is derived through algebraic operations. arrive Laplace transform of initial control quantity transfer function :

[0189] ;

[0190] in, and These are the control quantity u and the initial control quantity. The Laplace transform of these two variables is their representation in the complex frequency domain; the initial control quantity This is a "theoretical" or "preliminary" control command calculated entirely from the deviation between the target and the current state, without considering real-time system disturbances. The final control quantity *u* is a physically executable control signal, processed by the dynamic fusion compensator H2, and actually output to the BUCK topology DC / DC converter F and the H-bridge inverter A; *s* is a complex frequency variable; this transfer function... The initial control quantity was accurately characterized under the disturbance feedforward compensation structure. To the total disturbance estimate The dynamic transmission relationship.

[0191] To address the complexity of active disturbance rejection controller (ADRC) parameter tuning, the bandwidth method is commonly used in control theory for parameter design. Its core idea is to correlate the dynamic characteristics of the observer and controller with a specific bandwidth to simplify the tuning process. This invention establishes a key parameter constraint: setting the bandwidth of the usk function used in the dynamic fusion compensator H2 as... It is always equal to the bandwidth corresponding to the nonlinear function used in FSE. N times, that is N is the bandwidth scaling factor, which was optimized based on a large amount of experimental data. Through repeated experiments, it was found that when N is in the range of [3, 6], a better control effect can be obtained. In this embodiment, it is set as follows: The value is 300, and N is 3, that is ,therefore The value is 100;

[0192] This invention addresses the problem of drastic changes in mutual inductance in wireless charging systems for electric vehicles caused by vehicle displacement or sudden load changes, and makes significant improvements at the controller structure level. The core of this improvement lies in constructing a control architecture with a clear dynamic priority relationship, where key controller parameters are systematically designed using a bandwidth parameterization method. Specifically, the bandwidth of controller H2... With the bandwidth of the observer Set as a multiple ratio, that is This constraint establishes a timing mechanism that prioritizes the observer's dynamic process over the execution of the control law. Under this mechanism, the focus state estimator H3 can estimate the system's overall disturbances (including mutual inductance changes) in real time at a faster speed, providing forward-looking information input for subsequent control law calculations. This design ensures that when faced with rapid disturbances, the controller can generate control commands based on accurate and timely estimates, thereby effectively avoiding overshoot or oscillations caused by information lag and significantly improving the system's response speed and anti-interference stability.

[0193] Regarding controller parameter tuning, this invention achieves significant improvements through bandwidth parameterization technology. In traditional active disturbance rejection control, the gain parameters of the observer and controller need to be tuned independently, a cumbersome process that relies on experience. This invention correlates all gain parameters of the focus state estimator H3 to the bandwidth of a single observer. Simultaneously, the gain of the improved state error feedback control law is correlated to the bandwidth of a single controller. Furthermore, by introducing This optimized and verified proportional relationship transforms the process of adjusting multiple parameters simultaneously into tuning only one dominant parameter. The problem is that the observer bandwidth only needs to be determined based on the basic requirements of the wireless charging system for electric vehicles for disturbance suppression speed. The value of the controller bandwidth It can be automatically determined proportionally, which not only greatly simplifies the complexity of engineering implementation, but also ensures the inherent coordination of parameter combinations and system robustness, and improves the portability of control methods.

[0194] This invention significantly improves the nonlinear function in a nonlinear state error feedback controller by replacing the traditional nonlinear function fal with a high-precision smoothing function usk, which possesses high-order continuous characteristics. This function is smooth and differentiable throughout its entire domain, fundamentally eliminating derivative discontinuities in the control signal and resulting in an extremely smooth output of the feedback control quantity. This improvement prevents high-frequency jitter in the modulation signal of the BUCK topology DC / DC converter, thereby significantly reducing current and voltage ripple caused by switching actions and improving output power quality. In terms of control performance, this function provides more accurate nonlinear gain through high-order polynomial design in the small error range, enhancing the compensation capability for small deviations and improving steady-state accuracy; in the large error range, it maintains smooth transition characteristics, avoiding shocks during setpoint switching and ensuring the stability of the dynamic process. The smoothing of the control signal also optimizes electromagnetic compatibility and acoustic performance, reducing high-frequency harmonic injection into the power loop, lowering electromagnetic interference levels, and eliminating control pulsations in the audible frequency band, thus improving the user experience.

[0195] At the hardware implementation level, the improvements to this nonlinear state error feedback controller demonstrate excellent engineering friendliness. Although the nonlinear function expression is highly analytical, its calculation involves only basic operations, requiring no complex logical judgments or table lookup operations. This results in high computational determinism and execution efficiency on a digital signal processor, which is beneficial for the precise stability of the control cycle. The controller's output is regulated within a reasonable range of 0 to 1, and the switching frequency has been optimized to ensure good timing matching with the digital control chip, avoiding hardware execution problems caused by excessively rapid changes in control input or excessively high switching frequencies.

[0196] In this embodiment, as Figure 5 As shown, in a step response test with an initial output voltage of 0V and a target output of 20V, the bilateral cooperative disturbance suppression control device reaches 20V in 39.4 milliseconds without overshoot, and the steady-state fluctuation range is controlled within 0.3V, demonstrating excellent tracking accuracy and stability. Under severe conditions such as a 50% drastic fluctuation of 12.5uH inductance or a 50% sudden load drop, it can complete adaptive adjustment within 0.2 milliseconds, and the output change range is controlled within 0.05V during the adjustment process. In contrast, the traditional proportional-integral-derivative controller has a large maximum overshoot (20%), and the output of the electric vehicle battery load G will have a large deviation (2.5V) when external disturbances occur. The traditional active disturbance rejection controller has a large maximum overshoot (37.5%) and oscillation number (3), and the output of the electric vehicle battery load G will have a large deviation (1.6V) when external disturbances occur. In comparison, the bilateral cooperative disturbance suppression control device of this invention has strong anti-disturbance capability and engineering practical value.

[0197] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit them. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some or all of the technical features therein. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the scope defined by the present invention.

Claims

1. A wireless charging system based on focus state estimation and dynamic fusion compensation, characterized in that: It consists of two main parts: a ground power supply unit and an on-board charging unit. The two are connected and work together through a two-way wireless communication link and a wireless energy transmission channel based on magnetic field coupling. The ground power supply device is connected to a DC bus converted from AC mains power to receive DC power input, converts DC power into high-frequency AC power, and after resonance optimization, transmits energy to the on-board charging device through magnetic field coupling; at the same time, it receives communication signals from the on-board charging device through a two-way wireless communication circuit. The on-board charging device receives energy from the ground power supply device through magnetic field coupling and converts it into electrical energy. It then rectifies the electrical energy to output DC power, regulates the voltage of the DC power, and supplies it to the vehicle's electrical load. At the same time, it adopts an observer-based control method to estimate and dynamically compensate for the total disturbance of the system in real time to stabilize the output voltage and transmit control commands to the ground power supply device. The on-board charging device includes a secondary receiving coil D, a secondary series compensation and rectification network E, a BUCK topology DC / DC converter F, an electric vehicle battery load G, a bilateral disturbance suppression control device H, and an on-board main control communication module I. The secondary receiving coil D receives energy from the primary transmitting coil C of the ground power supply device through magnetic field coupling and outputs electrical energy. The secondary-side series compensation and rectification network E receives the electrical energy output from the secondary-side receiving coil D and outputs rectified DC electrical energy. The BUCK topology DC / DC converter F receives DC power from the secondary-side series compensation and rectifier network E, and outputs DC power after voltage regulation; at the same time, it receives control signals from the bilateral disturbance suppression control device H. The electric vehicle battery load G receives DC power regulated by the output voltage of the BUCK topology DC / DC converter F, supplies power to the vehicle's electrical loads, and provides output voltage information to the bilateral disturbance suppression control device H. The bilateral disturbance suppression control device H estimates and compensates for disturbances caused by the mutual inductance changes due to the offset of the primary side transmitting coil C and the secondary side receiving coil D in real time, sends control signals to the BUCK topology DC / DC converter F, compensates for the output voltage error caused by the disturbance, calculates the control quantity u, and sends control signals to the primary side execution controller K through the vehicle main control communication module I and the ground controlled communication module J. The vehicle-mounted main control communication module I is connected to the bilateral disturbance suppression control device H to receive control signals, and establishes a two-way wireless communication link with the ground-controlled communication module J of the ground power supply device to exchange communication status information and control signals. The bilateral disturbance suppression control device H consists of a tracking differentiator H1, a dynamic fusion compensator H2, and a focal state estimator H3. The focus state estimator H3 receives the output voltage information provided by the electric vehicle battery load G, and outputs the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate to the dynamic fusion compensator H2. The tracking differentiator H1 receives a voltage reference signal from an external signal source, smooths the input voltage reference signal into a continuously changing target tracking signal with the current electric vehicle battery load G output voltage as the initial value and the voltage reference signal as the final value, and simultaneously generates its differential signal. The dynamic fusion compensator H2 receives the target tracking signal output by the tracking differentiator H1. and differential signal The system outputs the estimated values ​​of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate from the focus state estimator H3, outputs control signals to drive the BUCK topology DC / DC converter F, and outputs control signals to the vehicle main control communication module I. The bilateral disturbance suppression control device H compensates for the output voltage error caused by disturbances. The specific method for calculating the control quantity u is as follows: Step 1: The focus state estimator H3 samples the actual output voltage of the electric vehicle battery load G in real time. ; Step 2: The focus state estimator H3 combines the currently sampled output voltage. It performs state observation and uses its internally constructed ultra-smooth kernel function to calculate the estimated value of the electric vehicle battery load output voltage. and the estimated value of the rate of change of output voltage of electric vehicle battery load and will and Output to dynamic fusion compensator H2; Step 3: The tracking differentiator H1 receives the voltage reference signal provided by the external signal source. ; Step 4: Tracking differentiator H1 receives voltage reference signal Furthermore, a discrete iterative algorithm based on the fastest control synthesis function is used to perform transient process arrangement calculations on the voltage reference signal that changes abruptly. The process involves creating a smooth trajectory curve from the current output voltage value of the electric vehicle battery load G to the target value, while simultaneously generating a target tracking signal for this trajectory. and its differential signal And transmit it to the dynamic fusion compensator H2; Step 5: The dynamic fusion compensator H2 receives the target tracking signal from the tracking differentiator H1. and its differential signal and the output from the focus state estimator H3 and Calculate the tracking error of the output voltage. Tracking error of output voltage change rate ,in ; Step 6: The dynamic fusion compensator H2 tracks the output voltage based on the calculated tracking error. Tracking error of output voltage change rate The initial control quantity is calculated by applying a nonlinear proportional element. ; Step 7: Dynamically fuse compensator H2 to the initial control quantity Dynamic fusion compensation calculations are performed to obtain the final control quantity u, which is then output to the BUCK topology DC / DC converter F to adjust its switching action. Step 8: The dynamic fusion compensator H2 simultaneously outputs the final control quantity u to the vehicle main control communication module I, and sends the control command to the ground power supply device through the wireless communication link; Step 9: Return to step 4 and continue with output voltage sampling and adjustment.

2. The wireless charging system based on focus state estimation and dynamic fusion compensation according to claim 1, characterized in that: The ground power supply device includes an H-bridge inverter A, a primary-side LCC resonant network B, a primary-side transmitting coil C, a primary-side execution controller K, and a ground-controlled communication module J; The H-bridge inverter A is connected to the DC bus converted from AC mains power to receive DC power input and output high-frequency AC power. The primary-side LCC resonant network B receives the high-frequency AC power output from the H-bridge inverter A and outputs power optimized by resonance. The primary-side transmitting coil C receives the resonant optimized electrical energy output by the primary-side LCC resonant network B, and simultaneously transmits energy to the on-board charging device through magnetic field coupling. The primary-side execution controller K receives the communication signal transmitted by the ground-controlled communication module J and sends control commands to the H-bridge inverter A; The ground-controlled communication module J is connected to the primary-side execution controller K and is used to transmit communication signals to the primary-side execution controller K, while simultaneously receiving communication signals from the on-board charging device through a bidirectional wireless communication circuit.

3. The wireless charging system based on focus state estimation and dynamic fusion compensation according to claim 2, characterized in that: The H-bridge inverter A has ports A1, A2 and A3. Port A1 is configured to connect to the DC bus converted from AC mains power to receive DC power input. Port A2 receives control commands from the primary side execution controller K. Port A3 is configured to output high-frequency AC power. The primary-side LCC resonant network B has ports B1 and B2. Port B1 is connected to port A3 of the H-bridge inverter A to receive high-frequency AC power, and port B2 is used to output the power after resonance optimization. The primary-side transmitting coil C has a port C1, which is connected to port B2 of the primary-side LCC resonant network B to receive the resonant optimized electrical energy. At the same time, the primary-side transmitting coil C transmits energy to the on-board charging device through magnetic field coupling. The primary-side execution controller K has ports K1 and K2. Port K1 is configured to send control commands to the H-bridge inverter A, and port K2 is used to receive communication signals transmitted by the ground-controlled communication module J. The ground-controlled communication module J is equipped with port J1, which is connected to port K2 of the primary-side execution controller K to transmit communication signals and receive communication signals from the on-board charging device through a bidirectional wireless communication circuit.

4. The wireless charging system based on focus state estimation and dynamic fusion compensation according to claim 3, characterized in that: The secondary receiving coil D is provided with a port D1, which receives energy from the primary transmitting coil C of the ground power supply device through magnetic field coupling. The port D1 is configured to output electrical energy. The secondary-side series compensation and rectification network E has ports E1 and E2. Port E1 is connected to port D1 of the secondary-side receiving coil D to receive electrical energy, and port E2 is used to output rectified DC power. The BUCK topology DC / DC converter F has ports F1, F2 and F3. Port F1 is connected to port E2 of the secondary-side series compensation and rectification network module E to receive DC power. Port F2 is used to output DC power after voltage regulation. Port F3 is used to receive the control signal output by the bilateral disturbance suppression control device H. The electric vehicle battery load G has ports G1, G2 and G3. Port G1 is connected to port F2 of the BUCK topology DC / DC converter F to receive DC power after voltage regulation. Port G3 is used to supply power to the vehicle's electrical load. Port G2 is used to provide output voltage information to the bilateral disturbance suppression control device H. The tracking differentiator H1 has ports H1-1 and H1-2. Port H1-1 is configured to receive a voltage reference signal from an external signal source, and port H1-2 is connected to the dynamic fusion compensator H2 to output a smoothed signal. The focus state estimator H3 has ports H3-1 and H3-2. Port H3-2 is configured to receive the output voltage information of the electric vehicle battery load port G2. Port H3-1 is used to output the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate to port H2-2 of the dynamic fusion compensator H2. The dynamic fusion compensator H2 has ports H2-1, H2-2, H2-3 and H2-4. Port H2-1 is connected to port H1-2 of the tracking differentiator H1 to receive signals. Port H2-2 is connected to port H3-1 of the focus state estimator H3 to receive the estimated value of the electric vehicle battery load output voltage and the estimated value of the electric vehicle battery load output voltage change rate. Port H2-3 is configured to output control signals to drive the BUCK topology DC / DC converter F. Port H2-4 is configured to output control signals to the vehicle main control communication module I. The vehicle-mounted main control communication module I is provided with port I1, which is connected to port H2-4 of the dynamic fusion compensator H2 to receive control signals, and establishes a two-way wireless communication link with the ground controlled communication module J of the ground power supply device to exchange communication status information and control commands.

5. The wireless charging system based on focus state estimation and dynamic fusion compensation according to claim 4, characterized in that: The specific method by which the primary-side execution controller K sends control commands to the H-bridge inverter A is as follows: Step S1: The primary-side execution controller K receives the control quantity u transmitted from the ground-controlled communication module J; Step S2: The primary-side execution controller K generates a reference square wave signal with a duty cycle of 50% as the drive signal for the switching transistors Q1 and Q4 of the H-bridge inverter A. Step S3: The primary-side execution controller K uses the control quantity u as a relative delay. This relative delay is the dimensionless ratio of the time delay of the square wave signals used to control the switches Q2 and Q3 of the H-bridge inverter A relative to the square wave signals used to control the switches Q1 and Q4 of the H-bridge inverter A, to the duty cycle of the square wave signals used to control the switches Q2 and Q3 of the H-bridge inverter A. This relative delay is applied to the reference square wave signal generated in step S2, thereby generating another drive signal to control the switches Q2 and Q3 of the H-bridge inverter A. The phase difference between the two drive signals constitutes the regulated phase shift angle. The phase shift angle is used to control the effective value of the high-frequency AC power output by the H-bridge inverter A by adjusting the phase difference between the two drive signals of the H-bridge inverter, thereby regulating the energy transmitted to the vehicle side to stabilize the output voltage. Step S4: The primary-side execution controller K sends the phase shift angles corresponding to the two generated drive signals to the H-bridge inverter A; Step S5: The primary side execution controller K continuously monitors the communication port of the ground controlled communication module J to determine whether a new control quantity u has been received. If a new control quantity is received, it immediately returns to step S3 and updates the relative delay to adjust the phase shift angle. If no new signal is received, it maintains the output of the current drive signal.

6. The wireless charging system based on focus state estimation and dynamic fusion compensation according to claim 5, characterized in that: The focal state estimator H3 is structurally and functionally reconstructed based on the nonlinear extended state observer in traditional active disturbance rejection control, removing the input channel of the control quantity u and the channel specifically used to output the total disturbance estimate. The port is used, and the nonlinear function fal is replaced with the ultra-smooth kernel function usk to calculate the estimated value of the electric vehicle battery load output voltage. and the estimated value of the rate of change of output voltage of electric vehicle battery load The specific method is as follows: Obtain the actual output voltage of the electric vehicle battery load G. ; Based on actual output voltage and Calculate the state observation error ; Update the state observations based on the following dynamic equation: ； ； in, For observer bandwidth parameters, for The first derivative, for The first derivative, This is a super-smoothing kernel function used to map the state observation error e. Output updated and ; The super-smoothing kernel function It is configured to map the observation error e using the following functional relationship: when hour, ； when hour, ； Where e is the state observation error, To meet The nonlinear parameters, The width parameter of the linear interval is greater than 0.

7. The wireless charging system based on focus state estimation and dynamic fusion compensation according to claim 6, characterized in that: The dynamic fusion compensator H2 is based on the nonlinear state error feedback controller in traditional active disturbance rejection control, and undergoes structural and functional reconstruction to calculate the initial control quantity. and the initial control quantity The specific method for performing dynamic fusion compensation calculations to obtain the final control quantity u is as follows: Receive target tracking signal from tracking differentiator H1 and its differential and the estimated output voltage of the electric vehicle battery load from the focus state estimator H3. and the estimated value of the rate of change of output voltage of electric vehicle battery load ; Set the actual output voltage of the input in the nonlinear extended state observer Equal to 0 simplifies the state equation of the nonlinear extended state observer. Simultaneously, it sets the control quantity u to the initial control quantity. compositional relationship Substituting the simplified state equations and using the bandwidth method to simplify the observer gain parameters, while linearizing the nonlinear function fal, we obtain the modified state equations; then, we compare the modified state equations with the composition relation. Perform a Laplace transform to obtain the Laplace transform of the control variable u. To the initial control quantity Laplace transform transfer function To achieve the initial control quantity Perform dynamic fusion compensation; ； in, For controller bandwidth, and , This is the bandwidth ratio factor. This is the estimated control gain value; It is a complex frequency.

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