RECEIVER AND METHOD FOR DETERMINING PHASE INFORMATION OF A RECEIVED SIGNAL AND SYSTEM AND METHOD FOR MEASURING INFORMATION ABOUT A DISTANCE TO AN OBJECT

Abstract

A receiver (100) for determining phase information (125) (φ R ) of a received signal (105) (A R ), with the following characteristics: a sample supplyer (110) for providing at least one pair of samples (115) (A, B; A1, B1), wherein the sample supply (110) is designed to receive the signal (105) (A R ) with a sampling rate (111) (f Abt ) to sample frequencies smaller than a mean frequency (101) (f CW ) of the received signal (105) (A R ) is, wherein the sample provider (110) is designed to provide as a pair of samples (115) (A, B; A1, B1) two samples which have a time interval (T Abt ) from 1 4 TCW ( 4 m + 1 ) or from 1 4 TCW ( 4 m + 3 ) exhibit where m is an integer greater than or equal to 1, and where T CWa mean period of the received signal (105) (A R ) is, wherein the sampler (110) includes an analog-to-digital converter (112) (ADC) for sampling the received signal (105) (A R ) and a pair determiner (114) for determining at least one pair of samples (115) (A, B; A1, B1) based on a sample signal (113) sampled by the analog-to-digital converter (112) (ADC); wherein the analog-to-digital converter (112) (ADC) is designed to convert the received signal (105) (A R ) with the sampling rate (111) (f Abt ) to scan, wherein the value pair determiner (114) is designed to decimate (312) the sampled signal (113) from the analog-to-digital converter (112) (ADC) by a decimation factor of 4 in order to obtain a first sample (A) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to adjust the sampled signal (113) from the analog-to-digital converter (112) (ADC) by the time interval (T Abt ) to delay (322) and by a time interval (T) Abt ) to decimate the delayed delay signal (323) by the decimation factor 4 (324) in order to obtain a second sample (B) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is configured to determine a first pair (A, B) and a second pair (A1, B1) of the at least one pair of sample values ​​(115) (A, B; A1, B1) based on a sample signal (113) sampled by the analog-to-digital converter (112) (ADC); wherein the pair determiner (114) is designed to decimate (312) the sampled signal (113) from the analog-to-digital converter (112) (ADC) by a decimation factor of 4 in order to obtain a first sample (A) of the first pair (A, B) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to adjust the sampled signal (113) from the analog-to-digital converter (112) (ADC) by the time interval (T Abt ) to delay (322) and by a time interval (T) Abt ) to decimate the delayed delay signal (323) by the decimation factor 4 (324) to obtain a second sample (B) of the first pair (A, B) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to adjust the sampled signal (113) from the analog-to-digital converter (112) (ADC) by twice the time interval (T Abt) to delay (332) and by twice the time interval (T Abt ) to decimate the delayed delay signal (333) by the decimation factor 4 (334) to obtain a first sample (A1) of the second pair (A1, B1) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to adjust the sampled signal (113) from the analog-to-digital converter (112) (ADC) by three times the time interval (T) Abt ) to delay (342) and by three times the time interval (T Abt ) to decimate the delayed delay signal (343) by the decimation factor 4 (344) to obtain a second sample (B1) of the second pair (A1, B1) of the at least one pair of samples (115) (A, B; A1, B1).

Description

[0001] Embodiments of the present invention relate to a receiver and a method for determining phase information of a received signal. Further embodiments of the present invention relate to a system and a method for measuring information about a distance to an object. Further embodiments of the present invention relate to a method for measuring the distance and the reflectivity properties of an object using a CW ("Continuous Wave") LIDAR ("Light Detection and Ranging") method, for example, using a real-time capable algorithm.

[0002] In a well-known CW ("Continuous Wave") LIDAR ("Light Detection and Ranging") method, an intensity-modulated light signal is sent to the object, which is then reflected back to a detector. The time it takes for the light beam to travel to the object and back to the detector reveals a phase shift between the emitted, intensity-modulated light and the detected signal. Distance=Speed⋅Time R∗= c⋅tL where R* is the distance, c is the speed and t L the time is.

[0003] If the phase shift between both signals can be determined, a conclusion can be drawn about the distance to the object (cf. Fig. 12, Fig. 13). In Fig. Figure 12 shows an illustration of a CW method (outgoing wave 2, reflected wave 4). Fig. Figure 13 shows the structure of a known CW system.

[0004] If the intensity of the laser A T (T = “Transmit”) for example with a sine function of period T CW modulated AT(t)=α^T⋅sin(2πTcwt+φT) the received detector signal A contains R (R = “Receive”) is also a sinusoidal function of period T CW AR(t)=α^R⋅sin(2πTCWt+φR)

[0005] The phase difference Δφ between emitted signal A T (t) and received signal A R (t) is determined by subtracting both phases Δφ=φT−φR

[0006] From the relationship between the propagation time of the signal t L and the period T CW and the phase difference Δφ / 2π results in the ratio tLTcw=Δφ2π

[0007] From this relationship, the distance R (“range”) to the object can be calculated R=c2⋅Tcw2π⋅Δφ

[0008] From equation 6 above, the speed of light c and the period T are known CW of the intensity-modulated light signal. The factor 1 / 2 is explained by the fact that the light travels twice the distance 2R (round trip). However, since only the distance to the object is measured, it is divided by a factor of 2. To determine the distance, the phase difference Δφ between the emitted and the received, intensity-modulated light wave should be determined.

[0009] In Fig. Figure 13 shows an example of a known CW LIDAR system. As shown in Fig. As shown in Figure 13, a laser diode (LD) with the sinusoidal function A T (t), which modulates the laser intensity sinusoidally. A photodiode (PD) converts the reflected light into electrical signals. Subsequent signal conditioning generates the received signal A R(t). The functions A T (t) and A R (t) differ in amplitude and phase, but not in frequency.

[0010] One difficulty with a scanning CW LIDAR system is the constantly changing phase at the detector. Imagine a comb structure (see Fig. 14). In Fig. Figure 14 shows a scan of a ridge peak (left) and a scan of a valley (right). If the light beam points toward a ridge peak, the detector experiences a different phase shift than if the light beam were radiating exactly between two ridge peaks. The faster the light beam is scanned across a ridge peak and the ridge valley, the faster the analysis algorithm should be able to detect and evaluate the different phase signals.

[0011] In Fig. 15 shows two exemplary measurements that have a different phase difference (resulting from the distance between the transmitter and the object). Fig. 15 shows a phase relationship between the outgoing and detected signal.

[0012] If the intensity is modulated with a sine function as described above, the emitted (outgoing), intensity-modulated light wave is band-limited. If a filter stage is also present in the return path, i.e., from the detector to the receiver, such as the ADC (analog-to-digital converter), a bandpass signal is sampled at the receiver or ADC.

[0013] The phase information can be extracted from a bandpass signal, for example, by a quadrature downconversion. In this case, an ideal BP 2 -Signal accepted UBP(t)=u^⋅sin(2πƒMt+φ)

[0014] In quadrature downconversion, the real BP signal (bandpass signal) is multiplied by two sine waves that are 90° out of phase with each other. This is Fig. 16 shown. In Fig. 16 shows an example of quadrature demodulation. From the trigonometric functions sin(a)⋅sin(b)=12⋅(cos(a−b)−cos(a+b)) sin(a)⋅cos(b)=12⋅(sin(a−b)+sin(a+b)) It can be seen that as soon as the arguments (a and b) are identical, two signal components are created. The component at twice the frequency should be removed by a suitable filter. Applied to equation 7, after a TP (low-pass) filter, 12cos(φ) 12sin(φ)

[0015] Thus, after the TP filter, we obtain a complex pair of values ​​(equations 10 and 11) with two different values ​​from which the phase relationship and amplitude can be calculated.

[0016] A disadvantage of quadrature demodulation according to Fig. 16a, however, is that half of the amplitude is lost during demodulation and a group delay is added by the TP filter.

[0017] DE 100 01 015 A1 describes a method for measuring the distance of objects, atmospheric particles, and the like using LIDAR or laser radar signals. This known method is essentially based on an FFT ("Fast Fourier Transform").

[0018] US 2011 / 0317147 A1 describes a known method using time-shifted PN (“pseudorandom noise”) codes for CW LIDAR, radar, and sonar. This known method performs FM (frequency modulation) demodulation for a CW signal. This FM demodulation occurs as described in Fig. 10 of this document, before data acquisition. In particular, this known method requires additional resources.

[0019] In Bogatscher, S. et al., “Laser rangefinder based on MEMS mirrors for adaptive robotics,” Microsystems Technology Congress 2013, October 14-16, 2013 in Aachen, another well-known LIDAR system is described. In particular, a block diagram of Fig. 3 describes a LIDAR system and its analysis. Among other things, a laser can be seen operating at the frequency f m modulated. Furthermore, it can be seen that an ADC is sampled at 8 times the modulation frequency. Subsequently, a conventional demodulation is performed using two multipliers and low-pass filters.

[0020] Further range finders are from the document Poujouly S., Journet B.: “A twofold modulation frequency laser range finder”, In: Journal of Optics A: Pure and Applied Optics, No. 4, 2002, pages 356 to 363, from the document US 7 177 014 B2 and from the document Zhi-yuan Song, Shao-lan Zhu, Li-jun Dong, Li Feng, Hao-dong He, “Applications of digital IF receivers and under-sampling techniques in ladar,” International Symposium on Photoelectronic Detection and Imaging 2011: Laser Sensing and Imaging; and Biological and Medical Applications of Photonics Sensing and Imaging, Proc. Of SPIE Vol. 8192, 819249, 2011 SPIE, doi: 10.1117 / 12.901928.

[0021] A general problem with the LIDAR systems described above is that they are relatively complicated and at the same time involve a comparatively high computational effort and the use of additional resources.

[0022] The object of the present invention is therefore to provide a concept for determining phase information of a received signal that enables a simpler, more efficient and more resource-saving implementation.

[0023] This object is achieved by a receiver according to claim 1, a system according to claim 8, a method according to claim 13, a method according to claim 14 and a computer program according to claim 15.

[0024] Embodiments of the present invention provide a concept for determining phase information of a received signal using a sample value provider, wherein the sample value provider is designed to sample the received signal at a sampling rate that is lower than a mean frequency of the received signal. Here, the sample value provider is designed to provide two samples as a pair of samples that are spaced apart by a time interval of 14TCW(4m+1) or from 14TCW(4m+3) where m is an integer greater than or equal to (≥) 1, and where T CW is an average period of the received signal.

[0025] The core idea of ​​the present invention is that the aforementioned simpler, more efficient, and more resource-saving implementation can be achieved if the received signal is sampled at a sampling rate that is lower than a mean frequency of the received signal, and if two sample values ​​are provided as a pair of samples that have the aforementioned time interval. This avoids a high level of implementation complexity and, at the same time, a comparatively high computational effort or the use of additional resources. Thus, the simpler, more efficient, and more resource-saving implementation can be enabled.

[0026] In further embodiments of the present invention, the receiver comprises a parameter calculator for calculating the phase information of the received signal based on the pair of sample values. Thus, the phase information of the received signal can be determined for calculating information about the distance to an object.

[0027] Further embodiments of the present invention provide a system for measuring information about a distance to an object, comprising a transmitter, a receiver, and a control unit. The transmitter is configured to transmit a transmitted signal. The receiver is configured to receive a received signal. As described above, the receiver is configured to determine phase information of a received signal. The control unit is configured to control a mean frequency of the transmitted signal and the sampling rate as a function of a clock frequency of the system.

[0028] In further embodiments of the present invention, the receiver of the system comprises a parameter calculator for calculating the phase information of the received signal based on the pair of sample values. Furthermore, the parameter calculator is configured to calculate the information about the distance to the object based on phase information of the transmitted signal and the phase information of the received signal.

[0029] Thus, according to embodiments, the phase information of the received signal or the information about the distance to the object can be determined.

[0030] Embodiments of the present invention are explained in more detail below with reference to the accompanying figures, in which identical or equivalent elements are designated by identical reference numerals. They show: Fig. 1a shows a receiver for determining phase information of a received signal according to an embodiment of the present invention; Fig. 1b an exemplary illustration of providing at least one pair of samples with a sample provider of the embodiment of the receiver according to Fig. 1a; Fig. 2 a block diagram of an embodiment of the sample provider of the embodiment of the receiver according to Fig. 1a; Fig. 3 a block diagram of another embodiment of the sample provider of the embodiment of the receiver 100 according to Fig. 1a; Fig. 4 a block diagram of another embodiment of the sample value provider of the embodiment of the receiver according to Fig. 1a; Fig. 5 a block diagram of an embodiment of the value pair determiner of the embodiment of the sample value provider according to Fig. 4; Fig. 6a is a block diagram of an embodiment of a parameter calculator of the embodiment of the receiver according to Fig. 1a; Fig. 6b, c a schematic diagram and a block diagram respectively to illustrate an exemplary time-shifted sampling; Fig. 6d an exemplary equivalent circuit diagram of an ideal scanner; Fig. 7a-d show embodiments of a time-shifted sampling with various exemplary parameters; Fig. 8a an illustration of an exemplary power distribution during scanning; Fig. 8b shows an illustration of an exemplary scan area of ​​a 2D scanner without adjustment or with adaptive scan area adjustment; Fig. 8c an illustration of an exemplary measurement according to the embodiment of Fig. 4; Fig. 9 shows a system for measuring information about a distance to an object according to an embodiment of the present invention; Fig. 10a-c Details of the Fig. 9 according to further embodiments of the present invention; Fig. 11 shows a further embodiment of a time-shifted sampling; Fig. 12, Fig. 13 an illustration or a structure of a known CW system; Fig. Figure 14 is an illustration of a ridge top scan and a valley scan for a known CW system; Fig. 15 a phase relationship between outgoing and detected signal in a known CW system; Fig. 16 is a block diagram of a known quadrature demodulation; Fig. 17 a table summarising various terms used in the description; and Fig. 18a-f inventive algorithms with legend for simulating an exemplary time-shifted sampling.

[0031] Before the present invention is explained below with reference to the figures, it should be noted that in the exemplary embodiments presented below, identical elements or functionally identical elements are provided with the same reference numerals in the figures. A description of elements with the same reference numerals is therefore interchangeable and / or applicable to one another in the various exemplary embodiments.

[0032] Fig. 1a shows a receiver 100 for determining phase information 125, φ R a received signal 105, A R , according to an embodiment of the present invention. As shown in Fig. 1a, the receiver 100 comprises a sample provider 110 for providing at least one pair of samples 115 (A, B; A1, B1). The sample provider 110 is designed to convert the received signal 105, A R , with a sampling rate of 111, f Abt , which is less than a mean frequency (f CW ) of the received signal 105, A R , is.

[0033] In embodiments, the received signal 105, A R , a sinusoidal signal a^Rsin(2πƒCWt+φR) where â R and φ R the amplitude information or the phase information and f CW the mean frequency of the received signal 105, A R , represent.

[0034] Fig. 1b illustrates, by way of example, providing at least one pair of sample values ​​115 (A, B; A1, B1) with the sample value provider 110 of the embodiment of the receiver 100 according to Fig. 1a. In Fig. 1b shows exemplary diagrams 10, 20, 30 and 40. The exemplary diagrams 10, 20 each illustrate a time interval T Abt two sample values ​​or the pair of sample values ​​115 (A, B; A1, B1), while the exemplary diagrams 30, 40 each illustrate the corresponding pair of sample values ​​115 (A, B; A1, B1).

[0035] In the exemplary diagram 10, a time interval T Abt from 14TCW(4m+1) where m is equal to 1, for example, and where T CW the average period of the received signal 105, A R , is. In the exemplary diagram 20, a time interval T Abt from 14TCW(4m+3) where m is an integer, for example 1, and where T CW the average period of the received signal 105, A R , is.

[0036] Furthermore, in the exemplary diagram 10, the time interval T Abt a 5 / 4 times the average period of the received signal 105, A R , while in the exemplary diagram 20 the time interval T Abt a 7 / 4 times the average period of the received signal 105, A R , corresponds to. As shown in the exemplary diagrams 10, 20, the time interval T Abt each greater than the mean period T CW of the received signal 105, A R . The respective time interval T Abt In the exemplary diagrams 10, 20, for example, the sampling rate 111, f Abt , while the mean period T CW of the received signal 105, AR , the mean frequency (f CW ) of the received signal 105, A R , Thus, the exemplary diagrams 10, 20 essentially represent a sampling of the received signal 105, A R , with a sampling rate of 111, f Abt , where the sampling rate is 111, f Abt , smaller than the mean frequency (f CW ) of the received signal 105, A R , In other words, the exemplary diagrams 10, 20 essentially illustrate a subsampling of the received signal 105, A R .

[0037] In exemplary embodiments, m can be chosen as an integer greater than or equal to (≥) 1. By adjusting m, the time interval T Abt increased accordingly or the sampling rate 111, f Abt , be reduced accordingly.

[0038] In the exemplary diagrams 10, 20, four samples A, B; A1, B1 are shown, wherein the samples A, B form a first pair and the samples A1, B1 form a second pair of samples. The time offset between the first pair A, B and the second pair A1, B1 of samples is determined by the time interval T. Abt given.

[0039] The exemplary diagrams 30, 40 each show the corresponding sample values ​​A, B; A1, B1 as a sequence s(n) of sample values. Here, n denotes an index (n = 0, 1, 2, 3, ...) of the respective sample value A, B; A1, B1 of the sequence s(n).

[0040] Fig. 2 shows a block diagram of an embodiment of the sample provider 110 of the embodiment of the receiver 100 according to Fig. 1a. In the Fig. 2, the sample value provider 110 comprises an analog-to-digital converter 112 (ADC) for sampling the received signal 105, A R , and a value pair determiner 114 for determining the at least one pair of sample values ​​115 (A, B; A1, B1) based on a sample signal 113 sampled by the analog-to-digital converter 112 (ADC). Here, the analog-to-digital converter 112 (ADC) is designed, for example, to determine the received signal 105, A R , with sampling rate 111, f Abt , to scan.

[0041] Fig. 3 shows a block diagram of another embodiment of the sample value provider 110 of the embodiment of the receiver 100 according to Fig. 1a. In Fig. 3 shows a time-shifted sampling with an ADC. In the Fig. 3, the sample value provider 110 comprises an analog-to-digital converter 112 (ADC) for sampling the received signal 105, A R , and a value pair determiner 114 for determining the at least one pair of sample values ​​115 (A, B; A1, B1) based on a sample signal 113 sampled by the analog-to-digital converter 112 (ADC). Here, the analog-to-digital converter 112 (ADC) is designed to convert the received signal 105, A R , with sampling rate 111, f Abt , to sample. For example, the value pair determiner 114 is designed to decimate the sampled signal 113 sampled by the analog-to-digital converter 112 (ADC) with a decimation factor of 4 (block 312) in order to obtain a first sampled value A of the at least one pair of sampled values ​​115 (A, B; A1, B1). Furthermore, the value pair determiner 114 can be designed to decimate the sampled signal 113 sampled by the analog-to-digital converter 112 (ADC) by the time interval T Abtto delay (Block 322) and to delay by the time interval T Abt to decimate the delayed delay signal 323 with the decimation factor 4 (block 324) in order to obtain a second sample value B of the at least one pair of samples 115 (A, B; A1, B1).

[0042] Block 322 in Fig. 3 is, for example, a delay unit Z -TAbt to delay by the time interval T Abt , while blocks 312, 324 in Fig. 3, for example, are decimators for decimating with the decimation factor 4.

[0043] Fig. 4 shows a block diagram of another embodiment of the sample provider 110 of the embodiment of the receiver 100 according to Fig. 1a. In Fig. 4 shows an extended setup of the time-shifted sampling or a block diagram of a “LIDAR Core Algorithm”. Fig. 4, the sample value provider 110 comprises an analog-to-digital converter 112 (ADC) for sampling the received signal 105, A R , and a value pair determiner 114 for determining a first pair A, B and a second pair A1, B1 of the at least one pair of sample values ​​115 (A, B; A1, B1) based on a sample signal 113 sampled by the analog-to-digital converter 112 (ADC). Here, the analog-to-digital converter 112 (ADC) is designed to convert the received signal 105, A R , with sampling rate 111, f Abt, to sample. For example, the value pair determiner 114 is designed to decimate the sampled signal 113 sampled by the analog-to-digital converter 112 (ADC) with a decimation factor of 4 (block 312) in order to obtain a first sampled value A of the first pair A, B of the at least one pair of sampled values ​​115 (A, B; A1, B1). Furthermore, the value pair determiner 114 can be designed to decimate the sampled signal 113 sampled by the analog-to-digital converter 112 (ADC) by the time interval T Abt to delay (Block 322) and to delay by the time interval T Abt to decimate the delayed delay signal 323 with the decimation factor 4 (block 324) in order to obtain a second sample value B of the first pair A, B of the at least one pair of samples 115 (A, B; A1, B1).

[0044] Furthermore, the value pair determiner 114 can be designed to delay the sampling signal 113 sampled by the analog-to-digital converter 112 (ADC) by twice the time interval T Abt to be delayed (Block 332) and by a time interval twice the time interval T Abt Delayed delay signal 333 is decimated by the decimation factor 4 (block 334) to obtain a first sample value A1 of the second pair A1, B1 of the at least one pair of sample values ​​115 (A, B; A1, B1). Furthermore, the value pair determiner 114 can be designed to decimate the sample signal 113 sampled by the analog-to-digital converter 112 (ADC) by three times the time interval T Abt to be delayed (Block 342) and by a time interval three times the time interval T Abtto decimate the delayed delay signal 343 with the decimation factor 4 (block 344) in order to obtain a second sample value B1 of the second pair A1, B1 of the at least one pair of samples 115 (A, B; A1, B1).

[0045] For example, blocks 322, 332, 342 in Fig. 4 a delay unit Z -TAbt to delay by the time interval T Abt , a delay unit Z -2TAbt to delay by twice the time interval T Abt or a delay unit Z -3TAbt to delay by three times the time interval T Abt , while blocks 312, 324, 334, 344 in Fig. 4 are each a decimator for decimating with the decimation factor 4.

[0046] Optionally, the value pair determiner 114 is configured to multiply the first sample A1 of the second pair A1, B1 of the at least one pair of sample values ​​115 (A, B; A1, B1) by a factor (-1) (block 335) to obtain a modified first sample A1'. Furthermore, the value pair determiner 114 is configured (optionally) to multiply the second sample B1 of the second pair A1, B1 of the at least one pair of sample values ​​115 (A, B; A1, B1) by a factor (-1) (block 345) to obtain a modified second sample B1'.

[0047] Blocks 335, 345 in Fig. 4, for example, are multipliers for multiplying by the factor (-1).

[0048] Referring to Fig. 4, an extension of the subsampling can be realized. According to embodiments, the subsampling can be extended, for example, to improve the signal-to-noise ratio (SNR). For example, compare Fig. 3 with the corresponding simulation in Fig. 7a. The real or imaginary part is determined, for example, from one of four sample values ​​( Fig. 7a). In embodiments, the last two samples (A1, B1) are discarded. However, it can be seen that the last two samples (A1, B1) correspond to the previous samples (A, B) when multiplied by the factor -1. In Fig. The extended setup is shown schematically in Figure 4. All other boundary conditions, such as the ratio of the sampling frequency f Abt to the applied frequency f CW are identical to those according to the embodiment of Fig. 3. The first two branches are made of Fig. 3. In the third branch, the signal is delayed by two clock cycles (block 332) and then decimated by a factor of 4 (block 334). In the fourth branch, it is delayed by three clock cycles (block 342) and also decimated by a factor of 4 (block 344). In this structure, in addition to the already known complex value pair (A + jB), we receive another complex value pair (A1 + jB1) or (A1' + jB1'). In embodiments, the addition (e.g., blocks 510, 520 in Fig. 5) combining both value pairs into one value pair, the signal quality can be improved. Furthermore, in further embodiments, by retaining the value pairs, a pixel can now be divided into two subpixels.

[0049] The extension according to the embodiment of Fig. 4 allows compared to the embodiment of Fig. 3 a doubling of the pixel resolution or an improvement of the SNR by a double measurement within a pixel.

[0050] For this purpose, Fig. 8c shows an example of how the dual measurement further improves the image quality (amount) of a noisy image. Fig. 8c is a comparison of both methods (according to the block diagrams of Fig. 3 and Fig. 4). Above is shown an ideal magnitude image, bottom left the original method (according to the example of Fig. 3), bottom right the modified method (according to the example of Fig. 4), with an SNR of -10 dB each.

[0051] The extension of subsampling described above according to Fig. 4 is also based on the simulation result of Fig. 7a. For a complex pair of values ​​(“Real Value” / “Imag Value”), Fig. 3 two samples are taken from the “Sampled Values” row (first and second sample A, B). The third and fourth samples (A1, B1) are calculated according to Fig. 3, for example, was rejected.

[0052] However, it was recognized that the discarded sample points (A1, B1) can be multiplied by a factor of -1 to assume the values ​​of the first and second sample values ​​(A, B). The embodiment of Fig. 4.

[0053] Fig. 5 shows a block diagram of an embodiment of the value pair determiner 114 of the embodiment of the sample value provider 110 according to Fig. 4. In the Fig. 5, the value pair determiner 114 is designed to combine the first sample A of the first pair A, B of the at least one pair of sample values ​​115 (A, B; A1, B1) and the first sample A1 of the second pair A1, B1 of the at least one pair of sample values ​​115 (A, B; A1, B1) (block 510) to obtain a first combination value C. Furthermore, the value pair determiner 114 can be designed to combine the second sample B of the first pair A, B of the at least one pair of sample values ​​115 (A, B; A1, B1) and the second sample B1 of the second pair A1, B1 of the at least one pair of sample values ​​115 (A, B; A1, B1) (step 520) to obtain a second combination value C1.

[0054] In Fig. 5, the sample signal 113 sampled by the analog-to-digital converter 112 (ADC) is also referred to as “ADC_Sample”. Furthermore, in Fig. 5, the first combination value C is designated "real," while the second combination value C1 is designated "imag." The blocks 322, 332, 342, 312, 324, 334, 344 in Fig. 5 essentially correspond to the corresponding blocks in Fig. 4. Blocks 322, 332, 342 in Fig. 5 are each marked with the symbol 501, Z -n , Here, the symbol 501 represents an n-th order delay element or a delay unit for delaying by n times the time interval T Abt Furthermore, blocks 312, 324, 334, 344 are in Fig. 5 is marked with the symbol 502. Here, the symbol 502 represents a “downsampling” by a factor of 4 or a decimator for decimating by a decimation factor of 4. Furthermore, Fig. 5, blocks 510, 520 are labeled with the symbol 503. Symbol 503 represents a combiner (e.g., adder).

[0055] The Fig. The embodiments shown in Figures 3 to 5 essentially represent different implementations for a value pair determination with the value pair determiner 114. For example, the sample values ​​A, B, A1, B1, A1', B1' or the combination values ​​C, C1 can be determined flexibly. Thus, in the embodiments according to Fig. 3 to 5 a flexible implementation for the sample value or combination value determination can be realized.

[0056] Fig. 6a shows a block diagram of an embodiment of a parameter calculator 120 of the embodiment of the receiver 100 according to Fig. 1a. As in Fig. 6a, the receiver 100 comprises, for example, a parameter calculator 120 for calculating the phase information 125, φ R , of the received signal 105, A R , based on the at least one pair of samples 115 (A, B; A1, B1).

[0057] For example, the parameter calculator 120 is designed to calculate the phase information 125, φ R , of the received signal 105, A R , using the equation φR=tan−1(B / A) where A and B are two samples of the pair of samples 115 (A, B; A1, B1) which have the time interval T Abt have.

[0058] Thus, according to embodiments, the phase information of the received signal can be determined for measuring information about a distance to an object.

[0059] The sampling of the received signal with the sample value provider 110 described above can also be referred to as a time-shifted sampling of the received signal or the detector signal. Embodiments of the present invention have the advantage that the time-shifted sampling of the detector signal makes it possible to select a specific sampling frequency or sampling rate (f Abt ) to be able to perform a direct downmixing (folding) to the zero position (DC component).

[0060] The time-delayed scanning according to the invention can be visualized with a wheel rotating at a known speed. This wheel is illuminated by a strobe light. Since the wheel's speed is known, a frequency for the strobe light can be calculated at which one gets the impression that the wheel is not rotating.

[0061] We further imagine that the wheel has only one spoke and that this spoke corresponds to a pointer in the unit circle (cf. Fig. 6b). Fig. Figure 6b shows various positions in the unit circle. If the real and imaginary components of the pointer can be determined, the phase position of the wheel can be calculated.

[0062] The following describes the mathematical derivation of how the appropriate sampling frequency can be determined at which the wheel comes to a standstill, so to speak.

[0063] Fig. Figure 6d shows an equivalent circuit diagram of an exemplary ideal scanner. As in Fig. As can be seen in Figure 6d, in an ideal ADC a BP signal (rotational speed of the wheel) can be multiplied by a Dirac pulse.

[0064] In the following derivation, the multipliers are Fig. 16, for example, is replaced by two ADCs. These ADCs sample the bandpass signal U BP(t) with frequency f Abt The clock of one ADC is, for example, 90° out of phase with the other ADC (cf. Fig. 6c). Fig. Figure 6c shows a schematic setup of a time-shifted sampling with two ADCs.

[0065] The Dirac impulse as a Fourier series representation is δ(t)=12π+∑i=−∞∞1π⋅cos(2πfAbt⋅i⋅t)|⋅2π δ(t)=1+∑i=−∞∞2⋅cos(2πfAbt⋅i⋅t) and whose Fourier transform is Δ(f)=∑i=−∞∞δ(f−ifAbt)

[0066] The spectrum of a Dirac pulse time-shifted by φ is obtained by multiplying by e -j2πƒφ to Δ(f)=∑i=−∞∞δ(f−ifAbt)⋅e−j2πfφ Δ(f)=∑i=−∞∞δ(f−ifAbt)⋅e−j2πifAbtφ

[0067] From the ideal bandpass signal (signal of one frequency, ie bandwidth = 0), cf. Fig. 16, UBP(t)=cos(2πfCWt) and its Fourier transform UBP(f)=12δ(f−fCW)+12δ(f+fCW) The spectra of the sampled signals are created by convolution with the spectrum of the time-shifted Dirac pulse.

[0068] For channel A, which is generated by sampling the signal U BP (t) arises, results UBPA(f)=12∑i=−∞∞δ(f−ifAbt−fCW)+δ(f−ifAbt+fCW)

[0069] For channel B, which is generated by sampling the signal U BP (t), but with the sampling phase shifted by 90° to channel A, we get UBPB(f)=12∑i=−∞∞[δ(f−ifAbt−fCW)+δ(f−ifAbt+fCW)]⋅e−j2πifAbtφ

[0070] To determine the center frequency f CW To be able to mix to the zero position, the sampling frequency f Abt be chosen so that the center frequency f CW an integer multiple of the sampling frequency f Abt corresponds, so fCW=x⋅fAbt from which the spectra of the two channels then look as follows.

[0071] For channel A, UBPA(f)=12∑i=−∞∞δ(f−(i+x)fAbt)+δ(f−(i−x)fAbt) while for channel B UBPB(f)=12∑i=−∞∞[δ(f−(i+x)fAbt)+δ(f−(i−x)fAbt)]⋅e−j2πifAbtφ The channels contain the spectral components for the zero position for i = ± x.

[0072] This results in channel A UA(f)=12δ(f−0)+12δ(f+0) while for channel B UB(f)=12δ(f−0)⋅e−j2πxfAbtφ+12δ(f+0)⋅e−j2πxfAbtφ It was recognized that U A (f) and U B (f) are then phase-shifted by 90° to each other if the condition 2πxfAbtφ=π2+2π⋅m|m=integer is fulfilled. Furthermore, it was recognized that this condition is fulfilled by a suitable choice of the time offset φ, namely φ=TAbt41+4mx

[0073] A time offset of π / 2 can be achieved, among other things, by operating the ADC at 4 times the sampling frequency. For example, the input signal or the received signal can be 105, A R , in two branches by a factor of 4, whereby in one of the two branches there is initially a delay of one clock cycle (see the embodiment of Fig. 3). In this case, the time offset φ can be chosen, for example, so that φ=TAbt4 applies and 1+4mx=1 According to embodiments, a suitable sampling frequency f Abt can be calculated from the following formula. This formula is fAbt=4⋅fcw(1+4m) with m=1,2,3,4

[0074] In the following, examples of implementation of a time-shifted sampling are presented using exemplary diagrams ( Fig. 7a-d), where the time-shifted sampling is based on the equation just mentioned. In other words, Fig. 7a-d illustrate by way of example the provision of the at least one pair of sample values ​​115 (A, B; A1, B1) with the sample value provider 110 of the embodiment of the receiver 100 according to Fig. 1a.

[0075] Examples of time-shifted sampling can be found in “Attachment A” of Fig. 18a-f can be found.

[0076] Fig. 7a shows an embodiment of a time-shifted sampling with the exemplary parameters f CW = 1 MHz, m = 1 and f Abt = 4 / 5 MHz = 800 kHz. In Fig. 7a, an exemplary oversampled 1 MHz cosine signal with phase 0 can be seen in the first diagram 710. The amplitude was set to 1 for example. To display the second diagram 720, the 1 MHz cosine signal was oversampled with the sampling rate f Abt= 800 kHz, whereby in the second diagram 720 a total of four samples with the exemplary values ​​of [1 0 -1 0] can be seen. The third diagram 730 and the fourth diagram 740 correspond to the signal decimated by a factor of 4, whereby the imaginary part (“Imag”) is only delayed by 1 clock cycle. In the first to fourth diagrams 710, 720, 730, 740 of Fig. 7a, the 1 MHz cosine signal essentially corresponds to the received signal 105, A R , while the four samples with the exemplary values ​​of [1 0 -1 0] essentially correspond to the at least one pair of samples 115 (A, B; A1, B1).

[0077] In embodiments, the phase position φ R out of φR=tan−1(BA) be determined (see the example of Fig. 6a), where B = “Imag” (imaginary part) and A = “Real” (real part), while the amplitude â R of the received signal 105, A R, or the absolute value of the complex pair of values ​​from α^R=A2+B2 can be determined.

[0078] Fig. 7b shows an embodiment of a time-shifted sampling with the exemplary parameters f CW = 1 MHz, m = 2 and f Abt = 4 / 9 MHz = 444.44 kHz. The 1 MHz cosine signal in the first diagram corresponds to 710 of Fig. 7b essentially corresponds to the received signal 105, A R , from Fig. 1a, while the four samples with the exemplary values ​​of [1 0 -1 0] in the second to fourth diagrams 720, 730, 740 of Fig. 7b essentially corresponds to at least one pair of samples 115 (A, B; A1, B1) of Fig. 1a.

[0079] Fig. 7c shows an embodiment of a time-shifted sampling with the exemplary parameters f CW = 1 MHz, m = 6 and f Abt= 4 / 25 MHz = 160 kHz. Here, the 1 MHz cosine signal in the first diagram corresponds to 710 of Fig. 7c essentially corresponds to the received signal 105, A R , from Fig. 1a, while the four samples with the values ​​of [1 0-1 0] in the second to fourth diagrams 720, 730, 740 of Fig. 7c essentially the at least one pair of samples 115 (A, B; A1, B1) of Fig. 1a.

[0080] Fig. 7d shows an embodiment of a time-shifted sampling with the exemplary parameters f CW = 1 MHz, m = 1 and f Abt = 4 / 5 MHz = 800 kHz and with an exemplary 45° phase shift. Here, the 1 MHz cosine signal in the first diagram, which is phase-shifted by 45°, corresponds to 710 of Fig. 7d essentially corresponds to the received signal 105, A R , from Fig. 1a, while the four samples with the exemplary values ​​of [1 / 2 −1 / 2 −1 / 2 1 / 2] in the second to fourth diagram 720, 730, 740 of Fig. 7d essentially corresponds to at least one pair of samples 115 (A, B; A1, B1) of Fig. 1a.

[0081] In relation to Fig. 7a-d it should be noted that the time interval T Abt of the respective sample values. The time interval T corresponds to Abt the sampling rate or sampling frequency f Abt according to the equation derived above (Equation 30). Thus, according to embodiments, the time interval T Abt or the sampling frequency f Abt depending on m (m = 1, 2, 3, 4, ...).

[0082] It was therefore recognized that the time interval T Abt or the sampling frequency f AbtFor example, it can be flexibly adjusted depending on the ambient conditions of the receiver. Embodiments of the present invention thus enable a more flexible adjustment of the sampling rate for sampling the received signal in the receiver.

[0083] Furthermore, it was recognized that if the frequency position of a bandpass signal is known, a direct downmix to the zero position can be achieved by the sampling itself by choosing a specific sampling frequency.

[0084] In the following, with reference to the embodiments described above, it is explained that according to these embodiments, various received signal parameters, such as the phase information φ R or the amplitude information a R of the received signal A R and reflectivity information of the object p can be determined.

[0085] Referring to the embodiment of Fig. 3 there is a signal 105 at the input of the ADC 112, A R , whose frequency f CW (or mean period T CW ) with the modulation frequency of the laser (or with the modulation frequency of a transmission signal A emitted by a transmitter T ), where the signal 105, A R with an unknown phase position or phase information φ R As in Fig. 3, the digital signal 113 is split into two branches after sampling with the ADC 112. In the first branch, the sampled signal 113 is decimated by a factor of 4 (block 312). In the second branch, the sampling clock (T Abt ) (block 322) and then decimated by a factor of 4 (block 324). After decimation, the pixel clock (f clk ) continued working (cf. Fig. 3).

[0086] Referring to the embodiment of Fig. 6, the phase information 125, φ R , of the received signal 105, A R , or the phase position of the detector signal can be calculated using the following equation φR=tan−1(BA)

[0087] Since the phase position of the laser intensity or the phase information φ T of the signal A transmitted by a transmitter T is known, the phase difference Δφ can be calculated according to embodiments. In further embodiments, the distance R to the object can be further calculated from this phase difference Δφ (see equation 6).

[0088] In further embodiments, the amplitude information â Rof the received signal 105, A R , or the amplitude of the signal reflected from an object can be calculated as the magnitude of the complex pair of values ​​(A, B) using the following equation α^R=A2+B2

[0089] Furthermore, according to embodiments, the amplitude of the detector signal (â R ) Conclusions can be drawn about the reflectivity properties of the object. In a LIDAR system, the optical design and the sensitivity of the photodiode are given. Fig. Figure 8a shows a schematic representation of a power distribution during detection or scanning with a LIDAR system. Fig. 8a are exemplary equations for the power transmitted by a transmitter (e.g. aircraft), P T , for the power received by the transmitter (e.g. aircraft), Pr=Ar2πR2⋅M⋅ρ⋅M⋅PT the power received at an object (e.g. ground surface), M · P T, as well as the power reflected from the object (e.g. ground surface) ψ2π⋅ρ⋅M⋅PT assuming a Lambertian surface. Here, P R the received light power, P T the transmitted light power, M the atmospheric transmission, A r the area of ​​the receiving optics, p the reflectivity of the surface and R the distance to the surface.

[0090] If the equation for the power received by the transmitter is Fig. 8a is rearranged to p, we get p=PRPT⋅2π⋅R2M2⋅Ar

[0091] The P factor R / P T can be adjusted with the amplitude of the outgoing signal â T and the received signal â R Thus, according to embodiments, the reflectivity of the object can be calculated using the following formula p=α^Rα^T⋅2π⋅R2M2⋅Ar

[0092] Fig. Figure 9 shows a system 200 for measuring information about a distance to an object 129, Δ φ , R. As in Fig. 9, the system 200 comprises a transmitter 10, a receiver 100 and a control unit 210. The transmitter 10 is designed to transmit a transmission signal 15, A T , to transmit. The receiver 100 is designed to receive a signal 105, A R , to receive. Here, the receiver 100 of the embodiment of Fig. 9 essentially the recipient of the embodiment of Fig. 1a. The control unit 210 is designed to provide a mean frequency 11, f CW ', of the transmission signal 15, A T , and the sampling rate 111, f Abt , depending on a clock frequency 205, f clk , to control the system 200.

[0093] In embodiments, the transmission signal 15, A T , a sinusoidal signal α^Tsin(2πfCW,t+φT) where âT and φ T the amplitude information or the phase information and f CW ' the mean frequency of the transmitted signal 15, A T , represent.

[0094] Furthermore, it was recognized that in the system 200 the functions A T (t) and A R (t) for the transmitted signal 15 and the received signal 105 differ only with respect to their amplitude information and their phase information. These functions A T (t) and A R (t) essentially not in their frequency (f CW '=f CW ). In other words, if a light signal (e.g., the transmitted signal 15) is intensity modulated with a band-limited function, we also obtain a band-limited signal (signal 105) at the detector.

[0095] In embodiments, the control unit 210 comprises a center frequency provider 212 and a sampling rate provider 214. Here, the center frequency provider 212 is designed, for example, to provide the center frequency 11, f CW ', of the transmission signal 15, A T , depending on the clock frequency 205, f clk , to provide. Furthermore, the sampling rate provider 214 can be designed to provide the sampling rate 111, f Abt , depending on the clock frequency 205, f clk , to provide.

[0096] In embodiments, the control unit 210 is designed to control the clock frequency 205, f clk , multiply by a factor of 4 to get the sampling rate 111, f Abt , to provide.

[0097] In embodiments, the system 200 includes a clock frequency provider 201 for providing the clock frequency 205, f clk, for the control unit 210. For example, the clock frequency provider 201 is designed to provide the clock frequency 205, f clk , for the control unit 210 based on a desired resolution (N px ) of a scanning range of the system 200 and at an oscillation frequency (f s ) of a scanning mirror of the system 200.

[0098] In embodiments, the receiver 100 comprises a parameter calculator 120 for calculating the phase information 125, φ R , of the received signal 105, A R , based on the at least one pair of sample values ​​115 (A, B; A1, B1). Here, the parameter calculator 120 for the system 200 according to Fig. 9 essentially the parameter calculator 120 of Fig. 6. For example, the parameter calculator 120 is designed to calculate the information about the distance to the object 129, Δ φ , R, based on a phase information φ Tof the transmission signal 15, A T , and the phase information 125, φ R , of the received signal 105, A R , to calculate.

[0099] For example, a phase difference Δφ results from a difference between the phase information φ T of the transmission signal 15 and the phase information φ R of the received signal 125.

[0100] Furthermore, the reflectivity information of the object p can be calculated based on the amplitude information â T of the transmission signal 15 and the amplitude information â R of the received signal 105 using equation 34.

[0101] For embodiments according to Fig. 9, the receiver 100 has knowledge of the mean frequency 11, f CW ', of the transmission signal 15, A T . For example, the parameter calculator 120 is designed to calculate the distance to the object R based on the mean frequency f CW' of the transmission signal 15, A T , to calculate. For example, equation 6 can be used.

[0102] Fig. 10a-c show further details of the system 200 with the control unit 210 according to further embodiments of the present invention. Here, the embodiments of Fig. 10a-c the elements already described above have the same reference numerals. In Fig. 10a shows a block diagram of a scanning system. Fig. 10a, the receiver 100 further comprises a detector 1010 for providing a detection signal as input signal 1015. Furthermore, the receiver 100 in Fig. 10a, a filter 1020 for filtering the input signal 1015, wherein the received signal 105 is provided. The filter 1020 (“AA”) is, for example, an anti-aliasing filter designed to apply low-pass filtering to the input signal 1015. Furthermore, the parameter calculator 120 in Fig. 10a two blocks 1022, 1024 which are designed to receive the phase information φ R or the amplitude information â R of the received signal 105. Furthermore, Fig. 10a, the control unit 210 is designated “Clock Gen.” The system 200 according to Fig. 10a further includes a control logic 1002 configured to control various parameters of the system 200. The control logic 1002 and the control unit 210 of Fig. 10a are in turn controlled by a mirror driver 1001, so that parameters of a scanning mirror of the system 200 can be set. The transmitter 10 of Fig. 10a further includes a block 1030 ("DAC") and a block 1040 ("laser"). Block 1030 corresponds to a digital-to-analog converter (DAC), while block 1040 corresponds to a laser for transmitting the transmission signal.

[0103] Referring to Fig. Figure 10a shows a simplified block diagram of a scanning LIDAR system. The phase φ R corresponds, for example, to the result of equation 31, while the amplitude â R for example, corresponds to the result from equation 32. Furthermore, the detector signal 1015 is band-limited, for example, by a suitable anti-aliasing filter (AA).

[0104] In Fig. 10b shows a block diagram of an entire system. In Fig. 10b, the detector 1010 is, for example, an avalanche photodiode (APD). The avalanche photodiode 1010 is configured to provide the input signal 1015. Furthermore, the receiver 100 comprises Fig. 10b a block 1012 (“Analog Back End”). The block 1012 represents an analog interface for receiving the received signal 105. The control unit 210 is in Fig. 10b is labeled “LIDAR Core Clock Gen.” The Fig. The control unit 210 shown in Figure 10b sets the average frequency 11 (“f_cw”), the sampling rate 111, f Abt , and another frequency 1111 (“fs_dac”) as the control frequency for block 1030 (DAC). As shown in Fig. 10b, the control unit 210 can be controlled by a block 1001 (“Mirror IP”) by means of a frequency f_reso 1,2 For example, block 1001 is designed to control the frequency f_reso 1,2for the control unit 210 to achieve a desired resolution of the scanning area of ​​the system. Furthermore, the Fig. 10b, the transmitter 10 comprises a block 1032 (“Analog Front End”) as an interface between the DAC 1030 and the laser 1040. Furthermore, the transmitter 10 comprises Fig. 10b shows a mirror 1050 that can be controlled by the block 1001. The mirror 1050 and the block 1001 are connected to each other via a communication link 1055. For example, the oscillation frequency (f s ) of the mirror 1050 can be adjusted with the block 1001. The mirror 1050 is designed to deflect the transmission signal sent by the laser 1040 onto an object. As shown in Fig. As shown in Figure 10b, the transmitted signal deflected onto the object is reflected in the direction of the avalanche photodiode 1010 after being reflected from the object.

[0105] Fig. 10c shows an embodiment of a control unit 210 of the embodiment of the system 200 according to Fig. 10b. In Fig. 10b shows a block diagram of a “LIDAR Core Clock Generator.” As in Fig. 10c, the control unit 210 comprises the center frequency provider 212 and the sampling rate provider 214. Here, the blocks 212, 214 are designed to generate the signals 11 (f_cw), 111 (f Abt ) and 1111 (fs_dac) depending on the input frequency f_reso 1,2 Furthermore, Fig. 10c shows that the blocks 212, 214 can be set by a block labeled “Register” by means of a control signal (“Resolution”).

[0106] Embodiments of the present invention enable a static LIDAR measurement. In a static measurement (i.e., the intensity-modulated laser radiates constantly in one direction), any reasonable frequency f CW for the intensity modulation. For example, using equation 30, a sampling rate of the ADC 112 (f Abt ) can be calculated, with which the “intensity-modulated” detector signal (or the received signal 105) is divided by the sampling rate f Abt is mixed into the zero position. According to embodiments, a new pair of values ​​or at least one pair of sample values ​​115 (A, B; A1, B1) with a frequency of f Abt / 4. This was done using the example diagrams of Fig. 7a-d (see also “Attachment A” of Fig. 18a-f). It was recognized that the sampling frequency of the measurement can be influenced by the factor m from equation 30. Furthermore, it was recognized that with the frequency f CW The resolution of the measurement result can be affected because at a lower frequency, the modulated wavelength becomes longer, allowing for further measurement distances. At a higher frequency, the wavelength of the modulated signal becomes shorter.

[0107] Embodiments of the present invention provide a concept for a time-shifted sampling, wherein the advantage of the time-shifted sampling in a static LIDAR measurement is realized in the freely selectable repetition time of a measurement (or in the freely selectable time interval T Abt). This repetition time or time interval can be selected from the hour range down to the microsecond range. In comparison, the outputs at the measurement time for a typical laser rangefinder are, for example, < 0.5 seconds and a maximum of 4 seconds. Furthermore, according to embodiments, the intensity frequency (f CW ') can be freely varied. This has the further advantage that the measuring range can be varied. In addition, according to exemplary embodiments, the required tasks can be performed by a very small µ-controller, since no large calculations or complex data handling are required.

[0108] Further embodiments of the present invention provide a scanning LIDAR measurement (1D / 2D). With a scanning system, the challenge lies in the ability to select the right measurement speed, depending on the scanning speed (Hz, kHz, ...) and type of scanner (raster scanner, Lissajous scanner, etc.), as well as the resolution of the area to be scanned (cf. Fig. 8b), to accurately capture the phase of each pixel. In Fig. Figure 8b shows a scan area of ​​a 2D scanner without adjustment (left) and with adaptive scan area adjustment (right). Embodiments create a concept that is advantageous in that it is independent of the type and scanning speed of the scanner. For this purpose, for example, a grid is projected over the area to be scanned. The spacing in the grid corresponds to a pixel clock (or the clock frequency f clk). After each pulse of the pixel clock, the laser from the scanner will, for example, irradiate a different area of ​​the raster. The pixel clock will now be set to f Abt / 4, i.e. a quarter of the sampling rate f Abt of the ADC 112. If, for example, the pixel clock is used to determine the sampling rate f Abt of the ADC 112 and adjust equation 30 for f CW This gives a frequency that matches the sampling rate and automatically allows for downmixing to zero, with each pixel clock pulse receiving a new complex pair of values ​​for a new pixel.

[0109] The prerequisite for a time-shifted scanning according to the invention in a scanning system is that the oscillation frequency of the scanner (f Mirror or f s) is known, the spatial resolution of the scanner is known, the required resolution of the scan area (pixels in x or x / y direction or the parameter "resolution") is given, the intensity modulation is a sinusoidal function, the frequency of the intensity-modulated CW signal is known (f CW ) and the detector signal is band-limited.

[0110] Based on the given requirements, a variable (frequency) output signal can be calculated according to exemplary embodiments. The frequency f CW for example from equation fCW=fpclk∗(4∗m+1) It consists of the frequency of the pixel clock fpclk=2⋅resolution⋅fMirror (or f clk ) with a factor. This factor results from the transformation of equation 30.

[0111] Since the frequency of the intensity modulation is known, a clock can be generated corresponding to this frequency, with which the sampler of the ADC 112 can be operated. This clock can also be adjusted during operation like the modulation frequency and corresponds, for example, to four times the pixel clock fAbt=4∗fpclk

[0112] Increasing the m-factor in Equation 35 increases the modulation frequency of the CW signal (which corresponds to a shorter wavelength), which in turn enables more accurate phase determination. It is also possible to perform multiple measurements of the same pixel by determining the position of, for example, the MEMS mirror. Thus, the accuracy of the phase determination can be further increased by averaging the measured values ​​across the same pixel.

[0113] This measuring method according to the invention has the advantage, for example in scanners, that no data needs to be stored for further signal processing (as is the case when using FFT, digital filters, etc.), but that per pixel clock in the scan (f Abt / 4) there is exactly one complex measurement pair (A + jB). From this measurement pair, the phase shift and amplitude of the sampled signal can be calculated. From these values, the distance and brightness of the pixel can be determined.

[0114] In Fig. 11 shows an example of such a scan, in which the phase position of the detector signal changes linearly from pixel to pixel. The following example scan of Fig. 11 explained in more detail. In Fig. 11 the MATLAB script “Simulation - Time-shifted sampling LIDAR-” was developed by Fig. 18a-f. In the first diagram 1110, a cosine-shaped signal with a frequency of 68 MHz is sampled at 54.4 MHz, for example. The respective samples can be seen in the first diagram 1110 ("Function Values"). In the second and third diagrams 1120, 1130, the sampled signal was decimated by a factor of 4 or first delayed by one clock cycle and then decimated ("Real Value", "Imag Value"). In the fourth diagram 1140, the phase relationship of the sampled signal was calculated ("Phase [deg]"). As in Fig. As can be seen in Figure 11, these samples have a spacing of 1 / fclk. Thus, each pixel can be assigned a phase value; more precisely, an individual distance can be calculated for each pixel.

[0115] The concept just described is advantageous in that it can be applied regardless of the scanner used (1D / 2D scanner, raster scanner / Lissajous scanner, etc.). According to embodiments, it is possible to vary the resolution during operation and to perform multiple measurements of a pixel.

[0116] Further embodiments provide a concept for adaptive scan resolution. According to embodiments, the resolution of the scan area can be adjusted regardless of the scanning method (static or scanning measurement). For example, if the scan area is increased, this means that the grid becomes more tightly meshed. This in turn means that the pixel clock (equation 36) is increased in this area, thereby increasing the frequency of the CW signal (equation 35) and the sampling rate of the ADC 112 (equation 37). For example, if there is an area that is to be scanned with less high resolution, the pixel clock can be reduced in this area, which results in the frequency of the CW signal (f CW ) and the sampling rate (f Abt) of the ADC 112 can also be reduced. This adaptive scan resolution according to the invention can be of interest, for example, in edge detection or edge inspection. Here, for example, an edge / column is detected in a first scan. Furthermore, in a subsequent scan, the resolution in the area of ​​the edge / column can be increased in order to better resolve the edge / column.

[0117] Embodiments of the present invention provide a real-time evaluation algorithm that uses a continuous wave method to determine the distance and reflectivity of the area to be scanned. The accuracy of the measured value can be continuously increased by averaging multiple measurements.

[0118] Applications of the embodiments described above include static measurements (e.g., methods for monitoring mechanical movement) and scanning measurements. Static measurements include, for example, Doppler measurements, while scanning measurements include variable adjustment to a scan frequency, adaptive pixel resolution, possible multiple measurements of a pixel, and 3D photography (inspection applications, biometric applications, etc.).

[0119] In contrast to known methods, embodiments of the present invention create a concept that is significantly more resource-efficient. Furthermore, the algorithm according to the invention does not require any data collection (as is the case with an FFT, for example).

[0120] Furthermore, the present invention is advantageous in that, in contrast to known methods in which FM demodulation is carried out for a CW signal, this demodulation being carried out before the data acquisition, no further resources are required.

[0121] Based on Fig. 16 it was shown that demodulation is achieved, for example, by multiplication. Since in the equivalent circuit of a scanner (cf. Fig. 6d) also contains a multiplier, it is possible to select a specific sampling frequency (f Abt ) to perform quadrature demodulation. It was recognized that the multipliers from Fig. 16 can be replaced by ADC. Furthermore, it was recognized that this eliminates the need for subsequent TP filtering.

[0122] According to embodiments, the computer program (algorithm) according to the invention can be implemented, for example, with the simulation program MATLAB, as described in Fig. 18 is listed.

[0123] Although some aspects have been described in the context of a device, it should be understood that these aspects also represent a description of the corresponding method, so that a block or component of a device can also be understood as a corresponding method step or as a feature of a method step. Analogously, aspects described in the context of or as a method step also represent a description of a corresponding block, detail, or feature of a corresponding device. Some or all of the method steps may be performed by (or using) a hardware apparatus, such as a microprocessor, a programmable computer, or an electronic circuit. In some embodiments, some or more of the key method steps may be performed by such an apparatus.

[0124] A signal according to the invention, such as a signal comprising the sample values ​​115 (A, B; A1, B1), may be stored on a digital storage medium or may be transmitted on a transmission medium such as a wireless transmission medium or a wired transmission medium, e.g. the Internet.

[0125] The signal according to the invention can be stored on a digital storage medium, or can be transmitted on a transmission medium, such as a wireless transmission medium or a wired transmission medium, such as the Internet.

[0126] Depending on specific implementation requirements, embodiments of the invention may be implemented in hardware or software. The implementation may be performed using a digital storage medium, such as a floppy disk, a DVD, a Blu-ray Disc, a CD, a ROM, a PROM, an EPROM, an EEPROM, or a FLASH memory, a hard disk, or other magnetic or optical storage device storing electronically readable control signals that can interact or cooperate with a programmable computer system to perform the respective method. Therefore, the digital storage medium may be computer-readable.

[0127] Some embodiments according to the invention thus comprise a data carrier having electronically readable control signals capable of interacting with a programmable computer system such that one of the methods described herein is carried out.

[0128] In general, embodiments of the present invention may be implemented as a computer program product having a program code, wherein the program code is effective to perform one of the methods when the computer program product is run on a computer.

[0129] The program code can, for example, also be stored on a machine-readable medium.

[0130] Other embodiments include the computer program for performing one of the methods described herein, wherein the computer program is stored on a machine-readable medium. In other words, one embodiment of the method according to the invention is thus a computer program that has program code for performing one of the methods described herein when the computer program is executed on a computer.

[0131] A further embodiment of the method according to the invention is thus a data carrier (or a digital storage medium or a computer-readable medium) on which the computer program for carrying out one of the methods described herein is recorded.

[0132] A further embodiment of the method according to the invention is thus a data stream or a sequence of signals that represents the computer program for carrying out one of the methods described herein. The data stream or the sequence of signals can be configured, for example, to be transferred via a data communication connection, for example, via the Internet.

[0133] A further embodiment comprises a processing device, for example a computer or a programmable logic device, which is configured or adapted to carry out one of the methods described herein.

[0134] A further embodiment comprises a computer on which the computer program for performing one of the methods described herein is installed.

[0135] A further embodiment according to the invention comprises a device or a system designed to transmit a computer program for carrying out at least one of the methods described herein to a receiver.

[0136] The transmission may, for example, be electronic or optical. The recipient may, for example, be a computer, a mobile device, a storage device, or a similar device. The device or system may, for example, include a file server for transmitting the computer program to the recipient.

[0137] In some embodiments, a programmable logic device (e.g., a field-programmable gate array, an FPGA) may be used to perform some or all of the functionalities of the methods described herein. In some embodiments, a field-programmable gate array may interact with a microprocessor to perform any of the methods described herein. In general, in some embodiments, the methods are performed by any hardware device. This may be general-purpose hardware such as a computer processor (CPU) or method-specific hardware such as an ASIC.

[0138] One aspect according to the invention is therefore configuration data for a programmable logic device or for a field-programmable gate array (FPGA), for carrying out the method described herein when the configuration data is loaded into the programmable logic device or the field-programmable gate array.

[0139] The above-described embodiments are merely illustrative of the principles of the present invention. It is understood that modifications and variations of the arrangements and details described herein will be apparent to others skilled in the art. Therefore, it is intended that the invention be limited only by the scope of the following claims and not by the specific details presented in the description and explanation of the embodiments herein.

[0140] Embodiments of the present invention provide a method for determining phase information 125, φ R , a received signal 105, A R . The method comprises providing at least one pair of samples 115 (A, B; A1, B1), sampling the received signal 105, A R , with a sampling rate of 111, f Abt which is less than an average frequency 101, f CW , of the received signal 105, A R , and providing two samples as a pair of samples 115 (A, B; A1, B1) which have a time interval (T Abt ) from 14TCW(4m+1) or from 14TCW(4m+3) where m is an integer greater than or equal to (≥) 1, and where T CW an average period of the received signal 105, A R , is.

[0141] Further embodiments of the present invention provide a method for measuring information about a distance to an object 129 (Δφ, R) with a system 200. The method comprises transmitting a transmission signal 15, A T , with a transmitter 10, receiving a received signal 105, A R , with a receiver as described above, and controlling a medium frequency 11, f CW ', of the transmission signal 15, A T , and the sampling rate 111, f Abt , depending on a clock frequency 205, f clk of the System 200.

Claims

[1] A receiver (100) for determining phase information (125) (φ R ) of a received signal (105) (A R ), with the following features: a sample value provider (110) for providing at least one pair of samples (115) (A, B; A1, B1), wherein the sample provider (110) is designed to receive the received signal (105) (A R ) with a sampling rate (111) (f Abt ) which is less than a mean frequency (101) (f CW ) of the received signal (105) (A R ) is, wherein the sample value provider (110) is designed to provide, as a pair of samples (115) (A, B; A1, B1), two samples which are spaced apart in time (T Abt ) from 14TCW(4m+1) or from 14TCW(4m+3) have, where m is an integer greater than or equal to 1, and where T CWan average period of the received signal (105) (A R ) is, wherein the sample value provider (110) comprises an analog-to-digital converter (112) (ADC) for sampling the received signal (105) (A R ) and a value pair determiner (114) for determining the at least one pair of sample values ​​(115) (A, B; A1, B1) based on a sample signal (113) sampled by the analog-to-digital converter (112) (ADC); wherein the analog-to-digital converter (112) (ADC) is designed to convert the received signal (105) (A R ) with the sampling rate (111) (f Abt ) to scan, wherein the value pair determiner (114) is designed to decimate (312) the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) with a decimation factor of 4 in order to obtain a first sampling value (A) of the at least one pair of sampling values ​​(115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to separate the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by the time interval (T Abt ) and to delay (322) a time interval (T Abt ) to decimate (324) the delayed delay signal (323) with the decimation factor 4 in order to obtain a second sample value (B) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to determine a first pair (A, B) and a second pair (A1, B1) of the at least one pair of sample values ​​(115) (A, B; A1, B1) based on a sample signal (113) sampled by the analog-to-digital converter (112) (ADC); wherein the value pair determiner (114) is designed to decimate (312) the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) with a decimation factor of 4 in order to obtain a first sampling value (A) of the first pair (A, B) of the at least one pair of sampling values ​​(115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to separate the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by the time interval (T Abt ) and to delay (322) a time interval (T Abt ) to decimate (324) the delayed delay signal (323) with the decimation factor 4 in order to obtain a second sample value (B) of the first pair (A, B) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to multiply the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by twice the time interval (T Abt) (332) and by a time interval twice the time interval (T Abt ) to decimate (334) the delayed delay signal (333) with the decimation factor 4 in order to obtain a first sample value (A1) of the second pair (A1, B1) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is designed to multiply the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by three times the time interval (T Abt ) (342) and by a time interval three times the time interval (T Abt ) delayed delay signal (343) with the decimation factor 4 to obtain a second sample value (B1) of the second pair (A1, B1) of the at least one pair of samples (115) (A, B; A1, B1). [2] The receiver (100) according to claim 1, wherein the value pair determiner (114) is designed to multiply (335) the first sample value (A1) of the second pair (A1, B1) of the at least one pair of sample values ​​(115) (A, B; A1, B1) by a factor (-1) to obtain a modified first sample value (A1'); wherein the value pair determiner (114) is designed to multiply (345) the second sample value (B1) of the second pair (A1, B1) of the at least one pair of sample values ​​(115) (A, B; A1, B1) by a factor (-1) in order to obtain a modified second sample value (B1'). [3] The receiver (100) according to claim 2, wherein the value pair determiner (114) is designed to combine (510) the first sample value (A) of the first pair (A, B) of the at least one pair of sample values ​​(115) (A, B; A1, B1) and the first sample value (A1) of the second pair (A1, B1) of the at least one pair of sample values ​​(115) (A, B; A1, B1) to obtain a first combination value (C); wherein the value pair determiner (114) is designed to combine (520) the second sample value (B) of the first pair (A, B) of the at least one pair of sample values ​​(115) (A, B; A1, B1) and the second sample value (B1) of the second pair (A1, B1) of the at least one pair of sample values ​​(115) (A, B; A1, B1) to obtain a second combination value (C1); [4] The receiver (100) according to any one of claims 1 to 3, wherein the receiver (100) comprises a parameter calculator (120) for calculating the phase information (125) (φ R ) of the received signal (105) (A R) based on the at least one pair of samples (115) (A, B; A1, B1). [5] The receiver (100) according to any one of claims 1 to 4, wherein the parameter calculator (120) is designed to calculate the phase information (125) (φ R ) of the received signal (105) (A R ) using the equation φR=tan−1(B / A) to calculate, where A and B are two samples of the pair of samples (115) (A, B; A1, B1) which have the time interval (T Abt ) have. [6] A system (200) for measuring information about a distance to an object (129) (Δφ, R), having the following features: a transmitter (10) for transmitting a transmission signal (15) (A T ), a receiver (100) for receiving a received signal (105) (A R ) according to one of claims 1 to 3; and a control unit (210) for controlling a medium frequency (11) (fCW ') of the transmission signal (15) (A T ) and the sampling rate (111) (f Abt ) depending on a clock frequency (205) (f clk ) of the system (200). [7] The system (200) according to claim 6, wherein the control unit (210) comprises a center frequency provider (212) and a sampling rate provider (214), wherein the center frequency provider (212) is designed to provide the center frequency (11) (f CW ') of the transmission signal (15) (A T ) depending on the clock frequency (205) (f clk ) provide; wherein the sampling rate provider (214) is designed to determine the sampling rate (111) (f Abt ) depending on the clock frequency (205) (f clk ) provide; wherein the control unit (210) is designed to control the clock frequency (205) (f clk ) by a factor of 4 to obtain the sampling rate (111) (f Abt ). [8] The system (200) according to claim 6 or 7, wherein the system (200) comprises a clock frequency provider (201) for supplying the clock frequency (205) (f clk ) for the control unit (210); wherein the clock frequency provider (201) is designed to set the clock frequency (205) (f clk ) for the control unit (210) based on a desired resolution (N px ) of a scanning range of the system (200) and at an oscillation frequency (f s ) of a scanning mirror of the system (200). [9] The system (200) according to any one of claims 6 to 8, wherein the receiver (100) comprises a parameter calculator (120) for calculating the phase information (125) (φ R ) of the received signal (105) (A R ) based on the at least one pair of samples (115) (A, B; A1, B1); wherein the parameter calculator (120) is designed to calculate the information about the distance to the object (129) (Δφ, R) based on phase information (φ T ) of the transmission signal (15) (A T ) and the phase information (125) (φ R ) of the received signal (A R ) to calculate. [10] The system (200) according to claim 9, wherein the receiver (100) has knowledge of the mean frequency (11) (f CW ') of the transmission signal (15) (A T ) has; wherein the parameter calculator (120) is designed to calculate the distance to the object (R) based on the mean frequency (f CW ') of the transmission signal (15) (A T ) to calculate. [11] A method for determining phase information (125) (φ R ) of a received signal (105) (A R ), with the following steps: Providing at least one pair of samples (115) (A, B; A1, B1), Sampling the received signal (105) (A R ) with a sampling rate (111) (f Abt ) which is smaller than a mean frequency (101) (f CW ) of the received signal (105) (A R ) is, Providing two samples as a pair of samples (115) (A, B; A1, B1) which have a time interval (T Abt ) from 14TCW(4m+1) or from 14TCW(4m+3) have, where m is an integer greater than or equal to 1, and where T CW an average period of the received signal (105) (A R ) is, wherein a sample value provider (110) is used which comprises an analog-to-digital converter (112) (ADC) which is used to sample the received signal (105) (A R) is used, and a value pair determiner (114) which is used to determine the at least one pair of sample values ​​(115) (A, B; A1, B1) based on a sample signal (113) sampled by the analog-to-digital converter (112) (ADC); wherein the analog-to-digital converter (112) (ADC) is used to convert the received signal (105) (A R ) with the sampling rate (111) (f Abt ) to scan, wherein the value pair determiner (114) is used to decimate (312) the sampled signal (113) sampled by the analog-to-digital converter (112) (ADC) with a decimation factor of 4 in order to obtain a first sample value (A) of the at least one pair of sample values ​​(115) (A, B; A1, B1); wherein the value pair determiner (114) is used to separate the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by the time interval (T Abt ) and to delay (322) a time interval (TAbt ) to decimate (324) the delayed delay signal (323) with the decimation factor 4 in order to obtain a second sample value (B) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is used to determine a first pair (A, B) and a second pair (A1, B1) of the at least one pair of sample values ​​(115) (A, B; A1, B1) based on a sample signal (113) sampled by the analog-to-digital converter (112) (ADC); wherein the value pair determiner (114) is used to decimate (312) the sampled signal (113) sampled by the analog-to-digital converter (112) (ADC) with a decimation factor of 4 in order to obtain a first sample value (A) of the first pair (A, B) of the at least one pair of sample values ​​(115) (A, B; A1, B1); wherein the value pair determiner (114) is used to separate the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by the time interval (T Abt ) and to delay (322) a time interval (T Abt ) to decimate (324) the delayed delay signal (323) with the decimation factor 4 in order to obtain a second sample value (B) of the first pair (A, B) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is used to multiply the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by twice the time interval (T Abt ) (332) and by a time interval twice the time interval (T Abt ) to decimate (334) the delayed delay signal (333) with the decimation factor 4 in order to obtain a first sample value (A1) of the second pair (A1, B1) of the at least one pair of samples (115) (A, B; A1, B1); wherein the value pair determiner (114) is used to multiply the sampling signal (113) sampled by the analog-to-digital converter (112) (ADC) by three times the time interval (T Abt ) (342) and by a time interval three times the time interval (T Abt ) delayed delay signal (343) with the decimation factor 4 to obtain a second sample value (B1) of the second pair (A1, B1) of the at least one pair of samples (115) (A, B; A1, B1). [12] A method for measuring information about a distance to an object (129) (Δφ, R) with a system (200), the method comprising the following steps: Sending a transmission signal (15) (A T ) with a transmitter (10); Receiving a received signal (105) (A R ) with a receiver (100) according to one of claims 1 to 3; and Controlling a medium frequency (11) (f CW') of the transmission signal (15) (A T ) and the sampling rate (111) (f Abt ) depending on a clock frequency (205) (f clk ) of the system (200). [13] Computer program with a program code for carrying out the method according to claim 11 or 12 when the computer program runs on a computer. [14] Configuration data for a programmable logic device or for a field-programmable gate array (FPGA), for carrying out the method according to claim 11 or 12 when the configuration data are loaded into the programmable logic device or the field-programmable gate array.

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