DEVICE AND METHOD FOR CONTROLLING A SYNCHRONOUS MACHINE AND FOR ESTIMATING THE ROTOR POSITION OF START-UP AT A PREDECIDED LOW SPEED

Description

DOMAINE TECHNIQUE DE L'INVENTION

[0001] The technical field of the invention is that of the control of synchronous rotating electrical machines without an angular position sensor by an estimation of an angular position of a rotor relative to a stator.

[0002] The present invention relates to the method of controlling the synchronous motor without an angular position sensor from start-up to low speed. ARRIERE-PLAN TECHNOLOGIQUE DE L'INVENTION

[0003] An electrical machine is equipped with an inductor stator comprising windings and a rotor comprising an armature body and a rotating shaft.

[0004] An example of relevant prior art is described in document CN 110 429 886 B.

[0005] To control a synchronous electric machine, it is necessary to know the position and speed of the rotor. A rotor angle sensor (encoder or resolver) is used to detect the exact position of the rotor and its rotational speed, and send this rotor position signal to a control unit. In a self-controlled electric machine, this signal is used to regulate the currents in the stator windings. This allows the converter to adjust the supply frequency according to the rotor position, maintaining an angle θ of 90° between the rotating stator field Hs and the rotor field Hr, so that the motor torque is always at its maximum.

[0006] However, in certain high mass density applications, particularly in aeronautics, integrating position sensors onto the rotor shaft of the electric machine is becoming increasingly difficult, especially when the environment is bulky or maintenance is challenging. For example, such position sensors must withstand high thermal stress; in particular, there is a need to use synchronous motors on aircraft that can experience temperatures ranging from 45°C, for example on a tarmac, to a negative temperature of -70°C when the aircraft is at high altitude. Furthermore, these position sensors are expensive and can be subject to other stresses, such as dust, humidity, etc. All these stresses lead to a reduction in MTBF (Mean Time Between Failures), which is the average time between failures of a repairable system.

[0007] Furthermore, when the torque is very high, the rotor shaft can slip relative to the rotor body of the position sensor. An absolute position error arises between the reference axis of the position sensor and the magnetic axis of the electric machine. Consequently, the initial angular setting of the angular sensor is altered. In addition, the moving part of the angular sensor can also slip on the rotor shaft, resulting in a shift in the angular position. This shift leads to degradation that can cause a malfunction in the control of the electric machine. Indeed, in a synchronous machine, for example, a three-phase permanent magnet machine, the torque produced depends on the interaction between the rotor flux and the stator flux in a plane known as the rotor plane, which is specific to the electric machine.Since the rotor flux is produced by the permanent magnets, the torque is regulated by adjusting the stator flux, for which two parameters are accessible: the flux amplitude, itself determined by the amplitude of the currents in the three-phase power supply system, and the phase of the stator flux relative to the rotor flux. This phase is itself determined by the phase of the stator currents. For a smooth-pole synchronous machine, at a given current amplitude, the maximum torque is obtained when the rotor flux is, to a very small extent, in quadrature phase with respect to the stator flux.

[0008] There are therefore control methods without position sensors which increase reliability and reduce the cost of the control device.

[0009] One solution exists for estimating the angular position of the motor using an approach that estimates the back electromotive force (EMF) between stator phases. However, the magnitude of the back EMF is proportional to the rotational speed of the electric machine. Therefore, the detectability of the electrical position naturally diminishes as the magnitude of the back EMF decreases with increasing rotational speed. This solution is thus inapplicable at low rotational speeds and at standstill.

[0010] A control device is known that uses an open-loop starting sequence of the I / f (current / frequency) or V / f (voltage / frequency) type for starting, without using measured rotor position information. In fact, the stator field's electrical frequency is imposed by the user and gradually increased until it is high enough to estimate the electromotive forces. The slope of the electrical frequency variation must be slow to ensure that the actual rotor flux of the electric machine matches the virtual rotor flux created by this open-loop approach. Only then can the control loop, using the sensorless control algorithm, be used in closed loop.This method requires a perfect knowledge of the inertia and the load driven by the electric machine as well as specific management, on the one hand, of the acceleration of the speed to be introduced as a function of the torque and, on the other hand, of the transition between the open loop to the closed loop under penalty of stalling of the electric machine.

[0011] Furthermore, a phenomenon that adds limitations to sensorless operation in the low-speed range stems from the power converter's semiconductors, which operate non-linearly due to the dead time applied to the rising edges of the PWM commands, the durations of the PWM commands' ON and OFF states, and the voltage drop across the semiconductors. All of these phenomena contribute to the creation of an error known as the voltage error, present at the microscopic level, whose impact is averaged at the macroscopic level (at the level of a fundamental electrical cycle) and is dominant in the low-speed range. Indeed, the fundamental error-to-amplitude ratio is high at low speeds, which is not the case at high speeds. The frequency value of the first-order voltage or current (h1) is called the "fundamental."

[0012] This voltage error induces a distortion of the fundamental frequency of the control voltage applied to the input of the electric machine (by introducing low-frequency harmonics). This is reflected identically in the stator currents, due to the low impedance of the electric machine which, at low rotational speeds, is unable to filter these low-frequency harmonics. Thus, as mentioned previously, the useful signal-to-noise ratio degrades, which negatively impacts the rotor position estimation process at low rotational speeds (since the latter is designed under the assumption of the first fundamental frequency).

[0013] Thus, one of the problems is to start a synchronous type electric motor in a closed loop and without a position sensor, from starting with a torque up to a specific rotation speed. RESUME DE L'INVENTION

[0014] The invention offers a solution to the problems mentioned above by enabling sensorless, closed-loop starting, thus preventing instability at low rotational speeds and under maximum load torque, and allowing operation in a wide flux deflux zone of the electric machine. The invention proposes a new solution based on a position sensor-free control algorithm, which ensures reliable and robust closed-loop starting from the moment the electric machine stops until it reaches maximum rotational speed.

[0015] One aspect of the invention relates to a control device for an inverter converter for starting a multi-phase synchronous electric motor up to a predetermined threshold speed, of an electrical machine, the control device comprising: a control loop, comprising: a current regulator for delivering a voltage setpoint including a quadratic and direct voltage with an angle, from a current setpoint for regulation; a calculation unit for calculating a direct and indirect Park transformation; the calculation unit comprising a current return output in a Park frame, from measured values ​​of the received phase currents, transformed in a Park frame into quadratic return current and direct return current, taking into account an estimated angular position value of the received rotor; characterized in that the control device further comprises: a closed-loop adaptive angle estimator, for estimating the estimated angular position value of the rotor, based on a difference between at least one data point of a reference stator flux vector calculated as a function of the return currents;and a datum of an adaptive stator flux vector, calculated from the voltage setpoint, the return currents, and an estimated electrical velocity calculated as a function of this difference, a setpoint current modifier, in which, when an absolute value of the estimated electrical velocity received by the adaptive angle estimator is less than the predetermined threshold velocity, the modifier calculates a direct setpoint current for the regulator having a non-zero value, thus modifying a direct setpoint current by a received setpoint current.

[0016] Thanks to the invention, the control device allows for an estimation of angular position for a start-up up to at least a predetermined rotor rotation speed and then optionally beyond this predetermined rotation speed to switch to another calculation of the angular position, for example the calculation of the angular position of the rotor by means of the electromotive force EMF.

[0017] Indeed, since the torque is normally equal to a gain multiplied by the quadratic current: T em = K t i q Without angular position error, by injecting a direct current at startup, if there is no position error, the rotor torque is equal to the estimated torque, but if there is a position error, the rotor torque is different from the estimated torque, since the torque depends on the sine of the estimated angular position error: T em ≈ T ^ em + K t i ^ s sin Δ θ .

[0018] Therefore, as long as the sine of the angular position difference Δθ is not equal to 0, the actual torque Tem is not equal to the estimated torque T̂ em because it is dependent on the direct current multiplied by the sine of the angular position difference Δθ.

[0019] Thus, by injecting a direct current I d into the closed loop, the adaptive angle estimator will, through calculations, adapt the angular angle so that the latter corresponds until the sine of the difference in angular position Δ θ is equal to 0.

[0020] In prior art, angle estimators calculate angular position from the electromotive force (EMF). However, since the magnitude of the EMF is directly related to the rotational speed of the electric machine, the detectability of the electrical position diminishes as the magnitude of the EMF decreases with the rotational speed of the electric machine. At zero rotational speed (at rest), the EMF is zero (undetectable), and the EMF of the electric machine is therefore unobservable. In other words, methods based on the EMF are not useful from a standstill until the speed allows a ratio between the fundamental EMF and the measurement noise to be sufficiently high to enable the identification of angular position.

[0021] The rotor flux remains constant and is non-zero, influencing the stator flux according to its rotation. Thus, the invention allows for an estimation of the rotor position based on the stator flux expressed in the rotor frame of reference. The invention enables an approach to estimating the rotor position of the permanent magnet synchronous machine, based on a non-linear stator flux observer that adapts to changes in rotational speed. At low speeds and when stationary, the estimation is enhanced by injecting a non-zero current onto the direct axis in the Park frame of reference.

[0022] The invention thus has a logic for managing the starting of the electric machine, which is achieved by injecting a non-zero value of the direct component of the current vector into the rotor space of the electric machine. This management logic thus makes it possible to add a current setpoint on the direct current axis, denoted i d as different from zero (non-zero), allowing the angle estimator to calculate the angular position from the stator flux and not from the EMF as in the prior art and to obtain the angular position by eliminating an angular position error between the actual rotor plane of the electric machine and the virtual rotor plane of the control.

[0023] Another component of the control algorithm proposed in this invention consists of an adaptive angle estimator, which cancels the angular error between a real machine reference frame and a virtual control reference frame. This angular error, denoted ε, is simply the difference between a reference stator flux and an estimated stator flux. The angle estimator calculates at least one value for the reference stator flux vector from the measured stator currents. The estimated stator flux is calculated adaptively by the angle estimator, which calculates at least one value for the stator flux vector based on the control voltages. v dq # , measured stator currents and as a function of an estimated rotational speed feedback action. The estimated rotational speed feedback action is deduced by an adaptive mechanism for eliminating rotor position error.

[0024] In other words, a correct electrical rotation speed is only obtained if the estimated stator flux vector tends towards the reference stator flux vector.

[0025] A simple integration of the rotational speed information then allows us to deduce the rotor position, denoted θ̂ elec .

[0026] Furthermore, such a process works in both directions of rotation of the electric machine, whether in the positive trigonometric direction or negative (clockwise) since one can inject a direct current I d with a positive or negative sign depending on the direction of rotation of the rotor.

[0027] In addition to the characteristics mentioned in the preceding paragraph, the method according to one aspect of the invention may have one or more complementary characteristics among those described in the following paragraphs, considered individually or according to all technically possible combinations: According to one embodiment, at least one data point of the reference stator flux vector is the quadratic value and at least one data point of the quadratic stator flux vector is the quadratic value.

[0028] According to another embodiment, the closed-loop adaptive angle estimator is configured to estimate the estimated angular position value of the rotor based on a difference between the reference stator flux vector (direct and / or quadratic and / or angle data) calculated as a function of the return currents, and the adaptive stator flux vector (direct and / or quadratic and / or angle data) calculated from the voltage setpoint, the return currents, and an estimated electrical speed calculated as a function of this difference.

[0029] According to one embodiment, the adaptive angle estimator comprises: a first stator flux calculator comprising: a first calculation block for the reference quadratic stator flux, a second calculation block for the adaptive quadratic stator flux, a second calculator for an estimated electrical speed based on a comparison of the adaptive quadratic stator flux and the reference quadratic stator flux, a third calculator for estimating an estimated angular position value of the rotor based on the calculated estimated electrical speed.

[0030] According to one example of this embodiment, the first block for calculating the reference stator flux is modeled in a rotor reference frame.

[0031] According to an example of this embodiment, the second adaptive calculation block of the stator flux model is modeled in the rotor reference frame.

[0032] According to an example of this embodiment, the first calculation block calculates the quadratic reference flux according to the formula: LqIq where Lq is the quadratic value of the stator inductance on the q axis, Iq is the quadratic return current.

[0033] According to an example of this embodiment, the second calculation block calculates the adaptive quadratic stator flux according to the integral of the following formula: v q − R s i q − ω ^ elec λ ^ d v + k i ˜ q in which Rs is the stator resistance, λ̂ dv is the adaptive direct stator flux equal to the integral of the following formula: v d − R s i d + ω ^ elec λ ^ q v + k i ˜ d and K is a positive gain matrix, vq is the squared voltage of the setpoint voltage at the output of the current regulator, I q (sometimes denoted Iq) is the squared return current, ω̂ elec is the estimated electrical speed, ĩ d And ĩ q are the direct and quadratic components of the error in currents with i ˜ d = i d − i ^ d i ˜ q = i q − i ^ q and in which the estimated direct current î d and the estimated quadratic current î q are calculated based on: i ^ d = λ ^ d v − φ PM L d i ^ q = λ ^ q v L q

[0034] According to one embodiment, the direct setpoint current modifier includes a setpoint current control input.

[0035] According to one embodiment, the setpoint forward current modifier calculates a setpoint forward current for the regulator equal to the square root of the sum of a value of the maximum squared current with the value of the setpoint squared current, received in the setpoint current: I d ′ # = Iqmax 2 − I <none / > # q <none / > <none / > 2 .

[0036] According to an example of this embodiment, the calculation of the modified setpoint direct current is imposed with the same sign as the estimated electrical velocity received.

[0037] According to one embodiment, the setpoint forward current modifier includes a comparator for comparing the absolute value of the estimated electrical speed to the predetermined threshold speed.

[0038] According to one embodiment, the control device is capable of controlling the converter for a rotational speed beyond the predetermined threshold speed, in which if the absolute value of the estimated electrical speed is greater than the predetermined threshold speed, the direct current setpoint modifier transmits the setpoint current for regulation according to only the setpoint current.

[0039] According to an example of this embodiment, when the absolute value of the estimated electrical speed is greater than the predetermined threshold speed, the setpoint forward current modifier transmits the setpoint current with a setpoint forward current equal to zero, unless a deflux setpoint is sent.

[0040] According to an example of this embodiment, if the absolute value of the estimated electric velocity is greater than the predetermined threshold velocity, the angular position estimator estimates the position and angular velocity from a measured electromotive force Fem.

[0041] According to one embodiment, the computing unit includes a PWM control output capable of connecting to the converter to control its electronic switches and in that the computing unit is capable of transforming the inverse Park transform signal of the received voltage setpoint into a pulse width modulation signal.

[0042] According to one embodiment, the second computer includes a comparator comparing the reference stator quadratic flux to the adaptive quadratic flux and in that the estimated calculated electrical velocity is transmitted to the first computer of the stator flux, forming a closed loop.

[0043] According to one embodiment, the second computer includes a phase-locked loop (PLL) which, from the comparison value between the reference quadratic stator flux and the adaptive quadratic stator flux, calculates an electrical speed value.

[0044] In one embodiment, the third computer calculates an estimated rotor angle by integralizing the estimated electrical speed. For example, the estimated angular position of the rotor is an electrical angular position.

[0045] According to one embodiment, the control device further includes a current measurement input per phase of the electric machine, intended to be connected to a current sensor measuring the current on the corresponding supply phase line.

[0046] According to one embodiment, the calculation unit includes a voltage setpoint input received from the current regulator and is adapted to be connected to the inverter converter to transmit to it a pulse width modulation control which allows the electronic power switches of the converter to be driven at a specific switching frequency and thus drive the fundamental of the stator voltage seen at the input of the electric machine for the control of the electric machine.

[0047] According to one embodiment, the adaptive angle estimator transmits the estimated electrical speed to the current regulator, in particular in a block of PI current regulators.

[0048] According to one embodiment, the control device includes a processor and a memory and in that the control loop, the adaptive angle estimator and the setpoint modifier are integrated into a program implemented by the processor from the memory.

[0049] Another aspect of the invention relates to a machine comprising: a synchronous electric motor comprising a rotor, a stator, a current measuring sensor, an inverter converter comprising power switches and the control device described above with or without the different embodiments described above, in which the computing unit transmits a command to the converter from the voltage setpoint to drive the electronic power switches at a specific switching frequency and thus drive the fundamental of the stator voltage seen at the input of the electric machine for the control of the electric motor.

[0050] Another aspect of the invention relates to a closed-loop control method without a position sensor for a synchronous machine, from start-up to maximum rotation speed, comprising the following steps: modification of a received current setpoint by imposing a non-zero modified direct current setpoint as long as an estimated speed is less than a predetermined threshold speed value; calculation of a voltage setpoint including a quadratic and direct voltage with an angle, from a current setpoint for regulation including a modified direct current setpoint; calculation of a control to drive electronic power switches of the inverter converter at a specific switching frequency and thus drive the fundamental of the stator voltage seen at the input of the electrical machine, by an inverse Park transformation of the voltage setpoint and an estimated angular position value of the rotor; measurement of the phase currents and transformation them in a Park frame, into quadratic return current and direct return current, taking into account the estimated angular position value of the rotor.Calculation of a reference quadratic stator flux as a function of the quadratic return current, the forward return current, the return current output, and a calculated electrical frequency; calculation of an adaptive quadratic stator flux from the voltage setpoint, the quadratic return current, the forward return current, and the estimated calculated electrical speed; calculation of a rotor electrical speed from a comparison of the adaptive quadratic stator flux and the reference quadratic stator flux; calculation of the estimated angular position value of the rotor as a function of the calculated estimated electrical speed.

[0051] The invention and its various applications will be better understood by reading the following description and examining the accompanying figures. BREVE DESCRIPTION DES FIGURES

[0052] The figures are presented for illustrative purposes only and are in no way limiting to the invention. [ Fig. 1 [ ] shows a representation of a schematic diagram with functional blocks of an electrical machine comprising a control device according to an embodiment of the invention. Fig. 2 [ ] shows a schematic representation of an adaptive angle estimator of the control device according to an example of an embodiment of the invention. ] Fig. 3 [ ] shows a schematic representation of a modifier of a direct current setpoint of a control device according to an example of a first embodiment of the invention. ] Fig. 4a [ ] shows a schematic representation of different graphs representing various measurements of a motor with decreasing rotational speed. Fig. 4b ] shows a schematic representation of an enlargement of one of the graphs of the figure 4a representing the actual and estimated position measurement of an engine. DESCRIPTION DETAILLEE

[0053] The figures are presented for illustrative purposes only and are in no way limiting to the invention.

[0054] There figure 1 represents a schematic diagram with functional blocks of an electric machine M comprising a synchronous electric motor 1, an inverter converter C supplying phases of the synchronous electric motor 1, a battery B and a control device according to an embodiment of the invention controlling the inverter converter C.

[0055] The electric machine M can be a turbomachine, in which the electric motor 1 comprises a stator with X phases and a rotor surrounded by the stator. The electric motor 1 can operate in motor mode and / or generator mode. The electric machine M is, for example, a multi-phase synchronous electric machine for an aircraft propulsion system. The rotor comprises a shaft and an active part mounted on the shaft. The active part of the rotor comprises magnets forming P pole pairs. The number X in this example is three, but could be greater, for example, five or six. The stator, therefore, in this example, comprises three windings forming three phase outputs U, V, W of three phases, which are represented in the diagram. figure 1 , such as star-coupled but could, for example, be triangle-coupled.

[0056] In this example, the inverter C is a DC input inverter, meaning it's a DC / AC converter or a reversible inverter (AC / DC and DC / AC) if the electrical machine M also allows for generator mode. The inverter C has N outputs, specifically three outputs, each connected to one of the corresponding phase outputs U, V, and W. The inverter C also includes power inputs, specifically two DC voltage inputs connected to the terminals of a DC voltage bus B.The inverter converter C includes electronic power switches and a control setpoint input, in this case a pulse width modulation control which allows the electronic power switches to be driven at a specific switching frequency and thus control the fundamental of the stator voltage seen at the input of the electric machine for the control of the electric motor 1 of the electric machine.

[0057] The electrical machine M further includes a measuring means 3 for measuring phase currents Iu, Iv, Iw flowing on the phase outputs U, V, W. The measuring means 3 includes, for example, a current sensor per phase for measuring the current in the corresponding phase.

[0058] The control device 2 includes a control loop R, schematically represented by an example, comprising a current regulator 4 with a current setpoint input of a current setpoint for regulation I#< dq' comprising a modified direct current setpoint I#< d', explained below. The current setpoint for regulation I#< dq' further includes a quadratic current and an angle α.

[0059] Current regulator 4 includes a voltage setpoint output V#< dq comprising a direct setpoint voltage and a quadratic setpoint voltage.

[0060] The control loop R further includes a calculation unit 5 using a known mathematical method, namely the so-called "Park transform", directly to go from a three-phase reference frame U; V; W linked to the stator to a two-phase rotating reference frame d; q, knowing the angular position θ of the rotor of the electric motor 1 relative to its stator, as well as the inverse Park transform to go from the Park d;q frame to the three-phase U;V;W frame, also using an angular position of the rotor. The angular position θ The position of the rotor is the position of the axis d in the dq frame, relative to the magnetic reference axis (which is the horizontal axis in (abc) and is fixed). The dq frame is rotational. Thus, between the two frames, there is an angle that varies from 0 to 360°. This is the position of the rotor in space.

[0061] The calculation unit 5 includes a voltage setpoint input connected to the output of the regulator 4 to transform the voltage setpoint Vdq, with an inverse transformation and control calculator, and a control output connected to the control input of the inverter converter C, to transmit a PWM command calculated by the forward transformation calculator, in this case using pulse-width modulation. The inverse Park transformation and control calculator is therefore configured to transform the forward and quadratic setpoint voltage vectors into a Park plane from an estimated angular position value of the rotor (θ̂). elec ) explained below, in a pulse-width modulation (PWM) control. The pulse-width modulation (PWM) control includes electrical voltage signals to control each phase using pulse-width modulation, thus generating a balanced three-phase AC voltage system.

[0062] The calculation unit 5 further includes measured current inputs receiving the measured phase currents iu, iv, iw, the measured current inputs are connected to the measuring means 3, and a quadratic current return output Iq (sometimes noted I q) and direct current return output Id (sometimes noted I d) connected to the return input of the current regulator 4.

[0063] The calculation unit 5 further includes an estimated angular position input receiving the estimated angular position value of the rotor θ̂ elec explained below, and a Park transformer calculator configured to transform the measured currents lu, Iv, Iw according to the estimated angular position value of the rotor θ̂ elec received as a component on the quadratic axis of the current vector called in the following quadratic current Iq, and the component on the direct axis of the direct current vector Id in a Park plane, also called in the following direct current Id of return.

[0064] In the request, each reference in the sequence with "^" is an estimated value.

[0065] The direct return current Id is therefore in the following the direct component of the stator current in the Park plane calculated from the measured stator currents.

[0066] The quadratic return current Iq is therefore in the following the quadratic component of the stator current in the Park plane calculated from the measured stator currents.

[0067] The current regulator 4 includes summing junctions 41 (represented by a single summing junction on the figure 1 ), receiving at non-inverting input the setpoint quadratic current vector I #< q' and the modified setpoint direct current vector I #< d ' and at inverting input the return quadratic current vector Iq as well as the return direct current vector Id.

[0068] At the output of the summing junctions 41, the regulator 4 has one line carrying the difference between the setpoint quadratic current I<q' and the return quadratic current Iq, and another line carrying the setpoint forward current I<d' and the return forward current Id. Both lines are represented by a single line. The current regulator 4 includes a block of PI (proportional-integral) current regulators 42 connected to the difference lines and delivering the voltage setpoint V<dq, which includes quadratic and forward voltages.

[0069] The control device 2 further includes, in addition to the control loop R, a closed-loop adaptive angle estimator 6, to estimate an estimated angular position value of the rotor θ̂ elec, In particular, from the start-up of the electric machine (rotor speed = 0) to the predetermined threshold speed. The control loop R may include an angle estimator based on the electromotive force (not shown), which is then used beyond the predetermined speed.

[0070] In this example of this embodiment, the speed is the electrical speed (volts / s) but could be the rotor's rotational speed. The predetermined threshold speed is, for example, the speed required for the angle estimator to operate according to the electromotive force. For example, the predetermined threshold speed is an electrical speed corresponding to a rotor rotational speed of 500 rpm. An electrical angle = the mechanical angle of the rotor multiplied by the number of pole pairs.

[0071] The adaptive angle estimator 6, shown in detail on the figure 2 , in closed loop includes a first stator flux calculator 61. The first stator flux calculator 61 includes two calculation blocks to calculate a reference quadratic stator flux λq calculated from measured value and predetermined constant and an adaptive quadratic stator flux λqv different from the reference quadratic stator flux λq in that it is further dependent on the calculated setpoint voltage V#dq.

[0072] The first stator flux calculator 61 therefore includes a first calculation block, 611, of the reference squared stator flux λq as a function of the squared return current Iq. Specifically, in this example of this embodiment, the reference squared stator flux λq is calculated according to the formula: LqIq, where Lq is the stator inductance on the q-axis. In this example, the stator inductance Lq is a predetermined value; here, it is assumed that the variation of the inductances as a function of the current is negligible, that is, that the stator does not exhibit magnetic saturation (inductance that drops sharply when the stator current increases in amplitude). The inductances are considered as apparent inductances (therefore almost constant for a given current point), or the relationship between the flux and the current remains linear via this inductance: L = flux / current → an increasing straight line.

[0073] The first stator flux calculator 61 therefore also includes a second calculation block 612 of the adaptive quadratic stator flux λqv, which can also be called adaptive flux observer, from the voltage setpoint V# dq, the quadratic return current I q, the forward return current I d, and the estimated calculated electric speed ω̂ elec.

[0074] In particular, in this example of this embodiment, the adaptive quadratic stator flux λqv is calculated by the integral of the following formula: v q − R s i q − ω ^ elec λ ^ d v + k i ˜ q where Rs is the stator resistance, λ̂ dv is the adaptive direct stator flux equal to the integral of the following formula: v d − R s i d + ω ^ etec λ ^ q v + k l ˜ d and in that K is a positive gain matrix.

[0075] ĩ d And ĩ q are the direct and quadratic components of the error in currents with i ˜ d = i d − i ^ d i ˜ q = i q − i ^ q

[0076] and in which the estimated direct current î d and the estimated quadratic current î q are calculated based on: ι ^ d = λ ^ d v − φ PM L d ι ^ q = λ ^ q v L q

[0077] The first stator flux calculator 61 has the role of enabling the obtaining of values ​​of the reference quadratic flux λ q which has as a variable return currents which is a function of measured current and an estimated angular position, and the adaptive quadratic stator flux λqv which also has as a variable the voltage setpoint V #< dq and the calculated electrical speed in order to differentiate them in order to identify a positioning estimation error.

[0078] The adaptive angle estimator 6 therefore includes a second calculator 62 of an estimated electrical speed ω̂ The electric current is calculated based on a comparison of the adaptive quadratic stator flux λq and the reference quadratic stator flux λq. The second computer 62 therefore includes a comparator 620 which, in this example, has as its negative input the reference quadratic stator flux λq and as its positive input the adaptive quadratic stator flux λq, and as its output a comparison value ε representing the difference between the two calculated fluxes. The second computer 62 further includes a phase-locked loop (PLL) 621 which, based on the comparison value between the reference quadratic stator flux and the adaptive quadratic stator flux, calculates an estimated electric speed value. ω̂ elec.

[0079] The adaptive angle estimator 6 also includes a third calculator which, based on the estimated electrical speed values ω̂ elec , the calculation of the electrical position of an estimated angular position value of the rotor θ̂ elec. The estimated angular position value of the rotor θ̂ elec being transmitted to the calculation unit 5. The estimated angular position value of the rotor θ̂ elec is estimated at each instant t as a function of the integral of the estimated electrical velocity values ω̂ elec.

[0080] The control device 2 further includes a modifier 7 for a setpoint direct current I<dq received by an input. The modifier 7 thus allows, from start-up to the predetermined threshold speed, the modification of the setpoint direct current I<dq into a non-zero setpoint direct current for the regulator I<d' transmitted to the current regulator. In particular, the modifier 7 includes a comparator 70 for the absolute value of the estimated electrical speed ω̂ Electricity received by the adaptive angle estimator 6 with the predetermined threshold speed. If the estimated electric speed ω̂ elec is less than the predetermined threshold speed, modifier 7 includes a calculation block 71 that calculates the setpoint forward current for the regulator I#< d' having a non-zero value and transmits it to the current regulator 4. In this example, calculation block 71 calculates the setpoint forward current for the regulator I#< d' equal to the sign of the estimated electrical speed ω̂ elec (to determine the direction of rotation of the rotor) multiplied by one the maximum of two terms which are as follows:

[0081] The first term, noted μi q # , constitutes a proportion of the quadratic component of the current setpoint.

[0082] The second term is the square root of the difference between the square of the maximum amplitude of the stator currents that the voltage inverter can withstand (denoted Ipeak) and the square of the quadratic component of the current setpoint: I peak 2 − i q # 2

[0083] According to another example, the setpoint forward current for regulator I #< d' is a predetermined forward current.

[0084] The setpoint current I #< dq can come from a thruster control unit or from several thrusters transmitting the setpoint in direct setpoint current I #< d and in quadratic setpoint current I #< q with an angle a to the modifier 7.

[0085] In this example, modifier 7 also transmits the quadratic setpoint current for regulator I #< q' which is equal to the quadratic setpoint current I #< q, the latter can be transmitted directly to the current regulator 4.

[0086] In this embodiment, the control device is capable of controlling the converter for a rotational speed beyond the predetermined threshold speed. The setpoint current modifier 7 transmits the setpoint current for regulation I#< dq' according to only the setpoint current I#< dq if the absolute value of the estimated electrical speed ω̂ The electricity is higher than the predetermined threshold speed.

[0087] In this embodiment, the setpoint current I#< dq includes a direct setpoint current I#< d equal to zero. According to another example, in the case where the direct setpoint current I#< d is not zero, for a rotational speed beyond the predetermined threshold speed, the setpoint current modifier 7 sets the direct setpoint current for the controller I#< d to zero except in the case of flux deflux control.

[0088] There figure 4a represents different time graphs showing the performance of the control device according to the invention.

[0089] Column 1, row 2 represents the speed as a function of time in which the speed decreases from 0.15 (t) from 2000 rpm to 200 rpm stabilized from 0.2 to 0.35 (t), (t) being a unit of time.

[0090] Column 1, row 1 shows a curve of the setpoint torque Temc as a function of time and a curve of the measured torque Temmes, where the setpoint torque is 2500 N·m between 0(t) and 0.3(t), and a curve of the measured torque Temc equal to K Iq which begins to deviate from the actual torque between 0.2(t) and 0.3(t) and then again between 0.3(t). This deviation is related to an angular position error in the control system.

[0091] Column 2, row 1 represents the setpoint forward current for regulator I #< d' in which between 0.25(t) and 0.3 (t) the setpoint forward current for regulator I #< d' = 50Amp is modified, then from 0.3 (t) to 0.35 (t) = 200Amp.

[0092] Column 2, row 2 represents the quadratic setpoint current for regulator I #< q' which is regular around 550 Amperes.

[0093] Column 2, row 2 represents the angular position of the actual rotor θ relative to the angular position of the calculated rotor θ (a function of the electrical angular position). θelec and the number of poles in the rotor). The figure 4b This represents an enlargement of this graph in the Z region, showing the angular position of the actual rotor θ relative to the angular position of the calculated rotor θ̂ between 0.25(t) and 0.35(t). This figure shows a significant difference in angular position Δθ around 0.25, which decreases slightly around 0.3, and then decreases sharply from 0.3 to 0.35.

[0094] Thus, the fact that the forward current modifier 7 changes the setpoint forward current Id' from 0 to 50 Amperes only slightly reduces the angular position error (angular position difference Δθ), but by significantly changing the setpoint forward current Id' from 0 to 200 Amperes, the position error is significantly reduced. As explained previously, the actual torque Tem is equal to the sum of the estimated torque T̂ em = KI q and of K t î d sin(Δ θ ), but as sin(Δ θ ) tends towards 0 when there is no positional error, the product K t î d sin(Δ θ ), tends towards zero and thus T em = K t i q without angular position error as can be seen on line 1 column 1 from 0.3(t) to 0.35(t).

[0095] Thus, the control device of the invention makes it possible to estimate the angular position of a rotor of a synchronous motor coupled to a load with a potentially significant torque, from standstill (start-up) up to a predetermined threshold speed, for example, 1000 rpm. Beyond the predetermined threshold speed, the control device can calculate the angular position of the rotor and the rotational speed according to the flux using the adaptive angle estimator 6, but with a direct current setpoint for the controller equal to zero, or using another angular position estimator that calculates the position according to the electromotive force as in the previous example.

[0096] Unless otherwise specified, the same element appearing on different figures has a unique reference.

Claims

1. A device (2) for controlling an inverter converter (C) for starting a multiphase synchronous electric motor (1) of an electric machine (M) up to a predetermined threshold speed, the control device comprising: - a regulation loop (R), comprising: o a current regulator (4) for delivering a voltage setpoint (V#dq) comprising a quadratic and direct voltage with an angle (a), from a regulation current setpoint (I#dq'), o a calculation unit (5) for calculating a direct and indirect Park transformation, the calculation unit (5) comprising a current return output (Idq) in a Park reference frame, from measurement values of the phase currents (lu, Iv, 1w) received, transformed in a Park reference frame, into a return quadratic current (Iq) and into a return direct current (Id), taking account of a value of estimated rotor angular position (θ̂elec) received, characterised in that the control device further includes: - a closed loop adaptive angle estimator (6), for estimating the value of the estimated rotor angular position (θ̂elec), based on a difference between at least one piece of data of a reference stator flux vector (λq) calculated as a function of the return currents (Iq), and a piece of data of the adaptive stator flux vector (λqv), calculated from the voltage setpoint (V#dq), the return currents (Iq, Id), and an estimated electrical speed (ω̂elec) calculated as a function of this difference, - a modifier (7) of the setpoint current, wherein, when an absolute value of the estimated electrical speed (ω̂elec) received by the adaptive angle estimator (6) is less than the predetermined threshold speed, the modifier (7) calculates a regulator setpoint direct current (I#d') having a non- zero value, thus modifying a setpoint direct current (I#d) of a setpoint current (I#dq) received.

2. The device (2) for controlling an inverter converter (C) for starting an electric motor (1) according to the preceding claim, wherein the adaptive angle estimator (6) comprises: o a first stator flux calculator (61) comprising:

1. a first block (611) for calculating the reference stator quadratic flux (λq), 2. a second block (612) for calculating the adaptive quadratic stator flux (λqv), o a second calculator (62) for an estimated electrical speed (ω̂elec) as a function of a comparison between the adaptive quadratic stator flux (λqv) and the reference quadratic stator flux (λq), o a third calculator (63) for estimating a value for the estimated rotor angular position (θ̂elec) as a function of the estimated electrical speed (ω̂elec) calculated.

3. The device (2) for controlling an inverter converter (C) for starting an electric motor (1) according to the preceding claim, the first calculation block (611) calculates the reference quadratic flux (λq) according to the formula: LqIq wherein Lq is the stator inductance on the axis q.

4. The device (2) for controlling an inverter converter (C) for starting an electric motor (1) according to claim 2 or 3, wherein the second calculation block (612) calculates the adaptive quadratic stator flux (λqv) according to the integral of the following formula: vq - Rsiq - ωelecλ̂dv + kĩq wherein Rs is the stator resistance, λdv is the adaptive direct stator flux equal to the integral of the following formula: vd - Rsid + ωelecλ̂qv + kĩd and K is a positive gain matrix, l̃d and l̃q are the direct and quadratic components of the error in currents with: ∘ i ˜ d = i d − i ^ d ∘ i ˜ q = i q − i ^ q ∘ and wherein the estimated direct current l̃d and the estimated quadratic current l̃q are calculated as a function of: ∘ i ^ d = λ ^ d v − φ PM L d i ^ q = λ ^ q v L q 5. The device (2) for controlling an inverter converter (C) for starting an electric motor (1) according to one of the preceding claims 1 to 4, wherein the modifier (7) of the setpoint direct current (I#d) calculates a regulator setpoint direct current (I#d') equal to the square root of the sum of a value of the squared maximum quadratic current with the value of the setpoint quadratic current (I#q), received in the setpoint current (I#dq): I d ′ # = Iqmax 2 − I <none / > # q <none / > <none / > 2 .

6. The device (2) for controlling an inverter converter (C) for starting an electric motor (1) according to the preceding claim, wherein calculating the modified setpoint direct current ( I#d') is imposed with the same sign as the estimated electrical speed (ω̂elec) received.

7. The device (2) for controlling an inverter converter (C) for starting an electric motor (1) according to one of the preceding claims 1 to 6, wherein the modifier (7) of the setpoint direct current (I#d) comprises a comparator for comparing the absolute value of the estimated electrical speed (ω̂elec) with the predetermined threshold speed.

8. The device (2) for controlling an inverter converter (C) for starting an electric motor (1) according to one of the preceding claims 1 to 7, wherein the control device is able to control the converter for a rotation speed beyond the predetermined threshold speed, wherein if the absolute value of the estimated electrical speed (ω̂elec) is greater than the predetermined threshold speed, the setpoint direct current modifier transmits the regulation setpoint current (I#dq') according to only the setpoint current (I#dq).

9. The device (2) according to the preceding claim, wherein when the absolute value of the estimated electrical speed (ω̂elec) is greater than the predetermined threshold speed, the modifier (7) of the setpoint direct current (I#d) transmits the setpoint current (I#dq) with a setpoint direct current (I#d) equal to zero, unless a defluxing setpoint is sent.

10. The device (2) according to claim 8 or 9, wherein, if the absolute value of the estimated electrical speed (ω̂elec) is greater than the predetermined threshold speed, the angular position estimator estimates the position and angular speed from an electromotive force EMF measured.

11. A synchronous electric machine comprising: - an electric motor comprising a rotor and a stator, - a current measurement sensor, - an inverter converter comprising power switches and - the control device according to one of the preceding claims, wherein the calculation unit transmits a command to the converter from the voltage setpoint (V#dq) to drive the electronic power switches to a specific chopping frequency and thus drive the fundamental frequency of the stator voltage input to the electric machine for driving the electric motor (1).

12. A method for driving a synchronous machine in a closed loop without a position sensor, from start-up to maximum rotation speed, comprising the steps of: - modifying a received current setpoint (I#dq) by imposing a non-zero modified setpoint direct current (I#d') as long as an estimated speed (ω̂elec) is less than a predetermined threshold speed value, - calculating a voltage setpoint (V#dq) comprising a quadratic and direct voltage with an angle, from a regulation current setpoint (I#dq') comprising a modified setpoint direct current (I#d'), - calculating a command (PWM) for driving the electronic power switches of the inverter converter (C) at a specific chopping frequency and thus driving the fundamental frequency of the stator voltage input to the electric machine, by an inverse Park transformation of the voltage setpoint (V#dq) and of a rotor estimated angular position (θ̂elec), - measuring the phase currents (Iu, Iv, Iw) and transforming them in a Park reference frame, into a return quadratic current (Iq ) and a return direct current (Id ), taking account of the estimated rotor angular position (θ̂elec), - calculating a reference stator quadratic flux (λq) as a function of the return quadratic current (Iq), the return direct current (Id), the current return output (Idq) and a calculated electrical frequency (ω̂elec), - calculating adaptive quadratic stator flux (λqv), from the voltage setpoint (V#dq), the return quadratic current (Iq), the return direct current (Id), and the estimated electrical speed (ω̂elec) calculated, - calculating an electrical speed (ωelec) of the rotor from a comparison of the adaptive quadratic stator flux (λqv) with the reference quadratic stator flux (λq), - calculating the value of the estimated rotor angular position (θ̂elec) as a function of the estimated electrical speed (ω̂elec) calculated.