Switching power supply

Abstract

Provided is a current resonance type switching power supply device with a simple configuration that has a wide variable range of output voltage and can also achieve ZVS of the main switching element. 【Solution means】The switching power supply device 38 includes a ZVS auxiliary circuit 40 formed by a series circuit of an auxiliary diode 42, an auxiliary inductor 44, and an auxiliary DC power supply 46, which is connected in parallel across both ends of the main switching element 14(2). The switching control circuit 36 determines first and second on-times ton1 and ton2 based on the control voltage Vs of the output by the error amplification circuit 34, and provides a dead time Td to turn on and off the main switching elements 14(1) and 14(2) complementarily. When the first on-time ton1 is ton1 ≧ Tk, control of the first operation mode is performed to set the second on-time ton2 to ton2 = ton1. When the first on-time ton1 becomes ton1 < Tk, control of the second operation mode is performed to set the second on-time ton2 to ton2 > ton1.

Description

Technical Field

[0001] The present invention relates to a switching power supply device using a current resonance type half - bridge method or full - bridge method.

Background Art

[0002] This type of current resonance type switching power supply device has the advantages of high efficiency and low noise, but there is a problem that it is difficult to widen the variable range of the output voltage Vo. Therefore, the inventor has studied a new technology for widening the variable range of the output voltage Vo. <Configuration of Conventional Switching Power Supply Device 10> First, the conventional switching power supply device 10 shown in Fig. 26(a) will be described. The switching power supply device 10 is a slightly improved version of a general current resonance type half - bridge method switching power supply device.

[0003] The switching power supply device 10 has a first arm 14 connected between a pair of input terminals 12a and 12b to which an input voltage Vi is applied. The first arm 14 is composed of a series circuit of first and second main switching elements 14(1) and 14(2). The first main switching element 14(1) is arranged on the high - side, and the second main switching element 14(2) is arranged on the low - side. The first and second main switching elements 14(1) and 14(2) can use, for example, N - channel MOS - type FETs.

[0004] The main transformer 16 has an input winding 18 and an output winding 20 that are magnetically coupled to each other. One end of the input winding 18 is connected to the midpoint of the first arm 14, a resonant capacitor 22 is connected between the other end of the input winding 18 and the input terminal 12b, and a resonant inductor 24 is inserted in series with the input winding 18. The input winding 18, the resonant capacitor 22, and the resonant inductor 24 form a current-resonant type resonant circuit. The resonant inductor 24 may be made of discrete components or may be substituted with the leakage inductance generated between the input and output windings of the main transformer 16. When the main switching elements 14(1) and 14(2) are switched on or off, an AC voltage is applied across the input winding 18, and an AC voltage is generated in the output winding 20.

[0005] A so-called center-tap rectifier type output rectifier and smoothing circuit 26 is connected to both ends of the output winding 20. The output rectifier and smoothing circuit 26 consists of rectifier diodes 28x and 28y and a smoothing capacitor 30. The cathode of the rectifier diode 28x is connected to one end of the output winding 20, the cathode of the rectifier diode 28y is connected to the other end of the output winding 20, the anodes of the rectifier diodes 28x and 28y are connected to each other, and the smoothing capacitor 30 is connected between the anodes of the rectifier diodes 28x and 28y and the midpoint of the output winding 20. The output rectifier and smoothing circuit 26 rectifies and smooths the AC voltage generated in the output winding 26 to generate an output voltage Vo, and outputs the output voltage Vo and output current Io to the load 32 connected to the smoothing capacitor 30.

[0006] Furthermore, the switching power supply unit 10 includes an error amplification circuit 34 and a switching control circuit 36. The error amplification circuit 34 amplifies the difference between the output voltage Vo and the target voltage Vor, and outputs a control signal Vs that increases or decreases the output voltage Vo in the direction of bringing it closer to the target voltage Vor. This error amplification circuit 34 is configured as an inverting amplifier circuit, and operates by decreasing the control signal Vs when the output voltage Vo is higher than the target voltage Vor, and increasing the control signal Vs when the output voltage Vo is lower than the target voltage Vor. The target voltage Vor can be varied externally by the user, thereby allowing the output voltage Vo to be set or changed to a desired value.

[0007] The switching control circuit 36 ​​is a circuit that controls the on / off switching of the first and second main switching elements 14(1) and 14(2). Upon receiving a control signal Vs, it determines the first on-time ton1 [ton1≧0], which is the on-time of the first main switching element 14(1), and the second on-time ton2 [ton2≧0], which is the on-time of the second main switching element 14(2). Then, while providing a predetermined dead time Td so that the first and second main switching elements 14(1) and 14(2) do not turn on simultaneously, it turns the first and second main switching elements 14(1) and 14(2) on and off complementaryly. Therefore, the switching period Tsw is Tsw = ton1 + ton2 + 2·Td ≈ ton1 + ton2. Hereafter, the dead time Td will be described assuming that it is fixed to a constant length.

[0008] The switching control circuit 36 ​​outputs a drive pulse Vgs(1) for driving the first main switching element 14(1) and a drive pulse Vgs(2) for driving the second main switching element 14(2). Since the first main switching element 14(1) is located on the high-side, the drive pulse Vgs(1) is input to the first main switching element 14(1) through the high-side driver 36a. The high-side driver 36a is part of the switching control circuit 36.

[0009] The method for determining the first and second on-times ton1 and ton2 by the switching control circuit 36 will be described. As shown in the graph of FIG. 26(b), the switching control circuit 36 sets the first on-time ton1 to a value substantially proportional to the control voltage Vs. When the control voltage Vs increases, the first on-time ton1 is lengthened, and when the control voltage Vs decreases, it is shortened. When the determined first on-time ton1 satisfies ton1≧Tk (when the control voltage Vs is higher than a predetermined value), control in the first operation mode where the second on-time is set to ton2 = ton1 is performed. When the determined first on-time ton1 becomes ton1<Tk (when the control voltage Vs is lower than a predetermined value), control in the second operation mode where the second on-time ton2 satisfies ton2>ton1 is performed. Note that Tk is a reference time set in advance in the switching control circuit 36 [Tk>0].

[0010] FIG. 27(a) is a graph with the output voltage Vo and the target voltage Vor on the vertical axis and the output current Io on the horizontal axis, representing the area where the switching power supply device 10 performs control in the first operation mode as the first operation region DR1 and the area where it performs control in the second operation mode as the second operation region DR2. The boundary line separating the two operation regions DR1 and DR2 indicates the position where ton1 = ton2 = Tk.

[0011] Note that the position of the boundary line shown in FIG. 27(a) is only a typical example, and the position of the boundary line between the first operation region DR1 and the second operation region DR2 varies depending on the constant settings of the power section of the switching power supply device 10. <Features of the conventional switching power supply device 10> An improvement of the conventional switching power supply device 10 over a general switching power supply device is that when the first on-time ton1 becomes shorter than a certain level, the control in the first operation mode is stopped and the control in the second operation mode is performed.

[0012] A general switching power supply device stabilizes the output voltage Vo only by controlling the first operation mode, but there are limitations to the control of the first operation mode. For example, when the target voltage Vor is changed and the output voltage Vo is reduced from 100% (rated) to about 60 - 80%, the switching period Tsw becomes shorter to the range of 1 / 10 or less of the rated value, resulting in problems such as extremely large switching losses or unstable response of the switching control circuit. Therefore, it is extremely difficult to widen the variable range of the output voltage Vo only by controlling the first operation mode.

[0013] Therefore, in the conventional switching power supply device 10, when the first on - time ton1 becomes ton1 < Tk, the operation of the second control mode of making ton1 < ton2 is performed. In the current - resonance - type half - bridge method, when the switching period Tsw is fixed, the output voltage Vo is the highest when ton1 = ton2, and the output voltage Vo becomes lower as the difference between ton1 and ton2 becomes larger. That is, since the output voltage Vo can be changed by changing the ratio of ton1 and ton2, the switching period Tsw does not become so short.

[0014] Therefore, even when the target voltage Vor is changed and the output voltage Vo is reduced from 100% (rated) to about 10 - 20% in the conventional switching power supply device 10, the switching period Tsw only becomes shorter to about 1 / 2 - 1 / 3 of the rated value, and the above - mentioned problems are less likely to occur, so the variable range of the output voltage Vo can be significantly widened.

[0015] Note that in a current - resonance - type switching power supply device, the idea of making ton1 ≠ ton2 under certain conditions is not new. For example, Patent Document 1 describes controlling to make ton1 ≠ ton2 when the output current Io is small so that the switching period Tsw does not become too short.

Prior Art Documents

Patent Documents

[0016] [Patent Document 1] Japanese Patent Publication No. 2006-204044 [Overview of the project] [Problems that the invention aims to solve]

[0017] In conventional switching power supplies 10, under certain conditions, the first main switching element 14(1) may become unable to perform ZVS (Zero Voltage Switching), potentially leading to increased switching losses and switching noise. This will be explained below with reference to Figures 27 to 33. <Regarding the operating point and the signs of the current-voltage waveforms used for the operation explanation> In Figure 27(a), the second operating region DR2 described above is further divided into two regions, DR2(1) and DR2(2). Region DR2(2) is a region within the second operating region DR2 in which the output voltage Vo is relatively low and the output current Io is relatively small. The operating points P1, P2, and P3 of the switching power supply 10 are plotted on the graph in Figure 27(a). Operating point P1, located in the first operating region DR1, is the point where the target voltage Vor is set to 100% (rated), and the output voltage Vo=Vor and output current Io=100% (rated) are output. Similarly, operating point P2, also located in the first operating region DR1, is the point where the target voltage Vor is set to 100% (rated), and the output voltage Vo=Vor and output current Io≈30% are output. Furthermore, operating point P3, located in the second operating region DR2(2), is the point where the target voltage Vo is set to approximately 25%, and the output voltage Vo=Vor and output current Io≈30% are output. In the first operating region DR1 [operating points P1, P2] and the second operating region DR2(1), both the first and second switching elements 14(1) and 14(2) can perform ZVS (Zero Vital Stability). However, in the second operating region DR2(2) [operating point P3], only the first switching element 14(1) cannot perform ZVS. This point will be explained in detail later, but before that, the signs of the voltages and currents in each part indicated in the time charts, etc., in Figures 28 to 33 will be explained based on Figure 27(b).

[0018] Figure 27(b) is an equivalent circuit for clearly explaining the operation of the switching power supply 10. The first main switching element 14(1) is shown separately as the FET body (switch), the parasitic diode between the drain and source, and the parasitic capacitor between the drain and source. The drive pulse input between the gate and source of the FET body is denoted as Vgs(1), the voltage generated between the drain and source as Vds(1), the combined current obtained by combining the currents flowing through the FET body and the parasitic diode is denoted as Id(1), and the current flowing through the parasitic capacitor is denoted as Ic(1). Incidentally, if a snubber capacitor is connected in parallel to both ends of the first main switching element 14(1), that snubber capacitor can be considered as being included in the parasitic capacitor.

[0019] Similarly, the second main switching element 14(2) is represented separately as the FET body (switch), the parasitic diode between the drain and source, and the parasitic capacitor between the drain and source. The drive pulse input between the gate and source of the FET body is denoted as Vgs(2), the voltage generated between the drain and source as Vds(2), the combined current obtained by combining the currents flowing through the FET body and the parasitic diode is denoted as Id(2), and the current flowing through the parasitic capacitor is denoted as Ic(2). Incidentally, if a snubber capacitor is connected in parallel across both ends of the second main switching element 14(2), that snubber capacitor can be considered as being included in the parasitic capacitor.

[0020] Furthermore, the input winding 18 of the main transformer 16 is shown separately as the excitation inductor and the input winding of the ideal transformer, with the voltage across the excitation inductor being denoted as V(Lm) and the current flowing through the excitation inductor as I(Lm). The current flowing through the resonant inductor 24 is denoted as I(Lk), and the voltage across the resonant capacitor 22 is denoted as V(Ck). Additionally, the current flowing through the rectifier diode 28x is denoted as Ifx, and the current flowing through the rectifier diode 28y is denoted as Ify. The definition of the "positive direction" for each voltage and current is shown by the arrows in Figure 27(b). <Operation of the switching power supply unit 10 [Operating point P1]> First, the operation at the operating point P1 of the switching power supply device 10 will be described based on FIGS. 28 to 30. The time chart in FIG. 28 is a schematic diagram of the voltage and current waveforms of each part at the operating point P1, and the switching period Tsw(P1), which is one cycle of switching, is divided into periods t1 to t10 and shown.

[0021] The first main switching element 14(1) turns on during the periods t9, t10, t1 when the drive pulse Vgs(1) is high, and the total length of these three periods is the first on-time ton1. Similarly, the second main switching element 14(2) turns on during the periods t4, t5, t6 when the drive pulse Vgs(2) is high, and the total length of these three periods is the second on-time ton2. Also, during the periods t2, t3, both the drive pulses Vgs(1) and Vgs(2) are low, and the total length of these two periods is the dead time Td. Similarly, during the periods t7, t8, both the drive pulses Vgs(1) and Vgs(2) are low, and the total length of these two periods is the dead time Td.

[0022] At the operating point P1, the control of the first operation mode is performed, and ton1 = ton2. During the period of the first on-time ton1, the relationship V(Lm) = Vi - V(Ck) ≒ Vi / 2 is maintained and the current I(Lm) rises with a right shoulder upward. During the period of the second on-time ton2, the relationship V(Lm) = V(Ck) ≒ -Vi / 2 is maintained and the current I(Lm) decreases with a right shoulder downward. Therefore, the waveform of the current I(Lm) becomes a triangular waveform that oscillates symmetrically with positive and negative amplitudes centered on zero.

[0023] The current I(Lk) has a waveform in which the load current component is added to the current I(Lm). The load current component corresponds to the components of the currents Ifx and Ify. During the period of the first on-time ton1, a sawtooth-like load current component with a rounded peak (the component corresponding to the current Ifx) is generated in the positive direction. During the period of the second on-time ton2, a sawtooth-like load current component with a rounded peak (the component corresponding to the current Ify) is generated in the negative direction. These load current components become the output current Io and are output to the load 32.

[0024] For the second main switching element 14(2) to achieve ZVS, during the dead time Td from when the drive pulse Vgs(1) transitions from high to low until the drive pulse Vgs(2) transitions from low to high, the voltage Vds(2) needs to rapidly decrease from Vi to 0V, and the voltage Vds(1) needs to rapidly increase from 0V to Vi. Generally speaking, the rate of decrease of the voltage Vds(2) is determined by the combined occurrence of the L-component energy release operation shown in Fig. 29(a) and the free oscillation operation shown in Fig. 29(b).

[0025] The L-component energy release operation is an operation in which the energy stored in the resonance inductor 24 at the end of the first on-time ton1 serves as an energy source, and an energy release current (current I(Lk)) is generated at the start of the dead time Td. This current becomes the currents Ic1 and Ic2 and flows through the parasitic capacitors Coss(1) and Coss(2) of the first and second main switching elements 14(1) and 14(2), charging Coss(1) and discharging Coss(2). Since the value of the energy release current is approximately the value of the current I(Lk) at the end of the first on-time ton1, the greater the |I(Lk)| at the end of the first on-time ton1, the faster the rate of decrease of the voltage Vds(2).

[0026] Free oscillation is the operation in which, at the end of the first on-time ton1, the energy stored in the parasitic capacitor Coss(2) of the second main switching element 14(2) becomes the energy source, and during the dead time Td, an oscillating current flows through the resonant system formed by the parasitic capacitors Coss(1), Coss(2), etc. This current becomes Ic(1), Ic(2) and flows through the parasitic capacitors Coss(1), Coss(2), charging Coss(1) and discharging Coss(2). The center voltage of the free oscillation of voltage Vds(2) is approximately the voltage V(Ck) at the end of the first on-time ton1. If |Vds(2)-V(Ck)|=Vi-V(Ck) at the end of the first on-time ton1 is Va1, then the voltage Vds(2) decreases from Vi towards Vi-2·Va1 as the dead time Td begins. Therefore, the lower V(Ck) is at the end of the first on-time ton1, the higher the voltage Va1 becomes, and the faster the rate of decrease of the voltage Vds(2).

[0027] At the operating point P1, the switching period Tsw(P1) ≈ ton1 + ton2 is sufficiently long, so as shown in Figure 28, the current I(lk) at the end of the first on-time ton1 is sufficiently large. Also, the voltage Va1 at the end of the first on-time ton1 is a suitable value Vi / 2. Therefore, both the L component energy release operation and the free oscillation operation are strongly performed, the currents Ic1 and Ic2 become sufficiently large, the rate of change of voltages Vds(1) and Vds(2) accelerates, and the voltage Vds(2) ≈ 0V in a very short time, ending period t2. Then, as period t3 begins, a current I(Lk) ≈ I(Lm) starts flowing through the parasitic diode of the second main switching element 14(2), generating a current Id(2). Subsequently, the second main switching element 14(2) turns on, and period t4 begins, and the path of current Id(2) switches from the parasitic diode to the FET body.

[0028] Thus, at the operating point P1, the voltage Vds(2) can quickly drop to 0V before the start of period t4 (before the end of dead time Td), and the second main switching element 14(2) can easily be switched to ZVS.

[0029] For the first main switching element 14(1) to perform ZVS, during the dead time Td period from when the drive pulse Vgs(2) switches from high to low until when the drive pulse Vgs(1) switches from low to high, the voltage Vds(1) must rapidly decrease from Vi to 0V, and the voltage Vds(2) must rapidly increase from 0V to Vi. Generally speaking, the rate at which the voltage Vds(1) decreases is determined by a combination of the L component energy release operation shown in Figure 30(a) and the free oscillation operation shown in Figure 30(b).

[0030] The L-component energy release operation is an operation in which the energy stored in the resonant inductor 24 at the end of the second on-time ton2 becomes the energy source, and generates an energy release current (current I(Lk)) at the start of the dead time Td. This becomes currents Ic1 and Ic2 and flows through the parasitic capacitors Coss(1) and Coss(2), discharging Coss(1) and charging Coss(2). The value of the energy discharge current is approximately equal to the value of the current I(Lk) at the end of the second on-time ton2, so the larger |I(Lk)| at the end of the second on-time ton1, the faster the voltage Vds(1) decreases.

[0031] Free oscillation is the process in which the energy stored in the parasitic capacitor Coss(1) at the end of the second on-time ton2 becomes the energy source, and during the dead time Td, an oscillating current flows through the resonant system formed by the parasitic capacitors Coss(1), Coss(2), etc. This current, Ic(1) and Ic(2), flows through the parasitic capacitors Coss(1) and Coss(2), charging Coss(1) and discharging Coss(2). The center voltage of the free oscillation of voltage Vds(2) is approximately the voltage V(Ck) at the end of the second on-time ton2. If |Vds(2)-V(Ck)|=V(Ck) at the end of the second on-time ton2 is Va2, then voltage Vds(2) rises from 0V towards 2·Va2 at the start of the dead time Td, and voltage Vds(1) decreases from Vi towards Vi-2·Va2. Therefore, the higher V(Ck) is at the end of the second on-time ton2, the higher the voltage Va2 becomes, and the faster the rate of decrease of the voltage Vds(1).

[0032] At the operating point P1, the switching period Tsw(P1) ≈ ton1 + ton2 is sufficiently long, so as shown in Figure 28, the current I(Lk) at the end of the second on-time ton2 is sufficiently large. Also, the voltage Va2 at the end of the second on-time ton2 is a suitable value Vi / 2. Therefore, both the L component energy release operation and the free oscillation operation are strongly performed, the currents Ic1 and Ic2 become sufficiently large, the rate of change of voltages Vds(1) and Vds(2) accelerates, and the voltage Vds(1) ≈ 0V becomes very short, ending period t7. Then, as period t8 begins, a current I(Lk) ≈ I(Lm) starts flowing through the parasitic diode of the second main switching element 14(2), generating a current Id(1). Subsequently, the first main switching element 14(1) turns on, and period t9 begins, and the path of current Id(1) switches from the parasitic diode to the FET body.

[0033] Thus, at the operating point P1, the voltage Vds(1) can quickly drop to 0V before the period t9 begins (before the dead time Td ends), and the first main switching element 14(1) can easily perform ZVS.

[0034] As explained above, the operating point P1 of the switching power supply 10 is such that the first and second switching elements 14(1) and 14(2) can be easily switched using ZVS. <Operation of switching power supply unit 10 [Operating point P2]> Next, the operation of the switching power supply 10 at the operating point P2 will be explained based on Figure 31. The time chart in Figure 31 is a schematic diagram of the voltage and current waveforms of each part at the operating point P2, and the switching period Tsw(P2), which is one period of switching, is divided into periods t1 to t10.

[0035] At operating point P2, the first operating mode is controlled, resulting in ton1=ton2, and the overall shape of each waveform is similar to that of each waveform at operating point P1.

[0036] At operating point P2, the output voltage Vo = 100% (rated), similar to operating point P1. However, since the output current Io is reduced from 100% (rated) to approximately 30%, the switching period Tsw(P2) becomes shorter than at operating point P1. The extent of this reduction depends on the constant settings of the power section, but in Figure 31, the amplitude of the current I(Lm) is reduced to approximately 1 / 2, assuming that the switching period Tsw(P2) is Tsw(P1) / 2. Also, because the output current Io is smaller (approximately 30%), the load current component (a sawtooth wave component with rounded peaks) generated in the current I(Lk) is also reduced.

[0037] Looking at the ZVS of the second main switching element 14(2), at the operating point P2, the L-component energy release operation shown in Figure 29(a) and the free vibration operation shown in Figure 29(b) are performed in combination.

[0038] However, at the operating point P2, the amplitude of the current I(lk) is small, so the L component energy release operation at the end of the first on-time period ton1 is weak. On the other hand, the free oscillation operation at the end of the first on-time period ton1 is performed with the same strength as at the operating point P1. Therefore, at the operating point P2, the free oscillation operation is strong, which allows the voltage Vds(2) to drop to 0V before the start of period t4 (before the end of dead time Td), enabling the second main switching element 14(2) to perform ZVS.

[0039] Next, looking at the ZVS of the first main switching element 14(1), at the operating point P2, the L component energy release operation shown in Figure 30(a) and the free vibration operation shown in Figure 30(b) are performed in combination.

[0040] However, since the amplitude of the current I(lk) is small at the operating point P2, the L component energy release operation at the end of the second on-time period ton2 is weak. On the other hand, the free oscillation operation at the end of the second on-time period ton2 is performed with the same strength as at the operating point P1. Therefore, because the free oscillation operation is strong at the operating point P2, the voltage Vds(1) can drop to 0V before the start of period t9 (before the end of dead time Td), and the first main switching element 14(1) can perform ZVS.

[0041] Thus, the operating point P2 of the switching power supply 10 can be ZVS'd by the first and second switching elements 14(1) and 14(2). <Operation of switching power supply unit 10 [Operating point P3]> Next, the operation of the switching power supply 10 at the operating point P3 will be explained based on Figures 32 and 33. The time chart in Figure 32 is a schematic diagram of the voltage and current waveforms of each part at the operating point P3, and the switching period Tsw(P3), which is one period of switching, is divided into periods t11 to t16.

[0042] The first main switching element 14(1) turns on during the period t11 when the drive pulse Vgs(1) is high, and the length of the period t11 becomes the first on-time ton1. The second main switching element 14(2) turns on during the periods t14 and t15 when the drive pulse Vgs(2) is high, and the total length of these two periods becomes the second on-time ton2. Also, during the periods t12 and t13, both the drive pulses Vgs(1) and Vgs(2) are low, and the total length of these two periods is the dead time Td. Similarly, during the period t16, both the drive pulses Vgs(1) and Vgs(2) are low, and the length of the period t16 is the dead time Td.

[0043] The operating point P3 has an output current Io ≒ 30% similar to the operating point P2, but since the output voltage Vo has decreased from 100% (rated) to approximately 25%, the control of the second operating mode is performed, and the switching period Tsw(P3) becomes shorter than Tsw(P2). How much shorter it becomes depends on the constant setting of the power unit, but in Fig. 32, it is assumed that Tsw(P3) ≒ Tsw(P2) / 2.

[0044] Since ton1 < ton2 at the operating point P3, the voltage V(Ck) becomes lower than at the operating point P2 and V(Ck) < Vi / 2. Therefore, during the period of the first on-time ton1, the voltage V(Lm) becomes a predetermined value within the range of Vi / 2 < V(Lm) < Vi, and the current I(Lm) rises with a relatively steep slope and rises with a right shoulder. Also, during the period of the second on-time ton2, the voltage V(Lm) becomes a predetermined value within the range of -Vi / 2 < V(Lm) < 0, and the current I(Lm) decreases with a relatively gentle slope and decreases with a right shoulder. The waveform of the current I(Lm) becomes a triangular waveform that oscillates positive and negative around zero even when ton1 < ton2.

[0045] The current I(Lk) is represented by a waveform obtained by adding the load current component to the current I(Lm). The load current component corresponds to the currents Ifx and Ify, but at the operating point P3, current Ifx flows only through rectifier diode 28x, and current Ify does not flow through rectifier diode 28y. This is because the voltage V(Lm) during the second on-time period ton2 is lower than the voltage V(Lm) during the first on-time period ton1, and the terminals of rectifier diode 28y are always reverse-biased.

[0046] The current Ifx is generated during the period t11 to t14. During period t11 (the period of the first ON time ton1), the current is supplied from the input power supply and increases steadily. During period t12 to t14, the current Ifx is the current that releases the energy stored in the resonant inductor 24 at the end of period t11, and decreases steadily until period t14 ends when all the energy has been released. Therefore, a load current component (a component corresponding to the current Ifx) is generated in the current I(Lk) in the positive direction during the period t11 to t14. This load current component becomes the output current Io and is output to the load 32. Note that in Figure 32, the load current component generated in I(Lk) is drawn to be somewhat prominent, but since the output current Io is small at the operating point P3, the load current component generated in I(Lk) is not as large as it appears.

[0047] Looking at the ZVS of the second main switching element 14(2), at the operating point P3, the L component energy release operation shown in Figure 29(a) and the free vibration operation shown in Figure 33(a) are performed in combination.

[0048] At operating point P3, similar to operating point P2, the amplitude of the current I(lk) is small, so the L component energy release operation at the end of the first on-time ton1 is weak. On the other hand, the free oscillation operation at the end of the first on-time ton1 is stronger than at operating points P1 and P2. This is because, as shown in Figure 33(a), the voltage V(Ck) becomes lower than Vi / 2 and the voltage Va1 becomes higher. Therefore, at operating point P3, the free oscillation operation is strong, allowing the voltage Vds(2) to drop to 0V before the start of period t14 (before the end of dead time Td), and the second main switching element 14(2) can be switched to zero-voltage mode (ZVS).

[0049] Looking at the ZVS of the first main switching element 14(1), at the operating point P3, the L component energy release operation shown in Figure 30(a) and the free vibration operation shown in Figure 33(b) are performed in combination.

[0050] At operating point P3, similar to operating point P2, the amplitude of the current I(lk) is small, so the L component energy release operation at the end of the second on-time ton2 is weak. Also, the free oscillation operation at the end of the second on-time ton2 is weaker than at operating points P1 and P2. This is because, as shown in Figure 33(b), the voltage V(Ck) becomes lower than Vi / 2, and the voltage Va2 becomes lower. Therefore, both the L component energy release operation and the free oscillation operation are weaker, and the voltage Vds(1) cannot drop to 0V before the start of period t11 (before the end of dead time Td), and the second main switching element 14(2) cannot perform ZVS.

[0051] Thus, at the operating point P3, the first switching element 14(1) can no longer perform ZVS, resulting in increased switching losses and switching noise. <Summary> The conventional switching power supply device 10 controls the first operation mode when the first on-time ton1 is long, and controls the second operation mode when the first on-time ton1 becomes shorter than a certain value. Therefore, in principle, the variable range of the output voltage Vo can be widened. However, in the region DR2(2) [the region where the operating point P3 is located] within the second operation region DR2 where the second operation mode is controlled, the first main switching element 14(1) cannot achieve ZVS, resulting in problems such as increased switching losses and noise.

[0052] The present invention has been made in view of the above background art, and an object thereof is to provide a current resonance type switching power supply device with a simple configuration that has a wide variable range of output voltage and can also achieve ZVS of the main switching element.

Means for Solving the Problems

[0053] The present invention is a current resonance type half-bridge switching power supply device including: a first arm composed of a series circuit of first and second main switching elements connected between a pair of input terminals to which an input voltage Vi is applied; a main transformer having an input winding and an output winding; a current resonance type resonance circuit connected to the first arm and composed of the input winding, a resonance capacitor, and a resonance inductor; and an output rectifying and smoothing circuit that rectifies and smoothes an AC voltage generated in the output winding when the first and second main switching elements are turned on and off to generate a predetermined output voltage Vo, and supplies the output voltage Vo and an output current Io to a load.

[0054] And a circuit that amplifies the difference between the output voltage Vo and a target voltage Vor and outputs a control signal Vs that increases or decreases the output voltage Vo in a direction approaching the target voltage Vor. When Vo > Vor, the control signal Vs is changed in a direction to shorten the first on-time ton1 [ton1 ≧ 0], which is the on-time of the first main switching element. When Vo < Vor, the control signal Vs is changed in a direction to lengthen the first on-time ton1. A circuit for controlling the on / off of the first and second main switching elements, which receives the control signal Vs and determines the first on-time ton1 and the second on-time ton2 [ton2 ≧ 0] which is the on-time of the second main switching element, and provides a predetermined dead time Td so that the first and second switching elements do not turn on simultaneously, and complementarily turns on and off the first and second main switching elements; It is provided with a ZVS auxiliary circuit composed of a series circuit of an auxiliary diode, an auxiliary inductor, and an auxiliary DC power supply connected in parallel across both ends of the second main switching element; The direction of the auxiliary diode and the polarity of the auxiliary DC power supply are set such that when the second main switching element is on, the auxiliary diode is on, and the current flowing out from the auxiliary DC power supply flows through the second main switching element and the auxiliary diode and is supplied to the auxiliary inductor. The voltage value of the auxiliary DC power supply is set to a low value at which both ends of the auxiliary diode are reverse-biased when the second main switching element is off; A reference time Tk [Tk > 0] is set in advance in the switching control circuit. When the first on-time ton1 determined based on the control signal Vs satisfies ton1 ≧ Tk, the switching control circuit performs control in a first operation mode where the second on-time ton2 is set to ton2 = ton1. When the first on-time ton1 determined based on the control signal Vs becomes ton1 < Tk, the switching control circuit performs control in a second operation mode where the second on-time ton2 is set to ton2 > ton1.

[0055] Preferably, the ZVS auxiliary circuit has the auxiliary inductor and the auxiliary DC power supply adjacent to each other in series, and protective diodes are connected in parallel to both ends of the series circuit of the auxiliary inductor and the auxiliary DC power supply, and the orientation of the protective diodes is set to prevent current flowing from the positive terminal of the auxiliary DC power supply to the protective diodes. Furthermore, it is preferable that the switching control circuit is configured to shut off a specific part of the ZVS auxiliary circuit to prevent current from flowing to the ZVS auxiliary circuit, or to forcibly set the voltage value of the auxiliary DC power supply to zero, when the first on time ton1 determined based on the control signal Vs is ton1 ≥ Tk.

[0056] The switching control circuit may be configured to stop controlling the second operating mode when the first on-time ton1, determined based on the control signal Vs, becomes ton1 = 0, thereby maintaining the switching period at a constant length. It may also be configured to control a third operating mode in which, when the control signal Vs changes in a direction that shortens the first on-time ton1, the second on-time ton2 is shortened in accordance with the control signal Vs, and when the control signal Vs changes in a direction that lengthens the first on-time ton1, the second on-time ton2 is lengthened in accordance with the control signal Vs.

[0057] The switching control circuit may be configured such that when the first on-time ton1 determined based on the control signal Vs becomes ton1=0, it stops controlling the second operating mode and performs a fourth operating mode control that forcibly holds the second on-time ton2 at ton2=0.

[0058] When the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0, while continuing the control of the second operation mode, when the control signal Vs changes in a direction to shorten the first on-time ton1, the voltage value of the auxiliary DC power supply is decreased according to the control signal Vs, and when the control signal Vs changes in a direction to lengthen the first on-time ton1, the voltage value of the auxiliary DC power supply may be increased according to the control signal Vs.

[0059] Further, the present invention includes a first arm composed of a series circuit of first and second main switching elements connected between a pair of input terminals to which an input voltage Vi is applied, a second arm composed of a series circuit of third and fourth main switching elements connected in parallel with the first arm, a main transformer having an input winding and an output winding, a current resonance type resonance circuit composed of the input winding, a resonance capacitor, and a resonance inductor connected between the first arm and the second arm, and an output rectifying and smoothing circuit that rectifies and smooths an AC voltage generated in the output winding when the first to fourth main switching elements are turned on and off to generate a predetermined output voltage Vo and supplies the output voltage Vo and an output current Io to a load, which is a current resonance type full-bridge switching power supply device.

[0060] And a circuit that amplifies the difference between the output voltage Vo and the target voltage Vor and outputs a control signal Vs that increases or decreases the output voltage Vo in a direction approaching the target voltage Vor. When Vo > Vor, the control signal Vs is changed in a direction to shorten the first on-time ton1 (ton1 ≧ 0), which is the on-time of the first main switching element, and when Vo < Vor, the control signal Vs is changed in a direction to lengthen the first on-time ton1, an error amplification circuit. A switching control circuit for controlling the on / off states of the first to fourth main switching elements, wherein, upon receiving the control signal Vs, it determines the on-time ton1 of the first main switching element and the on-time ton2 [ton2≧0] of the second main switching element, provides a predetermined dead time Td so that the first and second switching elements do not turn on simultaneously, and turns the first and second main switching elements on and off complementaryly, turns the third main switching element on and off in the same phase as the second main switching element, and turns the fourth main switching element on and off in the same phase as the first main switching element, A first ZVS auxiliary circuit, consisting of a series circuit of a first auxiliary diode, a first auxiliary inductor, and a first auxiliary DC power supply, is connected in parallel to both ends of the second main switching element. The third main switching element is connected in parallel to both ends of the third main switching element and comprises a second ZVS auxiliary circuit consisting of a series circuit of a second auxiliary diode, a second auxiliary inductor, and a second auxiliary DC power supply. The orientation of the first auxiliary diode and the polarity of the first auxiliary DC power supply are set such that when the second main switching element is on, the first auxiliary diode is on, and the current flowing out from the first auxiliary DC power supply flows to the second main switching element and the first auxiliary diode and is supplied to the first auxiliary inductor; and the voltage value of the first auxiliary DC power supply is set to a low value such that when the second main switching element is off, both ends of the first auxiliary diode are reverse-biased. The orientation of the second auxiliary diode and the polarity of the second auxiliary DC power supply are set such that when the third main switching element is on, the second auxiliary diode turns on, and the current flowing out from the second auxiliary DC power supply flows to the third main switching element and the second auxiliary diode and is supplied to the second auxiliary inductor; and the voltage value of the second auxiliary DC power supply is set to a low value such that when the third main switching element is off, both ends of the second auxiliary diode are reverse-biased. A reference time Tk [Tk > 0] is set in advance in the switching control circuit. When the first on-time ton1 determined based on the control signal Vs satisfies ton1 ≥ Tk, the switching control circuit performs control in a first operation mode to set the second on-time ton2 to ton2 = ton1. When the first on-time ton1 determined based on the control signal Vs becomes ton1 < Tk, the switching control circuit performs control in a second mode to set the second on-time ton2 to ton2 > ton1.

[0061] In the first ZVS auxiliary circuit, the first auxiliary inductor and the first auxiliary DC power supply are adjacent to each other and in a series position. First protection diodes are connected in parallel across both ends of the series circuit of the first auxiliary inductor and the first auxiliary DC power supply. Preferably, the direction of the first protection diode is set to prevent the current flowing out from the positive electrode of the first auxiliary DC power supply from flowing through the first protection diode. In the second ZVS auxiliary circuit, the second auxiliary inductor and the second auxiliary DC power supply are adjacent to each other and in a series position. Second protection diodes are connected in parallel across both ends of the series circuit of the second auxiliary inductor and the second auxiliary DC power supply. Preferably, the direction of the second protection diode is set to prevent the current flowing out from the positive electrode of the second auxiliary DC power supply from flowing through the second protection diode. Further, when the first on-time ton1 determined based on the control signal Vs satisfies ton1 ≥ Tk, the switching control circuit preferably cuts off a specific part of the first ZVS auxiliary circuit to prevent current from flowing through the first ZVS auxiliary circuit, or forcibly sets the voltage value of the first auxiliary DC power supply to zero, and cuts off a specific part of the second ZVS auxiliary circuit to prevent current from flowing through the second ZVS auxiliary circuit, or has a configuration to forcibly set the voltage value of the second auxiliary DC power supply to zero.

[0062] The switching control circuit may be configured to stop controlling the second operating mode when the first on-time ton1, determined based on the control signal Vs, becomes ton1 = 0, thereby maintaining the switching period at a constant length. It may also be configured to control a third operating mode, in which the second on-time ton2 is shortened according to the control signal Vs when the control signal Vs changes in a direction that shortens the first on-time ton1, and the second on-time ton2 is lengthened according to the control signal Vs when the control signal Vs changes in a direction that lengthens the first on-time ton1.

[0063] The switching control circuit may be configured such that when the first on-time ton1, determined based on the control signal Vs, becomes ton1=0, it stops controlling the second operating mode and performs a fourth operating mode control that forcibly holds the second on-time ton2 at ton2=0.

[0064] The switching control circuit may be configured such that when the first on-time ton1 determined based on the control signal Vs becomes ton1=0, it continues to control the second operating mode, and when the control signal Vs changes in a direction that shortens the first on-time ton1, it lowers the voltage values ​​of the first and second auxiliary DC power supplies in accordance with the control signal Vs, and when the control signal Vs changes in a direction that lengthens the first on-time ton1, it raises the voltage values ​​of the first and second auxiliary DC power supplies in accordance with the control signal Vs. [Effects of the Invention]

[0065] According to the switching power supply device of the present invention, by simply making minor improvements such as adding a ZVS auxiliary circuit to a conventional switching power supply device, it is possible to obtain a current-resonant type switching power supply device with a wide variable output voltage range and a simple configuration that also realizes ZVS of the main switching element. [Brief explanation of the drawing]

[0066] [Figure 1] This is a circuit diagram showing a first embodiment of the switching power supply device of the present invention. [Figure 2] Figure 1 shows graph (a) illustrating the control of the first and second operating modes performed by the switching control circuit, and graph (b) plotting the output voltage-output current at the operating point of the switching power supply of the first embodiment. [Figure 3] This is an equivalent circuit diagram for illustrating the operation of the switching power supply device according to the first embodiment. [Figure 4] This is a time chart showing the operation of the operating point P3 of the switching power supply in the first embodiment. [Figure 5] This diagram shows the operation of the switching power supply of the first embodiment after the second ON time ton2 has ended, and includes an equivalent circuit and time chart (a) showing the L component energy release operation performed by the resonant inductor, and an equivalent circuit (b) showing the L component energy release operation performed by the auxiliary inductor of the ZVS auxiliary circuit. [Figure 6] Figure 1 is a circuit diagram showing specific examples of the error amplification circuit and switching control circuit. [Figure 7] Figure 6 is a time chart showing the operation [operation of operating points P1 and P2] of the switching control circuit when it controls the first operating mode. [Figure 8] Figure 6 is a time chart showing the operation of the switching control circuit when it controls the second operating mode [operation at operating point P3]. [Figure 9] (a) is a circuit diagram showing one modified example of the switching power supply device of the first embodiment, and (b) is a circuit diagram showing another modified example. [Figure 10] This is a circuit diagram showing another variation of the switching power supply device of the first embodiment. [Figure 11] (a) is a circuit diagram showing a second embodiment of the switching power supply device of the present invention, and (b) is a graph showing the control content of the first, second, and third operating modes performed by the switching control circuit. [Figure 12]This figure illustrates potential problems that may occur in the switching power supply of the first embodiment, and shows an equivalent circuit (a) showing the current flow during the period when the second main switching element is ON at the operating point P3a, and an equivalent circuit (b) showing the current flow during the period when the second main switching element is OFF at the operating point P3a. [Figure 13] (a) is a graph of the output voltage-output current plotted at the operating point of the switching power supply of the second embodiment, and (b) is a time chart showing the waveform of the auxiliary inductor current at operating points P3a and P4. [Figure 14] This is a circuit diagram showing specific examples of the error amplification circuit and switching control circuit of the switching power supply device according to the second embodiment. [Figure 15] Figure 14 is a time chart showing the operation of the switching control circuit when it controls the second operating mode [operation at operating point P3]. [Figure 16] Figure 14 is a time chart showing the operation of the switching control circuit when it controls the third operating mode [operation at operating point P4]. [Figure 17] This figure shows the operation of a modified example of the switching power supply device of the second embodiment, and is a time chart showing the waveforms of the auxiliary inductor current at operating points P3a and P4. [Figure 18] (a) is a circuit diagram showing a third embodiment of the switching power supply device of the present invention, and (b) is a graph showing the control content of the first, second, and fourth operating modes performed by the switching control circuit. [Figure 19] (a) is a graph of the output voltage-output current plotted at the operating point of the switching power supply of the third embodiment, and (b) is a time chart showing the operation of the operating point P4. [Figure 20] This is a circuit diagram showing a fourth embodiment of the switching power supply device of the present invention. [Figure 21] This is a circuit diagram showing a modified example of the switching power supply device of the fourth embodiment. [Figure 22]This figure shows modified examples of the ZVS auxiliary circuits of the switching power supply devices of the first, second, and third embodiments, with equivalent circuit (a) showing the current flow immediately after the first main switching element is turned ON, and equivalent circuit (b) showing the current flow thereafter. [Figure 23] This is a circuit diagram showing a configuration in which an output current limiting circuit for overcurrent protection is added to the switching power supply device of the first embodiment. [Figure 24] Figure 23 shows a circuit diagram and graph (a) illustrating the first specific example of the output current limiting circuit, and a circuit diagram and graph (b) illustrating the second specific example. [Figure 25] The circuit diagram (a) and the graph (b) show a configuration in which an output current limiting circuit for constant current control is added to the switching power supply device of the second embodiment. [Figure 26] (a) is a circuit diagram showing a conventional switching power supply, and (b) is a graph showing the control of the first and second operating modes performed by the switching control circuit. [Figure 27] (a) is a graph of output voltage-output current plotted at the operating point of a conventional switching power supply, and (b) is an equivalent circuit for explaining its operation. [Figure 28] This is a time chart showing the operation of the operating point P1 of a conventional switching power supply. [Figure 29] (a) shows the equivalent circuit and time chart of a conventional switching power supply, which shows the L component energy release operation that occurs after the first ON time ton1 has ended, and (b) shows the equivalent circuit and time chart of the free oscillation operation. [Figure 30] (a) shows the equivalent circuit and time chart of a conventional switching power supply, which shows the L component energy release operation that occurs after the second ON time ton2 has finished, and (b) shows the equivalent circuit and time chart of the free oscillation operation. [Figure 31] This is a time chart showing the operation of the operating point P2 of a conventional switching power supply. [Figure 32] This is a time chart showing the operation of the operating point P3 of a conventional switching power supply. [Figure 33](a) shows the equivalent circuit and time chart of a conventional switching power supply that takes place after the first on-time ton1 has finished, and (b) shows the equivalent circuit and time chart of a conventional switching power supply that takes place after the second on-time ton2 has finished. [Modes for carrying out the invention]

[0067] <<<1. Switching power supply device 38 of the first embodiment>>> Hereinafter, a switching power supply device 38, which is a first embodiment of the switching power supply device of the present invention, will be described with reference to Figures 1 to 5. Here, components similar to those in the conventional switching power supply device 10 are denoted by the same reference numerals and their description is omitted. <<1.1 Configuration of Switching Power Supply Unit 38>> The switching power supply unit 38 is a conventional switching power supply unit 10 with the addition of a ZVS auxiliary circuit 40, and the other configurations and power section constant settings are almost the same as those of the switching power supply unit 10. The ZVS auxiliary circuit 40 is a circuit that assists the first main switching element 14(1) in performing ZVS.

[0068] As shown in Figure 1, the ZVS auxiliary circuit 40 consists of a series circuit of an auxiliary diode 42, an auxiliary inductor 44, and an auxiliary DC power supply 46, and is connected in parallel to both ends of the second main switching element 14(2). The orientation of the auxiliary diode 42 and the polarity of the auxiliary DC power supply 46 are set so that when the second main switching element 14(2) is ON, the auxiliary diode 42 is ON, and the current flowing out from the auxiliary DC power supply 46 flows to the second main switching element 14(2) and the auxiliary diode 42 and is supplied to the auxiliary inductor 44. In addition, the voltage value of the auxiliary DC power supply 46 is set to a low value so that both ends of the auxiliary diode 42 are reverse-biased when the second main switching element 14(2) is OFF.

[0069] The series order of the auxiliary diode 42, auxiliary inductor 44, and auxiliary DC power supply 46 is arbitrary. For example, as shown in Figure 1, if the auxiliary DC power supply 46 is placed on the input terminal 12b side, a DC power supply that supplies an operating DC voltage (approximately 10-20V) to the switching control circuit 36 ​​can be used as the auxiliary DC power supply 46.

[0070] The switching control circuit 36 ​​is the same as the switching control circuit 36 ​​of the first switching power supply unit 10, and the method for determining the first and second on times ton1 and ton2 shown in Figure 2(a) is almost the same as that described earlier in Figure 26(a). <<1.2 Regarding the operating point and the signs of current-voltage waveforms used for operation explanation>> Figure 2(b) is a graph with the output voltage Vo and target voltage Vor on the vertical axis and the output current Io on the horizontal axis. It shows the first operating region DR1 where the switching power supply 38 controls the first operating mode, and the second operating region DR2 where it controls the second operating mode. Furthermore, the second region DR2 is divided into region DR2(1) and region DR2(2). The positions of the boundary lines between the first operating region DR1 and the second operating region DR2, and between region DR(1) and region DR(2), are almost the same as those shown in Figure 27(a) of the switching power supply 10 described earlier.

[0071] The operating points P1, P2, and P3 of the switching power supply 38 are plotted in the graph in Figure 2(b). The positions of the operating points P1 to P3 are the same as the operating points P1 to P3 of the switching power supply 10 shown in Figure 27(a). The hatched region in Figure 2(b), that is, the region where the output voltage Vo is relatively lower or the output current Io is relatively smaller than region DR2(2), will be described later when explaining the switching power supply 54 of the second embodiment.

[0072] According to the switching power supply 38, the problem with the switching power supply 10, namely the problem that the first switching element 14(1) cannot be zero-switched in the second operating region DR2(2) [operating point P3], is solved, and both the first and second main switching elements 14(1) and 14(2) can be zero-switched in the first operating region DR1 and the second operating regions DR2(1) and DR(2) as shown in Figure 2(b).

[0073] This point will be explained in detail later, but before that, the signs of the voltages and currents in each part of the time chart in Figure 4 will be explained based on the equivalent circuit in Figure 3. Figure 3 is almost the same as Figure 27(b) of the switching power supply 10, and the current flowing through the newly added ZVS auxiliary circuit 40 is denoted as I(D), and the "positive direction" of the current I(D) is defined as the direction in which the auxiliary diode 42 becomes the forward current.

[0074] <<1.3 Operation of the Switching Power Supply 38>> <1.3.1 Operation of operating point P3> First, the operation of the switching power supply unit 38 at its operating point P3 will be explained based on Figures 4 and 5. The time chart in Figure 4 is a schematic diagram of the voltage and current waveforms of each part at the operating point P3, and the switching period Tsw(P3), which is one period of switching, is divided into periods t11 to t17.

[0075] The first main switching element 14(1) turns on during the period t11 when the drive pulse Vgs(1) is high, and the length of this period t11 is the first on-time ton1. The second main switching element 14(2) turns on during the periods t14 and t15 when the drive pulse Vgs(2) is high, and the sum of these two periods is the second on-time ton2. During periods t12 and t13, both drive pulses Vgs(1) and Vgs(2) are low, and the sum of these two periods is the dead time Td. Similarly, during periods t16 and t17, both drive pulses Vgs(1) and Vgs(2) are low, and the sum of these two periods is the dead time Td.

[0076] The operating waveforms of the switching power supply 38 shown in Figure 4 are almost identical to those of the conventional switching power supply 10 shown in Figure 32, in terms of the currents Ifx, Ify, I(Lm), I(Lk), Id(1), and V(Ck).

[0077] The main difference is that, due to the inclusion of the ZVS auxiliary circuit 40, a triangular wave current I(D) is generated during the second on-time ton2 period (periods t14, t15). The triangular wave current I(D) flows from the auxiliary DC power supply 46 through the auxiliary inductor 44, auxiliary diode 42, and second main switching element 14(2). When the voltage Vgs(2) turns low and period 15 ends, the rise stops, and it decreases to 0A during the dead time Td period (periods t16, t17).

[0078] Looking at the ZVS of the second main switching element 14(2), at the operating point P3, similar to the conventional switching power supply 10, the L component energy release operation shown in Figure 29(a) and the free vibration operation shown in Figure 33(a) are performed in combination. Therefore, similar to the conventional switching power supply 10, the free vibration operation is performed strongly, allowing the voltage Vds(2) to drop to 0V before the start of period t14 (before the end of dead time Td), and the second main switching element 14(2) is able to perform ZVS.

[0079] Looking at the ZVS of the first main switching element 14(1), at the operating point P3, similar to the conventional switching power supply 10, the L-component energy release operation shown in Figure 30(a) and the free vibration operation shown in Figure 33(b) are performed in combination. The L-component energy release operation shown in Figure 30(a) is performed by the resonant inductor 24 and, as shown in Figure 5(a), is performed regardless of the presence or absence of the ZVS auxiliary circuit 40. Similarly, the free vibration operation shown in Figure 33(b) is also performed regardless of the presence or absence of the ZVS auxiliary circuit 40. Both this L-component energy release operation and the free vibration operation are weak.

[0080] However, since the switching power supply 38 has an additional ZVS auxiliary circuit 40, the L component energy release operation by the auxiliary inductor 44 is performed in parallel, as shown in Figure 5(b). In other words, the energy stored in the auxiliary inductor 44 at the end of the second on-time ton2 becomes an energy source, and generates an energy release current at the start of the dead time Td, which flows as currents Ic1 and Ic2 to the parasitic capacitors Coss(1) and Coss(2), discharging Coss(1) and charging Coss(2). The value of this energy discharge current is approximately equal to the value of the current I(D) at the end of the second on-time ton2, so increasing |I(D)| at the end of the second on-time ton2 can increase the rate at which the voltage Vds(1) decreases.

[0081] The switching power supply 38 adjusts the L value of the auxiliary inductor 44 and the voltage value of the auxiliary DC power supply 44 as appropriate to increase the peak value of the current I(D) above a certain level. As a result, the L component energy release operation is strongly performed, which allows the voltage Vds(1) to drop to 0V before the start of period t11 (before the end of dead time Td), enabling the second main switching element 14(2) to perform ZVS.

[0082] In this way, the operating point P3 of the switching power supply unit 38 allows the first and second switching elements 14(1) and 14(2) to be easily subjected to ZVS (Zero-Variable Switching). Furthermore, since much of the energy output by the ZVS auxiliary circuit 40 is ultimately transmitted to the load 32 and becomes output power, power loss within the device is kept to a minimum.

[0083] <1.3.2 Operation of operating points P1 and P2> As explained earlier, even in a conventional switching power supply 10 that does not have a ZVS auxiliary circuit 40, the first and second switching elements 14(1) and 14(2) can be ZVS at operating points P1 and P2. In the switching power supply 38, since a ZVS auxiliary circuit 40 is added, the first and second switching elements 14(1) and 14(2) can be ZVS even more easily.

[0084] Furthermore, the switching period Tsw is longer at operating points P1 and P2 than at operating point P3, resulting in a longer second on-time ton2 during which the current I(D) rises steadily. Consequently, the peak and RMS values ​​of the current I(D) become considerably larger, potentially making it impossible to ignore the power loss in the auxiliary inductor 44 of the ZVS auxiliary circuit 40, as well as the power loss in circuit elements along the path from the energy output by the ZVS auxiliary circuit 40 to the load 32. Additionally, the current stress on the second main switching element 14(2) may also become impossible to ignore. Therefore, if these power losses and current stresses become problematic, it is advisable to disconnect a specific part of the ZVS auxiliary circuit 44 with a switch when the first on-time ton1, determined based on the control signal Vs, is ton1 ≥ Tk (when operating in the first control region DR1 [the region where operating points P1 and P2 are located]). Alternatively, the voltage of the auxiliary DC power supply 46 can be forcibly reduced to OV. In either case, the ZVS auxiliary circuit 40 prevents current I(D) from flowing, thus preventing unnecessary power loss and current stress.

[0085] <<1.4 Summary of Switching Power Supply Unit 38>> As explained above, with the switching power supply 38, by simply adding the ZVS auxiliary circuit 40 to the conventional switching power supply 10, it is possible to obtain a current-resonant type switching power supply with a wide variable range for the output voltage Vo and the ability to achieve ZVS for the first and second main switching elements 14(1) and 14(2).

[0086] <<1.5 Specific Examples of Error Amplifier Circuit 34 and Switching Control Circuit 34>> Here, specific examples of the error amplification circuit 34 and the switching control circuit 36, namely the error amplification circuit 34x and the switching control circuit 36x, will be explained based on Figures 6 to 8. <1.5.1 Circuit Configuration> First, the configurations of the error amplification circuit 34x and the switching control circuit 36x will be explained based on Figure 6.

[0087] The error amplification circuit 34x is an inverting amplifier circuit using an operational amplifier, which obtains a detected voltage Vok obtained by resistively dividing the output voltage Vo, and outputs a control signal Vs which is a voltage obtained by amplifying the difference between the reference voltage Vref corresponding to the target voltage Vor and the detected voltage Vok. Therefore, the control signal Vs changes in a direction to decrease when Vok > Vref (i.e., when Vo > Vor), and changes in a direction to increase when Vok < Vref (i.e., when Vo < Vor). The reference voltage Vref can be varied from the outside, and thereby the output voltage Vo can be set to a desired value.

[0088] The switching control circuit 36x is composed of an RS-FF100, a first PWM circuit 102, a second control voltage generation circuit 104, a second PWM circuit 106, dead time generation circuits 108(1), 108(2) and a high-side driver 36a.

[0089] The RS-FF100 is a set-reset flip-flop having an S terminal to which an S signal is input, an R terminal to which an R signal is input, a Q terminal for outputting a Q signal, and a Q-bar terminal for outputting a Q-bar signal. Here, a type combining two NOR circuits is used.

[0090] The first PWM circuit 102 is composed of a sawtooth wave voltage generation circuit 110(1) and a comparator 112(1). The sawtooth wave voltage generation circuit 110(1) includes a timer capacitor Ct(1) whose positive terminal is connected to the non-inverting input terminal of the comparator 112(1) and whose negative terminal is connected to the ground, a constant current source J(1) for supplying current to the positive terminal of the timer capacitor Ct(1), and a discharge diode Dr(1) whose anode is connected to the positive terminal of the timer capacitor Ct(1) and whose cathode is connected to the Q-bar terminal of the RS-FF100, and generates a sawtooth wave voltage Vosc(1) across the timer capacitor Ct(1). The comparator 112(1) has the sawtooth wave voltage Vosc(1) input to its non-inverting input terminal, the control signal Vs input to its inverting input terminal, and its output terminal outputs an S signal toward the S terminal of the RS-FF100. Details will be described in the subsequent operation explanation.

[0091] The second control voltage generation circuit 104 is composed of a plurality of resistors, two transistors, and a constant voltage source Vcc, and generates a second control signal Vs2 at the midpoint of voltage dividing resistors R1 and R2 (both ends of the lower resistor R2) connected in parallel to the constant voltage source Vcc. Specifically, when the control signal Vs satisfies Vs ≧ R2 / (R1 + R2)·Vcc, Vs2 = Vs, and the second control signal Vs2 is changed according to the control signal Vs. When the control signal Vs satisfies Vs < R2 / (R1 + R2)·Vcc, the second control signal Vs2 is set to Vs2 = R2 / (R1 + R2)·Vcc = Vs2(min), and the second control signal Vs2 is fixed to Vs2(min) regardless of the control signal Vs.

[0092] The second PWM circuit 106 is composed of a sawtooth wave voltage generation circuit 110(2) and a comparator 112(2). The sawtooth wave voltage generation circuit 110(2) is composed of a timer capacitor Ct(2) with its positive terminal connected to the non-inverting input terminal of the comparator 112(2) and its negative terminal connected to the ground, a constant current source J(2) that supplies current to the positive terminal of the timer capacitor Ct(2), and a discharge diode Dr(2) with its anode connected to the positive terminal of the timer capacitor Ct(2) and its cathode connected to the Q terminal of the RS-FF100, and generates a sawtooth wave voltage Vosc(2) across the timer capacitor Ct(2). The comparator 112(2) has the sawtooth wave voltage Vosc(2) input to its non-inverting input terminal, the second control signal Vs2 input to its inverting input terminal, and its output terminal outputs an R signal toward the R terminal of the RS-FF100. Details will be described in the subsequent operation explanation.

[0093] The dead time generation circuit 108(1) generates a drive pulse Vgs(1) based on the Q-bar signal output by the RS-FF100 and inputs it between the gate and source of the first main switching element 14(1) via the high-side driver 36a. Comparing the Q-bar signal and the drive pulse Vgs(1), both are square waves that alternate between high and low in almost the same phase. However, this dead time generation circuit 108(1) incorporates an RC integrator, so after the Q-bar signal changes from low to high, the drive pulse Vgs(1) changes from low to high with a slight delay. This delay time is the dead time Td, which will be described later. The dead time Td can be adjusted by changing the time constant of the RC integrator.

[0094] The dead time generation circuit 108(2) generates a drive pulse Vgs(2) based on the Q signal output by the RS-FF100 and inputs it between the gate and source of the second main switching element 14(2). Comparing the Q signal and the drive pulse Vgs(2), both are square waves that alternate between high and low in almost the same phase. However, this dead time generation circuit 108(2) incorporates an RC integrator, so after the Q signal switches from low to high, the drive pulse Vgs(2) switches from low to high with a slight delay, and this delay time becomes the dead time Td. The dead time Td can be adjusted by changing the time constant of the RC integrator. <1.5.2 Operation of the Switching Control Circuit 36x (First Operating Mode)> Next, the operation of the switching control circuit 36x when it controls the first operating mode [operation of operating points P1 and P2] will be explained based on Figure 7. The control of the first operating mode is performed when the second control signal Vs2 is Vs2 = Vs, and Vs2 changes according to the control signal Vs.

[0095] During the first ON time ton1 [ton1≧Tk], the drive pulse Vgs(1) is high, and the Q-bar signal is held high, so the discharge diode Dr(1) is held off, current from the constant current source J(1) flows into the timer capacitor Ct(1), and the sawtooth voltage Vosc(1) rises steadily.

[0096] Also, during the period of the first on-time ton1, since the drive pulse Vgs(2) goes low and the Q signal is also held low, the discharge diode Dr(2) is kept on, the current of the constant current source J(2) is bypassed by the discharge diode Dr(2), and the timer capacitor Ct(2) is not charged. Therefore, the sawtooth wave voltage Vosc(2) is held at a predetermined low voltage (the forward voltage VF of the discharge diode Dr(2)).

[0097] After that, when the sawtooth wave voltage Vosc(1) rises and slightly exceeds the control signal Vs, the S signal output by the comparator 112(1) turns from low to high. Then, the Q-bar signal output by the RE-FF100 turns from high to low, the drive pulse Vgs(1) instantaneously turns from high to low, and the period of the first on-time ton1 ends. Also, when the Q-bar signal turns low, the discharge diode Dr(1) turns on and the timer capacitor Ct(1) is instantaneously discharged, and the sawtooth wave voltage Vosc(1) drops to a predetermined low voltage (the forward voltage VF of the discharge diode Dr(1)). Also, since Vosc(1) < Vs, the S signal output by the comparator 112(1) also instantaneously returns to low.

[0098] Also, when the S signal output by the comparator 112(1) turns from low to high, the Q signal output by the RE-FF100 turns from low to high, the discharge capacitor Dr(2) turns off, the current of the constant current source J(2) flows through the timer capacitor Ct(2) and charges it, and the sawtooth wave voltage Vosc(2) starts to rise with a right shoulder. Then, when the dead time Td elapses, the drive pulse Vgs(2) turns from low to high, and the period of the second on-time ton2 starts.

[0099] The period of the second on-time ton2 is the period during which the drive pulse Vgs(2) is high, and since the Q signal is held high, the discharge diode Dr(2) is held off, and the current of the constant current source J(2) flows into the timer capacitor Ct(2), causing the sawtooth voltage Vosc(2) to rise with a right shoulder. The slope when the sawtooth voltage Vosc(2) rises is set to be the same as the slope when the sawtooth voltage Vosc(1) rises.

[0100] Also, during the period of the second on-time ton2, the drive pulse Vgs(1) becomes low and the Q bar signal is also held low, so the discharge diode Dr(1) is held on, and the current of the constant current source J(1) is bypassed by the discharge diode Dr(1), and the timer capacitor Ct(1) is not charged. Therefore, the sawtooth voltage Vosc(1) is held at a predetermined low voltage (the forward voltage VF of the discharge diode Dr(1)).

[0101] After that, when the sawtooth voltage Vosc(2) rises and slightly exceeds the second control signal Vs2 (= Vs), the R signal output by the comparator 112(2) changes from low to high. Then, the Q signal output by the RE-FF100 changes from high to low, and the drive pulse Vgs(2) instantaneously changes from high to low, ending the period of the second on-time ton2 [ton2 = ton1]. Also, when the Q signal changes to low, the discharge diode Dr(2) turns on and the timer capacitor Ct(2) is instantaneously discharged, causing the sawtooth voltage Vosc(2) to drop to a predetermined low voltage (the forward voltage VF of the discharge diode Dr(1)). Also, since Vosc(2) < Vs2, the R signal output by the comparator 112(2) also instantaneously returns to low.

[0102] Also, when the R signal output by the comparator 112(2) transitions from low to high, the Q-bar signal output by the RE-FF 100 transitions from low to high, the discharge capacitor Dr(1) turns off, the current of the constant current source J(1) flows through the timer capacitor Ct(1) and charges it, and the sawtooth wave voltage Vosc(1) starts to rise with a right shoulder upward. Then, when the dead time Td elapses, the drive pulse Vgs(1) transitions from low to high, and the period of the first on-time ton1 starts. Then, the periods of ton1, Td, ton2, and Td described above are repeated in order.

[0103] As described above, the switching control circuit 36x determines the first on-time ton1 based on the control signal Vs output by the error amplification circuit 34x, and when ton1≧Tk, determines the second on-time ton2 as ton2 = ton1 (control of the first operation mode).

[0104] <1.5.3 Operation of the Switching Control Circuit 36x (Second Operation Mode)> Next, the operation [operation at the operating point P3] when the switching control circuit 36x performs control in the second operation mode will be described based on FIG. 8. In the control of the second operation mode, the first on-time ton1 is determined according to the control voltage Vs in the same manner as the control of the first operation mode, and varies within a range shorter than the reference time Tk. On the other hand, when ton1<Tk, the second control signal Vs2 is fixed at Vs2 = Vs2(min), so the second on-time ton2 cannot be shorter than the reference period Tk and is fixed at the reference time Tk.

[0105] Thus, the switching control circuit 36x determines the first on-time ton1 based on the control signal Vs output by the error amplification circuit 34x, and when ton1<Tk, determines the second on-time ton2 as ton2 = Tk (control of the second operation mode).

[0106] <1.5.4 Summary of the Error Amplification Circuit 34x and the Switching Control Circuit 36x> As explained above, the error amplification circuit 34x and the switching control circuit 36x enable reliable control of the first and second operating modes, and allow for smooth switching between the two operating modes. While this type of complex control could also be implemented using digital control with an expensive digital processor, it can be achieved very inexpensively using the switching control circuit 36x.

[0107] <<1.6 Modified Version of Switching Power Supply 38>> Next, several modified versions of the switching power supply 38 will be described in order. The first modified version, the switching power supply 48, as shown in Figure 9(a), divides the resonant capacitor 22 into two resonant capacitors 22(1) and 22(2). Resonant capacitor 22(1) is connected between one end of the input winding 18 and the input terminal 12a, and resonant capacitor 22(2) is connected between one end of the input winding 18 and the input terminal 12b, with each set to the same capacitance value. Even when the resonant capacitor 22 is divided into the two 22(1) and 22(2) described above, the same operation as the switching power supply 38 is performed and the same effect is obtained.

[0108] The second modified switching power supply unit 50, as shown in Figure 9(b), has the position where the resonant capacitor 22 is connected changed to be between one end of the input winding 18 and the input terminal 12a. Even with the position where the resonant capacitor 22 is connected changed as shown in Figure 9(b), the switching power supply unit 38 operates in almost the same way and achieves the same effect.

[0109] As shown in Fig. 10, the switching power supply device 52 of the third modification example arranges the first main switching element 14(1) on the low side and the second main switching element 14(2) on the high side. Even in this connection case, when in the second operation mode [ton1 < ton2], it is difficult to achieve ZVS for the first main switching element 14(1). Therefore, similar to the switching power supply device 38, the ZVS auxiliary circuit 40 is connected in parallel across both ends of the second main switching element 14(2). As a result, an operation substantially similar to that of the switching power supply device 38 is performed, and a similar effect is obtained.

[0110] <<<2. Second Switching Power Supply Device 54 of the Second Embodiment>>> Next, the switching power supply device 54, which is the second embodiment of the switching power supply device of the present invention, will be described based on Figs. 11 to 16. Here, the same components as those of the switching power supply device 38 in the above embodiment are denoted by the same reference numerals, and the description thereof is omitted. <<2.1 Problems of the Switching Power Supply Device 38>> Before explaining the switching power supply device 54 shown in Figs. 11(a) and (b), first, the problems that may occur in the above switching power supply device 38 will be explained.

[0111] The switching power supply device 38 may not be able to operate in the hatched area in the graph of Fig. 2(b). That is, when the output current Io is small, even if the target voltage Vor is adjusted to the operating point in the hatched area, the output voltage Vo may not drop to the target voltage Vor.

[0112] Fig. 12(a) shows the operation of the operating point P3a on the boundary line between the hatched area and the second operating area DR2(2) of the switching power supply device 38. At the operating point P3a, ton1 = 0 and ton2 = Tk, the first main switching element 14(1) is fixed off, and only the second main switching element 14(2) turns on and off.

[0113] During the period when the second main switching element 14(2) is ON, as shown in Figure 12(a), a triangular wave current I(D) flows from the auxiliary DC power supply 46 through the path of the auxiliary inductor 44, auxiliary diode 42, and second main switching element 14(2), and energy is stored in the auxiliary inductor 44. Then, when the second ON time ton2 ends and the second main switching element 14(2) turns OFF, as shown in Figure 12(b), the energy in the auxiliary inductor 44 becomes the energy source and an energy release current is generated, a portion of which flows to the output side and increases the output voltage Vo. In addition, although not shown in Figure 12(b), after the stored energy in the auxiliary inductor 44 is transferred to the parasitic capacitors of the first and second switching elements 14(1) and 14(2), that energy is further transferred to the output side and increases the output voltage Vo.

[0114] Thus, even if the first on-time ton1 becomes zero, the energy stored in the auxiliary inductor 44 during the second on-time ton2 is released to the output side after the second on-time ton2 ends. Therefore, when the output current Io is relatively large, the smoothing capacitor 30 is discharged, allowing the output voltage Vo to be reduced to almost OV. However, when the output current Io is small, it may not be possible to reduce the output voltage Vo to 0V.

[0115] <<2.2 Configuration of Switching Power Supply Unit 54>> The switching power supply unit 54 of the second embodiment is an improved version of the switching power supply unit 38, solving the problem that "when the output current Io is small, it may not be possible to reduce the output voltage Vo to 0V." The change in configuration is that, as shown in Figure 11(a), the switching control circuit 36 ​​of the switching power supply unit 38 is replaced with a new switching control circuit 56.

[0116] The method for determining the first and second on-times ton1 and ton2 by the switching control circuit 56 will be described. As shown in the graph of FIG. 11(b), the switching control circuit 56 sets the first on-time ton1 to a value approximately proportional to the control voltage Vs. When the control voltage Vs increases, the first on-time ton1 is lengthened, and when the control voltage Vs decreases, it is shortened. When the determined first on-time ton1 satisfies ton1≧Tk (when the control voltage Vs is higher than a predetermined value), the control of the first operation mode is performed, where the second on-time is set to ton2 = ton1. Also, when the determined first on-time ton1 becomes ton1 < Tk (when the control voltage Vs is lower than a predetermined value), the control of the second operation mode is performed, where the second on-time ton2 is set to ton2 > ton1. Tk is a reference time set in advance in the switching control circuit 56 [Tk > 0]. So far, it is the same as the above-described switching control circuit 36.

[0117] In the case of the switching control circuit 56, further, when the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0, the control of the second operation mode is stopped, and the switching period is maintained at a constant length. When the control signal Vs changes in the direction of shortening the first on-time ton1, the second time ton2 is shortened according to the control signal Vs. When the control signal Vs changes in the direction of lengthening the first on-time ton1, the control of the third operation mode is performed, where the second on-time ton2 is lengthened according to the control signal Vs.

[0118] <<2.3 Operating Point for Operation Explanation>> FIG. 13(a) is a graph with the output voltage Vo and the target voltage Vor on the vertical axis and the output current Io on the horizontal axis. The first operation area DR1 where the switching power supply device 54 performs the control of the first operation mode and the areas for performing the control of the second operation mode are represented as the second operation areas DR2(1) and DR(2). So far, it is the same as FIG. 2(b) of the switching power supply device 38. However, in the case of the switching power supply device 56, the area where the output voltage Vo is relatively low or the output current Io is relatively small compared to the area DR2(2) becomes the third operation area DR3 where the control of the third operation mode is performed.

[0119] The positions of the operating points P1 to P3 plotted on the graph in Figure 13(a) are the same as the operating points P1 to P3 of the switching power supply 38 shown in Figure 2(b). The operating point P4 added in the third operating region DR3 is the point where the target voltage Vor is set to approximately 10%, and the output voltage Vo=Vor and output current Io≈30% are output. The switching power supply 54 can operate without problems even in the third operating region DR3 where the switching power supply 38 cannot operate.

[0120] <<2.4 Operation of the Switching Power Supply 54>> <2.4.1 Operation of operating points P1, P2, and P3> The operation of the operating points P1, P2 (first operating mode) and operating point P3 (second operating mode) of the switching power supply 54 is the same as that of the switching power supply 38. <2.4.2 Operation of operating points P3a and P4> The operation of the switching power supply unit 54 at its operating point P3a is the same as that of the switching power supply unit 38, and the operation shown in the equivalent circuits of Figures 12(a) and (b) is performed. The switching period at the operating point P3 is Tsw(P3a), and the current I(D) of the auxiliary inductor 44 has a waveform as shown in the upper part of Figure 13(b).

[0121] When the target voltage Vor is set to an operating point P4 lower than the operating point P3a, the third operating mode is controlled, ton1 becomes 0, and the switching period Tsw(P4) = Tsw(P3a) is fixed. Then, the control voltage Vs becomes even lower (changing to a method that shortens the first on time ton1), so the second on time ton2 is shortened according to the control voltage Vs, and the peak value of the current I(D) in the auxiliary inductor 44 becomes lower, as shown in the lower waveform of Figure 13(b). In other words, the energy stored in the auxiliary inductor 44 during the second on time ton2 period becomes smaller, so the energy released to the output side after the second on time ton2 ends gradually decreases. Therefore, it becomes possible to control the output voltage Vo based on the control voltage Vs.

[0122] <<2.5 Summary of Switching Power Supply Unit 54>> As explained above, the switching power supply 54 can achieve the same effects as the switching power supply 38, and furthermore, it can operate without problems even in the third operating region DR3 where the switching power supply 38 cannot operate. In other words, it is possible to reduce the output voltage Vo to almost 0V regardless of the magnitude of the output current Io.

[0123] <<2.6 Specific Examples of Error Amplifier Circuit 34 and Switching Control Circuit 56>> Here, specific examples of the error amplification circuit 34 and the switching control circuit 56, namely the error amplification circuit 34x and the switching control circuit 56x, will be explained based on Figures 14 to 16.

[0124] <2.6.1 Circuit Configuration> The error amplifier circuit 34x in Figure 14 is the same as the error amplifier circuit 34x shown in Figure 6, so its explanation will be omitted. Here, we will explain the switching control circuit 56x.

[0125] The switching control circuit 56x, like the switching control circuit 36x shown in Figure 6, includes an RS-FF100, a first PWM circuit 102, a second control voltage generation circuit 104, a second PWM circuit 106, dead time generation circuits 108(1) and 108(2), and a high-side driver 36a. Furthermore, a new configuration of a third PWM circuit 114 and an AND gate 116 has been added. The following explanation will focus on the changes from the switching control circuit 36x.

[0126] The third PWM circuit 114 consists of a sawtooth voltage generation circuit 110(3) and a comparator 112(3). The sawtooth voltage generation circuit 110(3) has a timer capacitor Ct(3) whose positive terminal is connected to the inverting input terminal of comparator 112(3) and whose negative terminal is connected to ground, a discharge resistor Rh connected in parallel across both ends of the timer capacitor Ct(3), and an NPN transistor TR connected between a constant voltage source Vcc and the positive terminal of the timer capacitor Ct(3), with the base of the transistor TR connected to the output terminal of comparator 112(2) via a resistor.

[0127] Transistor TR conducts when the R signal output by comparator 112(2) becomes high, supplying current to the positive terminal of timer capacitor Ct(3) and instantaneously raising the voltage at the positive terminal of timer capacitor Ct(3) to a value equal to the forward voltage VF of the discharge diodes Dr(1) and Dr(2). Subsequently, the R signal quickly turns low, and transistor TR turns off, causing timer capacitor Ct(3) to discharge slowly through the discharge resistor Rh. This operation generates a sawtooth wave voltage Vosc(3) with a peak value of VF across the terminals of timer capacitor Ct(3). Further details will be described in the operation description below.

[0128] Comparator 112(3) receives a sawtooth wave voltage Vosc(3) at its inverting input terminal and a control signal Vs at its non-inverting input terminal, and outputs a voltage Vx to one of the input terminals of AND gate 116 at its output terminal.

[0129] The connection point between the Q terminal of the RF-FF100 and the discharge diode Dr(2) is disconnected from the dead time generation circuit 108(2), and an AND gate 116 is inserted between them. The AND gate 116 receives a voltage Vx at one input terminal and a Q signal at the other input terminal, and outputs a voltage Vy from its output terminal toward the dead time generation circuit 108(2). Voltage Vy is high when both the Q signal and voltage Vx are high, and low otherwise.

[0130] <2.6.2 Operation of the Switching Control Circuit 56x (Second Operation Mode)> Next, the operation when the switching control circuit 56x performs control in the second operation mode [operation at the operating point P3] will be described based on FIG. 15. When performing control in the second operation mode, the control voltage Vs is in the range of VF < Vs < Vs2(min). Therefore, the situation of Vosc(3) < Vs always occurs, the voltage Vx output by the third PWM circuit 114 is always high, and the voltage Vy output by the AND gate 116 becomes the same as the Q signal. This is because there is no difference from the case where the Q signal is directly input to the dead time generation circuit 108(2), so it is substantially the same operation as the operation of the switching control circuit 36x described above (the operation shown in FIG. 8).

[0131] Therefore, similar to the switching control circuit 36x, the switching control circuit 56x determines the first on-time ton1 based on the control signal Vs output by the error amplification circuit 34x, and when the first on-time ton1 is less than the reference time Tk, the second on-time ton2 is set to ton2 > ton1 (ton2 = Tk) (control in the second operation mode).

[0132] Although the description of the operation when the switching control circuit 56x performs control in the first operation mode is omitted, the situation of Vosc(3) < Vs always occurs even when performing control in the first operation mode. Therefore, for the same reason as above, the operation when the switching control circuit 56x performs control in the first operation mode is substantially the same as the operation of the switching control circuit 36x (the operation shown in FIG. 7).

[0133] <2.6.3 Operation of the Switching Control Circuit 56x (Third Operation Mode)> Next, the operation when the switching control circuit 56x performs control in the third operation mode [operation at the operating point P4] will be described based on FIG. 16.

[0134] When controlling the third operation mode, the control voltage Vs is in the range of Vs < VF. Therefore, the situation of Vs < Vosc(1) always occurs, the S signal output by the first PWM circuit 102 is always high, and the Q-bar signal is always low. Accordingly, the drive pulse Vgs(1) is always low and the period of the first on-time ton1 does not occur (ton1 = 0), and the first switching element 14(1) is fixed to be off.

[0135] On the other hand, since the second control voltage Vs2 is fixed at Vs2 = Vs2(min) > VF, the second PWM circuit 106 outputs an impulse-like R signal at a constant period. When the R signal is high, the Q signal becomes low, and when the R signal is low, the Q signal becomes high. This is the same as in the second control mode, and the time when the Q signal is high is fixed at approximately the sum of the reference time Tk and the dead time Td.

[0136] Also, since the control voltage Vs is in the range of Vs < VF, the waveform of the control voltage Vs intersects the waveform of the sawtooth wave voltage Vosc(3). The voltage Vx output by the third PWM circuit 114 becomes low during the period of Vosc( > Vs and becomes high during the period of Vosc(3) < Vs. During the period when the voltage Vx is low, the voltage Vy output by the AND gate 116 becomes low regardless of the Q signal. On the other hand, during the period when the voltage Vx is high, since the Q signal is high, the voltage Vy also becomes high.

[0137] [[ID=ID=ID=12]]Therefore, as shown in FIG. 16, the first on-time ton1 becomes zero, and the switching period Tsw(P4) is fixed at the time when the Q signal is high (≒ Tk + Td). The second on-time ton2 is the time obtained by subtracting the dead time Td from the time when the voltage Vx is high (< Tk) and changes according to the control voltage Vs.

[0138] <id= Thus, the switching control circuit 56x determines the first on-time ton1 based on the control signal Vs output by the error amplification circuit 34x. When the determined first on-time ton1 becomes zero or less, it stops the second operating mode and controls the third operating mode. That is, it maintains the switching period Tsw(P4) at a constant length (≒Tk+Td) and changes the second on-time ton2 according to the control signal Vs.

[0139] <2.6.4 Summary of Switching Control Circuit 56x> As explained above, the switching control circuit 56x can reliably control the first, second, and third operating modes, and can also smoothly switch between the three operating modes. Furthermore, while this type of complex control could also be implemented using digital control with an expensive digital processor, it can be achieved at a very low cost using the switching control circuit 56x.

[0140] <<2.7 Modified Switching Control Circuit 54>> As shown in Figure 13(b), when the switching control circuit 56 shortens the second on-time ton2 in the third operating mode, the switching power supply 54 fixes the end point of the second on-time ton2 and changes the start point. Alternatively, as shown in the switching power supply 54-1 in Figure 17, the operation may be to fix the start point of the second on-time ton2 and change the end point, and almost the same effect can be obtained.

[0141] <<<3. Switching power supply device 58 of the third embodiment>>> Next, a switching power supply device 58, which is a third embodiment of the switching power supply device of the present invention, will be described with reference to Figures 18 and 19. Here, components similar to those in the switching power supply device 38 of the above embodiment are denoted by the same reference numerals and their description is omitted. <<3.1 Configuration of Switching Power Supply Unit 58>> The switching power supply device 58 of the third embodiment improves on the switching power supply device 38 and solves the problem of "there may be cases where the output voltage Vo cannot be reduced to 0V when the output current Io is small" by a method different from that of the switching power supply difference 54. The point of configuration change is that, as shown in Fig. 18(a), the switching control circuit 36 of the switching power supply device 38 is replaced with a new switching control circuit 60.

[0142] To explain the method for determining the first and second on-times ton1 and ton2 by the switching control circuit 60, as shown in the graph of Fig. 18(b), the switching control circuit 60 sets the first on-time ton1 to a value approximately proportional to the control voltage Vs. When the control voltage Vs increases, the first on-time ton1 is lengthened, and when the control voltage Vs decreases, it is shortened. When the determined first on-time ton1 satisfies ton1≧Tk (when the control voltage Vs is higher than a predetermined value), control in the first operation mode where the second on-time is set to ton2 = ton1 is performed. Also, when the determined first on-time ton1 becomes ton1 < Tk (when the control voltage Vs is lower than a predetermined value), control in the second operation mode where the second on-time ton2 is set to ton2 > ton1 is performed. Tk is a reference time set in advance in the switching control circuit 56 [Tk > 0]. So far, it is the same as the above-mentioned switching control circuit 36.

[0143] In the case of the switching control circuit 60, further, when the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0, the control in the second operation mode is stopped, and control in the fourth operation mode where the second on-time ton2 is forcibly held at ton2 = 0 is performed.

[0144] <<3.2 Operating Points for Operation Explanation>> Figure 19(a) is a graph with the output voltage Vo and target voltage Vor on the vertical axis and the output current Io on the horizontal axis. The first operating region DR1 is where the switching power supply 58 controls the first operating mode, and the areas where it controls the second operating mode are represented as the second operating regions DR2(1) and DR(2). Up to this point, it is almost the same as Figure 2(b) for the switching power supply 38. However, in the case of the switching power supply 58, the region where the output voltage Vo is relatively lower or the output current Io is relatively smaller than region DR2(2) becomes the third operating region DR3, where the second and fourth operating modes are alternately repeated.

[0145] The positions of the operating points P1 to P3 plotted on the graph in Figure 19(a) are the same as the operating points P1 to P3 of the switching power supply 38 shown in Figure 2(b). The operating point P4 added in the third operating region DR3 is the point where the target voltage Vor is set to approximately 10%, and the output voltage Vo=Vor and output current Io≈30% are output. The switching power supply 58 can operate without problems even in the third operating region DR3 where the switching power supply 38 cannot operate.

[0146] <<3.3 Operation of the Switching Power Supply 58>> <3.3.1 Operation of operating points P1, P2, and P3> The operation of the operating points P1, P2 (first operating mode) and operating point P3 (second operating mode) of the switching power supply 58 is the same as that of the switching power supply 38.

[0147] <3.3.2 Operation of operating point P4> When the user reduces the target voltage Vor from the value at operating point P3 to the value at operating point P4, Vo temporarily becomes greater than Vor, so the error amplification circuit 34 performs an action to reduce the control voltage Vs. Then, when the control voltage Vs falls below the threshold Vs(th), the switching control circuit 60 starts the fourth operating mode control, which sets ton1=ton2=0. Since the fourth operating mode control is a control that stops the switching operation, during the period of the fourth operating mode control, the output voltage Vo decreases toward 0V, as shown in Figure 19(b).

[0148] Subsequently, when the output voltage Vo falls below the target voltage Vor, the error amplification circuit 34 increases the control voltage Vs. Then, when the control voltage Vs rises above the threshold Vs(th), the switching control circuit 60 performs control in a second operating mode, setting ton1>0 and ton2=Tk, and the switching operation begins. If the control in the second operating mode continues, the operation described above, "when the output current Io is small, it may not be possible to lower the output voltage Vo to 0V" (shown in Figure 12), occurs, and during the period of control in the second operating mode, the output voltage Vo rises towards a value higher than the target voltage Vor, as shown in Figure 19(b).

[0149] Subsequently, when the output voltage Vo becomes higher than the target voltage Vor, the error amplification circuit 34 lowers the control voltage Vs. Then, when the control voltage Vs becomes lower than the threshold voltage Vs(th), the switching control circuit 60 starts controlling the fourth operating mode, setting ton1=ton2=0.

[0150] Thus, at the operating point P4, the second and fourth operating modes are alternately repeated, i.e., burst operation is performed, and the average value of the output voltage Vo becomes approximately equal to the target voltage Vor. In burst operation, the two operating modes are smoothly switched by the error amplifier circuit 34 changing the control voltage Vs, and the duration of each operating mode is also automatically controlled, making it possible to control the output voltage Vo based on the control voltage Vs.

[0151] <<3.4 Summary of Switching Power Supply Unit 58>> As explained above, the switching power supply 58 can achieve the same effects as the switching power supply 38, and furthermore, it can operate without problems even in the third operating region DR3 where the switching power supply 38 cannot operate. In other words, it is possible to reduce the output voltage Vo to almost 0V regardless of the magnitude of the output current Io.

[0152] <<<4. Switching power supply 62 of the fourth embodiment>>> Next, a switching power supply 62, which is a fourth embodiment of the switching power supply of the present invention, will be described with reference to Figure 20. Here, components similar to those in the switching power supply 38 described above are denoted by the same reference numerals and their description is omitted. <<4.1 Configuration of Switching Power Supply Unit 62>> The switching power supply unit 62 is a so-called current-resonant full-bridge switching power supply unit. The switching power supply 62 has a first arm 14-1 connected between a pair of input terminals 12a and 12b to which the input voltage Vi is applied. The first arm 14-1 consists of a series circuit of first and second main switching elements 14(1) and 14(2), with the first main switching element 14(1) positioned on the high-side and the second main switching element 14(2) positioned on the low-side.

[0153] Furthermore, a second arm 14-2 is connected in parallel to the first arm 14-1. The second arm 14-2 consists of a series circuit of the third and fourth main switching elements 14(3) and 14(4), with the third main switching element 14(3) positioned on the high-side and the fourth main switching element 14(4) positioned on the low-side. The four main switching elements 14(1) to 14(4) can be, for example, N-channel MOS-type FETs.

[0154] The main transformer 16 has an input winding 18 and an output winding 20 that are magnetically coupled to each other. The input winding 18 is connected between the midpoint of the first arm 14-1 and the midpoint of the second arm 14-2. A resonant capacitor 22 and a resonant inductor 24 are inserted in series with the input winding 18, and the input winding 18, resonant capacitor 22, and resonant inductor 24 form a current-resonant type resonant circuit. When the main switching elements 14(1) to 14(4) are switched on or off, an AC voltage is applied across the input winding 18, and an AC voltage is generated in the output winding 20.

[0155] An output rectifier and smoothing circuit 26, similar to the output rectifier and smoothing circuit 26 of the switching power supply 38, is connected to the output winding 20. This circuit rectifies and smooths the AC voltage generated in the output winding 20 to generate an output voltage Vo, and outputs the output voltage Vo and output current Io to the load 32.

[0156] Furthermore, the switching power supply unit 62 is equipped with an error amplification circuit 34 and a switching control circuit 36. The error amplification circuit 34 has the same configuration as the error amplification circuit 34 of the switching power supply unit 38 (inverting amplifier circuit), and amplifies the difference between the output voltage Vo and the target voltage Vor, and outputs a control signal Vs that increases or decreases the output voltage Vo in the direction of bringing it closer to the target voltage Vor.

[0157] The switching control circuit 36 ​​is a circuit that controls the on / off switching of the first to fourth main switching elements 14(1) to 14(4). The switching control circuit 36 ​​is a circuit that operates in much the same way as the switching control circuit 36 ​​of the switching power supply unit 38. Upon receiving a control signal Vs, it determines the first on-time ton1 [ton1≧0], which is the on-time of the first main switching element 14(1), and the second on-time ton2 [ton2≧0], which is the on-time of the second main switching element 14(2). Then, while providing a predetermined dead time Td so that the first and second main switching elements 14(1) and 14(2) do not turn on simultaneously, it turns the first and second main switching elements 14(1) and 14(2) on and off complementaryly. The method for determining the first and second on-times ton1 and ton2 is as shown in the graph in Figure 2(a), and the first and second operating modes are controlled according to the control voltage Vs.

[0158] Furthermore, the switching control circuit 36 ​​is configured to turn the third main switching element 14(3) on and off in the same phase as the second main switching element 14(2), and to turn the fourth main switching element 14(4) on and off in the same phase as the first main switching element 14(1). Therefore, the fourth main switching element 14(4) turns on during the first on-time period ton1, and the third main switching element 14(3) turns on during the second on-time period ton2.

[0159] Furthermore, the switching power supply unit 62 is equipped with two sets of ZVS auxiliary circuits similar to the ZVS auxiliary circuit 40 of the switching power supply unit 38. The first ZVS auxiliary circuit 40-1 consists of a series circuit of a first auxiliary diode 42-1, a first auxiliary inductor 44-1, and a first auxiliary DC power supply 46-1, and is connected in parallel to both ends of the second main switching element 14(2). The orientation of the first auxiliary diode 42-1 and the polarity of the first auxiliary DC power supply 46-1 are set so that when the second main switching element 14(2) is on, the first auxiliary diode 42-1 is on, and the current flowing out from the first auxiliary DC power supply 46-1 flows to the second main switching element 14(2) and the first auxiliary diode 42-1 and is supplied to the first auxiliary inductor 44-1. Furthermore, the voltage value of the first auxiliary DC power supply 46-1 is set to a low value such that the terminals of the first auxiliary diode 42-1 are reverse-biased when the second main switching element 14(2) is off.

[0160] The second ZVS auxiliary circuit 40-2 consists of a series circuit of a second auxiliary diode 42-2, a second auxiliary inductor 44-2, and a second auxiliary DC power supply 46-2, and is connected in parallel to both ends of the third main switching element 14(3). The orientation of the second auxiliary diode 42-2 and the polarity of the second auxiliary DC power supply 46-2 are set so that when the third main switching element 14(3) is ON, the second auxiliary diode 42-2 is ON, and the current flowing out of the second auxiliary DC power supply 46-2 flows to the third main switching element 14(3) and the second auxiliary diode 42-2 and is supplied to the second auxiliary inductor 44-2. In addition, the voltage value of the second auxiliary DC power supply 46-2 is set to a low value so that when the third main switching element 14(3) is OFF, both ends of the second auxiliary diode 42-2 are reverse-biased.

[0161] <<4.2 Operation of the Switching Power Supply 62>> The operation of the switching power supply 62 is almost the same as that of the switching power supply 38 described above. That is, as shown in the graph in Figure 2(b), at operating points P1 and P2, the first operating mode is controlled, and the first to fourth main switching elements 14(1) to 14(4) can be easily controlled via ZVS. At operating point P3, the second operating mode is controlled, and the second and third main switching elements 14(2) and 14(3) can be easily controlled via ZVS. The first main switching element 14(1) can be controlled via ZVS when the first ZVS auxiliary circuit 40-1 is activated, and the fourth main switching element 14(4) can also be controlled via ZVS when the second ZVS auxiliary circuit 40-2 is activated.

[0162] <<4.3 Summary of Switching Power Supply Unit 62>> As explained above, the switching power supply 62, like the switching power supply 38 described above, provides a current-resonant type switching power supply with a wide variable range for the output voltage Vo and the ability to achieve ZVS for the first to fourth main switching elements 14(1) to 14(2).

[0163] However, since the switching power supply 62 uses a switching control circuit 36 ​​that controls only the first and second operating modes, it may not be able to operate in the hunting region shown in the graph of Figure 2(b). However, by replacing the switching control circuit 36 ​​with a switching control circuit 56 that controls the first, second and third operating modes, or with a switching control circuit 60 that controls the first, second and fourth operating modes, it becomes possible to operate in the hunting region as well.

[0164] <<4.4 Modified Examples of Switching Power Supply 62>> Next, a switching power supply device 64, which is a modified example of the switching power supply device 62, will be described based on FIG. 21. In the switching power supply device 64, in the first arm 14-1, the first main switching element 14(1) is arranged on the low side and the second main switching element 14(2) is arranged on the high side. In the second arm 14-2, the third main switching element 14(3) is arranged on the low side and the fourth main switching element 14(4) is arranged on the high side. Even in this connection mode, when in the second operation mode [when ton1 < ton2], it is difficult to achieve ZVS for the first and fourth main switching elements 14(1) and 14(4). Therefore, similar to the switching power supply device 62, the first ZVS auxiliary circuit 40-1 is connected in parallel across the second main switching element 14(2), and the second ZVS auxiliary circuit 40-2 is connected in parallel across the third main switching element 14(3). Thereby, an operation substantially similar to that of the switching power supply device 62 is performed, and a similar effect is obtained.

[0165] <<<5. Other Embodiments and Modified Examples>>> Note that the switching power supply device of the present invention is not limited to the above embodiments and modified examples. Other embodiments and modified examples will be described below. <<5.1 Modified Example of ZVS Auxiliary Circuit>> The ZVS auxiliary circuit 40 included in the current resonance type half-bridge switching power supply devices 38, 54, and 58 can be changed to the configuration of the ZVS auxiliary circuit 66 shown in FIGS. 22(a) and (b). In the ZVS auxiliary circuit 66, the auxiliary inductor 44 and the auxiliary DC power supply 46 are adjacent to each other in a series position, and a protection diode 68 is connected in parallel across both ends of the series circuit of the auxiliary inductor 44 and the auxiliary DC power supply 46. The direction of the protection diode 68 is set to prevent the current flowing out from the positive electrode of the auxiliary DC power supply 46 from flowing through the protection diode 68.

[0166] When the first main switching element 14(1) turns on after the second main switching element 14(2) has turned off, a large surge current Ia may flow through the junction capacitance of the auxiliary diode 42, or a large surge current Ia may flow through the auxiliary diode 42 during recovery operation, and the energy of the surge current Ia may be stored in the auxiliary inductor 44.

[0167] In that case, if there were no protective diode 68, when the surge current Ia stops, a large surge voltage would be generated across the auxiliary diode 42 due to the back electromotive force of the auxiliary inductor 44, potentially damaging the auxiliary diode 42. However, by providing the protective diode 68, as shown in Figure 22(b), a path is created for the auxiliary inductor 44 to release energy (a path through which current Ib flows), preventing a large surge voltage from being generated across the auxiliary diode 42 and ensuring reliable protection of the auxiliary diode 42.

[0168] Note that the switching power supply unit 52 in Figure 10 has a slightly different configuration from the switching power supplies units 38, 54, and 58, as the ZVS auxiliary circuit 40 is connected to both ends of the main switching element 14(2) on the high-side. However, since similar problems may occur with a similar mechanism, in that case, the auxiliary diode 42 can be reliably protected by changing the configuration of the ZVS auxiliary circuit 40 to that of the ZVS auxiliary circuit 66. Similarly, in the current-resonant full-bridge switching power supplies units 62 and 64, the first and second auxiliary diodes 42-1 and 42-2 can be reliably protected by changing the configuration of the first and second ZVS auxiliary circuits 40-1 and 40-2 to that of the ZVS auxiliary circuit 66, respectively.

[0169] <<5.2 Output current limiting circuit for overcurrent protection>> Figure 23 shows a configuration in which an output current limiting circuit 70 is added to the switching power supply 38 to provide overcurrent protection. The output current limiting circuit 70 consists of an output current detection circuit 72 and a Vs forced variable circuit 74. The output current detection circuit 72 detects the output current Io or a current corresponding to the output current Io and outputs an output current detection value V(Io). When the output current detection value V(Io) is about to exceed the threshold voltage V(Ith), the Vs forced variable circuit 74 forcibly changes the control voltage Vs output by the error amplifier circuit 34 in the direction that shortens the first on time ton1, thereby lowering the output voltage Vo so that the output current detection value V(Io) does not exceed the threshold voltage V(Ith). Incidentally, since the error amplifier circuit 34 of the switching power supply 38 is configured as an inverting amplifier circuit, the Vs forced variable circuit 74 is configured to forcibly lower the control voltage Vs output by the error amplifier circuit 34 when the output current detection value V(Io) is about to exceed the threshold voltage V(Ith). Diode 74a is part of the Vs forced variable circuit 74 and is a reverse current blocking diode that prevents the operation of the Vs forced variable circuit 74 from acting in a direction that increases the control voltage Vs.

[0170] Figure 24(a) shows a specific example of the output current limiting circuit 70. The output current detection circuit 72 consists of a current detection resistor inserted in the path through which the output current Io flows, and outputs the output current detection value V(Io) generated at the current detection resistor. The Vs forced variable circuit 74 consists of an integrating circuit using a so-called operational amplifier, and the threshold voltage V(th), generated by dividing the DC voltage Vcc with fixed resistors Ra and Rb, is input to the non-inverting input terminal of the operational amplifier. When the average value of the output current detection value V(Io) generated at the inverting input terminal is about to exceed the threshold voltage V(th), the control voltage Vs is forcibly reduced via the diode 74a, and the output voltage Vo is reduced. Since the output current limiting circuit 70 is for overcurrent protection, the threshold voltage V(Ith) is set so that the drooping current value Ith is greater than 100% (rated). By using this output current limiting circuit 70, the drooping current value Ith when the output voltage Vo decreases is kept at a nearly constant value, resulting in a so-called inverted L-type drooping characteristic.

[0171] Figure 24(b) shows an output current limiting circuit 76, which is a modified version of the output current limiting circuit 70. The output current limiting circuit 76 is the output current limiting circuit 70 shown in Figure 24(a) with the addition of a resistor 74b. The resistor 74b is connected between the non-inverting input terminal and the output terminal of the operational amplifier of the Vs forced variable circuit 74, and when the control voltage Vs decreases, it works to lower the threshold voltage V(Ith) accordingly. In the case of the output current limiting circuit 76, the drooping current value Ith gradually decreases when the output voltage Vo decreases, resulting in a so-called "F" shaped drooping characteristic.

[0172] As shown in Figure 2(b), the switching power supply 38 reduces the output voltage Vo to 0V in accordance with the control voltage Vs when the output current Io is in the range of approximately 40% or more. Therefore, by adding output current limiting circuits 70 or 76, a clean inverted L-shaped or V-shaped droop characteristic can be obtained, and an automatic reset overcurrent protection function that is easy for the user to use can be easily realized. Other embodiments and modified switching power supply devices can also easily realize an automatic reset overcurrent protection function by adding the above-mentioned output current limiting circuits 70 or 76.

[0173] <<5.3 Output current limiting circuit for constant current control>> Figure 25 shows a configuration in which an output current limiting circuit 70-1 is added to the switching power supply 54 so that it can also be used as a constant current power supply. The output current limiting circuit 70-1 has basically the same configuration as the output current limiting circuit 70, but the difference is that the threshold voltage V(Ith) can be externally varied by the user. For example, in the specific circuit shown in Figure 24(a), the threshold voltage V(Ith) can be externally varied by adding a variable resistor to the voltage divider circuit consisting of fixed resistors Ra and Rb.

[0174] As shown in Figure 13(a), the switching power supply 54 exhibits a voltage drop of Vo to 0V depending on the control voltage when the output current Io is in the range of 0% to 100%. Therefore, by adding the output current limiting circuit 70, a clean inverted L-shaped droop characteristic can be obtained, and it can be used as a constant current power supply. However, in the case of the switching power supply 38, in the range where the output current Io is approximately 40% or less, the output voltage Vo may not drop to Vo=0V even if the control voltage Vs is changed, so it can be used as a constant current power supply if the droop current Ith > 40%. Similarly, other embodiments and modified switching power supply devices can also be easily made usable as constant current power supplies by adding the above-described output current limiting circuit 70-1.

[0175] <<5.4 Switching control circuit enabling operation in the third operating region, DR3>> The above-mentioned switching power supply 38 may not be able to operate in the hatched area in Figure 2(b), that is, in the third operating area DR3 in Figure 13(a). To solve this problem, the switching power supply 54 enables operation in the third operating region DR3. When the first on-time ton1 becomes ton1=0, the switching control circuit stops controlling the second operating mode and starts controlling the third operating mode. In other words, in the third operating region DR3, the second on-time ton2 is changed according to the control voltage Vs, and this second on-time ton2 is used to control the peak value of the current I(D), thereby making it possible to control the output voltage Vo (see Figure 13(b)).

[0176] As another method, when the first on-time ton1 becomes ton1 = 0, a method of controlling the voltage value of the auxiliary DC power supply 46 of the ZVS auxiliary circuit 40 while the switching control circuit continues to control the second operation mode can be considered. Specifically, when the first on-time ton1 becomes ton1 = 0, when the control signal Vs changes in the direction of shortening the first on-time ton1, the voltage value of the auxiliary DC power supply 46 is decreased according to the control signal Vs, and when the control signal Vs changes in the direction of lengthening the first on-time ton1, the voltage value of the auxiliary DC power supply 46 is increased according to the control signal Vs. This method changes the voltage value of the auxiliary DC power supply 46 according to the control voltage Vs in the third operation region DR3, uses the voltage value of this auxiliary DC power supply 46 to control the peak value of the current I(D), and thereby enables the output voltage Vo to be controlled, obtaining the same effect as the switching power supply device 54. Also, the concept of controlling the voltage value of the auxiliary DC power supply 46 can be applied to the full-bridge type switching power supply device 62 (the type using the switching control circuit 36), obtaining the same effect.

[0177] <<5.5 Others>> When the above switching control circuits 36, 56, 60 perform the control of the second operation mode, under the condition of ton1 < ton2, the second on-time ton2 is set to a constant length regardless of the control voltage Vs. However, under the condition of ton1 < ton2, the second on-time ton2 may be lengthened or shortened according to the control voltage Vs. Also, the above error amplification circuit 34 is an inverting amplification circuit, but it can be changed to a non-inverting amplification circuit configuration. In this case, when Vo > Vor, the control voltage Vs rises, and when Vo < Vor, the control voltage Vs falls, so it should be noted that the configuration of the switching control circuit also needs to be changed accordingly.

[0178] Furthermore, in each switching control circuit of the above embodiment, the dead time Td, which prevents the first and second switching elements from turning on simultaneously, is fixed to a certain length. However, if the dead time Td is longer than necessary, the time during which current flows through the parasitic diode of the main switching element increases, which may lead to increased conduction losses in the main switching element and a slight decrease in efficiency. Therefore, it is preferable to configure the dead time Td to be automatically shortened as much as possible within the range where ZVS is possible, while still specifying a maximum length.

[0179] Furthermore, the switching power supply device of the present invention is not limited to DC input power supplies, but can also be applied to AC input power supplies. In the case of an AC input power supply, a rectifier circuit is provided to rectify the AC voltage output by the input power supply, and the output terminal of the rectifier circuit is connected to the input terminals 12a and 12b. This allows the DC voltage output from the rectifier circuit to be considered as the input voltage Vi. [Explanation of Symbols]

[0180] 10, 38, 48, 50, 52, 54, 54-1, 58, 62, 64 Switching power supply 12a, 12b Input terminals 14,14-1 First Arm 14-2 Second Arm 14(1) First main switching element 14(2) Second main switching element 14(3) Third main switching element 14(4) Fourth main switching element 16 Main transformer 18 Input winding 20 Output windings 22 Resonant capacitor 24 Resonant Inductors 26 Output rectifier and smoothing circuit 32 load 34,34x Error Amplifier Circuit 36, 36x, 56, 56x, 60 switching control circuit 40 ZVS auxiliary circuit 40-1 First ZVS auxiliary circuit 40-2 Second ZVS auxiliary circuit 42 Auxiliary diodes 42-1 First auxiliary diode 42-2 Second auxiliary diode 44 Auxiliary Inductor 44-1 First auxiliary inductor 44-2 Second auxiliary inductor 46 Auxiliary DC power supply 46-1 First Auxiliary DC Power Supply 46-2 Second auxiliary DC power supply 68 Protective diodes Io Output Current Td Dead Time Tk reference time ton1 First ON time ton2 Second On Time Tsw switching period Vi Input Voltage Vo Output Voltage Vs control signal

Claims

1. A current-resonant half-bridge switching power supply device comprising: a first arm consisting of a series circuit of first and second main switching elements connected between a pair of input terminals to which an input voltage Vi is applied; a main transformer having an input winding and an output winding; a current-resonant resonant circuit connected to the first arm and consisting of the input winding, a resonant capacitor, and a resonant inductor; and an output rectifier and smoothing circuit that rectifies and smooths the AC voltage generated in the output winding by the switching on and off of the first and second main switching elements to generate a predetermined output voltage Vo, and supplies the output voltage Vo and output current Io to the load, A circuit that amplifies the difference between the output voltage Vo and the target voltage Vor, and outputs a control signal Vs that increases or decreases the output voltage Vo in a direction that brings it closer to the target voltage Vor, wherein when Vo > Vor, the control signal Vs is changed in a direction that shortens the first on-time ton1 [ton1≧0] which is the on-time of the first main switching element, and when Vo < Vor, the control signal Vs is changed in a direction that lengthens the first on-time ton1, A switching control circuit for controlling the on / off state of the first and second main switching elements, which receives the control signal Vs to determine the first on time ton1 and the second on time ton2 [ton2≧0] which is the on time of the second main switching element, and provides a predetermined dead time Td so that the first and second switching elements do not turn on simultaneously, thereby causing the first and second main switching elements to turn on and off complementaryly. The second main switching element is connected in parallel to both ends of the ZVS auxiliary circuit, which consists of a series circuit of an auxiliary diode, an auxiliary inductor, and an auxiliary DC power supply. The orientation of the auxiliary diode and the polarity of the auxiliary DC power supply are set such that the auxiliary diode turns on when the second main switching element is on, and the current flowing out from the auxiliary DC power supply flows to the second main switching element and the auxiliary diode and is supplied to the auxiliary inductor; and the voltage value of the auxiliary DC power supply is set to a low value such that the terminals of the auxiliary diode are reverse-biased when the second main switching element is off. A switching power supply device characterized in that the switching control circuit has a reference time Tk [Tk > 0] set in advance, and when the first on time ton1 determined based on the control signal Vs is ton1 ≥ Tk, the switching control circuit controls a first operating mode to set the second on time ton2 to ton2 = ton1, and when the first on time ton1 determined based on the control signal Vs becomes ton1 < Tk, it controls a second operating mode to set the second on time ton2 to ton2 > ton1.

2. The switching power supply device according to claim 1, wherein the ZVS auxiliary circuit has the auxiliary inductor and the auxiliary DC power supply adjacent to each other in series, a protective diode is connected in parallel to both ends of the series circuit of the auxiliary inductor and the auxiliary DC power supply, and the orientation of the protective diode is set to prevent current flowing out from the positive terminal of the auxiliary DC power supply from flowing to the protective diode.

3. The switching power supply device according to claim 1 or 2, wherein the switching control circuit blocks a specific part of the ZVS auxiliary circuit to prevent current from flowing to the ZVS auxiliary circuit, or forces the voltage value of the auxiliary DC power supply to zero, when the first on-time ton1 determined based on the control signal Vs is ton1 ≥ Tk.

4. The switching power supply device according to claim 1, wherein the switching control circuit stops controlling the second operating mode when the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0, thereby maintaining the switching period at a constant length, and controls a third operating mode in which the second on-time ton2 is shortened in accordance with the control signal Vs when the control signal Vs changes in a direction that shortens the first on-time ton1, and the second on-time ton2 is lengthened in accordance with the control signal Vs when the control signal Vs changes in a direction that lengthens the first on-time ton1.

5. The switching power supply device according to claim 1, wherein the switching control circuit stops controlling the second operating mode and controls a fourth operating mode to forcibly hold the second on-time ton2 at ton2 = 0 when the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0.

6. The switching power supply device according to claim 1, wherein the switching control circuit, when the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0, continues to control the second operating mode, and when the control signal Vs changes in a direction that shortens the first on-time ton1, it lowers the voltage value of the auxiliary DC power supply in accordance with the control signal Vs, and when the control signal Vs changes in a direction that lengthens the first on-time ton1, it raises the voltage value of the auxiliary DC power supply in accordance with the control signal Vs.

7. A current-resonant full-bridge switching power supply device comprising: a first arm consisting of a series circuit of first and second main switching elements connected between a pair of input terminals to which an input voltage Vi is applied; a second arm consisting of a series circuit of third and fourth main switching elements connected in parallel to the first arm; a main transformer having an input winding and an output winding; a current-resonant resonant circuit connected between the first arm and the second arm and consisting of the input winding, a resonant capacitor, and a resonant inductor; and an output rectifier and smoothing circuit that rectifies and smooths the AC voltage generated in the output winding by the on / off switching of the first to fourth main switching elements to generate a predetermined output voltage Vo, and supplies the output voltage Vo and output current Io to the load, A circuit that amplifies the difference between the output voltage Vo and the target voltage Vor, and outputs a control signal Vs that increases or decreases the output voltage Vo in a direction that brings it closer to the target voltage Vor, wherein when Vo > Vor, the control signal Vs is changed in a direction that shortens the first on-time ton1 [ton1≧0] which is the on-time of the first main switching element, and when Vo < Vor, the control signal Vs is changed in a direction that lengthens the first on-time ton1, A switching control circuit for controlling the on / off states of the first to fourth main switching elements, wherein, upon receiving the control signal Vs, it determines the on-time ton1 of the first main switching element and the on-time ton2 [ton2≧0] of the second main switching element, provides a predetermined dead time Td so that the first and second switching elements do not turn on simultaneously, and turns the first and second main switching elements on and off complementaryly, turns the third main switching element on and off in the same phase as the second main switching element, and turns the fourth main switching element on and off in the same phase as the first main switching element, A first ZVS auxiliary circuit, consisting of a series circuit of a first auxiliary diode, a first auxiliary inductor, and a first auxiliary DC power supply, is connected in parallel to both ends of the second main switching element. The third main switching element is connected in parallel to both ends of the third main switching element and comprises a second ZVS auxiliary circuit consisting of a series circuit of a second auxiliary diode, a second auxiliary inductor, and a second auxiliary DC power supply. The orientation of the first auxiliary diode and the polarity of the first auxiliary DC power supply are set such that when the second main switching element is on, the first auxiliary diode is on, and the current flowing out from the first auxiliary DC power supply flows to the second main switching element and the first auxiliary diode and is supplied to the first auxiliary inductor; and the voltage value of the first auxiliary DC power supply is set to a low value such that when the second main switching element is off, both ends of the first auxiliary diode are reverse-biased. The orientation of the second auxiliary diode and the polarity of the second auxiliary DC power supply are set such that when the third main switching element is on, the second auxiliary diode turns on, and the current flowing out from the second auxiliary DC power supply flows to the third main switching element and the second auxiliary diode and is supplied to the second auxiliary inductor; and the voltage value of the second auxiliary DC power supply is set to a low value such that when the third main switching element is off, both ends of the second auxiliary diode are reverse-biased. A switching power supply device characterized in that the switching control circuit has a reference time Tk [Tk > 0] set in advance, and the switching control circuit controls a first operating mode in which the second on time ton2 is set to ton2 = ton1 when the first on time ton1 determined based on the control signal Vs is ton1 ≥ Tk, and controls a second mode in which the second on time ton2 is set to ton2 > ton1 when the first on time ton1 determined based on the control signal Vs becomes ton1 < Tk.

8. The first ZVS auxiliary circuit is configured such that the first auxiliary inductor and the first auxiliary DC power supply are adjacent to each other in series, and a first protective diode is connected in parallel to both ends of the series circuit of the first auxiliary inductor and the first auxiliary DC power supply, and the orientation of the first protective diode is set to prevent current flowing out from the positive terminal of the first auxiliary DC power supply from flowing to the first protective diode. The switching power supply device according to claim 7, wherein the second ZVS auxiliary circuit has the second auxiliary inductor and the second auxiliary DC power supply adjacent to each other in series, a second protective diode is connected in parallel to both ends of the series circuit of the second auxiliary inductor and the second auxiliary DC power supply, and the orientation of the second protective diode is set to prevent current flowing out from the positive terminal of the second auxiliary DC power supply from flowing to the second protective diode.

9. The switching control circuit, when the first on-time ton1 determined based on the control signal Vs is ton1 ≥ Tk, shuts off a specific part of the first ZVS auxiliary circuit to prevent current from flowing to the first ZVS auxiliary circuit, or forcibly sets the voltage value of the first auxiliary DC power supply to zero, and also shuts off a specific part of the second ZVS auxiliary circuit to prevent current from flowing to the second ZVS auxiliary circuit, or forcibly sets the voltage value of the second auxiliary DC power supply to zero, according to claim 7 or 8.

10. The switching power supply device according to claim 7, wherein the switching control circuit stops controlling the second operating mode when the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0, thereby maintaining the switching period at a constant length, and controls a third operating mode in which the second time ton2 is shortened in accordance with the control signal Vs when the control signal Vs changes in a direction that shortens the first on-time ton1, and the second time ton2 is lengthened in accordance with the control signal Vs when the control signal Vs changes in a direction that lengthens the first on-time ton1.

11. The switching power supply device according to claim 7, wherein the switching control circuit stops controlling the second operating mode and controls a fourth operating mode in which the second on time ton2 is forcibly held at ton2 = 0 when the first on time ton1 determined based on the control signal Vs becomes ton1 = 0.

12. The switching power supply device according to claim 7, wherein when the first on-time ton1 determined based on the control signal Vs becomes ton1 = 0, the switching control circuit continues to control the second operating mode, and when the control signal Vs changes in a direction that shortens the first on-time ton1, it lowers the voltage values ​​of the first and second auxiliary DC power supplies in accordance with the control signal Vs, and when the control signal Vs changes in a direction that lengthens the first on-time ton1, it raises the voltage values ​​of the first and second auxiliary DC power supplies in accordance with the control signal Vs.

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