Output voltage generation method using HSC converter

Abstract

To provide an HSC converter which includes a transformer without a primary winding and supplies an output voltage that is one quarter of an input voltage.SOLUTION: An HSC voltage conversion module includes an input voltage, a transformer without a primary winding and including secondary windings magnetically coupled by a magnetic core, first and second switch bridges connected to the transformer and to the input voltage, and an output capacitor that is connected to a single node of the transformer and that supplies an output voltage that is one quarter of the input voltage.SELECTED DRAWING: Figure 2

Description

[Technical Field]

[0001] The present invention relates to DC-DC converters, and more particularly to hybrid switched-capacitor (HSC) converters that provide an output voltage that is one-quarter of the input voltage. [Background technology]

[0002] It is known to use a two-stage conversion where the first stage is an LLC resonant converter and the second stage is a point-of-load (POL) converter. The LLC resonant converter is used as an intermediate bus converter with zero-voltage switching (ZVS) on the primary or high-voltage (HV) side and zero-current switching (ZCS) on the secondary or low-voltage (LV) side. If isolation is not required, the first stage may be an unregulated zero-voltage switching switched-capacitor converter.

[0003] To reduce the converter's footprint, high switching frequency operation is required in the first and second stages. However, high frequency operation, especially in the first stage (i.e., 1 MHz to 2 MHz), results in higher losses (i.e., switching losses, gate drive losses, and conduction losses). In some applications, a high step-down ratio in the first stage may be desirable to reduce losses, including switching losses and conduction losses. In a 48V power delivery application, the first stage has an 8:1 step-down ratio, providing a 6V intermediate bus. Reducing the 48V input voltage to a 6V intermediate bus voltage enables higher power density while maintaining high efficiency. Switched-capacitor (SC) converters offer high conversion ratios, but SC converters are complex and bulky due to the floating driver requirements and the number of switches and ceramic capacitors required. LLC converters with a center-tapped rectifier offer high conversion ratios with reduced complexity, but non-isolated topologies are preferred because they increase power density when isolation is not required. For low output voltage applications, an LLC converter with a center-tapped rectifier can be used, but this topology has suboptimal copper utilization because the secondary winding is conducting for half the switching cycle.

[0004] For high step-down ratios, it is known to use hybrid switched capacitor (HSC) converters, which overcome the limitations of switched capacitor and LLC converters. HSC converters combine the advantages of switched capacitor converters with the high step-down ratio capability of transformer-based converters. By transferring energy through capacitors and magnetic devices, efficiency and power density can be significantly improved. Summary of the Invention [Problem to be solved by the invention]

[0005] However, known HSC converters cannot provide an output voltage that is one-fourth of the input voltage. For applications where the output voltage is one-fourth of the input voltage, a different topology is required, such as the LLC topology. In LLC converters, the current in the secondary winding is half-wave rectified, resulting in large current peaks and losses. [Means for solving the problem]

[0006] To solve the above problems, an exemplary embodiment of the present invention provides an HSC converter that includes a transformer without a primary winding and provides an output voltage that is one-fourth of the input voltage. According to an exemplary embodiment, an HSC voltage conversion module includes an input voltage, a transformer having no primary winding and including a secondary winding magnetically coupled by a magnetic core, first and second switch bridges connected to the transformer and the input voltage, and an output capacitor connected to a single node of the transformer and providing an output voltage that is one-fourth the input voltage.

[0007] The HSC voltage conversion module may further include a first resonant circuit connected between the first switch bridge and the transformer, and a second resonant circuit connected between the second switch bridge and the transformer, wherein the first resonant circuit may include a first resonant capacitor, and the second resonant circuit may include a second resonant capacitor, and the first resonant circuit and the second resonant circuit may utilize leakage inductance of the transformer.

[0008] The first switch bridge and the second switch bridge may each include a first switch, a second switch, and a third switch connected in series. The secondary winding of the transformer may include a first winding and a second winding. The HSC voltage conversion module may be bidirectional.

[0009] According to an exemplary embodiment of the present invention, a converter includes an input terminal for receiving an input voltage, a first switch bridge connected in parallel across the input voltage, the first switch bridge including a first switch, a second switch, and a third switch connected in series, a first node between the first switch and the second switch, and a third node between the second switch and the third switch, and a second switch bridge including a fourth switch, a fifth switch, and a sixth switch connected in series, a second node between the fourth switch and the fifth switch, and a third node between the fifth switch and the sixth switch. a second switch bridge including a fourth node; an output capacitor; a transformer including a single winding including a first secondary winding and a second secondary winding physically connected to each other to define a secondary winding group and magnetically coupled by a magnetic core; and a single tap connected to the output capacitor; a first resonant circuit connected between the first node and a first end of the single winding; a second resonant circuit connected between the second node and a second end of the single winding opposite the first end of the single winding; and an output terminal connected to the first output capacitor and providing a first output voltage that is one-fourth of the input voltage.

[0010] The first resonant circuit may include a first resonant capacitor, the second resonant circuit may include a second resonant capacitor, and the first resonant circuit and the second resonant circuit may utilize leakage inductance of a transformer.

[0011] The converter may be bidirectional so that power can flow from the input terminals to the output terminals and from the output terminals to the input terminals. These and other features, elements, characteristics, steps, and advantages of the present invention will become more apparent from the following detailed description of exemplary embodiments of the invention, which proceeds with reference to the accompanying drawings. [Brief explanation of the drawings]

[0012] [Figure 1] 1 shows a related art hybrid switched capacitor (HSC) converter. [Figure 2] 1 shows an HSC converter with no primary winding and two connected secondary windings. [Figure 3] 3 shows various current paths in the HSC converter of FIG. 2. [Figure 4A] 3 shows graphs of various waveforms of the HSC converter of FIG. 2. [Figure 4B] 3 shows graphs of various waveforms of the HSC converter of FIG. 2. [Figure 5] A table comparing the characteristics of LLC and HSC converters is shown below. [Figure 6] Graphs of various waveforms of an LLC converter are shown. [Figure 7] Graphs of various waveforms of an HSC converter are shown. [Figure 8] 7 and 8 show tables of various characteristics of the LLC and HSC converters used in FIGS. [Figure 9A] 3 shows a waveform graph illustrating simulation results of the HSC converter of FIG. 2. [Figure 9B] 3 shows a waveform graph illustrating simulation results of the HSC converter of FIG. 2. [Figure 9C] 3 shows a waveform graph illustrating simulation results of the HSC converter of FIG. 2. DETAILED DESCRIPTION OF THE INVENTION

[0013] Figure 1 is a circuit diagram of a related art hybrid switched capacitor (HSC) converter including a single output. The HSC converter of Figure 1 includes an input voltage Vin, a first bridge and a second bridge, a transformer connected between the first bridge and the second bridge, a first resonant circuit connected between the first bridge and the transformer, a second resonant circuit connected between the second bridge and the transformer, a first output capacitor Cout connected to the transformer, and an output terminal Vout connected to the output capacitor Cout.

[0014] The input voltage Vin of the HSC converter of Figure 1 can be any suitable DC voltage source, including, for example, 48 V. The output voltage Vout depends on the number of turns of windings L11-L22, as given by equation (1) below:

[0015] The first and second bridges of the HSC converter of FIG. 1 can be connected in parallel between an input voltage Vin and ground. The first bridge includes switches Q1, Q2, and Q3 connected in series, with a second switch Q2 connected between a first switch Q1 and a third switch Q3. The second bridge includes switches Q4, Q5, and Q6 connected in series, with a fifth switch Q5 connected between a fourth switch Q4 and a sixth switch Q6. A first node can be located between the first switch Q1 and the second switch Q2, a second node can be located between the second switch Q2 and the third switch Q3, a third node can be located between the fourth switch Q4 and the fifth switch Q5, and a fourth node can be located between the fifth switch Q5 and the sixth switch Q6. The switches Q1 through Q6 can be divided into a first switch group including the switches Q1, Q3, and Q5 and a second switch group including Q2, Q4, and Q6. The first and second switches may be controlled in a complementary manner using pulse-width modulators (PWMs) that are 180° phase-shifted. The first and second switches may be controlled with the same fixed duty cycle, which may be close to 50%, to reduce or minimize root-mean-square (RMS) current. The switches Q1-Q6 may be controlled by a controller that provides drive signals to the switches Q1-Q6. Any suitable controller may be used and may be implemented in hardware and / or software. The controller may be configured and / or programmed to provide the functionality described herein.

[0016] The transformer of the HSC converter of FIG. 1 may be a multi-tapped autotransformer including a single winding, including a first winding L12, a second winding L22, a third winding L21, and a fourth winding L11. The first winding L12 is physically connected to the second winding L22, the second winding L22 is physically connected to the third winding L21, and the third winding L21 is physically connected to the fourth winding L11. A first end of the single winding of the transformer (i.e., the end of the first winding L12 that is not connected to the second winding L22) may be connected to a first resonant circuit, and a second end of the single winding of the transformer (i.e., the end of the fourth winding L11 that is not connected to the third winding L21) may be connected to a second resonant circuit. The first winding L12, the second winding L22, the third winding L21, and the fourth winding L11 can be magnetically coupled to the magnetic core.

[0017] The single winding of the transformer of the HSC converter of FIG. 1 may include a first tap between the first winding L12 and the second winding L22, a second tap between the second winding L22 and the third winding L21, and a third tap between the third winding L21 and the fourth winding L11. The first tap may be connected to a fourth node between the fifth switch Q5 and the sixth switch Q6. The second tap may be connected to the output capacitor Cout. The third tap may be connected to a second node between the second switch Q2 and the third switch Q3.

[0018] In the HSC converter of FIG. 1, the first winding group or primary winding group of the transformer may include a winding between the first end of the transformer and the first tap of the transformer (i.e., the first winding L12) and a winding between the second end of the transformer and the third tap of the transformer (i.e., the fourth winding L11). The second winding group or secondary winding group of the transformer may include a winding between the first tap and the second tap of the transformer (i.e., the second winding L22) and a winding between the second tap and the third tap of the transformer (i.e., the third winding L21). Because the windings L11-L22 are physically connected to each other, the primary winding group and the secondary winding group are not isolated from each other. The HSC converter of FIG. 1 is a non-isolated DC-DC converter.

[0019] In the HSC converter of FIG. 1, the first resonant circuit may include a first resonant capacitor Cres1, and the first resonant circuit may be connected between the first winding L12 and a first node between the first switch Q1 and the second switch Q2. In the HSC converter of FIG. 1, the second resonant circuit may include a second resonant capacitor Cres2, and the second resonant circuit may be connected between the fourth winding L11 and a third node between the fourth switch Q4 and the fifth switch Q6. The first resonant circuit and the second resonant circuit may use the leakage inductance of a transformer. Alternatively, the first resonant circuit and the second resonant circuit may use discrete inductors.

[0020] The output terminal Vout of the HSC converter in Figure 1 provides a single output voltage, which may be an unregulated output voltage. The ratio of the input voltage Vin to the output voltage Vout is given by the following equation:

[0021]

number

[0022] Here, Vout is the output voltage at the output terminal Vout, Vin is the voltage of the input voltage Vin, N1 is the number of turns of the first and fourth windings L12 and L11, and N2 is the number of turns of the second and third windings L22 and L21. The output voltage Vout can be adjusted by adjusting the number of turns of the windings L11 to L22. However, no matter how the number of turns of the windings L11 to L22 is adjusted, the output voltage Vout cannot be adjusted to one-fourth of the input voltage.

[0023] Applications where the output voltage is a quarter of the input voltage usually require a different topology, such as the LLC topology, which uses half-wave rectification of the secondary winding current, resulting in high current peaks and losses.

[0024] FIG. 2 shows an HSC converter whose output voltage is one-fourth the input voltage. In the HSC converter of FIG. 2, the autotransformer includes a first winding L22 and a second winding L21, but does not include any primary winding. The autotransformer of FIG. 2 includes a single tap connected to the output capacitor Cout. Compared to the HSC converter of FIG. 1, the HSC converter of FIG. 2 does not include the first winding L12 or the fourth winding L11. The transformer of FIG. 2 includes only two windings (i.e., windings L22 and L21). The transformer of FIG. 2 includes only a secondary winding and does not include any primary winding. By not including a primary winding, copper loss, which is the loss caused by current flowing through the copper foil in the primary winding, is eliminated. The simplified transformer structure allows for more effective use of the space occupied by the primary winding.

[0025] Similar to the HSC converter of FIG. 1, the HSC converter of FIG. 2 includes an input voltage Vin, a first bridge and a second bridge, a transformer connected between the first bridge and the second bridge, a first resonant circuit connected between the first bridge and the transformer, a second resonant circuit connected between the second bridge and the transformer, a first output capacitor Cout connected to the transformer, and an output terminal Vout connected to the output capacitor Cout.

[0026] The first and second bridges of the HSC converter of FIG. 2 can be connected in parallel between an input voltage Vin and ground. The first bridge includes switches Q1, Q2, and Q3 connected in series, with a second switch Q2 connected between a first switch Q1 and a third switch Q3. The second bridge includes switches Q4, Q5, and Q6 connected in series, with a fifth switch Q5 connected between a fourth switch Q4 and a sixth switch Q6. A first node can be located between the first switch Q1 and the second switch Q2, a second node can be located between the second switch Q2 and the third switch Q3, a third node can be located between the fourth switch Q4 and the fifth switch Q5, and a fourth node can be located between the fifth switch Q5 and the sixth switch Q6. The switches Q1 through Q6 can be divided into a first switch group including the switches Q1, Q3, and Q5 and a second switch group including Q2, Q4, and Q6. The first and second groups of switches may be controlled in a complementary manner using pulse width modulators (PWM) with the phases shifted by 180°. The first and second groups of switches may be controlled with the same fixed duty cycle, which may be close to 50%, to reduce or minimize the root mean square (RMS) current. The switches Q1-Q6 may be controlled by a controller that provides drive signals to the switches Q1-Q6. Any suitable controller may be used and may be implemented in hardware and / or software. The controller may be configured and / or programmed to provide the functionality described herein.

[0027] The transformer of the HSC converter of FIG. 2 may be a single-tapped autotransformer including a single winding, including a first winding L22 and a second winding L21. The first winding L22 is physically connected to the second winding L21. A first end of the single winding of the transformer (i.e., the end of the first winding L22 that is not connected to the second winding L21) may be connected to a first resonant circuit, and a second end of the single winding of the transformer (i.e., the end of the second winding L21 that is not connected to the first winding L22) may be connected to a second resonant circuit. The first winding L22 and the second winding L21 may be magnetically coupled to a magnetic core.

[0028] The single winding of the transformer of the HSC converter of Figure 2 may include a single tap between the first winding L22 and the second winding L21, which may be connected to the output capacitor Cout.

[0029] The transformer of the HSC converter of Figure 2 includes a single winding group or secondary winding group. The single winding group or secondary winding group of the transformer may include windings between a first end of the transformer and a second end of the transformer (i.e., first winding L22 and second winding L21). Because windings L21 and L22 are physically connected to each other, the HSC converter of Figure 2 is a non-isolated DC-DC converter.

[0030] In the HSC converter of FIG. 2, the first resonant circuit may include a first resonant capacitor Cres1, and the first resonant circuit may be connected between the first winding L12 and a first node between the first switch Q1 and the second switch Q2. In the HSC converter of FIG. 2, the second resonant circuit may include a second resonant capacitor Cres2, and the second resonant circuit may be connected between the fourth winding L11 and a third node between the fourth switch Q4 and the fifth switch Q6. The first and second resonant circuits may use transformer leakage inductance. Alternatively, the first and second resonant circuits may use discrete inductors.

[0031] The output terminal Vout of the HSC converter in Figure 1 provides a single output voltage, which may be an unregulated output voltage. The ratio of the input voltage Vin to the output voltage Vout is given by the following equation:

[0032]

number

[0033] Figure 3 shows various current paths in the HSC converter of Figure 2. A first current path is from one terminal of the input voltage Vin through the switch Q1, which is on, and the first node, then through the first resonant circuit and the fourth node, then through the first winding L22 (i.e., winding 1) and the single tap, then through the output capacitor Cout, and then to the other terminal of the input voltage Vin connected to ground, creating a first current loop including the input voltage, the first switch Q1, the first resonant circuit, the first winding L22 (i.e., winding 1), and the output capacitor Cout. A second current path is from the second node through the second resonant circuit, then through the fifth switch that is on and the fourth node, then through the first winding L21 (i.e., winding 1) and the single tap, then through the output capacitor Cout, and then through the third switch Q3 back to the second node, creating a second current loop that includes the second resonant circuit, the fifth switch Q5, the first winding L22 (i.e., winding 1), the output capacitor Cout, and the third switch Q3. The first and second current paths in the first winding L22 (i.e., winding 1) induce a third current path in the second winding L21 (i.e., winding 2). The third current path passes through the second winding L21 (i.e., winding 2) and a single tap, then through the output capacitor Cout, and then through the third switch Q3 back to the second winding L21 (i.e., winding 2), forming a third current loop including the second winding L21 (i.e., winding 2), the output capacitor Cout, and the third switch Q3. The output terminal Vout is connected to the output capacitor Cout, the first current loop, the second current loop, and the third current loop. In FIG. 3, the capacitances of the first resonant capacitor Cres1 and the second resonant capacitor Cres2 are the same so that Cres1 = Cres2 = Cres. In addition, a first group of switches including switches Q1, Q3, and Q5 is turned on, and a second group of switches including switches Q2, Q4, and Q6 is turned off. The switches Q1 to Q6 can be switched at a frequency given by the following equation:

[0034]

number

[0035] where f sw is the switching frequency, L k is the leakage inductance of the winding, C cres is the capacitance of the resonant capacitors Cres1 and Cres2. The voltage across the resonant capacitors Cres1 and Cres2 is half the input voltage, as given by the following equation:

[0036]

number

[0037] where V cres is the voltage of the resonant capacitors Cres1 and Cres2, and V in is the voltage of the input voltage. The currents in the first and second resonant circuits are equal and are given by:

[0038]

number

[0039] where I cres1 , I cres2 are the currents in the first and second resonant circuits, and I cres is the current through either the first or second resonant circuit, and I outDC is the current output. The root mean square (RMS) current through switch Q1 is half the current through the first resonant circuit, as given by:

[0040]

number

[0041] where I Q1(rms)is the RMS current through the first switch Q1. The RMS current through the third switch Q3 is two-thirds of the current through the second resonant circuit, as given by the following equation:

[0042]

number

[0043] where I Q3(rms) is the RMS current through the first switch Q3. The current through the first winding L22 (shown as winding 1 in FIG. 3) and the second winding L21 (shown as winding 2 in FIG. 3) is the current through both the first resonant circuit and the second resonant circuit as given by the following equations:

[0044]

number

[0045] where I winding1 is the current through the first winding L22, and I winding2 is the current through the second winding L21. Figures 4A and 4B show simulated waveforms for the 1500W HSC converter of Figure 2, with a 54V input, a 13.5V output (=Vin / 4 = 54V / 4), and a 111A output current. The switches in the first and second bridges can be driven by two PWM signals with any frequency but a 180° phase difference. Thus, a resonant current Ires can be generated in a first resonant circuit and a second resonant circuit, each of which includes a resonant capacitor and a resonant inductor, which can be either the leakage inductance of a transformer or a discrete inductor. The resonant current Ires flows through each of switches Q1 and Q2. A current twice the resonant current Ires is induced in the second winding L21 of the transformer, and a current three times the resonant current Ires, which is the sum of the currents in the second and third current loops, flows through switch Q3. As shown in FIG. 4A, a current Is1 and Is2 twice the resonant current Ires flows through the first winding L22 (i.e., winding 1) and the second winding L21 (i.e., winding 2), respectively. FIG. 4B shows that the output voltage is one-fourth the input voltage (i.e., Vout = Vin / 4). The RMS current Iq1 through the first switch Q1 is approximately 22.26 A, the RMS current Iq2 through the second switch Q2 is approximately 21.78 A, and the RMS current Iq3 through the third switch Q3 is approximately 66.57 A. The RMS current Is1 through the first winding L22 (winding 1) is approximately 62.29 A, and the average current Is1 through the first winding L22 (winding 1) is approximately 55.65 A. The RMS current IS2 through the second winding L21 (winding 2) is approximately 62.16 A, and the average current IS1 through the first winding L22 (winding 1) is approximately 55.50 A. The RMS voltage of the output voltage is approximately 13.15 V, and the peak-to-peak voltage of the output voltage is approximately 115.74 mV.

[0046] Compared to an LLC converter, the HSC converter of Figure 2 full-wave rectifies the transformer winding current, improving current utilization of a single winding and reducing winding losses, including losses caused by the winding being contained on a printed circuit board (PCB).The effective current flowing through the output switches (i.e., switches Q3 and Q6 in Figure 2) can be reduced compared to the current flowing through the similar output switches of an LLC converter, reducing losses in the output switches.

[0047] Figure 5 shows a table comparing an equivalent LLC converter with the HSC converter of Figure 2, with both the LLC and HSC converters providing an output voltage that is one-quarter of the input voltage. As shown in the table, the LLC converter has a 4:1:1 turns ratio, while the HSC converter has a 1:1 turns ratio. Both the LLC and HSC converters contain four 80V transistors in the primary circuit and two 40V transistors in the secondary circuit. The current in the primary winding of the LLC converter is given by the following equation:

[0048]

number

[0049] An HSC converter does not have a primary winding. The half-wave current in the secondary winding of an LLC converter is given by the following equation:

[0050]

number

[0051] Also, the full wave current in the secondary winding of an HSC converter is half of the current in an LLC converter and is given by the following equation:

[0052]

number

[0053] The RMS current in the first and second secondary windings of the LLC converter is given by the following equations:

[0054]

number

[0055] Also, the RMS current in the first and second secondary windings of the HSC converter is about 29% of the current in the LLC converter and is given by:

[0056]

number

[0057] The RMS current in the primary switch of LLC and HSC converters is half-wave and is given by:

[0058]

number

[0059] The RMS current in the secondary switch of the LLC converter is half-wave rectified and is given by the following equation:

[0060]

number

[0061] Also, the RMS current in the secondary switch of the HSC converter is half-wave rectified and is three-quarters of the current in the LLC converter, and is given by the following equation:

[0062]

number

[0063] Figures 6 and 7 show graphs of current waveforms for equivalent LLC and HSC converters that deliver an output voltage equal to one-fourth the input voltage. Both the LLC and HSC converters can deliver 1500 W of power. The input voltage is 54 V, the output voltage is 13.5 V, and the output current is 111 A. For the LLC converter, Figure 6 shows graphs of current waveforms for switches Q1-Q3 at the top and graphs of current waveforms for the primary winding and the first and second secondary windings at the bottom. For the HSC converter, Figure 7 shows graphs of current waveforms for switches Q1-Q3 at the top and graphs of current waveforms for the first and second secondary windings at the bottom. The table in Figure 8 summarizes the peak and RMS currents for the graphs in Figures 6 and 7. In the LLC converter, the peak current through the first switch Q1 is 43.5 A and the RMS current is 21.8 A, the peak current through the second switch Q2 is 43.5 A and the RMS current is 21.8 A, the peak current through the third switch Q3 is 174 A and the RMS current is 87.2 A. The peak current through the primary winding is 43.5 A and the RMS current is 30.8 A, the peak current through the first secondary winding is 174 A and the RMS current is 87.2 A, and the peak current through the second secondary winding is 174 A and the RMS current is 87.2 A. In the HSC converter, the peak current through the first switch Q1 is 43.5 A and the RMS current is 21.8 A. The peak current through the second switch Q2 is 43.5 A and the RMS current is 21.8 A. The peak current through the third switch Q3 is 131 A and the RMS current is 65.4 A. The HSC converter has no primary winding. The peak current through the first secondary winding is 87.2 A and the RMS current is 61.6 A. The peak current through the second secondary winding is 87.2 A and the RMS current is 61.6 A.

[0064] 9A-9C show simulated waveforms of the bidirectional behavior of the HSC converter of FIG. 2 when power flows from the output terminal Vout to the input voltage Vin. FIGS. 9A-9C show simulated waveforms of a 1080W HSC converter with a 13.5V input at the output terminal Vout, a 54V output at the input voltage Vin (=Vin*4=13.5V*4), and a 20A output current. Similar to the normal operation shown in FIG. 4 above, FIGS. 9A-9C show that power can flow in the opposite direction with the switches driven by two PWM signals to cause resonant currents Ires to flow in the first and second resonant circuits. FIG. 9C shows that the output voltage (i.e., the voltage at the input voltage Vin) is four times the input voltage (i.e., the voltage at the output terminal Vout) (Vin=4×Vout). The peak-to-peak output voltage is approximately 230.42 mV. The RMS input voltage is 13.5V and the RMS output voltage is approximately 52.93V.

[0065] The above simulation results confirmed that an output voltage of 1 / 4 of the input voltage can be supplied by removing the primary winding, that the active current flowing in the secondary winding is reduced by 29% compared to an equivalent LLC converter system, and that the active current flowing in the secondary switch is reduced by 25% compared to an LLC converter system, and that the HSC converter in Figure 1 can operate in both directions and can be used as a bidirectional power supply.

[0066] It should be understood that the foregoing description is only illustrative of the present invention. Various alternatives and modifications may be devised by those skilled in the art without departing from the invention. Accordingly, the present invention is intended to embrace all such alternatives, modifications, and variations that fall within the scope of the appended claims.

Claims

1. 1. An HSC voltage conversion module, comprising: The input voltage, a transformer having no primary winding and including a secondary winding magnetically coupled by a magnetic core; a first switch bridge and a second switch bridge connected to the transformer and the input voltage; an output capacitor connected to a single node of the transformer to provide an output voltage that is one-fourth of the input voltage; a first resonant circuit connected between the first switch bridge and the transformer; a second resonant circuit connected between the second switch bridge and the transformer.

2. the first resonant circuit includes a first resonant capacitor; the second resonant circuit includes a second resonant capacitor; The HSC voltage conversion module of claim 1 , wherein the first resonant circuit and the second resonant circuit utilize leakage inductance of the transformer.

3. 3. The HSC voltage conversion module of claim 1, wherein each of the first switch bridge and the second switch bridge includes a first switch, a second switch, and a third switch connected in series.

4. The HSC voltage conversion module according to claim 1 or 2, wherein the secondary winding of the transformer includes a first winding and a second winding.

5. 3. The HSC voltage conversion module of claim 1, wherein the HSC voltage conversion module is bidirectional.

6. A converter comprising: an input terminal for receiving an input voltage; a first switch bridge connected in parallel across the input voltage, a first switch, a second switch, and a third switch connected in series; a first node between the first switch and the second switch; a first switch bridge including a third node between the second switch and the third switch; a second switch bridge, a fourth switch, a fifth switch, and a sixth switch connected in series; a second node between the fourth switch and the fifth switch; a second switch bridge including a fourth node between the fifth switch and the sixth switch; an output capacitor; A transformer, a single winding including a first secondary winding and a second secondary winding physically connected to each other to define a secondary winding group and magnetically coupled by a magnetic core; a transformer including a single tap connected to the output capacitor; a first resonant circuit connected between the first node and a first end of the single winding; a second resonant circuit connected between the second node and a second end of the single winding opposite the first end of the single winding; an output terminal connected to the output capacitor to provide a first output voltage that is one-quarter of the input voltage.

7. the first resonant circuit includes a first resonant capacitor; the second resonant circuit includes a second resonant capacitor; The converter of claim 6 , wherein the first resonant circuit and the second resonant circuit utilize leakage inductance of the transformer.

8. 8. A converter according to claim 6 or 7, wherein the converter is bidirectional so that power can flow from the input terminal to the output terminal and from the output terminal to the input terminal.

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