Current limited voltage mode control of multiple inputs

Abstract

Circuits and methods are provided that control voltage using current inputs. A first monitor receives multiple voltage inputs and selects a voltage input having a lowest voltage. A voltage control loop receives the selected voltage input and a reference voltage and generates a first signal. A second monitor receives multiple current inputs and selects a current input having a lowest current from the current inputs as a target current. A current control loop receives the first signal and the target current and generates a second signal, where the second signal corresponds to a lesser of the first signal and the target current.

Description

Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 CURRENT LIMITED VOLTAGE MODE CONTROL OF MULTIPLE INPUTS Inventor: Gregory Szczeszynski CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This patent application claims the benefit of and priority to in their entirety the following United States Provisional Patent Applications filed on January 12, 2024, which are all incorporated by reference in their entirety:

[0002] Application No.63 / 620,507 entitled “LEVEL SHIFTER AND BOOT CAPACITOR CIRCUITS, SYSTEMS, AND METHODS;”

[0003] Application No.63 / 620,623 entitled “LEVEL SHIFTER AND BOOT CAPACITOR CIRCUITS, SYSTEMS, AND METHODS;”

[0004] Application No.63 / 620,613 entitled “INTEGRATED CURRENT RESISTOR SENSING FOR MULTI-LEVEL CONVERTER;”

[0005] Application No.63 / 620,465 entitled “STARTUP INTERLOCK FOR POWER CONVERTER CIRCUITS;”

[0006] Application No.63 / 620,331 entitled “FULLY DIFFERENTIAL LEVEL SHIFT IN A NOISY ENVIRONMENT;”

[0007] Application No.63 / 620,450 entitled “CAPACITOR SENSING AND CAPACITOR BALANCING SYSTEMS AND METHODS;”

[0008] Application No.63 / 620,469 entitled “CAPACITOR SENSING AND CAPACITOR BALANCING SYSTEMS AND METHODS;”

[0009] Application No.63 / 620,678 entitled “RECONFIGURABLE MULTI-LEVEL POWER CONVERTER TO CHARGE PUMP MODE AND FRACTIONAL CHARGE PUMP MODE;”

[0010] Application No.63 / 620,417 entitled “INPUT CURRENT SLEW FOR A MULTI- LEVEL CONVERTER;”Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0011] Application No.63 / 620,726 entitled “ADJUSTING OVERVOLTAGE PROTECTION BASED ON MODE OF OPERATION SYSTEMS AND METHODS;”

[0012] Application No.63 / 620,737 entitled “HYBRID PEAK AVERAGE CURRENT MODE CONTROL;”

[0013] Application No.63 / 620,741 entitled “CURRENT LIMITED VOLTAGE MODE CONTROL OF MULTIPLE INPUTS;”

[0014] Application No.63 / 620,527 entitled “MULTI-FUNCTION COMP PIN SYSTEMS AND METHODS;”

[0015] Application No.63 / 620,488 entitled “LEVEL SHIFTER AND BOOT CAPACITOR CIRCUITS, SYSTEMS, AND METHODS;”

[0016] Application No.63 / 620,553 entitled “MULTI-LEVEL REVERSE CURRENT BLOCKING SYSTEMS AND METHODS;”

[0017] Application No.63 / 620,638 entitled “GENERAL STARTUP FOR MULTI- LEVEL POWER CONVERTER CIRCUITS;”

[0018] Application No.63 / 620,733 entitled “PRECISION ANALOG TO DIGITAL CIRCUIT TUNED VOLTAGE AND CURRENT MODE DC-DC CONVERTER;”

[0019] Application No.63 / 620,738 entitled “PREDICTIVE CONTROL LOOP PRE- CHARGING DURING A MULTI-LEVEL ZONE CHANGE;”

[0020] Application No.63 / 620,764 entitled “DETECTOR CIRCUIT FOR DETECTING ONE OF MULTI-INPUT CONTROLLING SIGNALS THAT CONTROLS A CONTROL LOOP CIRCUIT;”

[0021] Application No.63 / 620,607 entitled “STARTUP VOLTAGE SELECTION FOR MULTI-LEVEL POWER CONVERTER CIRCUITS;”

[0022] Application No.63 / 620,575 entitled “MULTI-LEVEL CAPACITOR FAULT DETECTION SYSTEMS AND METHODS;”

[0023] Application No.63 / 620,582 entitled “PARELLEL OPERATION OF MULTI- LEVEL POWER CONVERTERS;” andDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0024] Application No.63 / 620,763 entitled “AVERAGE AND PEAK CURRENT SENSE SYSTEMS AND METHODS.” BACKGROUND

[0025] This disclosure relates to electronic circuits, and more particularly for example to multi-level power converters.

[0026] Many electronic products, including mobile computing and / or communication products and components (e.g., notebook computers, ultra-book computers, tablet devices, LCD, LED displays, and the like) use multiple voltage levels for operation. For example, radio frequency (RF) transmitter power amplifiers may operate at relatively high voltages (e.g., 12V or more), whereas logic circuitry may operate at a relatively low voltage level (e.g., 1-3V) and other circuitry may operate at an intermediate voltage level (e.g., 5-10V).

[0027] Direct current power converters are often used to generate a lower or higher voltage from a common power source, such as a battery, solar cells, and rectified AC sources. Power converters which generate a lower output voltage level from a higher input voltage power source are commonly known as buck converters, so-called because the output voltage VOUT is less than the input voltage VIN, and hence the converter is “bucking” the input voltage. Power converters which generate a higher output voltage level from a lower input voltage power source are commonly known as boost converters, because VOUTis greater than VIN. Some power converters may be either a buck converter or a boost converter depending on which terminals are used for input and output. Some power converters may provide an inverted output.

[0028] One type of direct current power converter known as a multi-level power converter includes charge transfer capacitors as energy storage elements coupled by controlled switches to transfer charge from VIN to VOUT. Such charge transfer capacitors are commonly known as “fly capacitors” or “pump capacitors”. When a fly capacitor is used (i.e., not bypassed), the electrical energy flowing through that fly capacitor generally will either charge it or discharge it.

[0029] There is a continued need for improved circuits and methods for more effectively and efficiently operating and implementing various type of electrical circuits and devices, including for example multi-level converter circuits.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 SUMMARY

[0030] Embodiments of the present disclosure include systems, circuits, and methods for operating and implementing various electronics circuits, including multi-level converter circuits.

[0031] The embodiments are directed to a system comprising a first monitor configured to receive a plurality of voltage inputs and select a voltage input having a lowest voltage from the plurality of voltage inputs, a voltage control loop configured to receive the selected voltage input and a reference voltage, and generate a first signal, a second monitor configured to receive a plurality of current inputs and select a current input having a lowest current from the plurality of current inputs as a target current, and a current control loop configured to receive the first signal and the target current and generate a second signal, wherein the second signal corresponds to a lesser of the first signal and the target current.

[0032] The embodiments are directed to a system comprising a current control loop configured to receive a plurality of current inputs and select a first signal, wherein the first signal represents a current input having a lowest current, a voltage control loop configured to receive a plurality of voltage inputs and select a second signal, wherein the second signal represents a voltage input having a lowest voltage, a selector configured to combine the first signal and the second signal into a combined signal, and a mode control configured to select a mode based on the combined signal, wherein the mode corresponds to one or more voltage levels.

[0033] The embodiments are directed to a system comprising a multi-input error amplifier configured to receive a plurality of input current signals representing a plurality of target currents and a second input current signal, select an input current signal from the plurality of input current signals representing a lowest current, and generate an output signal representing an error between the selected input current signal and the second input current signal.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0034] The scope of the present disclosure is defined by the claims, which are incorporated into this section by reference. A more complete understanding of embodiments of the present disclosure will be afforded to those skilled in the art, as well as a realization of additional advantages thereof, by a consideration of the following detailed description of one or more embodiments. Reference will be made to the appended sheets of drawings that will first be described briefly. DESCRIPTION OF THE DRAWINGS

[0035] FIG.1A is an example power converter circuit with internal input current sense, in accordance with one or more embodiments of the present disclosure.

[0036] FIG.1B is an example power converter circuit with external input current sense, in accordance with one or more embodiments of the present disclosure.

[0037] FIG. 2A is an example dual integrated circuit (IC) power converter circuit with internal input current sense, in accordance with one or more embodiments of the present disclosure.

[0038] FIG. 2B is an example dual IC power converter circuit with external input current sense, in accordance with one or more embodiments of the present disclosure.

[0039] FIG. 3A is an example functional block diagram of a power converter circuit, in accordance with one or more embodiments of the present disclosure.

[0040] FIG. 3B is an example functional block diagram of a power converter circuit, in accordance with one or more embodiments of the present disclosure.

[0041] FIG. 4 is a diagram illustrating an example charging function in step down regulation mode of an example power converter circuit, in accordance with one or more embodiments of the present disclosure.

[0042] FIG. 5 is a diagram illustrating an example charging function in step down divide by 3 charge pump mode, in accordance with one or more embodiments of the present disclosure.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0043] FIG. 6 is a functional block diagram illustrating aspects of an example power converter circuit, in accordance with one or more embodiments of the present disclosure.

[0044] FIG. 7 is a block diagram illustrating an example system implementing a power converter circuit, in accordance with one or more embodiments of the present disclosure.

[0045] FIG. 8A is a circuit diagram illustrating an example 3-level converter circuit, in accordance with one or more embodiments of the present disclosure.

[0046] FIG. 8B is a circuit diagram illustrating an example 4-level converter circuit, in accordance with one or more embodiments of the present disclosure.

[0047] FIG. 8C is a circuit diagram illustrating an example M-level converter circuit, in accordance with one or more embodiments of the present disclosure.

[0048] FIG. 9 is an example M-level converter circuit, in accordance with one or more embodiments of the present disclosure.

[0049] FIG. 10 is a block diagram of an example embodiment of control circuitry for an M-level converter cell, in accordance with one or more embodiments of the present disclosure.

[0050] FIG. 11 is a block diagram of a controller, in accordance with one or more embodiments of the disclosure.

[0051] FIG. 12 is a flow chart of a method for operating a controller, in accordance with one or more embodiments.

[0052] FIG. 13 is a block diagram of a circuit that uses current to limit a voltage mode control, in accordance with one or more embodiments.

[0053] FIG.14 is a flow chart of a method of using current to limit a voltage mode control, in accordance with one or more embodiments.

[0054] FIG. 15 is a block diagram of an error amplifier, in accordance with one or more embodiments.

[0055] FIGs.16A-B are block diagrams of a multi-input error amplifier, in accordance with one or more embodiments.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0056] FIG. 16C are diagrams of graphs illustrating current flowing through transistors of a multi-input error amplifier, in accordance with one or more embodiments.

[0057] FIGs. 17A-C are block diagram illustrating an analog implementation of a voltage control loop and a current control loop, in accordance with one or more embodiments.

[0058] FIG.18 is an example method of using inputs from a multi-input error amplifier to limit an output from an error amplifier, in accordance with one or more embodiments.

[0059] Embodiments of the present disclosure and their advantages are best understood by referring to the detailed description that follows. It is noted that sizes of various compo- nents and distances between these components are not drawn to scale in the figures. It should be appreciated that like reference numerals are used to identify like elements illustrated in one or more of the figures. DETAILED DESCRIPTION

[0060] The present disclosure encompasses novel circuits, architectures, systems, and methods that more effectively and efficiently address the configuration and operation of multi- level converter circuits. It will be appreciated that various improvements disclosed herein encompass innovative circuits, hardware components, architectures, and related logic that are applicable to applications beyond multi-level converter circuits.

[0061] FIGs.1-6 illustrate various embodiments of a high efficiency 4-level step-down and step-up power converter for battery charging applications, such as single cell Li-ion and Li- polymer battery applications. In the illustrated embodiments, the power converter is configured to deliver up to 5 amperes (A) of charging current in regulation mode and in a divide-by-3 charge pump mode, though other configurations are within the scope of the present disclosure. The power converter can be configured, for example, into dual ICs operation for 9A charging current in regulation mode and in divide-by-3 charge pump mode. Although a 4-level power converter is illustrated, it will be appreciated that the embodiments described herein may be applicable to various M-level implementations, where M >= 3.

[0062] In some implementations, for example, the power converter may supply an input range of approximately 4.5 V to 18 V input to support both universal serial bus (USB) and wireless inputs, and in a reverse step-up mode, the output may be programmable from 4.8 V to 16 V in 100 mV step with a programmable output current limit up to 1.7 A. This input voltageDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 range may be used, for example, to support fast charging of single Li-Ion cells from USB and wireless input. It will be appreciated that other voltage and current ranges and limits may be implemented depending on the application. It will also be appreciated that while compatibility with USB is described herein, other wired interfaces and protocols may be implemented with the power converter of the present disclosure.

[0063] In various embodiments, the power converter may be implemented as a single integrated circuit (IC) (see, e.g., Figs.1A-B), dual-integrated circuits (see, e.g., Figs.2A-B), or in other configurations depending on the implementation. In various embodiments, the power converter may operate as a parallel charger along with a main charger, as shown in Fig.3B, to provide the desired functionality noted herein and, for example, as illustrated in Figs. 4 and 5 for the desired charging functionality for various applications, as would be understood by one skilled in the art. Fig. 3B may represent a system level point of view of a mobile architecture having a parallel charger and a main charger that accepts power from a wired port (e.g., a wired USB) or from a wireless interface. The parallel charger for one or more embodiments may represent an IC as illustrated in Figs.1-3A, for example, and may function to charge a battery for some portion of the charging profile (e.g., as shown in Figs.4 and 5), while the main charger charges the battery for other portions of the charging profile. In various embodiments, the parallel charger may also be configured to function as the main charger as well, depending upon the desired application. The novel architecture disclosed herein may be implemented to enable (i) improved efficiency (e.g., at 9A charging current) in a low-profile solution; (ii) low electromagnetic interference (EMI) fixed-frequency operation under heavy load conditions; (iii) input and output current and voltage, IC temperature monitoring and telemetry via inter- integrated circuit (I2C) technology; and / or (iv) full protection including input and output under voltage lockout (UVLO), input and output over voltage protection (OVP), input and output over current protection (OCP), and IC over-temperature with fault and warning status. In some implementations, the power converter supports divide-by-3, step-down and step-up regulating modes, dual external disconnect switch control, and / or paralleled operation.

[0064] In the illustrated embodiments, the power converter is implemented as a multi-level charge pump incorporating power switches and control circuitry. The power converter’s internal bias may be provided by the system battery through a VOUT connection (e.g., pin). The charging input can be USB (or other wired input) or wireless input by an external FET register control. In some implementations, the power converter may be programmed toDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 different operating modes, which may include a step-down regulation mode, a step-down divide-by-3 charge pump mode, and a reverse step-up mode.

[0065] In a step-down regulation mode, the power converter operates as a multi-level step- down regulator to support USB power delivery (USB-PD) (or other wired protocol) or fixed input charging. During a constant-current (CC) phase, the maximum charging current may be limited for example, by configuring registers. When the input current does not reach a predetermined maximum input setting, the charge current is set to a predetermined maximum output setting. If the input current reaches the input maximum setting, then the charge current throttles and maintains input current at the input maximum setting. This allows maximum charging current while ensuring that the charge current does not go above a battery maximum current rating and the input current does not trip adapter over-current protection.

[0066] During a constant-voltage (CV) phase, the CV regulation may be limited, for example, by configuring registers. In operation, a single-wire sense pin or other sensor is configured to sense the output voltage VOUT, which is compared to a predetermined value stored in a register, VOUT_REG. The voltage differential between the battery’s positive terminal and negative terminal is sensed and compared to a predetermined value stored in a register, VBATT_REG. In some implementations, a single-wire sense pin or other sensor senses VBATTP (battery voltage at positive terminal) and a single-wire sense pin or other sensor senses VBATTN (battery volage at negative terminal). The CV regulates to the lower of the two settings. If the VOUT sensed voltage reaches VOUT_REG first, then CV is regulated to VOUT_REG. If the VBATTP sensed voltage reaches VBATT_REG first, then CV is regulated to VBATT_REG. This provides a fast battery top off while preventing voltage above safety limit.

[0067] In a step-down divide-by-3 charge pump mode (which may be selected, for example, by setting a corresponding register), the power converter is configured as a divide- by-3 step-down charge divider to support USB-Programmable Power Supply (USB-PPS) or other charging protocol or programmable input charging. In some embodiments, the power converter allows the USB-PPS adapter to control voltage and current and ignores conflicting settings (e.g., settings stored in registers for IOUT_MAX, VOUT_REG and VBATT_REG). In this mode, the power converter monitors an IIN_MAX setting, shuts down the power train (which includes switches to configure, enable and disable various modes of operation) andDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 disconnects external FET when IIN current exceeds IIN_MAX setting. In the illustrated embodiment, the output current is up to 10A in dual IC operation and 5A in single IC operation.

[0068] In a reverse step-up mode (which may be selected, for example, by setting a corresponding register) the power converter is configured as a multi-level step-up regulator to power peripheral device(s) connected to USB (or other wired protocol or standard) or wireless input. The power converter draws power from the system battery and regulates VIN to the VOUT_REG programmable setting of 4.8V to 16V. The VIN output current limit may be set, for example, by an IIN_MAX register.

[0069] In some embodiments, to enable the IC, both an EN pin and an IC_EN bit are set to logic high (1). When either the EN pin or IC_EN bit is set to logic low (0), the IC is disa- bled. After the IC is enabled, the POR status bit sets to 1 to indicate the IC has a fresh power up.

[0070] In some embodiments, the power converter provides a gate driver to control two external N-channel MOSFETs and sense inputs to monitor source input voltage at each FET. The external FETs may be controlled by registers (e.g., 1-bit registers V_EXTG, EXTG_EN and EXTGX). The V_EXTG bit sets the gate drive voltage and can be set to 9V or 5V, in the illustrated embodiment. The EXTGX bits select which FET(s) to turn on. The EXTG_EN bit enables the gate driver to turn on the selected FET(s). In various embodiments, the external FET can be turned on or off independently from other IC operations except when the IC is disabled. The EXT_EN_IND status bit set to 1 when external FET is enabled. When a fault is detected and triggers a shutdown, the external FET may be turned off automatically. If EXT1 or EXT2 detects an OVP, then the respected FET would not turn on from the off mode.

[0071] In various embodiments, the power train is enabled after all the registers have been initialized and the target input external FET is turned on. Sufficient time based on ca- pacitance on the power path may be configured between the external FET on time and the power train on time to minimize in-rush current. Next, both PT_EN pin and PT_EN bit are set to logic high (1) to turn on the power train. When either PT_EN pin or PT_EN pin is logic low, the power train is off. In dual IC operation, the slave IC power train may be configured to turn on first before the master IC. The COMP, SYNC and SYNCH pins from two ICs gate the power train and synchronize the operation. The SYNC_SEL pin sets the IC to masterDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 mode or slave mode. IC internal fault and programmable fault detection shuts down the power train operation when fault is detected.

[0072] In a reverse step-up mode (which may be selected, for example, by setting a corresponding register), the power converter is configured as a multi-level step-up regulator to power peripheral device(s) connected to USB (or other wired port) or wireless input. The power converter draws power from the system battery and regulates VIN pin to a VOUT_REG programmable setting of 4.8V to 16V. The VIN output current limit is set by IIN_MAX register.

[0073] To enable the IC, both the EN pin and IC_EN bit are set to logic high (1). When either EN pin or IC_EN bit is set to logic low (0), the IC is disabled. After the IC enables, the POR status bit sets to 1 to indicate the IC has a fresh power up. The power converter provides a gate driver to control two external N-channel MOSFETs and sense inputs to monitor source input voltage at each FET. The external FETs are controlled by register bits, such as V_EXTG, EXTG_EN and EXTGX. The V_EXTG bit sets the gate drive voltage and can be set to 9V or 5V, for example. The EXTGX bits select which FET(s) to turn on. The EXTG_EN bit enables the gate driver to turn on the selected FET(s). The external FET can be turned on or off independently from other IC operation except when the IC is disabled. The EXT_EN_IND status bit set to 1 when external FET is enabled.

[0074] When a fault is detected and triggers a shutdown, the external FET may be turned off automatically. If EXT1 or EXT2 detects an OVP, then the respective FET would not turn on from off mode. The power train is enabled after all the registers have been initialized and the target input external FET is turned on. Sufficient time based on capacitance on the power path should be given between external FET on time to power train on time to minimize in-rush current. Next, both PT_EN pin and PT_EN bit are set to logic high (1) to turn on the power train. When either PT_EN pin or PT_EN pin is logic low, the power train is off. In dual IC operation, the slave IC power train is turned on before the master IC. The COMP, SYNC and SYNCH pins from the two ICs gate the power train and synchronize the operation. SYNC_SEL pin sets the IC to master mode or slave mode. IC internal fault and programmable fault detection shuts down power train operation when a fault is detected.

[0075] In accordance with various embodiments, an example power converter initialization, an example power up sequence, and an example fault handling will now beDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 described for the three different operating modes. In an example step-down regulation mode, the initialization and power up sequence uses EXT1 as an example. The same sequence may apply to EXT2 with the only change in EXTGX bit and related EXT2 register settings. First, pull EN to logic high and then set IC_EN bit=1 at 100us(TBD) after EN is logic high to enable IC. IC startup from POR stage, POR bit reports 1 indicating fresh IC startup. Next, the POR bit is read to confirm the IC is enabled. The FREQUENCY register is then set to a desired setting. In dual IC operation, both ICs are set to the same frequency setting. The VOUT_REG register is set to the target regulation voltage on the VOUT sense pin in CV operation. The VBATT_REG register is set to the target regulation voltage on the VBATTP sense pin in CV operation. The IOUT_MAX register is set to the target maximum charger current in CC operation, and the IIN_MAX register is set to a value below the adapter current limit. Next, the FAULT and WARNING registers was set to a desired setting. Each Fault and Warning enables at a different time based on IC status and operating mode. The WATCHDOG register is then set to a desired setting.

[0076] The MODE register and other related registers are set for step-down regulation mode, including power train setup and enablement of an external FET, while checking for faults. In a dual IC operation, the external FETs are controlled by the master IC. If a fault (e.g., OVP event) is detected, then a shutdown register may be set to “1” to indicate a fault shutdown event and a sequence to enable the external FET after the shutdown fault is initiated. Next, the power train is enabled. In a dual IC operation, the slave IC power train is turned on before the master IC. After the power train is enabled, a bit may be set to indicate that the power train is ready and charging the battery. In some embodiments, a watchdog timer may be set to periodically check the IC status during charging operation.

[0077] If a fault event is detected, then the IC determines which faults events were triggered, such as the power train may be set to enable but it is off due to fault(s), an external FET is set to enable but the FET is off due to fault(s). The shutdown procedure may include resetting register values and repeating setup steps of enabling the power train, external FET, or other component that is disabled due to a fault.

[0078] An example step-down divide-by-3 power converter mode initialization and power up sequence will now be described. The initialization and power up sequence uses EXT1 as an example, but it will be appreciated that the same sequence applies to EXT2 with a change in EXTGX bit and related EXT2 register settings. The EN is pulled to logic high and then IC_ENDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 bit=1 at 100us(TBD) after EN is logic high to enable IC. The IC starts up from POR stage, POR bit reports 1 indicating fresh IC startup. The POR bit is read to confirm the IC is enabled. The FREQUENCY register is set to a desired setting. In dual IC operation, both ICs are set to the same frequency setting. The IIN_MAX register is set to a value below the adapter current limit. VOUT_REG, VBATT_REG and IOUT_MAX registers are not used in step-down divide-by-3 charge pump mode. Voltage and current regulation in step-down divide-by-3 charge pump mode may be controlled by the PPS adapter. The FAULT, WARNING, and WATCHDOG registers are set to desired settings. Each Fault and Warning enables at different time based on IC status and operating mode.

[0079] The MODE register and other registers are set for step-down divide-by-three mode, including power train setup and external FET setup, while checking for faults. If a fault (e.g., OVP event) is detected, then a shutdown register may be set to “1” to indicate a fault shutdown event and a sequence to enable the power train or external FET, as appropriate, after the shutdown fault is initiated. Next, the power train is enabled. After the power train is enabled, a bit may be set to indicate that the power train is ready and charging the battery. In some embodiments, a watchdog timer may be set to periodically check the IC status during charging operation. Voltage and current regulation in step-down divide-by-3 charge pump mode may be controlled by the PPS adapter.

[0080] If a fault event is detected, then the IC determines which faults events were triggered, such as the power train may be set to enable but it is off due to fault(s), or an external FET is set to enable but the FET is off due to fault(s). The shutdown procedure may include resetting register values and repeating setup steps of enabling the power train, external FET, or other component that is disabled due to a fault.

[0081] An example reverse step-up mode initialization and power up sequence will now be described. This initialization and power up sequence uses EXT2 as an example, but the same sequence applies to EXT1 with the change in EXTGX bit and related EXT1 register setting. The value EN is pulled to logic high and then IC_EN bit is set to 1 at 100us(TBD) after EN is logic high to enable IC. The IC starts up from the POR stage, and the POR bit reports 1 indicating a fresh IC startup. The POR bit is read to confirm the IC is enabled. Next, the FREQUENCY register is set to a desired setting. In dual IC operation, both ICs are set to the same frequency setting. The VOUT_REG register is set to the target regulation voltage at VIN. Next, the IIN_MAX register is set to the target current limit. VBATT_REG and IOUT_MAXDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 registers are not used in reverse step-up mode. FAULT, WARNING, and WATCHDOG registers are set to desired settings. Each Fault and Warning enables at a different time based on IC status and operating mode.

[0082] The MODE register and other registers are set for reverse step-up mode, including power train setup and external FET setup, while checking for faults. If a fault (e.g., OVP event) is detected, then a shutdown register may be set to “1” to indicate a fault shutdown event and a sequence to enable the power train or external FET, as appropriate, after the shutdown fault is initiated. Next, the power train is enabled. After the power train is enabled, a bit may be set to indicate that the power train is ready and charging the battery. In some embodiments, a watchdog timer may be set to periodically check the IC status during charging operation. Voltage and current regulation in step-down divide-by-3 charge pump mode may be controlled by the PPS adapter. In dual IC operation, the slave IC power train is turned on before the master IC and is controlled by the master IC.

[0083] If a fault event is detected, then the IC determines which faults events were triggered, such as the power train may be set to enable but it is off due to fault(s), or an external FET is set to enable but the FET is off due to fault(s). The shutdown procedure may include resetting register values and repeating setup steps of enabling the power train, external FET, or other component that is disabled due to a fault. The EXT2 or VIN pins are not configured to detect OVP as it is set as the output in reverse step-up mode. But if EXT2 or VIN pin detects an OVP event, then IC_STATUS1 and IC_STATUS2 would report the fault event.

[0084] In an example system 700 illustrated in FIG. 7, a power converter 720 is implemented in a host 710 (e.g., a device or system) that includes a battery 730 and various system components 740. The host 710 may be any system or device that implements a power converter as described herein, including but not limited to a smart phone, tablet, portable electronics, a mobile device, low power electronics, and other electronic systems. The battery 730 may include one or more batteries that store electricity for use by the host 710, such as single cell Li-ion and Li-polymer batteries.

[0085] The power converter 720 may be configured to convert electricity stored in the battery 730 to a desired system voltage, VSYS, for powering various system components 740, which may include one or more logic devices 742, memories 744, communications components 746, input / output (I / O) components 748, circuitry 750, and other components 752. The powerDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 converter 720 may also supply power to one or more external devices 760, such as a component connected to the host 710 through a wired or wireless connection, such as a USB compatible device. The power converter 720 may also be configured to receive power from an external power source 712 and convert the received power to the battery 730 for storage, or to the system components 740 and / or external device 760, as applicable.

[0086] In various embodiments, the one or more logic devices 742 and memories 744 may be configured to perform operations of the host 710. A logic device 742 may be imple- mented as a general-purpose processor, a digital signal processor (DSP), an application-spe- cific integrated circuit (ASIC), a microcontroller, a programmable logic device (PLD), a field-programmable gate array (FPGA), or other programmable logic device(s). The logic de- vice 742 and other components may be configured through hardwiring, software execution, or a combination of both. In various embodiments, the host 710 includes one or more memory devices designed to retain data, such as software instructions for execution by the logic device. The memory may include volatile and non-volatile memories, such as random- access memory (RAM), dynamic RAM (DRAM), static RAM (SRAM), non-volatile random- access memory (NVRAM), read-only memory (ROM), programmable read-only memory (PROM), erasable programmable read-only memory (EPROM), electrically-erasable pro- grammable read-only memory (EEPROM), flash memory, hard disk drives, or other memory types. The logic device may be configured to execute software instructions residing in the memory, thereby accomplishing method steps and operations.

[0087] Referring to FIGs. 8A-8C, the converter circuit may be configured to switch between two or more switch states. One or more PWM duty cycle controllers may be provided to set the time in each switch state based on the voltage at VOUT. For example, FIG. 8A is a schematic diagram of a 3-level DC-to-DC buck converter circuit 800 that may be used as the converter circuit 920 of FIG. 9. A set of four switches, S1-S4, is series-coupled between VIN and circuit ground. A fly capacitor C1 is coupled in series with switches S3 and S4, and in parallel with switches S1 and S2. An inductor L1 is coupled to an output capacitor COUT and to a node Lx between switches S1 and S2, and the voltage across the output capacitor COUT is VOUT.

[0088] In the illustrated example, the presence of the single fly capacitor C1 in the converter circuit 800 enables four switch states that each generate one of three voltage levels at node Lx. In a first switch state, S2 and S4 are closed and S1 and S3 are open, effectivelyDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 bypassing C1 and connecting Lx to circuit ground (voltage level at Lx = GND). In a second switch state, S2 and S4 are open and S1 and S3 are closed, effectively bypassing C1 and connecting Lx to VIN (voltage level at Lx = VIN). In a third switch state S1 and S4 are open and S2 and S3 are closed, connecting C1 from VIN to LX, and thus charging C1 with inductor L1 current flowing into a load. The voltage across C1 will be about VIN / 2 and the voltage level at Lx will also equal about VIN / 2. In a fourth switch state, S1 and S4 are closed and S2 and S3 are open, connecting C1 from Lx to GND and thus discharging C1 with inductor L1 current flowing to a load. The voltage across C1 will be about VIN / 2 and the voltage level at Lx will also equal about VIN / 2 (e.g., this may assume that C1 was previously charged in state three). Accordingly, the illustrated converter circuit 800 has two switch states that generate a voltage level of VIN / 2 at the Lx node.

[0089] If the converter circuit 800 is toggled between switch states three and four (avoiding switch state two that bypasses the fly capacitor C1), the inductor L1 sees small jumps in the voltage level at Lx, going from GND to only VIN / 2 and back to GND, which results in reduced voltage ripple across the inductor L1 and less filtering to smooth VOUT than a converter circuit with only S1 and S2 switches.

[0090] Adding additional series switches Sx and fly capacitors Cx to the 2-level converter circuit 800 increases the number of switch states and resulting voltage levels between VIN and circuit ground that can be applied to the Lx node, thus generating an even smaller voltage ripple across the inductor L. This reduces the filtering requirements to get a smooth output voltage. For example, a 4-level DC-to-DC buck converter circuit (see, e.g., FIG.8B) includes 6 series- coupled switches S1-S6 and two fly capacitors Cx (X = 2). Consequently, a 4-level converter circuit can define 4 voltage levels (VIN, GND, VIN, and VIN) at node LX from 8 switch states (3 switch states result in the VIN level at Lx, and 3 other switch states result in the VIN level at Lx). For some applications, VOUT is set low enough that the voltage level at node Lx alternates between GND and the next higher voltage level available. For higher output voltages, the switching pattern may never use GND. For example, in a 4-level converter circuit,an output VOUT set to 0.5 VIN can be achieved by alternating the Lx node between VINand V.

[0091] A different interpretation of a multi-level converter circuit is that the fly capacitors Cx create a charge-pump for the buck converter circuit. Unlike a standard charge-pump where the output is restricted to one output, a multi-level converter circuit allows the fly capacitorsDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 Cx to be coupled to create multiple intermediate voltages. For the 4-level example, the two fly capacitors each act as a charge-pump with the additional benefit that any input voltage that is a sum of ratios can be created, including VIN and GND.

[0092] A multi-level converter circuit couples the fly capacitors Cx in different combinations in order to bring the voltage level at the Lx node down or up. As noted above, when a fly capacitor is used (i.e., not bypassed), the electrical energy flowing through that fly capacitor generally will either charge it or discharge it, which creates a control problem in maintaining an average voltage.

[0093] Resolving the charge-balance problem so as to maintain an average voltage across the single capacitor in a 3-level converter circuit will now be described. For example, in a 3- level converter circuit, one way to generate the Level-1 (GND) and Level-3 (VIN) voltage levels at the Lx node is to not use the fly capacitors C1 for these Lx voltage levels. However, for the Level 2 (VIN / 2) voltage level at Lx, two separate switch states can be used: one switch state charges the capacitor (S3 and S2 closed, S1 and S4 open) and the other switch state discharges the capacitor (S3 and S2 open, S1 and S4 closed). The control of a 3-level converter circuit may operate such that each time the converter circuit switches states to Level-2, a controller can alternate between charging and discharging the single capacitor to maintain its voltage. A voltage comparator can be used to monitor the capacitor to help decide on a charging state or a discharging state. For instance, if the capacitor voltage is below VIN / 2, then a controller would select charge (the third switch state), and if the capacitor voltage is above VIN / 2, then the controller would select discharge (the fourth switch state).

[0094] Referring to FIGs.8B, a 4-level converter circuit 830 (X = 2) illustrates the charge- balance difficulty when more capacitors are present. A Level-1 voltage level (GND) and a Level-4 voltage level (VIN) at the Lx node are each determined by a single switch state.However, the Level-2 voltage level ( VIN) and Level-3 voltage level ( VIN) at Lx each canbe achieved by any of three different switch states. At higher orders of a multi-level converter circuit (X > 2), more switch states are possible for generating the intermediate levels between VIN and GND. The problem gets more complicated with a 5-level converter circuit (X= 3). A Level-1 voltage level (GND) and a Level-5 voltage level (VIN) at the Lx node are each determined by a single switch state. However, the Level-2 voltage level (¼VIN) and Level-4 voltage level (¾ VIN) at Lx each can be achieved by any of four different switch states, the Level-3 voltage level (2 / 4 VIN) at Lx can be achieved by any of six different switch states.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0095] As should be clear from these examples, determining a suitable charge-balance method can become exceedingly difficult as the complexity of a multi-level converter circuit increases. As previously noted, most conventional control methods rely on establishing a sequence of linked state-changes to try to achieve charge balance. Control systems based on long sequences of switch states generally assume that all system variables - such as input voltage and output current - are constant during the sequence. This is unrealistic for a real- world environment, where all system variables tend to be dynamic.

[0096] In a 2-Level example, the converter circuit switches between two switch states: S1 closed and S2 open (voltage level at LX= VIN), or S1 open and S2 closed (voltage level at LX= GND). A PWM duty cycle controller sets the time in each switch state based on the voltage at VOUT, which determines the amplitude of the average voltage at LX (noting that, the average LX voltage in theory is equal to the VOUT average voltage, but that, due to parasitics, the LX average voltage is higher and / or lower (for negative currents) than the VOUT average). As can be appreciated, the inductor L sees large jumps in the voltage level at LX, from GND to VIN and back to GND. The resulting voltage ripple across the inductor L necessitates a significant amount of filtering to smooth VOUT.

[0097] An alternative way of reducing the voltage ripple across the inductor L is to add more series switches as well as charge transfer capacitors as energy storage elements to transfer charge from VINto VOUT. As noted above, such charge transfer capacitors are commonly known as “fly capacitors” or “pump capacitors” and may be external components coupled to an integrated circuit embodiment of a converter circuit. The presence of X fly capacitors Cx defines a multi-level capacitive converter circuit capable of generating M = X + 2 voltage levels at node LXfromswitch states.

[0098] FIG. 8C is schematic diagram of a generalized M-level multi-level converter cell 870 that may be used as the converter circuit 920 of FIG.9. A set of switches, S1-S[2*(M 1)], is series-coupled between VIN and circuit ground. The set of switches are organized in switch pairs: S1 & S2, S3 & S4, … S[2*(M 2)+1] & S[2*(M 1)]. A set of M 2 fly capacitor Cx is coupled in series with certain respective switches, and in parallel with switches in between those switches. In terms of switch pairs, there are M 1 pairs of switches, or one more than the number of fly capacitors. An optional inductor L is coupled to an output capacitor COUT and to a node LX between switches S1 and S2, and again the voltage across the output capacitor COUT is VOUT. The inductor L doubles as a virtual current source that facilitates movement of chargeDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 between the fly capacitors Cx. This creates a very efficient form of charge transfer, but introduces the problem of charge-balancing the fly capacitors Cx.

[0099] In various embodiments, each fly capacitor Cx has a first terminal coupled betweenan outer high-side switch S[2*x + 1] and an inner high-side switch S[2*x 1], where “high-side” refers to the VINside of the converter circuit. Each fly capacitor Cx has a second terminal coupled between an outer low-side switch S[2*x + 2] and an inner low-side switch S[2*x], where “low-side” refers to the circuit ground (GND) side of the converter circuit. Thus, for an M = 3 multi-level converter cell, a first terminal of the single (X = 1) fly capacitor C1 would be coupled between outer high-side switch S3 and inner high-side switch S1, and a second terminal of the capacitor C1 would be coupled between inner low-side switch S2 and outer low-side switch S4. Accordingly, each fly capacitor Cx within the multi-level converter cell 870 has four switches that can affect current flow through that fly capacitor Cx.

[0100] In some embodiments, a voltage detector, which may be a simple comparator-type circuit, is provided to sense the voltage across a corresponding fly capacitor Cx with respect to a reference voltage, VREF, which represents a desired target voltage for the fly capacitor Cx. Every fly capacitor Cx may have a target average voltage in order to maintain proper output level. For an M-level converter and capacitor Cx, where x = 1, 2, … [M 2], its target voltage Vtarget[Cx] = VIN *

[0101] The voltage detector may be configured to output a HIGH / LOW status signal, CFx_H / L, indicating with the voltage across the corresponding fly capacitor Cx is greater than VREFor less than VREF. The CFx_H / Lstatus signal is coupled to control circuitry for the switches associated with the fly capacitor Cx.

[0102] The control circuitry for the four switches that can affect current flow through a fly capacitor Cx set states for those switches in part as a function of the voltage across the fly capacitor Cx as measured by the associated voltage detector and conveyed by the CFx_H / Lx status signal. Accordingly, for ease of understanding, it can be said that each fly capacitor Cx “controls” its own pairs of high-side and low-side switches. If it is assumed that current flow in the inductor is charging the output VOUT, there are four possible states that can be defined for the pairs of high-side and low-side switches for each fly capacitor Cx.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0103] In a switch state in which the outer high-side and inner low-side switches associated with fly capacitor Cx are closed and all other associated switches are open, fly capacitor Cx would be in a charging configuration (whether or not charging actually occurs may depend on the switch states for other fly capacitors Cx). In a switch state in which the inner high-side and outer low-side switches associated with fly capacitor Cx are closed and all other associated switches are open, fly capacitor Cx would be in a discharging configuration (whether or not discharging actually occurs may depend on the switch states for other fly capacitors Cx). In a switching state in which the inner low-side and outer low-side switches associated with fly capacitor Cx are closed and all other associated switches are open, fly capacitor Cx would be bypassed. In a switching state in which the outer high-side and inner high-side switches associated with fly capacitor Cx are closed and all other associated switches are open, fly capacitor Cx would again be bypassed.

[0104] While each fly capacitor Cx can control both of its own pairs of high-side and low- side switches, in general, methods of control disclosed herein may utilize either the outer switches or the inner switches controllable by each corresponding capacitor. For example, referring to FIG.8B, in “outer-switch” methods, fly capacitor C1 will control its outer switches S3 and S4, fly capacitor C2 will control its outer switches S5 and S6, etc. Conversely, for example, in “inner-switch” methods, fly capacitor C1 will control its inner switches S1 and S2, fly capacitor C2 will control its inner switches S3 and S4, etc. The switch states of either pair (inner or outer) of switches controlled by a fly capacitor Cx may be complementary – that is, no fly capacitor Cx closes or opens both of its high-side and low-side controlled switches at the same time. If each fly capacitor Cx controls its outer-switches, then no fly capacitor controls the left-over innermost switches S1 and S2. If instead each fly capacitor Cx controls its inner- switches, then no fly capacitor controls the left-over outermost switches S[2*(M-1)] and S[2*(M-2)+1]. Switch states for the left-over switches are also complementary.

[0105] FIG. 9 is a high-level block diagram of an example circuit that includes a power converter 900, in accordance with one or more embodiments of the present disclosure. In the illustrated example, the power converter 900 includes a converter circuit 920 and a controller 910. The converter circuit 920 and controller 910 may be configured to implement, for example, any of the multi-level power converter circuits as previously described with reference to FIGs. 1A-8C, and as described further herein. In the illustrated embodiment, the converter circuit 920 is configured to receive an input voltage VIN from a voltage source and transformDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 the input voltage VIN into an output voltage VOUT. In some embodiments of the power converter 900, auxiliary circuitry (not shown), such as a bias voltage generator(s), a clock generator, a voltage control circuit, etc., may also be present and coupled to the converter circuit 920 and the controller 910.

[0106] The controller 910 receives a set of input signals and produces a set of output signals. Some of these input signals arrive along a signal path connected to the converter circuit 920. These input signals carry information that is indicative of the operational state of the converter circuit 920. The controller 910 may also receive a clock signal CLK (for synchronous converter circuits 920) and one or more external input / output signals I / O that may be analog, digital (encoded or direct signal lines), or a combination of both. Based upon the received input signals, the controller 910 produces a set of control signals back to the converter circuit 920 that control the internal components of the converter circuit 920 (e.g., internal switches, such as low voltage FETs / MOSFETs) to cause the converter circuit 920 to boost or buck VIN to VOUT. In some embodiments, an auxiliary circuit (not shown) may provide various signals to the controller 910 (and optionally directly to the converter circuit 920), such as the clock signal CLK, the input / output signals I / O, as well as various voltages, such as a general supply voltage VDD and a transistor bias voltage VBIAS.

[0107] FIG. 10 is a block diagram of one embodiment of advanced control circuitry 1000 for an M-level converter cell 1000 such as the generalized version depicted in FIG.8B. The M- level converter cell 1020 is shown coupled to an output block 1001 comprising an inductor L and an output capacitor COUT(conceptually, the inductor L also may be considered as being included within the M-level converter cell 1020). The advanced control circuitry 1000 functions as a control loop coupled to the output of the M-level converter cell 1020 and to switch control inputs of the M-level converter cell 1020. In general, the advanced control circuitry 1000 is configured to monitor the output (e.g., voltage and / or current) of the M-level converter cell 1020 and dynamically generate a set of switch control inputs to the M-level converter cell 1020 that attempt to stabilize the output voltage and / or current at specified values, taking into account variations of VIN and output load. In alternative embodiments, the advanced control circuitry 1000 may be configured to monitor the input of the M-level converter cell 1020 (e.g., voltage and / or current) and / or an internal node of the M-level converter cell 1020 (e.g., the voltage across one or more fly capacitors or the current through one or more power switches). Accordingly, most generally, the advanced control circuitry 1000Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 may be configured to monitor the voltage and / or current of a node (e.g., input terminal, internal node, or output terminal) of the M-level converter cell 1020. The advanced control circuitry 1000 may be incorporated into, or separate from, the overall controller for a power converter 100 embodying the M-level converter cell 1020.

[0108] A first block comprises a feedback controller 1002, which may be a traditional controller such as a fixed frequency voltage mode or current mode controller, a constant-ON- time controller, a hysteretic controller, or any other variant. The feedback controller 1002 is shown as being coupled to VOUTfrom the M-level converter cell 1020. In alternative embodiments, the feedback controller 1002 may be configured to monitor the input of the M- level converter cell 1020 and / or an internal node of the M-level converter cell 1020. The feedback controller 1002 produces a signal directly or indirectly indicative of the voltage at VOUT that determines in general terms what needs to be done in the multi-level converter cell 1020 to maintain desired values for VOUT: charge, discharge, or tri-state (i.e., open, with no current flow).

[0109] In the illustrated example, the feedback controller 1002 includes a feedback circuit 1004, a compensation circuit 1006, and a PWM generator 1008. The feedback circuit 1004 may include, for example, a feedback-loop voltage detector which compares VOUT (or an attenuatedversion of VOUT) to a reference voltage which represents a desired VOUTtarget voltage (whichmay be dynamic) and outputs a control signal to indicate whether VOUTis above or below the target voltage. The feedback-loop voltage detector may be implemented with a comparison device, such as an operational amplifier (op-amp) or transconductance amplifier (gm amplifier).

[0110] The compensation circuit 1006 is configured to stabilize the closed-loop response of the feedback controller 1002 by avoiding the unintentional creation of positive feedback, which may cause oscillation, and by controlling overshoot and ringing in the step response of the feedback controller 1002. The compensation circuit 1006 may be implemented in known manner, and may include LC and / or RC circuits.

[0111] The PWM generator 1008 generates the actual PWM control signal which ultimately sets the duty cycle of the switches of the multi-level converter cell 1020. In addition, in some embodiments, the PWM generator 1008 may pass on additional optional control signals CTRL indicating, for example, the magnitude of the difference between VOUT and theDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 reference voltage (thus indicating that some levels of the M-level converter cell 1020 should be bypassed to get to higher or lower levels), and the direction of that difference (e.g., whether VOUT is greater than or less than the reference voltage). In other embodiments, the optional control signals CTRL can be derived from the output of the compensation circuit 1006, or from the output of the feedback circuit 1004, or from a separate comparator (not shown) coupled to, for example, VOUT. One purpose of the optional control signals CTRL is for advanced control algorithms, when it may be beneficial to know how far away VOUTis from a target output voltage, thus allowing faster charging of the inductor L if the VOUTis severely under regulated.

[0112] A second block comprises a multi-level controller 1010, the primary function of which is to select the switch states that generate a desired VOUTwhile maintaining a charge- balance state on the fly capacitors within the M-level converter cell 1020 every time an output voltage level is selected, regardless of what switch state or states were used in the past.

[0113] The multi-level controller 1010 includes a Voltage Level Selector 1012 which receives the PWM control signal and the additional control signals CTRL if available. In addition, the Voltage Level Selector 1012 may be coupled to VOUT and / or VIN, and, in some embodiments, to the HIGH / LOW status signals, CFx_H / L, from the voltage detectors coupled to corresponding fly capacitors Cx within the M-level converter cell 1020. A function of the Voltage Level Selector 1012 is to translate the received signals to an output voltage Target Level (e.g., on a cycle-by-cycle basis). The Voltage Level Selector 1012 typically will consider at least VOUTand VINto determine which Target Level should charge or discharge the output of the M-level converter cell 1020 with a desired rate. For example, in a 6-level converter circuit, the available Target Levels are Level-1 (GND), Level-2 (1 / 5VIN), Level-3 (2 / 5VIN), Level-4 (3 / 5VIN), Level-5 (4 / 5VIN), and Level-6 (VIN), which may be represented as a count value from 1-6 (or 0-5).

[0114] As an example, in a 4-Level converter circuit, if VIN = 12V and VOUT nominally should be 3V, then the Voltage Level Selector 1012 may indicate that a Target Level of “2” can be selected, which results in a 1 / 3VIN voltage level at LX (i.e., 4V). The PWM control signal sets a duty cycle between that Target Level and another Target Level (e.g., GND) so that the average voltage level at LX will be about 3V.

[0115] In general, for steady-state operations, the Target Level voltage closest to VOUT that either charges or discharges the inductor L may be selected for simplicity of the selectionDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 algorithm. In general, for transient response, a Target Level that is higher (for charging) or lower (for discharging) than the closest Target Level may be selected to quickly charge or discharge the inductor L. The Voltage Level Selector 1012 may be implemented, for example, as a look-up table (LUT) or as comparison circuitry and combinatorial logic or more generalized processor circuitry. In some embodiments, the Voltage Level Selector 1012 can implement advanced methods (described below) that try to speed up charging or discharging based on additional factors, such as inductor voltage drop, load transients, the magnitude of output deviations, and / or external input signals from external sources. The output of the Voltage Level Selector 1012 may include duty cycle information (e.g., derived from the input PWM control signal) as well as switch state.

[0116] The output of the Voltage Level Selector 1012 is coupled to a Multi-Level Switch State Selector 1014, which generally would be coupled to the status signals, CFx_H / L, from the voltage detectors for the fly capacitors Cx. Taking into account the Target Level generated by the Voltage Level Selector 1012, the Multi-Level Switch State Selector 1014 determines a pattern of switch states for the desired output level that generally achieves charge-balancing the fly capacitors Cx. The Multi-Level Switch State Selector 1014 may be implemented, for example, as comparison circuitry and combinatorial logic, as a look-up table (LUT), or as more generalized processor circuitry. The output of the Multi-Level Switch State Selector 1014 is coupled to the switches of the multi-level converter cell 1020 (through appropriate level-shifter circuits and drivers circuits, as may be needed for a particular converter cell) and includes a pattern of switch state settings determined by the Multi-Level Switch State Selector 1014. The pattern of switch state settings selects the configuration of the switches within the multi-level converter cell 1020.

[0117] In general (but not always), for PWM-based control systems, the Voltage Level Selector 1012 and the M-level Switch State Selector 1014 only change their states when the PWM signal changes. For example, when the PWM signal goes high, the Voltage Level Selector 1012 selects which level results in charging of the inductor L and the M-level Switch State Selector 1014 sets which version to use of that level. Then when the PWM signal goes low, the Voltage Level Selector 1012 selects which level can discharge the inductor L and the M-level Switch State Selector 1014 sets which version of that level to use. Thus, the Voltage Level Selector 1012 and the M-level Switch State Selector 1014 generally only change states when the PWM signal changes (the PWM signal is in effect their clock signal). However, thereDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 may be situations or events where it is desirable for the CTRL signal to change the state of the Voltage Level Selector 1012. Further, there may be situations or events where it is desirable for the CFx_H / L status signal(s) to cause the M-level Switch State Selector 1014 to select a particular configuration of power switch settings, such as when a severe mid-cycle imbalance occurs. In some embodiments, it may be useful to include a timing function that forces the M- level Switch State Selector 1014 to re-evaluate the optimal version of the state periodically, for example, in order to avoid being “stuck” at one level for a very long time, potentially causing charge imbalances.

[0118] One notable benefit of the control circuitry shown in FIG. 10 is that it enables generation of voltages in boundary zones between voltage levels, which represent unattainable output voltages for conventional multi-level DC-to-DC converter circuits.

[0119] In alternative unregulated charge-pumps embodiments, the feedback controller 1002 and the Voltage Level Selector 1012 may be omitted, and instead a clock signal CLK may be applied to the M-level Switch State Selector 1014. The M-level Switch State Selector 1014 would generate a pattern of switch state settings that periodically charge balances the fly capacitors Cx regardless of what switch state or states were used in the past (as opposed to cycling through a pre-defined sequency of states). This ensures that if VIN changes or anomalous evens occur, the system generally always seeks charge balance for the fly capacitors Cx.

[0120] In some embodiments, the M-level Switch State Selector 1014 may take into account the current ILflowing through the inductor L by way of an optional current- measurement input 1016, which may be implemented in conventional fashion.

[0121] In an M-level multi-level converter circuit, the configuration of switches that achieves Level-1 (e.g., GND) or Level-M (e.g., VIN) effectively bypasses the fly capacitors Cx. Conversely, for all intermediate voltage levels, at least one fly capacitor Cx is coupled to VOUT and there are always at least two configurations of switches that can achieve any intermediate voltage level. For any particular intermediate voltage level, at least one configuration of switches results in charging the associated fly capacitor and at least one other configuration of switches results in discharging the associated fly capacitor. One aspect of the present disclosure is the realization that any achievable output voltage VOUT requiring intermediate voltage levels can be attained by dynamically selecting patterns of switch configurations – that is, by selectingDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 switch configurations without regard to or memory of the switch configurations of any previous switching cycle – to select appropriate Levels, and doing so in a way that purposefully selects either charging or discharging switch configurations that also balance charge across the fly capacitors Cx.

[0122] Embodiments of the disclosure use the following approach for positive inductor L current (charging VOUT): (1) a fly capacitor Cx that needs charging will be set to close its charging switch (the outer high-side switch in outer-switch control methods, or the inner low-side switch for inner-switch control methods); and (2) a fly capacitor Cx that needs discharging will be set to close its discharging switch (the outer low-side switch for outer-switch control methods, or the inner high-side switch for inner-switch control methods).

[0123] For negative inductor L current (discharging VOUT), the selection of switches inverts. Accordingly: (1) a fly capacitor Cx that needs charging will be set to close its charging switch (the outer low-side switch in outer-switch control methods, or the inner high-side switch for inner-switch control methods); and (2) a fly capacitor Cx that needs discharging will be set to close its discharging switch (the outer high-side switch for outer-switch control methods, or the inner low-side switch for inner-switch control methods).

[0124] Note again that whether or not charging actually occurs for a particular fly capacitor Cx generally depends on the switch states for all other fly capacitors. For a fly capacitor C(x)to actually charge or discharge, the next inward (if one exists) fly capacitor C(x 1) (for outer-switch control methods) or the previous outward (if one exists) fly capacitor C(x+1) (for inner- switch control methods) must be set to the opposite state (i.e., discharge or charge) so that a bypass situation does not occur.

[0125] For any multi-level converter circuit of order M that can create M voltage levels – i.e., Level-1 (e.g., GND) through Level-M (e.g., VIN) – then the following switch count rules apply for any Level-m: (1) M m low-side switches must be set to be closed (ON);Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 (2) m 1 high-side switches must be set to be closed (ON); and (3) switches that are not required to be ON must be set to be OFF (open).

[0126] With these switch count rules in mind, the following generalized capacitor control method applies for each state change of the Multi-Level Switch State Selector 1014: Step 1) Select a fly capacitor that has not previously been selected; Step 2) If the voltage of the selected fly capacitor is above its Vtarget and there are remaining (i.e., not been set by this method in this cycle) low-side or high-side switches that can be set to be closed to enable a discharge path for the selected fly capacitor, then set those switches that enable a discharge path for the selected fly capacitor to be closed, decrement one or more appropriate counters (e.g., for the number of low-side switches set to be closed and the number of high-side switches set to be closed), and flag the current fly capacitor as “done” (i.e., as having been selected); otherwise (since the voltage of the selected fly capacitor is below its Vtarget) set the switches that enable a charging path for the selected fly capacitor to be closed and flag the current fly capacitor as “done”; Step 3) Loop to Step 1 until all fly capacitors have been selected; Step 4) For the remaining pair of left-over switches, set the high-side switch or the low-side switch to be closed based on the switch count rules and the counter values.

[0127] With the above generalized capacitor control method, more specific multi-level charge-balancing control methods can be created. Examples can be found, for example, in U.S. Patent Publication No.20230148059, which is incorporated by reference herein in its entirety.

[0128] FIG. 11 is a block diagram 1100 of a controller, according to some embodiments. An exemplary controller may be controller 1002. Controller 1002 may regulate and control different system variables, including output voltage (VOUT), average current (IAVE), and peak current (IPEAK). An output voltage (VOUT) may be an output voltage of the multi-level converter. An average current (IAVE) may be a current measured across one or more switches of the multi- level converter over a predetermined amount of time. Peak current (IPEAK) may be a current measured across one or more switches of the multi-level converter at certain time. Multiple peak current (IPEAK) measurements may be used to determine the average current (IAVE) over a predetermined amount of time. In particular, controller 1002 may regulate and control different system variables of the multi-level power converter during different stages of the battery charging application.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0129] Controller 1002 illustrated in FIG. 11 may include multiple amplifiers, such as a first amplifier and a second amplifier. The first amplifier may be a voltage error amplifier 1102 and a second amplifier may be an average current error amplifier 1104. Voltage error amplifier 1102 and average current error amplifier 1104 may be operational transconductance amplifiers. Voltage error amplifier 1102 may determine an error between voltage inputs, such as output voltage (VOUT) that may be an output voltage of the multi-level converter and a reference voltage that may be sensed using a sensor or a load. Average current error amplifier 1104 may be a type of current error amplifier that determines an error between two inputs representing currents. In case of average current error amplifier 1104, the inputs may represent an average current (IAVE) that passes through the multi-level converter and may be regulated and a reference average current that may be sensed using one of the sensors. Additionally, controller 1002 may include a comparator, such as pulse width modulated (PWM) comparator 1106. A comparator may compare two inputs and generate an output indicating which input is greater. The PWM comparator 1106 may be a comparator that compares the two inputs and generates a PWM signal that may be used to set a duty cycle. In case of PWM comparator 1106 the inputs may represent the peak current (IPEAK) that may be regulated and the sensed peak current (IPEAK) that may be sensed using one or more sensors. The voltage error amplifier 1102, the average current error amplifier 1104, and the PWM comparator 1106 may be arranged in sequence, such that an output signal of the voltage error amplifier 1102 may be an input signal into the average current error amplifier 1104, and an output signal of the average current error amplifier 1104 may be an input signal into PWM comparator 1106.

[0130] In some embodiments, voltage error amplifier 1102, average current error amplifier 1104, and PWM comparator 1106 may receive a signal representing a system variable to be regulated and a signal representing a reference value for the system variable that may be sensed using one or more sensors. For example, voltage error amplifier 1102 may receive a system variable that may be an output voltage (VOUT) 1108A and a reference voltage (VREF) 1110 and determine an error between output voltage (VOUT) 1108A and a reference voltage (VREF) 1110. Output voltage (VOUT) may be an output voltage to be regulated by the multi-level converter. The reference voltage (VREF) 1110 may be a reference voltage that is a voltage across a load, e.g., a battery. The voltage error amplifier 1102 may use output voltage (VOUT) 1108A and reference voltage (VREF) 1110 to generate an output signal. The output signal may correspond to an error between output voltage (VOUT) 1108A and reference voltage (VREF) 1110 and represent an average current (IAVG) and may be referred to as a Vamp_Iavg 1112 signal. TheDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 Vamp_Iavg signal 1112 may represent a minimum of the output voltage (VOUT) 1108A and reference voltage (VREF) 1110 or an error between output voltage (VOUT) 1108A and reference voltage (VREF) 1110, and may correspond to an average current (IAVG) to be regulated by the multi-level converter. The value corresponding to the Vamp_Iavg signal 1112 may be measured at the comparison node or comp node 1115. When comp node 1115 represents an average current, the error between output voltage (VOUT) 1108A and reference voltage (VREF) 1110 may be an indication to controller 1002 to increase the average current (IAVG).

[0131] In some embodiments, voltage error amplifier 1102 may be included in a circuit referred to as a voltage control loop 1124. Voltage control loop 1124 may control the output voltage (VOUT) generated by the multi-level converter. Typically, voltage control loop 1124 may be a more precise but a slow loop when compared to other loops in controller 1002.

[0132] As discussed above, Vamp_Iavg signal 1112 that is an output of voltage error amplifier 1102 may be an input to average current error amplifier 1104. For example, average current error amplifier 1104 may receive the Vamp_Iavg signal 1112 that may correspond to an average current (IAVG) to be regulated and an average sensed current (ISENSE_AVG) signal 1114 and determine an error between average current (IAVG) and average sensed current (ISENSE_AVG). The average sensed current (ISENSE_AVG) may be sensed using a sensor that senses an average current across switches of the multi-level converter. For example, switches S3 and S4 of the 3-level converter discussed in FIG.8A and switches S5 and S6 of the 4-level converter discussed in FIG.8B may be sensed to determine the average sensed current (ISENSE_AVG).

[0133] The average current error amplifier 1104 may use Vamp_Iavg signal 1112 and average sensed current (ISENSE_AVG) signal 1114 to generate an output signal that corresponds to an error between average current (IAVG) and average sensed current (ISENSE_AVG). The output signal may be a Vamp_Ipeak signal 1116 representing a peak current (IPEAK). The peak current (IPEAK) may be a peak current to be regulated by the multi-level converter. The Vamp_Ipeak signal 1116 may be a minimum of or an error between the average current (IAVG) corresponding to the Vamp_Iavg signal 1112 and an average sensed current (ISENSE_AVG) corresponding to average sensed current (ISENSE_AVG) signal 1114. The value corresponding to the Vamp_Ipeak signal 1116 may be measured at the comparison node or comp node 1118. The error between the average current (IAVG) and the sensed average current (ISENSE_AVG) may be an indication to controller 1002 to increase the peak current (IPEAK).Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0134] In some embodiments, average current error amplifier 1104 may be included in a circuit referred to as a current control loop 1126. The current control loop 1126 may regulate the average current (IAVG) used by a system, such as the multi-level converter. Typically, current control loop 1126 may be a less precise but a faster loop when compared to other loops in controller 1002, such as voltage control loop 1124.

[0135] As discussed above, Vamp_Ipeak signal 1116 that is an output of average current error amplifier 1104 may be an input to PWM comparator 1106. For example, PWM comparator 1106 may receive Vamp_Ipeak signal 1116 that may correspond to the peak current (IPEAK) to be regulated and a sensed peak current (ISENSE_PEAK) signal 1120. PWM comparator 1106 may compare the peak current (IPEAK) to the sensed peak current (ISENSE_PEAK) and generate a PWM signal based on the comparison. The pulse width modulation signal may set the duty cycle. The sensed peak current (ISENSE_PEAK) may be sensed using sensors that senses current across switches of multi-level converters discussed in FIGs. 8A and 8B. For example, switches S3 and S4 of the 3-level converter discussed in FIG. 8A and switches S5 and S6 of the 4-level converter discussed in FIG. 8B may be sensed to determine the average sensed current (ISENSE_PEAK). In some instances the sensed peak current (ISENSE_PEAK) may also be mixed with a slope compensation waveform. The slope compensation waveform may be a triangular or sawtooth waveform, ramp voltage waveform, or a portion of the ramp voltage waveform and may mixed with the sensed peak current (ISENSE_PEAK) to make the sensed peak current (ISENSE_PEAK) more stable.

[0136] The PWM comparator 1106 may use Vamp_Ipeak signal 1116 and sensed peak current (ISENSE_PEAK) signal 1120 to generate a PWM signal 1122. The PWM signal 1122 may be a pulse width modulation signal that indicates whether the peak current (IPEAK) corresponding to Vamp_Ipeak signal 1116 or sensed peak current (ISENSE_PEAK) corresponding to the sensed peak current (ISENSE_PEAK) signal 1120 is greater. PWM signal 1122 may be a digital signal that may set a duty cycle as discussed in FIG.10. For example, PWM signal 1122 may set a duty cycle when the multi-level converter circuit changes zones, such as from a first zone to a second zone, from the second zone to a third zone, from the third zone to a fourth zone, or vice versa.

[0137] In some embodiments, PWM comparator 1106 may be included in a circuit referred to as a peak current control loop 1128. The peak current control loop 1128 may regulate the peak current (IPEAK) used by a system, such as the multi-level converter.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0138] Going back to voltage error amplifier 1102 in voltage control loop 1124. As discussed above, the reference voltage (VREF) 1110 may be a target voltage to which voltage control loop 1124 may regulate input voltages, such as output voltage (VOUT) 1108A, a battery voltage (VBATTERY) 1108B, and an input voltage (VIN) 1108C. To select among multiple voltage inputs 1108A-C, controller 1002 may include a monitor 1132. Monitor 1132 may monitor voltage inputs 1108A-C and include a minimum select that may select a voltage input from voltage inputs 1108A-C having the lowest voltage. Monitor 1132 may pass the voltage input with the lowest voltage to voltage error amplifier in voltage control loop 1124. In some instances, voltage inputs 1108A-C may be digital voltage signals, in which case, monitor 1132 may include a digital-to-analog converter that may convert the digital voltage signals into analog voltage inputs 1108A-C.

[0139] Going back to the average current error amplifier 1104 in current control loop 1126. As discussed above, the average sensed current (ISENSE_AVG) may be received at current control loop 1126 using average sensed current (ISENSE_AVG) signal 1114. In some embodiments, controller 1002 may receive multiple input current signals that may represent average sensed current (ISENSE_AVG). Example input current signals may be input current (IIN) signal 1134A representing input current (IIN) or output current (IOUT) signal 1134B representing output current (IOUT). To select among multiple input current signals 1134A-B, controller 1002 may include a monitor 1136. Monitor 1136 may include a minimum selector which may select an input current signal from input current signals 1134A-B that corresponds to the lowest current. Monitor 1136 may pass the selected input signal that represents the lowest current as average sensed current (ISENSE_AVG) 1114 to average current error amplifier 1104 in current control loop 1126. In some embodiments, one or more input current signals 1134A-B may be digital signals. In this case, monitor 1136 may convert the digital current signals to analog input current signals 1134A-B and then select the current input that corresponds to the lowest current.

[0140] Selecting a voltage input that has a lowest voltage and a current input that has a lowest current has multiple benefits. In the multi-level converter, multiple voltage inputs 1108A-C and multiple input current signals 1134A-B are monitored. For example, battery voltage (VBATTERY) 1108B is monitored to make sure the battery is not overcharged, the output voltage (VOUT) 1108A of a converter is monitored, which typically has a higher voltage than battery voltage (VBATTERY) 1108B, to make sure it does not exceed a maximum allowable voltage threshold and damages components of the multi-level converter. In another example,Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 output current (IOUT) is monitored to make sure the output current (IOUT) does not exceed a maximum current that a battery can handle, and input current (IIN) is monitored to make sure the input current (IIN) does not overload a power supply unit. Accordingly, with multiple voltage inputs, the voltage may increase as long as the voltage is not above the maximum value of one of the voltage inputs 1108A-C. Similarly, the input current signals 1134A-B may increase the input current as long as the input current is not above the maximum value of one of the input current signals 1134A-B. Further, the input current signals 1134A-B may further limit the voltage inputs 1108A-C when Vamp_Iavg signal 1112 is higher than Isense_avg signal 1114.

[0141] FIG. 12 is an example method 1200 of operating a controller, in accordance with one or more embodiments. Method 1200 may be implemented using components and circuits discussed in FIGs.1-13.

[0142] In operation 1202, voltage inputs are received and selected. For example, monitor 1132 may receive voltage inputs 1108A-C, convert voltage inputs 1108A-C into analog voltage signals as needed, and select a voltage input corresponding to the lowest voltage as an input to voltage control loop 1124.

[0143] In operation 1204, a selected input voltage and a target voltage are received to generate a first signal. For example, first amplifier, such as voltage error amplifier 1102 of the voltage control loop 1124 may receive the input voltage selected in operation 1202 and reference voltage (VREF) 1110 and generate a first signal, such as Vamp_Iavg signal 1112. Vamp_Iavg signal 1112 may indicate an error between the input voltage selected in operation 1202 and reference voltage (VREF) 1110, and may be a minimum of or an error between the selected input voltage and reference voltage (VREF) 1110. In some instances, the Vamp_Iavg 1112 may represent an average current (IAVG).

[0144] In operation 1206, current inputs are received and selected. For example, monitor 1136 may receive input current signals 1134A-B. If input current signals are digital signals, a DAC may first convert the digital signals into analog input current signals 1134A-B. Monitor 1136 may include a minimum selector that selects an input current signal from input current signals 1134A-B corresponding to the lowest current as a signal representing target current, e.g., average sensed current (ISENSE_AVG) signal 1114.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0145] At operation 1208, a first signal and a signal representing a target current are received to generate a second signal. For example, a second amplifier, which may be average current error amplifier 1104 of current control loop 1126, may receive the first signal, which may be Vamp_Iavg signal 1112 that represents an average current (IAVG) and average sensed current (ISENSE_AVG) signal 1114 that represents an average sensed current (ISENSE_AVG) to generate a second signal. The second signal may be Vamp_Ipeak signal 1116. The second signal may indicate an error between the first signal and the average sensed current (ISENSE_AVG) signal 1114. An example error may be between the average current (IAVG) corresponding to the first signal and average sensed current (ISENSE_AVG). In some instances, the second signal, e.g., Vamp_Ipeak signal 1116, may represent a peak (IPEAK) current to be regulated by the multi- level converter.

[0146] At operation 1210, a second signal and a reference signal, such as Isense_peak signal 1120 are received to generate a third signal. For example, PWM comparator 1106 of peak current control loop 1128 may receive a second signal, e.g., Vamp_Ipeak signal 1116 which represents (IPEAK) and reference signal, which may be sensed peak current (ISENSE_PEAK) signal 1120 to generate a third signal, which may be PWM signal 1122. PWM signal 1122 may be a minimum of second signal 1116 and Isense_peak signal 1120, and may set a duty cycle.

[0147] FIG. 13 is a block diagram 1300 of a circuit for using current to limit a voltage mode control, according to some embodiments. FIG.13 illustrates a voltage control loop 1302 that may set a duty cycle. Voltage control loop 1302 may receive voltage inputs, such as V11304A through VN1304N. As discussed above, some of the voltage inputs may be input voltage (VIN), output voltage (VOUT), and battery voltage (VBATTERY). Voltage control loop 1302 may have similar circuits as voltage control loop 1124 discussed in FIG.11. In some embodiments, voltage control loop 1302 may select a voltage input from voltage inputs 1304A-1304N that corresponds to the lowest voltage. Accordingly, the output voltage signal 1305 may correspond to the lowest voltage input in voltage inputs 1304A-1304N.

[0148] FIG. 13 illustrates a current control loop 1306 that may be in parallel with the voltage control loop 1302. Current control loop 1306 may receive current inputs, such as I1 1308A through IN 1308N. As discussed above, some of the current inputs may be input current (IIN) and output current (IOUT). Current control loop 1306 may have similar circuits as current control loop 1126 discussed in FIG.11. In some embodiments, current control loop 1306 may select a current input from current inputs 1308A-1308N that corresponds to the lowest current.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 Accordingly, an output current signal 1310 may correspond to the lowest current from current inputs 1308A-1308N.

[0149] In some embodiments, current control loop 1306 may bring the duty cycle set by the voltage control loop 1302 down. For example, one of current inputs 1308A-1308N, such as current input 1308N may be set to a predefined current threshold, such as 5 amperes, which sets a maximum input current. Accordingly, as long as one of current inputs 1308A-1308M is below the predefined current threshold amperes, current control loop 1306 may pass the lowest current input through. However, if current inputs 1308A-1308M are above the predefined current threshold, current input 1308N may pass through, and be the maximum allowable current input that may pass through current control loop 1306. Accordingly, the output current signal 1310 may correspond to the lowest current input in current inputs 1308A-1308N as long as the lowest current input is below the predefined current threshold.

[0150] In some embodiments, output voltage signal 1305 and output current signal 1310 may be combined or “or”-ed at node 1312 into a combined output signal 1314. Because output voltage signal 1305 and output current signal 1310 are “or”-ed and the output current signal 1310 is limited by the predefined current threshold, the combined output signal 1314 may be limited by output current signal 1310.

[0151] The voltage mode control 1316 receives the combined output signal 1314. The voltage mode control 1316 may set a duty cycle and hence voltage levels (VIN, GND, VIN, and VIN) based on the combined output signal 1314 that is limited by the output current signal 1310 which may correspond to a maximum allowable current set using predefined current threshold.

[0152] FIG.14 is an example method 1400 of using current to limit a voltage mode control, in accordance with one or more embodiments. Method 1400 may be implemented using components and circuits discussed in FIGs.1-13.

[0153] In operation 1402, voltage inputs are received and selected. For example, voltage control loop 1302 may receive voltage inputs 1304A-N, and select a voltage input corresponding to the lowest voltage as output voltage signal 1305. In some cases, voltage control loop 1302 may set a duty cycle.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0154] In operation 1404, current inputs are received and selected. For example, current control loop 1306 that is in parallel with voltage control loop 1302 may receive input current signals 1308A-N and select an input current signal corresponding to the lowest current as output current signal 1310. As discussed above, one of input current signals 1308A-N, such as current input 1308N, may be set to a predefined current threshold that causes the output current signal 1310 to be limited to the predefined current threshold. In other words, output current signal 1310 may represent the lowest current from input current signals 1308A-N, which is lower or equal to the predefined current threshold.

[0155] At operation 1406, an output signal from the voltage control loop and an output signal from the current control loop are combined. For example, output voltage signal 1305 is “or”-ed with output current signal 1310 at node 1312 into combined output signal 1314. Because output current signal 1310 may correspond to a maximum allowable current, output voltage signal 1305 may be controlled using output current signal 1310.

[0156] At operation 1408, a mode is selected. For example, the voltage mode control 1316 may receive the combined output signal 1314 and set a duty cycle based on the combined output signal 1314. In some embodiments, setting the duty cycle may also set the voltage levels, such as VIN, GND, VIN, and VIN.

[0157] Going back to FIG.11, in some embodiments, error amplifiers, such as voltage error amplifier 102 and average current error amplifier 1104, may be implemented using analog circuits. FIG.15 is a block diagram 1500 of an error amplifier, according to some embodiments. FIG.15 illustrates error amplifier 1501. Error amplifier 1501 may include multiple transistors, such as mosfets, including p-mosfets. Error amplifier 1501 may receive input signals, shown as input signals VPOS1502 and VNEG1504, that may represent voltage inputs or current inputs, and generate output signal 1506 that may represent an output current (IOUT). Input signals VPOS1502 and VNEG 1504 may be received at the transistors, with the transistor associated with VNEG 1504 being flipped. The output current may be based on a difference between input signals VPOS 1502 and VNEG 1504. Additionally, error amplifier 1501 may regulate an output voltage (VOUT) 1508, which is limited to a voltage between voltage VDD 1512 and voltage VSS 1510. The output voltage (VOUT) 1508 may be driven up or down per the polarity of the output current (IOUT).Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0158] Graphs 1514-1518 illustrate the behavior of the error amplifier 1501 based on input signals VPOS 1502 and VNEG 1504. Graph 1514 illustrates input signals VPOS 1502 and VNEG 1504. Graph 1516 illustrates output current (IOUT). Graph 1518 illustrates output voltage (VOUT). As illustrated in graphs 1514-1518, when input signal VPOS1502 is greater than input signal VNEG1504, the output current (IOUT) is negative and output voltage (VOUT) is set to VSS, and when input signal VNEG1504 is greater than input signal VPOS1502, the output current (IOUT) is positive and output voltage (VOUT) is set to VDD.

[0159] FIGs. 16A-B are block diagrams 1600A-B of a multi-input error amplifier, according to some embodiments. FIG.16A illustrates multi-input error amplifier 1601. Multi- input error amplifier 1601 may include multiple transistors, such as mosfets, including p- mosfets. Multi-input error amplifier 1601 may receive input signals, shown as input signals VPOS11602A, VPOS21602B and VNEG 1604. Input signals VPOS11602A, VPOS21602B and VNEG 1604 may be received at the transistors, with the transistor associated with VNEG 1604 being flipped. Multi-input error amplifier 1601 may generate output signal 1606 that may represent an output current (IOUT). Unlike error amplifier 1501 above, multi-input error amplifier 1601 may receive multiple input signals VPOS11602A and VPOS21602B. Multi-input error amplifier 1601 may select input signal from input signals VPOS11602A or VPOS21602B that has the lower value (e.g., lower current or lower voltage), which corresponds to the lower charge Q across the transistors. Multi-input error amplifier 1601 then determines the error between input signal VNEG1604 and the selected input signal VPOS11602A or VPOS21602B that has the lower value. The output current (IOUT) signal 1606 may be based on a difference between input signal VPOS11502A or VPOS21502B and VNEG1504. Additionally, multi-input error amplifier 1601 may regulate an output voltage (VOUT) signal 1608, which is limited to a voltage between voltage VSS 1610 and voltage VDD 1612. The output voltage (VOUT) may be driven up or down per the polarity of the output current (IOUT) represented in output signal 1606.

[0160] FIG.16B illustrates a multi-input error amplifier 1601 operating when input signal VPOS11602A is greater than input signal VPOS21602B, and when input signal VPOS11602A is less than input signal VPOS2 1602B. For example, when input signal VPOS11602A is greater than input signal VPOS21602B, the current 1614 flows through the transistor corresponding to VPOS21602B, generating output current (IOUT) 1606A that is a difference between input signal VPOS21602B and input signal VNEG 1604. In another example, when input signal VPOS11602A is less than input signal VPOS2 1602B, the current 1614 flows through the transistorDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 corresponding to VPOS1 1602A, generating output current (IOUT) 1606B that is a difference between input signal VPOS11602B and input signal VNEG 1604.

[0161] FIG. 16C is a diagram of graphs illustrating current flowing through transistors corresponding to input signals VPOS11602A and VPOS21602B. Graph 1615 illustrates the multi- input error amplifier selecting between input signal VPOS11602A and input signal VPOS21602B that corresponds to a lower value, e.g. value for current or voltage, as illustrated by the dashed line 1616. Graph 1618 illustrates the current 1614A that flows through a mosfet corresponding to input signal VPOS1, and current 1614B that flows through the mosfet corresponding to input signal VPOS2. For example, when input signal VPOS11602A is greater than input signal VPOS21602B, current 1614B flows through transistor corresponding to input signal VPOS21602B, while current 1614A is or around zero. When input signal VPOS11602A is less than input signal VPOS2 1602B, current 1614A flows through transistor corresponding to input signal VPOS1 1602A, while current 1614B is or around zero. Currents 1614A and current 1614B begin to switch when the values of input signals VPOS11602A and VPOS21602B are close to each other.

[0162] FIGs.17A-C are block diagrams 1700A-C illustrating an analog implementation of a voltage control loop and a current control loop discussed in FIG. 11, according to some embodiments. FIG. 17A illustrates error amplifier 1501 coupled in sequence to multi-input error amplifier 1601. Error amplifier 1501 receives input voltages that are full battery voltage (VFULL_BATTERY) 1702 and battery voltage (VBATTERY) 1704. The full battery voltage (VFULL_BATTERY) 1702 may be reference voltage (VREF) 1110 and battery voltage (VBATTERY) 1704 may be output voltage (VOUT) 1108A discussed in FIG. 11. Error amplifier 1501 may generate an output current 1706 and output signal 1708 that may correspond to Vamp_Iavg signal 1112 in FIG.11.

[0163] Multi-input error amplifier 1601 may receive output signal 1708 (Vamp_Iavg signal 1112) as input signal VPOS11602A, and input current signal 1710 (shown as V(Iavg_target)) as input signal VPOS21602B. Additionally, multi-input error amplifier 1601 may receive input signal V(Isense_Iavg ) as input signal 1604. Multi-input error amplifier 1601 may select either input signal VPOS1 1602A or input signal VPOS2 1602B, whichever one represents a lower current value, and then compare the selected input signal with input signal VNEG 1604, to generate an output current (IOUT) signal 1606 and output signal 1608. The output signal 1608 may correspond to Vamp_Ipeak 1116 in FIG.11.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0164] FIG. 17B illustrates a scenario where input signal VPOS11602A, which is output signal 1708 (Vamp_Iavg signal 1112) is greater than input signal VPOS21602B (which is input current signal V(Iavg_target) 1710) causing multi-input error amplifier 1601 to select input signal VPOS21602B. Multi-input error amplifier 1601 then determines an error between input signal VPOS21602B and input signal VNEG1604 (input signal V(Isense_Iavg)), which is shown as output signal 1608.

[0165] FIG. 17C illustrates a scenario where input signal VPOS21602B, which is input current signal V(Iavg_target) 1710 is greater than input signal VPOS11602A, which is output signal 1708 (Vamp_Iavg signal 1112) causing multi-input error amplifier 1601 to select input signal VPOS11602A. Multi-input error amplifier 1601 then determines an error between input signal VPOS11602A and input signal VNEG 1604 (input signal V(Isense_Iavg) 1114), which is shown as output signal 1608. In this way, input signal VPOS2 1602B (input current signal V(Iavg_target) 1710) may limit output signal 1708 (Vamp_Iavg signal 1112) generated using error amplifier 1501.

[0166] Notably, although FIGs. 17A-C illustrate error amplifier 1501 coupled to multi- input error amplifier 1601, other embodiments may also be implemented, such as multi-input error amplifier 1601 coupled to error amplifier 1501, or multi-input error amplifier 1601 coupled to another multi-input error amplifier 1601.

[0167] FIG. 18 is an example method 1800 of using inputs from a multi-input error amplifier to limit an output from an error amplifier, in accordance with one or more embodiments. Method 1800 may be implemented using components and circuits discussed in FIGs.1-17.

[0168] In operation 1802, input voltages are received at an error amplifier and an output signal is generated. For example, input voltages 1702 and 1704 are received at error amplifier 1501, and output signal 1708 representing an error between input voltages 1702 and 1704 is generated. As discussed above, input voltages 1702 and 1704 may represent full battery voltage (VFULL_BATTERY) and battery voltage (VBATTERY) and output signal 1708 may represent Vamp_Iavg signal 1112.

[0169] At operation 1804, first input current signals and a second input current signal are received. For example, multi-input error amplifier 1601 may receive input signals 1602A and 1602B. Input signal VPOS11602A may be output signal 1708 (corresponding to Vamp_IavgDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 signal 1112) and input signal VPOS21602B may be input current signal V(Iavg_target) 1710. Multi-input error amplifier 1601 may also receive a second input, such as input current signal 1604 which may be average sensed current (ISENSE_AVG) signal 1114.

[0170] At operation 1806, one of the first input current signals is selected. For example, multi-input error amplifier 1601 may select input signal VPOS11602A or 1602B, depending on which input signal represents a lower amount of current.

[0171] At operation 1808, an error is generated. For example, multi-input error amplifier 1601 may determine an error between the selected input signal VPOS11602A or 1602B and the input current signal 1604.

[0172] Further aspects of the present disclosure include the following:

[0173] Aspect 1 includes a system, comprising: a first monitor configured to receive a plurality of voltage inputs and select a voltage input having a lowest voltage from the plurality of voltage inputs; a voltage control loop configured to receive the selected voltage input and a reference voltage, and generate a first signal; a second monitor configured to receive a plurality of current inputs and select a current input having a lowest current from the plurality of current inputs as a target current; and a current control loop configured to receive the first signal and the target current and generate a second signal, wherein the second signal corresponds to a lesser of the first signal and the target current.

[0174] Aspect 2 includes the system of aspect 1, wherein the first signal corresponds to an average current that is regulated at one or more switches of a power converter.

[0175] Aspect 3 includes the system of any of aspects 1-2, wherein the first monitor comprises a digital-to-analog converter and the selected voltage input is an analog voltage.

[0176] Aspect 4 includes the system of any of aspects 1-3, wherein the plurality of volt- age inputs are one or more of an output voltage of a power converter, an input voltage to the power converter, or a battery voltage.

[0177] Aspect 5 includes the system of any of aspects 1-4, wherein the second monitor comprises a digital-to-analog converter and the target current is an analog current.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0178] Aspect 6 includes the system of any of aspects 1-5, wherein the plurality of cur- rent inputs are an input current to a power converter and an output current of the power con- verter.

[0179] Aspect 7 includes the system of any of aspects 1-6, wherein the first monitor com- prises a minimum selector to select the voltage input that has the lowest voltage.

[0180] Aspect 8 includes the system of any of aspects 1-7, wherein the second signal cor- responds to a peak current that is regulated at one or more switches of a power converter.

[0181] Aspect 9 includes a system comprising: a current control loop configured to re- ceive a plurality of current inputs and select a first signal, wherein the first signal represents a current input having a lowest current; a voltage control loop configured to receive a plurality of voltage inputs and select a second signal, wherein the second signal represents a voltage input having a lowest voltage; a selector configured to combine the first signal and the second signal into a combined signal; and a mode control configured to select a mode based on the combined signal, wherein the mode corresponds to one or more voltage levels.

[0182] Aspect 10 includes the system of aspect 9, wherein the current control loop is par- allel to the voltage control loop.

[0183] Aspect 11 includes the system of any of aspects 9-10, wherein one of the plurality of current inputs includes a predefined current that corresponds to a maximum current that the first signal can represent.

[0184] Aspect 12 includes the system of any of aspects 9-11, wherein the plurality of voltage inputs are one or more of an output voltage of a power converter, an input voltage to the power converter, or a battery voltage.

[0185] Aspect 13 includes a system comprising: a multi-input error amplifier configured to: receive a plurality of input current signals representing a plurality of target currents and a second input current signal; select an input current signal from the plurality of input current signals representing a lowest current; and generate an output signal representing an error be- tween the selected input current signal and the second input current signal.

[0186] Aspect 14 includes the system of aspect 13, wherein the plurality of input current signals correspond to target currents to be regulated.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0187] Aspect 15 includes the system of any of aspects 13-14, wherein the second input current signal corresponds to a sensed average current that is sensed at one or more sensors of a power converter.

[0188] Aspect 16 includes the system of any of aspects 13-15, wherein the multi-input er- ror amplifier comprises a maximum output voltage and a minimum output voltage, and wherein the multi-input error amplifier is configured to generate an output voltage that is be- tween the maximum output voltage and the minimum output voltage.

[0189] Aspect 17 includes the system of any of aspects 13-16, wherein the multi-input er- ror amplifier is included in a current control loop.

[0190] Aspect 18 includes the system of any of aspects 13-17, further comprising: a volt- age control loop configured to: receive a plurality of voltage inputs; and generate an output signal representing an error between the plurality of voltage inputs, wherein the output signal is one of the plurality of input current signals to the multi-input error amplifier.

[0191] Aspect 19 includes the system of any of aspects 13-18, wherein the voltage con- trol loop includes a voltage error amplifier configured to generate the output signal.

[0192] Aspect 20 includes the system of any of aspects 13-19, wherein the output signal represents an average current to be regulated at one or more switches by a power converter.

[0193] General Benefits and Advantages of Multi-Level Power Converters

[0194] Embodiments of the current invention improve the power density and / or power efficiency of incorporating circuits and circuit modules or blocks. As a person of ordinary skill in the art should understand, a system architecture is beneficially impacted utilizing em- bodiments of the current invention in critical ways, including lower power and / or longer bat- tery life. The current invention therefore specifically encompasses system-level embodiments that are creatively enabled by inclusion in a large system design and application.

[0195] More particularly, multi-level power converters provide or enable numerous bene- fits and advantages, including:

[0196] – adaptability to applications in which input and / or output voltages may have a wide dynamic-range (e.g., varying battery input voltage levels, varying output voltages);Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0197] – efficiency improvements on the run-time of devices operating on portable elec- trical energy sources (batteries, generators or fuel cells using liquid or gaseous fuels, solar cells, etc.);

[0198] – efficiency improvements where efficiency is important for thermal management, particularly to protect other components (e.g., displays, nearby ICs) from excessive heat;

[0199] – enabling design optimizations for power efficiency, power density, and form- factor of the power converter – for example, smaller-size multi-level power converters may allow placing power converters in close proximity to loads, thus increasing efficiency, and / or to lower an overall bill of materials;

[0200] – the ability to take advantage of the performance of smaller, low voltage transis- tors;

[0201] – adaptability to applications in which power sources can vary widely, such as batteries, other power converters, generators or fuel cells using liquid or gaseous fuels, solar cells, line voltage (AC), and DC voltage sources (e.g., USB, USB-C, power-over Ethernet, etc.);

[0202] – adaptability to applications in which loads may vary widely, such as ICs in gen- eral (including microprocessors and memory ICs), electrical motors and actuators, transduc- ers, sensors, and displays (e.g., LCDs and LEDs of all types);

[0203] – the ability to be implemented in a number of IC technologies (e.g., MOSFETs, GaN, GaAs, and bulk silicon) and packaging technologies (e.g., flip chips, ball-grid arrays, wafer level scale chip packages, wide-fan out packaging, and embedded packaging).

[0204] The advantages and benefits of multi-level power converters enable usage in a wide array of applications. For example, applications of multi-level power converters include portable and mobile computing and / or communication products and components (e.g., note- book computers, ultra-book computers, tablet devices, and cell phones), displays (e.g., LCDs, LEDs), radio-based devices and systems (e.g., cellular systems, WiFi, Bluetooth, Zigbee, Z- Wave, and GPS-based devices), wired network devices and systems, data centers (e.g., for battery-backup systems and / or power conversion for processing systems and / or electronic / op- tical networking systems), internet-of-things (IOT) devices (e.g., smart switches and lights,Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 safety sensors, and security cameras), household appliances and electronics (e.g., set-top boxes, battery-operated vacuum cleaners, appliances with built-in radio transceivers such as washers, dryers, and refrigerators), AC / DC power converters, electric vehicles of all types (e.g., for drive trains, control systems, and / or infotainment systems), and other devices and systems that utilize portable electricity generating sources and / or require power conversion.

[0205] Radio system usage includes wireless RF systems (including base stations, relay stations, and hand-held transceivers) that use various technologies and protocols, including various types of orthogonal frequency-division multiplexing (“OFDM”), quadrature ampli- tude modulation (“QAM”), Code-Division Multiple Access (“CDMA”), Time-Division Mul- tiple Access (“TDMA”), Wide Band Code Division Multiple Access (“W-CDMA”), Global System for Mobile Communications (“GSM”), Long Term Evolution (“LTE”), 5G, and WiFi (e.g., 802.11a, b, g, ac, ax), as well as other radio communication standards and protocols.

[0206] Programmable Embodiments

[0207] Some or all aspects of the invention, particularly the Multi-Level Switch State Se- lector 1014 of FIG.10, may be implemented in hardware or software, or a combination of both (e.g., programmable logic arrays). Unless otherwise specified, the algorithms included as part of the invention are not inherently related to any particular computer or other appa- ratus. In particular, various general purpose computing machines may be used with programs written in accordance with the teachings herein, or it may be more convenient to use a special purpose computer or special-purpose hardware (such as integrated circuits) to perform partic- ular functions. Thus, embodiments of the invention may be implemented in one or more com- puter programs (i.e., a set of instructions or codes) executing on one or more programmed or programmable computer systems (which may be of various architectures, such as distributed, client / server, or grid) each comprising at least one processor, at least one data storage system (which may include volatile and non-volatile memory and / or storage elements), at least one input device or port, and at least one output device or port. Program instructions or code may be applied to input data to perform the functions described in this disclosure and generate output information. The output information may be applied to one or more output devices in known fashion.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3

[0208] Each such computer program may be implemented in any desired computer lan- guage (including machine, assembly, or high-level procedural, logical, or object-oriented pro- gramming languages) to communicate with a computer system, and may be implemented in a distributed manner in which different parts of the computation specified by the software are performed by different computers or processors. In any case, the computer language may be a compiled or interpreted language. Computer programs implementing some or all of the in- vention may form one or more modules of a larger program or system of programs. Some or all of the elements of the computer program can be implemented as data structures stored in a computer readable medium or other organized data conforming to a data model stored in a data repository.

[0209] Each such computer program may be stored on or downloaded to (for example, by being encoded in a propagated signal and delivered over a communication medium such as a network) a tangible, non-transitory storage media or device (e.g., solid state memory media or devices, or magnetic or optical media) for a period of time (e.g., the time between refresh periods of a dynamic memory device, such as a dynamic RAM, or semi-permanently or per- manently), the storage media or device being readable by a general or special purpose pro- grammable computer or processor for configuring and operating the computer or processor when the storage media or device is read by the computer or processor to perform the proce- dures described above. The inventive system may also be considered to be implemented as a non-transitory computer-readable storage medium, configured with a computer program, where the storage medium so configured causes a computer or processor to operate in a spe- cific or predefined manner to perform the functions described in this disclosure.

[0210] Fabrication Technologies & Options

[0211] In various embodiments of multi-level power converters, it may be beneficial to use specific types of capacitors, particularly for the fly capacitors. For example, it is generally useful for such capacitors to have low equivalent series resistance (ESR), low DC bias degra- dation, high capacitance, and small volume. Low ESR is especially important for multi-level power converters that incorporate additional switches and fly capacitors to increase the num- ber of voltage levels. Selection of a particular capacitor should be made after consideration of specifications for power level, efficiency, size, etc. Various types of capacitor technologies may be used, including ceramic (including multi-layer ceramic capacitors), electrolytic ca-Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 pacitors, film capacitors (including power film capacitors), and IC-based capacitors. Capaci- tor dielectrics may vary as needed for particular applications, and may include dielectrics that are paraelectric, such as silicon dioxide (SiO2), hafnium dioxide (HFO2), or aluminum oxide Al2O3. In addition, multi-level power converter designs may beneficially utilize intrinsic par- asitic capacitances (e.g., intrinsic to the power FETs) in conjunction with or in lieu of de- signed capacitors to reduce circuit size and / or increase circuit performance. Selection of ca- pacitors for multi-level power converters may also take into account such factors as capacitor component variations, reduced effective capacitance with DC bias, and ceramic capacitor temperature coefficients (minimum and maximum temperature operating limits, and capaci- tance variation with temperature).

[0212] Similarly, in various embodiments of multi-level power converters, it may be ben- eficial to use specific types of inductors. For example, it is generally useful for the inductors to have low DC equivalent resistance, high inductance, and small volume.

[0213] The controller(s) used to control startup and operation of a multi-level power con- verter may be implemented as a microprocessor, a microcontroller, a digital signal processor (DSP), register-transfer level (RTL) circuitry, and / or combinatorial logic.

[0214] The term “MOSFET”, as used in this disclosure, includes any field effect transis- tor (FET) having an insulated gate whose voltage determines the conductivity of the transis- tor, and encompasses insulated gates having a metal or metal-like, insulator, and / or semicon- ductor structure. The terms “metal” or “metal-like” include at least one electrically conduc- tive material (such as aluminum, copper, or other metal, or highly doped polysilicon, gra- phene, or other electrical conductor), “insulator” includes at least one insulating material (such as silicon oxide or other dielectric material), and “semiconductor” includes at least one semiconductor material.

[0215] As used in this disclosure, the term “radio frequency” (RF) refers to a rate of os- cillation in the range of about 3 kHz to about 300 GHz. This term also includes the frequen- cies used in wireless communication systems. An RF frequency may be the frequency of an electromagnetic wave or of an alternating voltage or current in a circuit.

[0216] With respect to the figures referenced in this disclosure, the dimensions for the various elements are not to scale; some dimensions have been greatly exaggerated verticallyDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 and / or horizontally for clarity or emphasis. In addition, references to orientations and direc- tions (e.g., “top”, “bottom”, “above”, “below”, “lateral”, “vertical”, “horizontal”, etc.) are rel- ative to the example drawings, and not necessarily absolute orientations or directions.

[0217] Various embodiments of the invention can be implemented to meet a wide variety of specifications. Unless otherwise noted above, selection of suitable component values is a matter of design choice. Various embodiments of the invention may be implemented in any suitable integrated circuit (IC) technology (including but not limited to MOSFET structures), or in hybrid or discrete circuit forms. Integrated circuit embodiments may be fabricated using any suitable substrates and processes, including but not limited to standard bulk silicon, high- resistivity bulk CMOS, silicon-on-insulator (SOI), and silicon-on-sapphire (SOS). Unless otherwise noted above, embodiments of the invention may be implemented in other transistor technologies such as bipolar, BiCMOS, LDMOS, BCD, GaAs HBT, GaN HEMT, GaAs pHEMT, and MESFET technologies. However, embodiments of the invention are particu- larly useful when fabricated using an SOI or SOS based process, or when fabricated with pro- cesses having similar characteristics. Fabrication in CMOS using SOI or SOS processes ena- bles circuits with low power consumption, the ability to withstand high power signals during operation due to FET stacking, good linearity, and high frequency operation (i.e., radio fre- quencies up to and exceeding 300 GHz). Monolithic IC implementation is particularly useful since parasitic capacitances generally can be kept low (or at a minimum, kept uniform across all units, permitting them to be compensated) by careful design.

[0218] Voltage levels may be adjusted, and / or voltage and / or logic signal polarities re- versed, depending on a particular specification and / or implementing technology (e.g., NMOS, PMOS, or CMOS, and enhancement mode or depletion mode transistor devices). Component voltage, current, and power handling capabilities may be adapted as needed, for example, by adjusting device sizes, serially “stacking” components (particularly FETs) to withstand greater voltages, and / or using multiple components in parallel to handle greater currents. Additional circuit components may be added to enhance the capabilities of the dis- closed circuits and / or to provide additional functionality without significantly altering the functionality of the disclosed circuits.

[0219] Circuits and devices in accordance with the present invention may be used alone or in combination with other components, circuits, and devices. Embodiments of the presentDocket No.61658.39WO01 Client Ref. No. PER-532-PCT3 invention may be fabricated as integrated circuits (ICs), which may be encased in IC pack- ages and / or in modules for ease of handling, manufacture, and / or improved performance. In particular, IC embodiments of this invention are often used in modules in which one or more of such ICs are combined with other circuit blocks (e.g., filters, amplifiers, passive compo- nents, and possibly additional ICs) into one package. The ICs and / or modules are then typi- cally combined with other components, often on a printed circuit board, to form part of an end product such as a cellular telephone, laptop computer, or electronic tablet, or to form a higher-level module which may be used in a wide variety of products, such as vehicles, test equipment, medical devices, etc. Through various configurations of modules and assemblies, such ICs typically enable a mode of communication, often wireless communication.

[0220] A number of embodiments of the disclosure have been described. It is to be understood that various modifications may be made without departing from the spirit and scope of the disclosure. For example, some of the steps described above may be order independent, and thus can be performed in an order different from that described. Further, some of the steps described above may be optional. Various activities described with respect to the methods identified above can be executed in repetitive, serial, and / or parallel fashion.

[0221] It is to be understood that the foregoing description is intended to illustrate and not to limit the scope of the disclosure, which is defined by the scope of the following claims, and that other embodiments are within the scope of the claims. In particular, the scope of the disclosure includes any and all feasible combinations of one or more of the processes, machines, manufactures, or compositions of matter set forth in the claims below. (Note that the parenthetical labels for claim elements are for ease of referring to such elements, and do not in themselves indicate a particular required ordering or enumeration of elements; further, such labels may be reused in dependent claims as references to additional elements without being regarded as starting a conflicting labeling sequence).

Claims

Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 CLAIMS WHAT IS CLAIMED IS:

1. A system, comprising: a first monitor configured to receive a plurality of voltage inputs and select a voltage input having a lowest voltage from the plurality of voltage inputs; a voltage control loop configured to receive the selected voltage input and a reference voltage, and generate a first signal; a second monitor configured to receive a plurality of current inputs and select a cur- rent input having a lowest current from the plurality of current inputs as a target current; and a current control loop configured to receive the first signal and the target current and generate a second signal, wherein the second signal corresponds to a lesser of the first signal and the target current.

2. The system of claim 1, wherein the first signal corresponds to an average current that is regulated at one or more switches of a power converter.

3. The system of claim 1, wherein the first monitor comprises a digital-to-analog converter and the selected voltage input is an analog voltage.

4. The system of claim 1, wherein the plurality of voltage inputs are one or more of an output voltage of a power converter, an input voltage to the power converter, or a battery voltage.

5. The system of claim 1, wherein the second monitor comprises a digital-to-analog converter and the target current is an analog current.

6. The system of claim 5, wherein the plurality of current inputs are an input current to a power converter and an output current of the power converter.

7. The system of claim 1, wherein the first monitor comprises a minimum selector to select the voltage input that has the lowest voltage.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 8. The system of claim 1, wherein the second signal corresponds to a peak current that is reg- ulated at one or more switches of a power converter.

9. A system comprising: a current control loop configured to receive a plurality of current inputs and select a first signal, wherein the first signal represents a current input having a lowest current; a voltage control loop configured to receive a plurality of voltage inputs and select a second signal, wherein the second signal represents a voltage input having a lowest voltage; a selector configured to combine the first signal and the second signal into a combined signal; and a mode control configured to select a mode based on the combined signal, wherein the mode corresponds to one or more voltage levels.

10. The system of claim 9, wherein the current control loop is parallel to the voltage control loop.

11. The system of claim 9, wherein one of the plurality of current inputs includes a prede- fined current that corresponds to a maximum current that the first signal can represent.

12. The system of claim 9, wherein the plurality of voltage inputs are one or more of an out- put voltage of a power converter, an input voltage to the power converter, or a battery volt- age.

13. A system comprising: a multi-input error amplifier configured to: receive a plurality of input current signals representing a plurality of target currents and a second input current signal; select an input current signal from the plurality of input current signals repre- senting a lowest current; and generate an output signal representing an error between the selected input cur- rent signal and the second input current signal.

14. The system of claim 13, wherein the plurality of input current signals correspond to target currents to be regulated.Docket No.61658.39WO01 Client Ref. No. PER-532-PCT3 15. The system of claim 13, wherein the second input current signal corresponds to a sensed average current that is sensed at one or more sensors of a power converter.

16. The system of claim 13, wherein the multi-input error amplifier comprises a maximum output voltage and a minimum output voltage, and wherein the multi-input error amplifier is configured to generate an output voltage that is between the maximum output voltage and the minimum output voltage.

17. The system of claim 13, wherein the multi-input error amplifier is included in a current control loop.

18. The system of claim 13, further comprising: a voltage control loop configured to: receive a plurality of voltage inputs; and generate an output signal representing an error between the plurality of voltage inputs, wherein the output signal is one of the plurality of input current signals to the multi-input error amplifier.

19. The system of claim 18, wherein the voltage control loop includes a voltage error ampli- fier configured to generate the output signal.

20. The system of claim 18, wherein the output signal represents an average current to be regulated at one or more switches by a power converter.

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