High-frequency power amplifier

Abstract

A high-frequency power amplifier (10) comprises an input terminal (1), a carrier amplifier (4), a peak amplifier (5), a composite point (8), and an output terminal (2). When A is the magnitude of a reflection coefficient with the peak amplifier (5) side being viewed from the composite point (8) while operation of the peak amplifier (5) is OFF, and θ is the phase of the reflection coefficient, the phase θ of the reflection coefficient at a first distortion frequency is θ <= arccos[-4×A / (3+A×A)] within the range of 0° or greater and less than 180°, or the phase θ of the reflection coefficient at a second distortion frequency is θ >= arccos[-4×A / (3+A×A)] within the range of -180° or greater and less than 0°.

Description

High-frequency power amplifier

[0001] The present disclosure relates to a high-frequency power amplifier used in a device that transmits high-frequency signals.

[0002] In recent years, wireless communication base stations and the like have required wideband and highly efficient radio frequency power amplifiers capable of amplifying signals in multiple frequency bands with a single radio frequency power amplifier. In fifth-generation mobile communication systems (5G), the difference between average power and peak power (Peak-To-Average Power Ratio: PAPR) is large, so Doherty amplifiers are commonly used to increase the efficiency of radio frequency power amplifiers. A Doherty amplifier is an amplifier that divides the power of a radio frequency signal input from an input terminal using a divider, constantly amplifies one signal using a carrier amplifier, and amplifies the other signal using a peak amplifier when the input power is above a certain level, combines the output signal of the carrier amplifier and the output signal of the peak amplifier at a combining point, and outputs the amplified radio frequency signal from an output terminal.

[0003] Performance indicators of a radio frequency power amplifier include the relationship between input amplitude and output amplitude (AM-AM characteristics) and the relationship between input amplitude and output phase (AM-PM characteristics). If the AM-AM characteristics and AM-PM characteristics are linear, no distortion occurs in the radio frequency power amplifier, but generally, AM-AM characteristics and AM-PM characteristics are nonlinear, so distortion occurs in the radio frequency power amplifier. Because this distortion becomes an interference wave, radio communication base stations are required to keep distortion below a certain level.

[0004] Various techniques have been proposed to suppress distortion (see, for example, Patent Documents 1 and 2). Patent Document 1 describes a Doherty amplifier that has a predistortion compensation circuit in a stage preceding a carrier amplifier that has characteristics for compensating for distortion, particularly amplitude-phase (AM-PM) distortion, that occurs characteristically during operation of the carrier amplifier, and a predistortion compensation circuit in a stage preceding a peak amplifier that has characteristics for compensating for distortion, particularly amplitude-phase (AM-PM) distortion, that occurs characteristically during operation of the peak amplifier. Also, as described in Patent Document 2, a DPD (Digital Pre-Distortion) method is known that applies the inverse characteristics of the amplifier's nonlinearity to a transmission signal in advance and reduces distortion through digital processing.

[0005] JP 2004-222151 A JP 2012-129806 A

[0006] However, in the technology of Patent Document 1, the predistortion compensation circuit is a distortion compensation circuit that brings the nonlinearity of the AM-PM characteristics of the amplifier relative to the operating frequency band closer to linearity, and does not describe the relationship between the AM-PM characteristics of the amplifier and the predistortion compensation circuit at frequencies of distortion that occur lower and higher than the operating frequency band. Also, it is difficult to compensate for distortion generated in the carrier amplifier over a wide band using a predistortion compensation circuit.

[0007] Patent Document 2 also does not mention the relationship between the frequencies of distortion occurring on the lower and higher frequency sides of the operating frequency band and the AM-PM characteristics of the amplifier.

[0008] Therefore, an object of the present disclosure is to solve the above-mentioned problems and to provide a high-frequency power amplifier capable of performing distortion compensation over a wide band using the DPD system.

[0009] In order to achieve the above object, a radio frequency power amplifier according to an embodiment of the present disclosure includes an input terminal to which a radio frequency signal is input, a carrier amplifier and a peak amplifier connected to the input terminal, a combining point at which an output signal of the carrier amplifier and an output signal of the peak amplifier are combined, and an output terminal connected to the combining point, wherein the frequency of the radio frequency signal includes frequencies within an operating frequency band defined by a lower limit frequency, an upper limit frequency, and an operating bandwidth, a frequency lower than the lower limit frequency by the operating bandwidth is a first distortion frequency, and a frequency higher than the upper limit frequency by the operating bandwidth is a second distortion frequency, and when the peak amplifier is in an OFF state, the magnitude of the reflection coefficient when viewed from the combining point toward the peak amplifier is defined as A and the phase of the reflection coefficient is defined as θ, wherein the phase θ of the reflection coefficient at the first distortion frequency is θ<=arccos[-4×A / (3+A×A)] within a range of 0 degrees or more and less than 180 degrees, or the phase θ of the reflection coefficient at the second distortion frequency is set to θ>=arccos[-4×A / (3+A×A)] within a range of -180 degrees or more and less than 0 degrees.

[0010] According to the radio frequency power amplifier according to the present disclosure, the amount of distortion suppression can be improved over a wide band using the DPD method.

[0011] FIG. 1 is a diagram illustrating a radio frequency power amplifier according to a first embodiment. FIG. 2 is a diagram illustrating an example of the relationship between the operating bandwidth (WA) and the signal bandwidth (WC) of the radio frequency power amplifier according to the first embodiment. FIG. 3 is a diagram illustrating an example of the relationship between the phase θ of the reflection coefficient as viewed from the combining point toward the peak amplifier when the peak amplifier according to the first embodiment is in OFF operation, and the pass phase φ of the output signal from the carrier amplifier at the combining point. FIG. 4 is a diagram illustrating the relationship between the magnitude A of the reflection coefficient as viewed from the combining point toward the peak amplifier when the peak amplifier according to the first embodiment is in OFF operation, and the phase θ of the reflection coefficient at which the pass phase φ is an inflection point. FIG. 5 is a diagram illustrating an example of distortion characteristics (ACLR) after distortion compensation by the DPD system, with respect to the difference between the pass phase and the tangent of the pass phase at θ = 0 degrees, with respect to the phase θ of the reflection coefficient according to the first embodiment. FIG. 6 is a diagram illustrating an example of the difference between the pass phase and the tangent of the pass phase at θ = 0 degrees with respect to the phase θ of the reflection coefficient according to the first embodiment. FIG. 7 is a diagram illustrating a radio frequency power amplifier according to a second embodiment. FIG. 8 is a diagram illustrating an example of the configuration of a carrier amplifier according to the second embodiment. FIG. 9 is a diagram illustrating an example of the configuration of a peak amplifier according to the second embodiment. FIG. 10 is a diagram illustrating an example of impedance when looking at the peak amplifier side from the combining point according to the second embodiment.

[0012] Hereinafter, radio frequency power amplifiers according to embodiments will be described in detail with reference to the drawings. Each of the embodiments described below represents a specific example of the present disclosure. The numerical values, shapes, materials, components, component arrangements, and connection configurations shown in the following embodiments are merely examples and are not intended to limit the present disclosure. Furthermore, each drawing is not necessarily an exact illustration. In each drawing, substantially identical components are denoted by the same reference numerals, and redundant descriptions may be omitted or simplified. Furthermore, in this specification, "connection" refers to an electrical connection, and includes not only a case where two circuit elements are directly connected, but also a case where two circuit elements are indirectly connected with another circuit element inserted between them.

[0013] (First Embodiment) A radio frequency power amplifier according to a first embodiment will be described with reference to Fig. 1. Fig. 1 is a diagram illustrating a radio frequency power amplifier 10 according to the first embodiment.

[0014] The radio frequency power amplifier 10 is a Doherty amplifier including an input terminal 1 to which a radio frequency signal is input, a divider 3 connected to the input terminal 1 and power-dividing the input radio frequency signal, a carrier amplifier 4 connected to the divider 3 and amplifying one of the divided signals, a peak amplifier 5 connected to the divider 3 and amplifying the other of the divided signals, a combining point 8 where an output signal of the carrier amplifier 4 and an output signal of the peak amplifier 5 are combined, and an output terminal 2 connected to the combining point 8. As shown in FIG. 1 , a first impedance conversion line 6 for adjusting impedance may be provided between the carrier amplifier 4 and the combining point 8. Also, as shown in FIG. 1 , a second impedance conversion line 7 for adjusting impedance may be provided between the combining point 8 and the output terminal 2. Each circuit element will be described in detail below.

[0015] A high-frequency signal including a frequency within the operating frequency band is input from input terminal 1. The operating frequency band is a frequency band that can be amplified by high-frequency power amplifier 10. The operating frequency band may be, for example, a frequency band in which the gain decreases by 3 dB. The operating frequency band may also be a frequency band for 5G new radio (5GNR), such as n77 (3300-4200 MHz), n78 (3300-3800 MHz), or n79 (4400-5000 MHz). For example, when the operating frequency band is n78, the lower limit frequency (FL), upper limit frequency (FH), and operating bandwidth (WA) are FL = 3300 MHz, FH = 3800 MHz, and WA = 500 MHz.

[0016] The input high-frequency signal is power-split by a splitter 3, with one signal being input to a carrier amplifier 4 and the other being input to a peak amplifier 5. The carrier amplifier 4 is, for example, a class AB amplifier, and performs constant amplification. On the other hand, the peak amplifier 5 is, for example, a class C amplifier, and performs amplification when the input power is above a certain level. When the peak amplifier 5 performs amplification, it is referred to as an ON operation, and otherwise it is referred to as an OFF operation. The carrier amplifier 4 generates distortion due to its nonlinear characteristics. If the bandwidth of the high-frequency signal is referred to as the signal bandwidth (WC), distortion occurs on the frequency side lower and higher than the signal bandwidth. A frequency lower than the lower limit frequency by the operating bandwidth is referred to as a first distortion frequency (FLD), and a frequency higher than the upper limit frequency by the operating bandwidth is referred to as a second distortion frequency (FHD).

[0017] FIG. 2 shows an example of the relationship between the operating bandwidth (WA) and the signal bandwidth (WC) of the high-frequency power amplifier 10 according to the first embodiment. FIGS. 2(a) and 2(b) show an example in which one high-frequency signal is input within the operating frequency band, while FIG. 2(c) shows an example in which multiple (two) high-frequency signals are input within the operating frequency band. FIG. 2(a) shows a case in which the signal bandwidth and the operating bandwidth are the same, with adjacent channel distortion occurring on the lower and higher frequency sides of the signal bandwidth. FIG. 2(b) shows a case in which the signal bandwidth is narrower than the operating bandwidth, with adjacent channel distortion occurring on the lower and higher frequency sides of the signal bandwidth, similar to FIG. 2(a). FIG. 2(c) shows a case in which the signal bandwidth is narrower than the operating bandwidth, with two high-frequency signals input, with adjacent channel distortion occurring on the lower and higher frequency sides of the signal bandwidth, and intermodulation distortion also occurring. FIG. 2 also shows a first distortion frequency (FLD) and a second distortion frequency (FHD).

[0018] The output signal of carrier amplifier 4 and the output signal of peak amplifier 5 are combined at combining point 8. The combined signal is output from output terminal 2. When high-frequency power amplifier 10 is at low output, that is, when peak amplifier 5 is in OFF operation, the passing phase (φ) of the output signal of carrier amplifier 4 as it passes through combining point 8 changes depending on the impedance seen from combining point 8 toward the peak amplifier 5. The impedance seen from combining point 8 toward the peak amplifier 5 is converted into a reflection coefficient using characteristic impedance Zo (for example, 50 Ω), and the magnitude of the reflection coefficient is defined as A and the phase of the reflection coefficient is defined as θ. The passing phase (φ) and the reflection coefficient are related to each other by the following equation (1):

[0019] φ=arctan[2×A×sinθ / (3+A×A+4×A×cosθ)] ...(1)

[0020] FIG. 3 is a diagram showing an example of the relationship between the phase θ of the reflection coefficient when viewed from the combining point 8 toward the peak amplifier 5, and the passing phase φ of the output signal from the carrier amplifier 4 at the combining point 8, when the peak amplifier 5 according to the first embodiment is in an OFF state. FIG. 3 illustrates Equation (1) when the magnitude A of the reflection coefficient is set to 0.6, 0.8, and 0.9. In FIG. 3, the horizontal axis represents the phase (θ) of the reflection coefficient, and the vertical axis represents the passing phase (φ). The solid line represents the case where A = 0.9, the dashed line represents the case where A = 0.8, and the dashed-dotted line represents the case where A = 0.6. In each case, the phase θ of the reflection coefficient has inflection points at positive and negative values. The phase (θ) of the reflection coefficient at the inflection point can be found by differentiating Equation (1) with respect to θ to find the θ at which dφ / dθ = 0.

[0021] θ=arccos[-4×A / (3+A×A)] ...(2)

[0022] FIG. 4 is a diagram showing the relationship between the magnitude A of the reflection coefficient when viewed from the combining point 8 toward the peak amplifier 5 and the phase θ of the reflection coefficient at which the passing phase φ becomes an inflection point when the peak amplifier 5 according to the first embodiment is in an OFF state. FIG. 4 illustrates equation (2). In FIG. 4, the horizontal axis represents the magnitude (A) of the reflection coefficient, and the vertical axis represents the phase (θ) of the reflection coefficient. The solid line represents the case where θ is in the positive range, and the dashed line represents the case where θ is in the negative range. For example, when A=0.9, the inflection points are θ=160.8 degrees and -160.8 degrees; when A=0.8, the inflection points are θ=151.5 degrees and -151.5 degrees; and when A=0.6, the inflection points are θ=135.5 degrees and -135.5 degrees.

[0023] The output signal from the carrier amplifier 4 contains not only an amplified high-frequency signal but also distortions occurring on the lower and higher frequency sides of the signal bandwidth of the high-frequency signal. When the output signal of the carrier amplifier 4 passes through the combining point 8, the impedance seen from the combining point 8 toward the peak amplifier 5 affects not only the high-frequency signal but also the distortion. The DPD system extracts a portion of the output signal from the output terminal 2 and performs digital signal processing to linearize the AM-AM and AM-PM characteristics of the high-frequency power amplifier 10. If the frequency at which distortion occurs includes an inflection point, the digital signal processing becomes more complex and also affects the amount of distortion suppression achieved by the DPD system. For this reason, when the peak amplifier 5 is in OFF mode, it is necessary to set the phase θ of the reflection coefficient seen from the combining point 8 toward the peak amplifier 5 so that the frequency at which distortion occurs falls within a range that does not include an inflection point.

[0024] Distortion occurring on the frequency side lower than the signal bandwidth affects the inflection point where the phase θ of the reflection coefficient is positive, while distortion occurring on the frequency side higher than the signal bandwidth affects the inflection point where the phase θ of the reflection coefficient is negative. Regarding adjacent channel distortion and intermodulation distortion, which are third-order nonlinear distortions occurring in the high-frequency power amplifier 10, distortion occurring on the frequency side lower than the signal bandwidth can be achieved by ensuring that the phase θ of the reflection coefficient at the first distortion frequency (FLD) is equal to or lower than the inflection point where the phase θ of the reflection coefficient at the second distortion frequency (FHD) is equal to or higher than the inflection point where the phase θ of the reflection coefficient is negative. In other words, the phase θ of the reflection coefficient at the first distortion frequency (FLD) should be θ<=arccos[-4×A / (3+A×A)] within the range of 0 degrees or more and less than 180 degrees. Furthermore, the phase θ of the reflection coefficient at the second distortion frequency (FHD) may be θ >= arccos[−4×A / (3+A×A)] in the range of −180 degrees or more and less than 0 degrees. More specifically, at the first distortion frequency (FLD), A > 0.9 and θ < 160 degrees may be satisfied, or A > 0.8 and θ < 150 degrees may be satisfied, or A > 0.6 and θ < 135 degrees may be satisfied. At the second distortion frequency (FHD), A > 0.9 and θ > −160 degrees may be satisfied, or A > 0.8 and θ > −150 degrees may be satisfied, or A > 0.6 and θ > −135 degrees may be satisfied.

[0025] FIG. 5 is a diagram showing an example of distortion characteristics (ACLR) after distortion compensation using the DPD system with respect to the difference between the pass phase φ and the tangent of the pass phase φ at θ = 0 degrees according to the first embodiment. The pass phase φ ideally has a linear characteristic with respect to the phase θ of the reflection coefficient, but as described in FIG. 3 , it has an inflection point. In FIG. 5 , the difference between the pass phase φ and the tangent of the pass phase φ at θ = 0 degrees is used as an index of the linearity of the pass phase φ. If the pass phase φ has a linear characteristic with respect to the phase θ of the reflection coefficient, the difference between the pass phase φ and the tangent of the pass phase φ at θ = 0 degrees is zero. The slope of the tangent of the pass phase φ at θ = 0 degrees can be obtained by differentiating Equation (1) with respect to θ and substituting θ = 0. In FIG. 5 , the horizontal axis represents the difference between the pass phase φ and the tangent of the pass phase φ at θ = 0 degrees, and the vertical axis represents the distortion characteristics (ACLR) after distortion compensation using the DPD system. The black circles represent the measurement results of distortion occurring at frequencies lower than the signal bandwidth, and the solid line represents a linear approximation line. For example, in order to make the ACLR after distortion compensation by the DPD system −45 dBc or less, the difference between the passing phase φ and the tangent of the passing phase φ at θ=0 degrees must be 13 degrees or less.

[0026] FIG. 6 is a diagram illustrating an example of the difference between the pass phase and the tangent of the pass phase at θ=0 degrees relative to the phase θ of the reflection coefficient according to the first embodiment. In FIG. 6, the horizontal axis represents the phase θ of the reflection coefficient, and the vertical axis represents the difference between the pass phase and the tangent of the pass phase at θ=0 degrees. The solid line represents the difference when A=0.9, the dashed line represents the difference when A=0.95, and the dashed line represents the difference when A=1. From FIG. 6, in order to make the difference between the pass phase φ and the tangent of the pass phase φ at θ=0 degrees 13 degrees or less, the phase θ of the reflection coefficient should be set to 125 degrees or less, regardless of the magnitude A of the reflection coefficient. By setting the phase θ of the reflection coefficient at the first distortion frequency (FLD) to 125 degrees or less, the ACLR after distortion compensation using the DPD method can be set to -45 dBc or less. Similarly, by setting the phase θ of the reflection coefficient at the second distortion frequency (FHD) to -125 degrees or more, the ACLR after distortion compensation using the DPD method can be set to -45 dBc or less.

[0027] According to the radio frequency power amplifier 10 of the first embodiment, distortion occurring in the radio frequency power amplifier 10 can be accurately compensated for over a wide band using the DPD method, and the amount of distortion suppression can also be improved. Furthermore, if the phase θ of the reflection coefficient at the first distortion frequency (FLD) is set to 125 degrees or less, the ACLR after distortion compensation using the DPD method can be set to −45 dBc or less.

[0028] As described above, the high frequency power amplifier 10 according to this embodiment has an input terminal 1 to which a high frequency signal is input, a carrier amplifier 4 and a peak amplifier 5 connected to the input terminal 1, a combining point 8 at which the output signal of the carrier amplifier 4 and the output signal of the peak amplifier 5 are combined, and an output terminal 2 connected to the combining point 8, the frequency of the high frequency signal includes frequencies within an operating frequency band set by a lower limit frequency (FL), an upper limit frequency (FH) and an operating bandwidth (WA), and a frequency lower than the lower limit frequency by the operating bandwidth is set as a first distortion frequency (FLD), When a frequency higher than the upper limit frequency by the operating bandwidth is defined as the second distortion frequency (FHD), and when the peak amplifier 5 is in OFF operation, the magnitude of the reflection coefficient when viewed from the combining point 8 toward the peak amplifier 5 is defined as A and the phase of the reflection coefficient is defined as θ, the phase θ of the reflection coefficient at the first distortion frequency is θ<=arccos[-4×A / (3+A×A)] within the range of 0 degrees or more and less than 180 degrees, or the phase θ of the reflection coefficient at the second distortion frequency is θ>=arccos[-4×A / (3+A×A)] within the range of -180 degrees or more and less than 0 degrees.

[0029] This realizes a high frequency power amplifier 10 that can perform distortion compensation with high precision over a wide band using the DPD method and can also improve the amount of distortion suppression.

[0030] Second Embodiment Next, a radio frequency power amplifier according to a second embodiment will be described with reference to Fig. 7. Fig. 7 is a diagram illustrating a radio frequency power amplifier 20 according to the second embodiment. The radio frequency power amplifier 20 according to the second embodiment has a configuration in which the carrier amplifier 4 of the radio frequency power amplifier 10 according to the first embodiment shown in Fig. 1 is replaced with a carrier amplifier 40, and the peak amplifier 5 is replaced with a peak amplifier 50, and a description of the same circuit configuration will be omitted.

[0031] 8 is a diagram showing an example of the configuration of a carrier amplifier 40 according to the second embodiment. The carrier amplifier 40 includes a carrier amplifier transistor 41, an input matching circuit 42 connected to the gate of the carrier amplifier transistor 41, and an output matching circuit 43 connected to the drain of the carrier amplifier transistor 41. A gate voltage VGCA is supplied to the gate of the carrier amplifier transistor 41, and a drain voltage VDCA is supplied to the drain of the carrier amplifier transistor 41. The source of the carrier amplifier transistor 41 is connected to the ground potential. The carrier amplifier 40 is, for example, a class AB amplifier.

[0032] 9 is a diagram showing an example of the configuration of a peak amplifier 50 according to the second embodiment. The peak amplifier 50 includes a peak amplifier transistor 51, an input matching circuit 52 connected to the gate of the peak amplifier transistor 51, and an output matching circuit 53 connected to the drain of the peak amplifier transistor 51. A gate voltage VGPA is supplied to the gate of the peak amplifier transistor 51, and a drain voltage VDPA is supplied to the drain of the peak amplifier transistor 51. The source of the peak amplifier transistor 51 is connected to the ground potential. The peak amplifier 50 is, for example, a class C amplifier.

[0033] The gate width, which is the device size of the carrier amplifier transistor 41, is defined as Wgca, and the gate width, which is the device size of the peak amplifier transistor 51, is defined as Wgpa. In FIG. 9 , Wgpa > Wgca, and the device sizes of the carrier amplifier transistor 41 and the peak amplifier transistor 51 are different. The device size ratio of the carrier amplifier transistor 41 to the peak amplifier transistor 51 is defined as n (n = Wgpa / Wgca). When the peak amplifier 50 is in ON mode, the impedance seen from the combining point 8 toward the carrier amplifier 40 is set to Zo, and the impedance seen from the combining point 8 toward the peak amplifier 50 is set to Zo / n. For example, when Zo = 50 Ω and n = 2, the impedance seen from the combining point 8 toward the peak amplifier 50 is 25 Ω.

[0034] FIG. 10 is a diagram showing an example of the impedance seen from the combining point 8 toward the peak amplifier 50 according to the second embodiment. FIG. 10 is a Smith chart normalized at Zo = 50Ω. Points P1, P2, and P3 indicate the impedance at the center frequency of the operating frequency band. When the peak amplifier 50 is in ON mode, the impedance at the drain end of the peak amplifier transistor 51 is several Ω (e.g., 5Ω), and the output matching circuit 53 converts the impedance at the drain end of the peak amplifier transistor 51 to point P1 (25Ω). In a conventional Doherty amplifier, when the peak amplifier 50 is in OFF mode, the impedance seen from the combining point 8 toward the peak amplifier 50 is set to near open. In the case of FIG. 10 , the impedance seen from the combining point 8 toward the peak amplifier 50 is point P2 (950Ω), and in terms of reflection coefficients, the magnitude A = 0.9 and the phase θ = 0 degrees. When the peak amplifier 50 is in OFF mode, the impedance at the drain end of the peak amplifier transistor 51 is near short, and the output matching circuit 53 converts the impedance at the drain end of the peak amplifier transistor 51 to point P2. The impedance seen from the combining point 8 toward the peak amplifier 50 is converted by the output matching circuit 53 to point P2 when the peak amplifier 50 is in OFF mode and to point P1 when the peak amplifier 50 is in ON mode. However, if the difference in impedance between OFF mode and ON mode becomes large, the frequency dispersion of the output matching circuit 53 increases, making it difficult to achieve a wide bandwidth. In addition, the circuit size increases, resulting in increased loss. When n = 3, the impedance when the peak amplifier 50 is in ON mode is 16.7 Ω, making designing the output matching circuit 53 more difficult.

[0035] In the radio frequency power amplifier 20 according to the second embodiment, the impedance when the peak amplifier 50 is in OFF operation is set to, for example, point P3 (5.2+j49.7Ω). Expressed as a reflection coefficient, the magnitude A is 0.9 and the phase θ is 90 degrees. In this manner, the output matching circuit 53 simply converts the impedance at the drain end of the peak amplifier transistor 51 to point P3 when the peak amplifier 50 is in OFF operation and to point P1 when the peak amplifier 50 is in ON operation. This simplifies the design of the output matching circuit 53, enables the peak amplifier 50 to operate in an optimal state in both OFF and ON operations, and enables the operating bandwidth of the radio frequency power amplifier 20 to be widened.

[0036] In the second embodiment, an example has been described in which the phase θ of the reflection coefficient at the center frequency of the operating frequency band is 90 degrees for the impedance when the peak amplifier 50 is in OFF operation, but the present invention is not limited to this and may include frequencies within the operating frequency band where the phase θ of the reflection coefficient is 90 degrees or greater.Furthermore, the present invention may include frequencies within the operating frequency band where the phase θ of the reflection coefficient is −90 degrees or less.

[0037] The peak amplifier transistor 51 of the second embodiment may be formed of a nitride semiconductor on a semiconductor substrate including a Si substrate, or a nitride semiconductor on a semiconductor substrate including a SiC substrate. Forming the peak amplifier transistor 51 from a nitride semiconductor enables higher voltage operation.

[0038] Furthermore, the carrier amplifier transistor 41 of the second embodiment may be formed of a nitride semiconductor on a semiconductor substrate including a Si substrate. By forming the carrier amplifier transistor 41 from a nitride semiconductor, higher voltage operation is possible.

[0039] Furthermore, the carrier amplifier transistor 41 of the second embodiment may be formed of a nitride semiconductor on a semiconductor substrate including a SiC substrate. By using a semiconductor substrate including a SiC substrate, higher temperature operation is possible.

[0040] As described above, the radio frequency power amplifier 20 according to this embodiment is set so that the impedance when looking toward the peak amplifier 50 from the combining point 8 when the peak amplifier 50 is in OFF operation includes frequencies within the operating frequency band at which the phase θ of the reflection coefficient is 90 degrees or more, or includes frequencies within the operating frequency band at which the phase θ of the reflection coefficient is −90 degrees or less.

[0041] This allows the peak amplifier 50 to operate in an optimal state in both OFF and ON operations, thereby achieving a wider operating bandwidth for the high frequency power amplifier 20.

[0042] The radio frequency power amplifier according to the present disclosure is capable of performing distortion compensation with high precision over a wide band using the DPD method, and can be used as a radio frequency power amplifier in a wireless communication base station.

[0043] REFERENCE SIGNS LIST 1 Input terminal 2 Output terminal 3 Distributor 4, 40 Carrier amplifier 5, 50 Peak amplifier 6 First impedance transformation line 7 Second impedance transformation line 8 Combining point 10, 20 High frequency power amplifier 41 Carrier amplifier transistor 42, 52 Input matching circuit 43, 53 Output matching circuit 51 Peak amplifier transistor

Claims

1. A radio frequency power amplifier having an input terminal to which a radio frequency signal is input, a carrier amplifier and a peak amplifier connected to the input terminal, a combining point where an output signal of the carrier amplifier and an output signal of the peak amplifier are combined, and an output terminal connected to the combining point, wherein the frequency of the radio frequency signal includes frequencies within an operating frequency band set by a lower limit frequency, an upper limit frequency, and an operating bandwidth, a frequency lower than the lower limit frequency by the operating bandwidth is defined as a first distortion frequency, and a frequency higher than the upper limit frequency by the operating bandwidth is defined as a second distortion frequency, and when the peak amplifier is in an OFF state, the magnitude of the reflection coefficient when viewed from the combining point toward the peak amplifier is defined as A and the phase of the reflection coefficient is defined as θ, and the phase θ of the reflection coefficient at the first distortion frequency is θ<=arccos[-4×A / (3+A×A)] within the range of 0 degrees or more and less than 180 degrees, or A radio frequency power amplifier, wherein a phase θ of the reflection coefficient at the second distortion frequency satisfies θ>=arccos[−4×A / (3+A×A)] within a range of −180 degrees or more and less than 0 degrees.

2. The high frequency power amplifier according to claim 1, wherein the phase θ of the reflection coefficient at the first distortion frequency is 125 degrees or less.

3. A high frequency power amplifier according to claim 1, characterized in that the operating frequency band includes a frequency at which the phase θ of the reflection coefficient is 90 degrees or greater.

4. A high frequency power amplifier according to claim 1, characterized in that the operating frequency band includes a frequency at which the phase θ of the reflection coefficient is −90 degrees or less.

5. The radio frequency power amplifier according to any one of claims 1 to 4, wherein the carrier amplifier has a carrier amplifier transistor, the peak amplifier has a peak amplifier transistor, and the device size of the peak amplifier transistor is larger than the device size of the carrier amplifier transistor.

6. The high frequency power amplifier according to claim 5, wherein the peak amplifier transistor is formed of a nitride semiconductor on a semiconductor substrate containing Si.

7. The radio frequency power amplifier according to claim 5, wherein the carrier amplifier transistor is formed of a nitride semiconductor on a semiconductor substrate containing Si.

8. The radio frequency power amplifier according to claim 5, wherein the carrier amplifier transistor is formed of a nitride semiconductor on a semiconductor substrate containing SiC.

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