Method of and apparatus for processing wireless communication signals for communication and environment sensing

Abstract

A method of processing wireless signals suggests separately optimising analogue and digital beamformers, notably first optimising the analogue beamformer in a conventional manner, and afterwards optimising the digital beamformer by mapping the optimisation problem into a least squares problem with quadratic constraints, which is convex and thus open to an algebraic solution. When designing the beamformers the target is to maintain the SI leakage below a predetermined threshold, such that the residual leakage can be satisfyingly cancelled in a conventional manner in the receiver's analogue and digital SI cancellers. The beamformer design further targets to optimise for maximum SNR. Designing the digital beamformer may comprise applying numerical methods or applying a CVX process. The combined efforts of optimising the beamformers for minimum SI leakage and maximum SNR result in optimised sensing and communication metrics.

Description

[0001] METHOD OF AND APPARATUS FOR PROCESSING WIRELESS COMMUNICATION SIGNALS FOR COMMUNICATION AND ENVIRONMENT SENSING

[0002] FIELD OF THE INVENTION

[0003] The present invention relates to the field of environment sensing or environment mapping, in particular to such sensing using wireless communication signals. More specifically, the invention relates to a method of processing communication signals for environment perception, to a computer program product implementing the method, to a computer-readable storage medium storing the computer program product, and to an apparatus configured to execute the method. Throughout this specification the term environment sensing will be used for the various expressions widely used for capturing information about an environment for creating a three- dimensional representation thereof.

[0004] NOTATIONS

[0005] Vectors and matrices are denoted by boldface lowercase and boldface capital letters, respectively, as in a and A. The transpose and Hermitian transpose of a matrix are denoted by superscript letters ( ■ )Tand ( ■ )H, respectively, and det(A) is the determinant of a matrix A, while I„ (n > 2) is the nXn identity matrix and 0„x„ («, m > 2) is a nXm matrix with zeros. || A ||Fis the Frobenius norm of A, || a || stands for a’s Euclidean norm, and diag{a} denotes a square diagonal matrix with a’s elements in its main diagonal. [A]li7- , [A](i,:) and [A](:,j) represent A’s (z,j)-th element, z-th row, and j-th column, respectively, while [a] / denotes the z-th element of a. !R and > represent the real and complex number sets, respectively, E{-} is the expectation operator, and |-| is the amplitude of a complex number.

[0006] BACKGROUND

[0007] Joint Communication and Sensing (JCAS), also referred to as Integrated Sensing and Communication (ISAC), is a technique in wireless communications with the objective of retrieving information about the environment from the signal scattering which is present in the effective channel state information (CSI), e.g., due to objects in the environment, blockage, user activity, etc., while simultaneously achieving data communication. Most known JCAS methods exploit radar technology to infer information about the environment. This is also known as joint radar and communication (JRC). In the present specification the abbreviations JCAS and ISAC are used interchangeably.

[0008] In band full duplex, also known shortly as Full Duplex (FD), is a candidate technology for the Release 17 of the fifth Generation (5G) New Radio (NR) standard enabling simultaneous UpLink (UL) and DownLink (DL) communication within the entire frequency band. A FD radio can transmit and receive at the same time and frequency resource units, consequently, it can double the spectral efficiency achieved by a Half Duplex (HD) radio. Current wireless systems exploit Multiple Input Multiple Output (MIMO) communication, where increasing the number of Transmitter (TX) and Receiver (RX) antennas can increase the spatial Degrees of Freedom (DoF), hence boosting rate performance. Combining FD with MIMO operation can provide further spectral efficiency gains.

[0009] FD radios suffers from Self Interference (SI), a term referring to the signal transmitted by the FD radio TX that leaks to the RX of the same FD radio. At the RX of the FD radio, the SI power can be many times stronger than the power of the received signal of interest. Consequently, SI can severely degrade the reception of the signal of interest, and thus SI mitigation is required in order to maximize the spectral efficiency gain of the FD operation. As the number of antennas increases, mitigating SI becomes more challenging, since more antennas naturally result in more SI components. Conventional SI suppression techniques in Single-Input Single-Output (SISO) systems include propagation domain isolation, analogue domain suppression, and digital cancellation. Although analogue SI cancellation in FD MIMO systems can be implemented through SISO replication, its hardware requirements scale with the number of TX / RX antennas.

[0010] In radio equipment having multiple antennas the issue of SI has been addressed by increasing the distance between the antennas, using cross polarization, or through the use of highly directional antennas. In MIMO radio systems in which an estimate of the SI is known, spatial suppressing techniques, i.e., transmit and / or receive beamforming can also be used for reducing the impact of SI. The known techniques for cancelling SI assume unrealistic SI channel models, which results in less-than-predicted performance in real-word applications. Notably, those techniques employing iterative cancellation processes do not provide satisfying performance at high transmit SNR. Further, even when transmit precoding is employed, e.g., for electronic beamforming in multiple-input multiple output (MIMO) systems, the known techniques use overly simplistic iterative SI cancellation methods that do not perform satisfyingly at high SNR or do not consider SI leakage at all when modelling the transmit precoder. Moreover, the known techniques for cancelling SI exclusively target the optimisation of the UL and DL communication performance, ignoring the specific requirements of JCAS.

[0011] Thus, there is a need for an improved approach for addressing SI in MIMO ISAC systems, which provides high performance for both, sensing and communication, in real-world scenarios.

[0012] SUMMARY OF THE INVENTION

[0013] This need is addressed by the methods of claims 1 and 7, and the apparatus of claim 8. A computer program product and a corresponding computer-readable medium are provided in claims 10 and 11 , respectively. Advantageous embodiments and developments are provided in respective dependent claims.

[0014] The present invention builds on a practical channel model, i.e., a channel model that better represents channels in real-world scenarios, and proposes a method to reduce SI leakage while at the same time optimising both the sensing and the communication performance. More particularly, the present invention optimises the transmit beamformers such that the SI leakage from the transmitter to the receiver is minimised and that both the communication and the sensing performance, respectively, are maximised.

[0015] For a better understanding of the method in the following an underlying MIMO radio equipment as shown in the left part of figure 1 will be discussed first. The MIMO radio equipment, referenced node £ in figure 1 , comprises an N-tap analogue cancellation for the SI signal as well as A / D precoding and combining for the outgoing and incoming signals. The MIMO radio equipment node k is capable of hybrid beamforming (HBF) through its partially connected beamforming architecture. As (RF) shown in the figure, the analogue canceller interconnects the Nkinputs of the analogue TX precoder to the Mkoutputs of the analogue RX combiner. The complexity of the analogue canceller expressed in the number of taps N is independent of the numbers Nkand Mkof the TX and RX antennas, respectively, and, as will be shown next, it scales with the product ANkRF)MkRF)with A < 1. The setup may be referenced to as partially connected HBF.

[0016] Each of the NkTX RF chains of node k is connected to a separate subset of the available TX antenna elements. As shown in figure 1 , the i-th TX RF chain with i = l,2,...,NkRF)is connected via phase shifters with NkA)TX antenna elements, each denoted as TX(i,j)Vj = 1, 2, ...,NkA). Clearly, it holds Nk= NkRF)NkA)for the total number of TX antennas at node k. Stacking the values of the Nkphase shifters that connect each i-th TX RF chain with its antenna elements in a complexvalued NkA)x 1 vector Vj , the complex-valued Nkx NkRF)analogue TX precoder can be formulated as follows:

[0017] The elements of each v, are assumed to have unit magnitude, i.e., also assumes that v, G FTXVi = l,2,...,NkRF), which means that all analogue TX precoding vectors belong in a predefined beam codebook FTXincluding card(FTX) distinct vectors, or analogue beams. Apart from applying Vkin the analogue domain to the information bearing signal before transmission, the symbol vector skis also processed in the baseband (BB) with the digital TX precoder VkBB)G CNkRF)xdk, recalling that dk< rnin{Mq,NkRF)}, before entering into the Nk TX RF chains, as shown in figure 1 . Similar to the uplink (UL) communication from node m to k, it is assumed that the downlink (DL) transmission being the total available TX power at node k.

[0018] The RX side of node k is composed of an analogue combiner connecting the RX antenna elements to the inputs of the RX RF chains, and a digital combiner that processes the outputs of the RX RF chains in BB before signal decoding. In particular, the n-th RX RF chain with n = l,2,...,M(kRF)is connected through phase shifters with MkA)distinct RX antennas; these phase shifters are denoted as RX(n,l )VI = l,2,...,MkA)It should hold that Mk= MkRF)MkA)for the total number of RX antennas at node k. Define the complex-valued Mkx analogue RX combiner (RF)

[0019] Ukhaving a block diagonal structure similar to the one shown in equation (1 ). In particular, Ukcontains un's with n = 1,2, in the diagonal, where each uncontains the unit magnitude values of the MkA)phase shifters, i.e., | [un]j |2= lVj = l,2,...,MkA), connecting each n th RX RF chain with its antenna elements. It is also assumed that unG FRXVn, i.e., all analogue RX combiners belong in a predefined beam codebook FRXhaving card(FRX) vectors. Finally,

[0020] U<BB)CM*RF) x dm with dm< min{MkRF),Nm} represents the digital RX combiner at node k.

[0021] The analogue canceller at node k consists of N taps with each tap connected via a NkRF)-to-1 Multiplexer (MUX) to all NkRF)outputs of the respective TX RF chains. A tap includes a fixed delay, a variable phase shifter, and a variable attenuator. To route the cancellation signal to one of the adders located just before the RX RF chains, the output of each tap is connected to a 1-to- MkDEMUItipleXer (DEMUX). There is a total of NMk such adders and the notation "Adder (i ,j) " is used to label the adder that connects DEMUX j to RX RF chain i, where i = l,2,...,MkRF)and j = 1,2, ...,N. The adders before the RX RF chains can be implemented via power combiners or directional couplers, while the analogue RF MUXs / DEMUXs can be implemented with RF switches. Clearly, the proposed analogue canceller interconnects the outputs of some of the available TX RF chains to the inputs of some of the RX RF chains. The size of each MUX / DEMUX depends on the number of TX / RF RF chains and not on the number of TX / RX antennas.

[0022] The analogue processing realized by the analogue canceller is modelled in BB as Ck

[0023] The elements [l_T]j and [ l_3]j with j = 1,2, (RF)

[0024] Mktake the binary values 0 or 1 , and it must hold that

[0025] The L2in Ckis a diagonal matrix whose complex entries represent the attenuation and phase shift of the canceller taps; the magnitude and phase of the element [ L2]j with i = 1,2,...,N specify the attenuation and phase of the i th tap. It is recalled that the tap delays in each canceller tap are fixed, hence, we the effects of the i-th tap delay are modelled as a phase shift that is incorporated to the phase of [L2]ifi.

[0026] Using the previously described system configuration, the BB received signal yqG CMq X1at node q in the DL communication can be mathematically expressed as where Hq>kG cMq XNkis the DL channel gain matrix, i.e., between nodes q and k, Hq>mG cMqXNmdenotes the channel gain matrix for inter-node interference, i.e., between nodes q and m, and nqG CMq X1represents the Additive White Gaussian Noise (AWGN) at node q with variance aq. In the UL communication, the symbol vector smG Cdm X 1used for the estimation of smat the FD HBF node k is derived as where Hk kG CMkXNkdenotes the SI channel seen at the RX antennas of node k due to its own DL transmission, Hk mG cMkXNmis the UL channel gain matrix, i.e., between nodes k and m, and nkG CMkX1denotes the received AWGN at node k with variance ok.

[0027] The first term in equation (4) describes the residual SI signal after analogue cancellation and A / D TX / RX beamforming, while its second term contains the A / D RX combined signal transmitted from node m plus AWGN. Ckneeds to cancel the SI channe , which is a matrix of dimension x Nkand not the actual Mkx NkSI channel Hk>k.

[0028] One approach for making the best use of the MIMO radio equipment would be to use a sum-rate optimisation, which tries to maximise the sums of the UL and DL communication data rates. This maximisation is discussed hereafter mainly for illustrative and completeness purposes, focusing on the FD HBF node k in figure 1 and presenting a sum-rate optimization framework for the joint design ,V(BB) . .(RF)o. | |(BB) Vk, Uk, and Uk.

[0029] Using the notation Vk- VkRF)VkBB)and assuming capacity-achieving combining at node q, the achievable DL rate that is a function of the A / D TX precoding matrices VkRF)and VkBB)of node k as well as the digital TX precoder Vmof node m, is given by

[0030] PDL= l092 (det where QqG CMq XMqdenotes the covariance matrix of the Interference-plus-Noise (IpN) at node q that is obtained as

[0031] Hereafter it is assumed for simplicity that there is no inter-node interference between the HD multi-antenna nodes q and m due to, for example, appropriate node scheduling for the FD operation at node k. The latter assumption translates to setting the channel matrix between those involved nodes in equation (3) as Hq m= 0MqxNk, which means that equation (6) simplifies to Qq= Oqliviq-

[0032] For the computation of the achievable UL rate, the notation Uk- UkRF)UkBB)is used to express this rate as a function of the A / D RX combiners UkRF)and UkBB), the A / D TX precoders Vk , and the analogue cancellation matrix Ckof node k as well as of the digital TX precoder Vmof node m. Using equation (4), the UL rate is given by where Qkcdm Xdmdenotes the IpN covariance matrix after A / D RX combining at node q, which can be expressed as

[0033] ' (RF) (RF)

[0034] In the latter expression, Hk kG CMk x Nkdenotes the effective SI channel after performing analogue TX / RX beamforming and analogue cancellation, which is defined as

[0035] Using the expressions in equations (5) and (7) for the achievable DL and UL rates, respectively, the sum-rate optimization problem for the joint design of the analogue canceller and the A / D TX / RX beamformers is mathematically expressed as where constraint (Cl) relates to the average TX power at node k and constraint ( C2 ) refers to the hardware capabilities of the analogue canceller. Constraint (C3) imposes the thresholdA€ R on the average power of the residual SI signal after analogue cancellation and analogue TX / RX beamforming. Finally, constraint ( C4) refers to the predefined TX and RX beam codebooks. To tackle On, which is a non- convex problem with non-convex constraints, a decoupled way is adopted that in this case requires at most amax- NkRF)-1 iterations including closed form expressions for the design parameters. After solving for ,VkBB), and UkRF), maximizing the DL rate, Ukmaximizing the UL rate must be found. This results in the following optimization subproblem for the design of Ck,VkRF),VkBB), and UkRF):

[0036] For the solution of the latter problem an alternating optimization approach is used. First, find VkRF)and UkRF)constrained as (C4) that boosts the DL rate, while minimizing the SI signal before any other form of cancellation. Particularly, perform the following exhaustive search:

[0037] Given the solution for OII2 and supposing that the available number of analogue

[0038] (BB) canceller taps N and a realization of Cksatisfying (C2) are given, a search for Vkmaximizing the DL rate while meeting (C1) and (C3) is next. The latter procedure is repeated for all allowable realizations of Ckfor the given N in order to find the best Vksolving OII1 ; this procedure is summarized in the method shown in pseudocode below:

[0039] The values for Ck,VkRF),VkBB), and UkRF)solving OII1 are finally substituted into the (BB) achievable UL rate expression pULin equation (7). The Ukmaximizing this point- to-point MIMO rate can be is obtained in closed form, e.g., by using the method described by G. C. Alexandropoulos and C. B. Papadias, in "A reconfigurable iterative algorithm for the K-user MIMO interference channel," Signal Process.

[0040] (Elsevier), vol. 93, no. 12, pp. 3353-3362, Dec. 2013. While the foregoing method provides beamformers that optimise the sum-rate, such optimisation is unlikely to also provide an optimised solution for ISAC applications. The present invention thus proposes a different optimisation problem, which optimises the communication parameters and the sensing parameters, while at the same time minimising the SI. This different optimisation problem leads to a different optimisation of the beamformers, as will be discussed in the following.

[0041] Prior to discussing the optimisation of the beamformers of a MIMO radio equipment having multiple antennas the system model of the SI channel will be discussed. The SI channel considered in this specification is designed by assuming a Rician fading channel i.e.,

[0042] Hsi with where p is the normalisation factor which ensures that E || HLOS|| p = NbMband rmnis the distance between m-th element of transmitter and n-th element of receiver and is given further below.

[0043] For simplification it is assumed that the SI channel is entirely deterministic and only a line-of-sight (LOS) component is present. Hence

[0044] HRefl=0

[0045] It is further assumed that each of the N transmit (TX) radio frequency (RF) chains, i.e., each of the N RF paths to a corresponding TX antenna, is connected with each receiver (RX) of the M RF chains, i.e., with each of the M RF paths from a corresponding RX antenna, via an analogue SI canceller circuit. The analogue SI canceller circuit is represented by a matrix Cbthat collects all parameters and properties of the analogue SI canceller circuit. The analogue SI canceller considered herein is tap-based and comprises N taps, each of the N taps being connected via a respective NbF-to-1 -multiplexer (MUX) to each of the NbFoutputs of the TX RF chains. A tap includes a fixed delay, a variable phase shifter and a variable attenuator. SI cancellation is achieved by subtracting each respective TX signal of the RF radio equipment from each of the respective RX signals such that only “foreign” RX signals, i.e., signals from other RF radio equipment, remains. For routing the cancellation signal from each of the taps to each of the RF chains a 1-to-MbFdemultiplexer (DEMUX) is provided. The subtraction can be implemented via power combiners while the analogue RF MUX / DEMUX can be implemented with RF switches.

[0046] The cancellation matrix can be represented as Cb = where LT

[0047] G RNxNgFL GRNxN, and L3G RM*FxN. The elements [LJ.. and [L . .with take the binary values 0 and 1, and it must hold that

[0048] The L2in Cb is a diagonal matrix whose complex entries represent the attenuation and phase shift of the canceller taps. The magnitude and phase of the element [L2]. . with i = 1, 2, 3,..., N specify the attenuation and phase of the z-th tap. Recall that the tap delays in each canceller tap are fixed, hence, the effects of the z’-th tap delay are modelled as a phase shift that is incorporated to the phase of [ L2] ...

[0049] In order to design the analogue SI canceller, Cb, imperfect channel estimation in the RF radio equipment is assumed, i.e., a uniformly distributed error is introduced in each entry of the SI channel matrix HS|, and this estimate is then used for designing Cb-

[0050] The Cb is designed by checking how many canceller taps are available. If the number of taps is zero then there will be no analogue cancellation. If the number of taps 0 < ntaps NbFMbFthen the particular entry for the channel is subtracted by imperfect estimation of that entry. Hsi.eff = (wgF)HHS|VgF

[0051] Where t is an all-zero vector except for ones at the respective locations of the RF TX chains taps and r is likewise an all-zero vector except for ones at the respective locations of the RF RX chains taps. Moreover, HS! effcan be defined as

[0052] Where Herris the error in the estimation of self-interference channel and is modelled as random matrix.

[0053] In the analogue SI canceller some error in magnitude and phase in the canceller taps may be present, resulting in imperfect SI cancellation. The remaining SI cancellation may be handled by digital SI cancellation, which is discussed hereafter.

[0054] An ideal digital canceller Dbis designed such that the residual SI power is cancelled, i.e.,

[0055] However, in practical systems an error in the realization of Dbwill remain, such that with the digital channel estimation error HErr>D, whose each entry is modelled as zero mean and variance, i.e., [ HErr>D]i ~ XN(o,aErr D).

[0056] Thus, the optimal solution for the practical system lies in finding the matrix Dbthat provides the minimum remaining SI, i.e.,

[0057] For minimising the remaining SI over the variable Dbthe solution is intuitively presented as

[0058] However, since the true HS|eff is unknown Dbwill be designed as

[0059] The combined output after both digital and analogue cancellation is

[0060] Hsi

[0061] Given the objective of optimising both the radar properties and the communication properties simultaneously, the RX and / or TX beamformers the invention proposes to determine analogue and digital beamformers that do not produce the optimum SI cancellation, but just minimise the SI signal to a certain extent. In other words, an initial SI leakage at the radio equipment is restricted but not minimised, i.e., remains below a predetermined threshold that still permits eliminating the SI leakage through the analogue and digital SI canceller stages. This allows for more leeway when designing the beamformers for optimised sensing and communication. The less-than- optimal beamforming for SI cancellation permits an improved sensing performance, which sensing relies on reflections of transmit signals off of objects. While a narrow BF reduces SI and optimises communication, the sensing performance is severely impaired due to fewer reflexions arriving at the RX antennas. In fact, objects located outside a narrow RF beam will not reflect any signal, and will go undetected.

[0062] The radar (Rad) SINR and DL SNR's are defined as where WbFis the ISAC RX beamformer (analogue RX beamformer), HRadis the channel matrix of the radar channel, VdFis the RF TX beamformer matrix (analogue TX beamformer) at the FD radio equipment, VdBis the TX precoder matrix (digital TX beamformer) at the FD radio equipment, HS| is the channel matrix of the SI channel, Cb is the analogue SI canceller matrix, D is the digital SI canceller matrix, is the RX RF combiner matrix (analogue RX beamformer) at the FD radio equipment, HDLis the channel matrix of the DL communication channel, and oband ouare the RX radar and communication noise, respectively, at the FD radio equipment.

[0063] The objective is thus to find such that SI is below a predetermined value, but not zero, optimise Cb + Db, then find V®8such that YDL and YRad are in accordance with a predetermined or desired ratio or within a ratio range. After finding the analogue beamformers, the problem of finding the digital precoder is redesigned into the least square with quadratic constraints, which is a convex problem. This objective can also be expressed as ctive with convex constraints

[0064] VFand Wy can be found through a linear search over the BF codebook. Then calculate G = HD|VD^: :s} )where VD, is the right singular vectors of HDhNow solve for the digital precoder by solving the convex problem depicted above. As the local minima of the convex problem will always be global minima, solvers such as CVX or other numerical methods can be employed to find the desired precoders.

[0065] The proposed method will in the following be evaluated through simulations. For the simulations it is assumed that the noise power at the receiver end of the considered ISAC transceiver and DL user are -90dBm. Both results are shown in figure 3 and figure 4, where the variation in noise power is corresponding to the estimation error in digital cancellation i.e., oErr,D ■

[0066] Figure 3 shows the radar SINR with respect to the transmit power.

[0067] Figure 4 shows the DL SNR with respect to the transmit power.

[0068] Figure 5 shows a Range Angular Map with perfect Digital Cancellation and three users at -45, 45 and 70 degrees with ranges 40, 50 and 60 meters

[0069] Figure 6 shows a Range Angular Map with aErr D= -70dBm and three users as previously 2

[0070] Figure 7 shows a Range Angular Map oErr D= -50dBm and three users as previously

[0071] Figure 8 shows the radar SNR over the TX power.

[0072] Figure 9 shows the DL SNR over the TX power for different SI saturation values, with values obtained for beamfoerms set in accordance with the prior art method discussed by M. A. Islam, G. C. Alexandropoulos and B. Smida, in "Integrated Sensing and Communication with Millimeter Wave Full Duplex Hybrid Beamforming" ICC 2022 - IEEE International Conference on Communications, Seoul, Korea, Republic of, 2022, pp. 4673-4678, doi: 10.1109 / ICC45855.2022.9838368 provided for comparison.

[0073] Figure 10 shows a schematic flow diagram of the method 100 of processing communication signals for wireless transmission via multiple antennas of a multiple- input-multiple-output (MIMO) radio equipment (node k) in accordance with the first aspect of the invention. Each of the multiple antennas is connected to a transmission (TX) and / or a reception (RX) signal chain. The method comprises estimating (110) channel matrices for communication channels (HDL) between the MIMO radio equipment (node k) and one or more target radio equipment (node q, node m), for sensing channels (HRad) between the MIMO radio equipment (node k) and a sector of space coverable by electronic beamforming of wireless signals transmitted by the multiple antennas, and for a self-interference (SI) channel (HS|) between the multiple antennas of the MIMO radio equipment (node k). In step 120 the method comprises determining analogue TX (VdF) and RX (WdF) beamforming parameters for the respective signals transmitted and / or received via the multiple antennas, targeted to balance the properties of the signals transmitted via the respective communication link between a maximum SINR for communication with a respective target radio equipment and a maximum SINR for sensing in accordance with a predetermined or user settable ratio or ratio range, while reducing self-interference (SI) below a predetermined first value. Finally, in step 130, the method comprises determining digital TX and / or RX precoding parameters (V®8, Wd8) for the respective signals transmitted and / or received via the multiple antennas targeted to balance properties of the signals transmitted via the respective communication link between a maximum SINR for communication and a maximum SINR for sensing in accordance with the predetermined or user settable ratio or ratio range, while further reducing SI below a predetermined second value.

[0074] Figure 12 shows an exemplary flow diagram of a method 200 of processing communication signals for use in a multiple-input-multiple-output (MIMO) radio equipment (node k) having multiple antennas in accordance with a second aspect of the invention. Each of the multiple antennas is connected to a transmission (TX) and / or a reception (RX) signal chain. The method comprises, for joint communication and sensing (JCAS), processing communication signals for wireless transmission using with the method (100) in accordance with the first aspect of the invention. The method further comprises, in step 140, parameterising an analogue SI canceller stage (Cb) that selectively connects each TX RF chain with each RX RF chain targeting to provide a first level of SI cancellation, and in step 150, parameterising a digital SI canceller stage (Db) targeting to provide a second level of SI cancellation. In step 160 the method comprises transmitting (160 / 230) the processed wireless signals as one or more directed TX RF beams. In step 170 the method comprises receiving wireless signals transmitted by one or more remote radio equipment and reflexions of signals transmitted by the multiple-input-multiple-output (MIMO) radio equipment (node k), reflected off of objects in a space covered by the one or more directed TX RF beams. In step 180 the the SI in the analogue (Cb) and digital SI canceller stages (Db) is cancelled, and in step 190 at least the signals transmitted by one or more remote radio equipment are provided to a digital RX beamformer stage ( UgB), for separating the communication signals from the one or more remote radio equipment. The separated communication signals from the one or more remote radio equipment and the remaining received signals are provided to a JCAS signal processing stage in step 191 .

[0075] In light of the discussion above, in accordance with a first aspect of the invention a method of processing communication signals for wireless transmission via multiple antennas of a multiple-input-multiple-output (MIMO) radio equipment is presented. In the MIMO equipment each of the multiple antennas is connected to a transmission (TX) and / or a reception (RX) signal chain. The method comprises estimating channel matrices for communication channels between the MIMO radio equipment and one or more target radio equipment, for sensing channels between the radio MIMO equipment and a sector of space coverable by electronic beamforming of wireless signals transmitted by the multiple antennas, and for the self-interference (SI) channel between the multiple antennas of the MIMO radio equipment. In a next step, analogue TX and RX beamforming parameters for the respective signals transmitted and / or received via the multiple antennas are determined. The beamforming targets to balance the fitness / suitability / properties of the signals transmitted via the respective communication link between a maximum SINR for communication purposes with a respective target radio equipment and a maximum SINR for sensing purposes in accordance with a predetermined or user settable ratio or ratio range, while at the same time reducing self-interference (SI) below a predetermined first value.

[0076] Determining the analogue TX and RX beamforming parameters may be based on the known locations of the antennae and the estimated channel matrix for the SI channel, and may further be based on estimated channel matrices for the DL communication channel and the radar channel. Determining the analogue TX and RX beamforming parameters may further or alternatively be targeted to maintain RX signal levels of the SI signal below a predetermined third value. The latter will ensure that the SI signal does not take up too much of the available dynamic range of the RX analogue- to-digital converter (ADC), which in turn results in a better signal quality of the non-SI signals, i.e., signals received from remote transmitters and reflected signals.

[0077] Precoding aligns the vector containing the transmit symbols, i.e., the transmit vector, with the eigenvector(s) of the channel, for each antenna and the corresponding channel. In simple terms, it transforms the transmit symbols' vector in such a way that the vector reaches the receiver in the strongest form that is possible in the given channel and under the respective optimisation target. In a way, precoding is similar to equalization, but the main difference is that the precoder is optimised for a remote decoder. While channel equalization aims to minimize channel errors, the precoder aims to minimize the error in the receiver output. Once the analogue TX and RX beamforming parameters are determined, digital TX precoding parameters for the respective signals transmitted and / or received via the multiple antennas are determined. Similar to the previous analogue beamforming determining the digital TX precoding parameters are determined targeted to balance properties of the signals transmitted via the respective communication link between a maximum SINR for communication purposes with a respective target radio equipment and a maximum SINR for sensing purposes in accordance with the predetermined or user settable ratio or ratio range, while further reducing SI below a predetermined second value.

[0078] In one or more embodiments of the method estimating channel matrices, determining analogue TX and RX beamforming parameters and / or determining digital TX precoding parameters is repeated after a transmission event has begun. The respective updated parameters may be applied for a remainder of signals to be sent during an ongoing transmission event or for signals to be sent in a subsequent transmission event. Repeating the estimation and / or the determining steps may use or be based on signals previously received via the multiple antennas. The received signals may comprise communication signals as well as channel status information signals transmitted in addition thereto. The estimation and / or determining steps may be based on received signals of a previously completed or terminated transmission event, but may also be dynamically updated during an ongoing transmission event, based on signals received so far.

[0079] In one or more embodiments of the method determining respective parameters for analogue TX and RX beamforming for each of the TX and RX signal chains, respectively, comprises determining respective phase and / or amplitude variation values for each of the TX and RX antennas. This may include performing a linear search over a beamforming codebook.

[0080] In one or more embodiments of the method determining digital TX precoding parameters V®Bcomprises invoking a numerical solving process.

[0081] In one or more embodiments of the method determining digital TX precoding parameters comprises employing a singular value decomposition (SVD) of an auxiliary downlink channel without considering the corresponding channel coefficient. Finding the candidate precoder may be limited or focused on the angular directions towards a communication user. The angular directions may be used for initializing the auxiliary downlink channel and perform the SVD thereon. Only use the singular vectors are used, not the constituting singular values.

[0082] In accordance with a second aspect of the invention a method of processing communication signals for joint communication and sensing (JCAS) in a MIMO radio equipment is presented. The MIMO radio equipment has multiple antennas. Each of the multiple antennas is connected to a transmission (TX) and / or a reception (RX) signal chain. The method comprises processing communication signals for wireless transmission exploiting the method in accordance with the first aspect of the invention as described above. The method further comprises parameterising an analogue SI canceller stage that selectively connects each TX RF chain with each RX RF chain, targeting to provide a first level of SI cancellation, and parameterising a digital SI canceller stage targeting to provide a second level of SI cancellation. Once the various stages are parameterised, the processed wireless signals are transmitted as one or more directed TX RF beams, and wireless signals transmitted by one or more remote radio equipment as well as reflexions of signals transmitted by the MIMO radio equipment, reflected off of objects in a space covered by the one or more directed TX RF beams, are received. Self-interference (SI) is cancelled in the analogue and digital SI canceller stages, in accordance with the previous parameterisation. At least the signals transmitted by one or more remote radio equipment remaining after the SI cancellation may then be provided to a digital RX beamformer stage, for separating the communication signals from the one or more remote radio equipment and, ultimately, for extracting messages contained therein.

[0083] In accordance with a third aspect of the present invention a wireless apparatus configured for processing wireless communication signals for communication and sensing is presented. The wireless apparatus comprise multiple antennas, associated circuitry for processing radio frequency signals, one or more microprocessors, and associated volatile and non-volatile memory. The elements or components are connected via one or more data and / or signal lines or buses. The non-volatile memory stores computer program instructions which, when executed by the one or more microprocessors, configure elements or components of the wireless apparatus to implement or carry out one or more embodiments of the method in accordance with the first aspect of the present invention.

[0084] In one or more embodiments the circuitry for processing radio frequency signals comprises a low noise amplifier and / or a mixer configured for providing a representation of a received signal at an intermediate frequency. The mixer preferably uses a same oscillator signal as a transmitter co-located with the receiver in the wireless apparatus. The latter may enable using signals transmitted by the entity comprising the receiver, which are reflected off objects, for environment perception.

[0085] The present invention suggests separately optimising the analogue and digital beamformers, notably first optimising the analogue beamformer in a conventional manner, and afterwards optimising the digital beamformer by mapping the optimisation problem into a least squares problem with quadratic constraints, which is convex and thus open to an algebraic solution. When designing the beamformers the target is to maintain the SI leakage below a predetermined threshold, such that the residual leakage can be satisfyingly cancelled in a conventional manner in the receiver’s analogue and digital SI cancellers. The beamformer design further targets to optimise for maximum SNR. Designing the digital beamformer may comprise applying numerical methods or applying a CVX process. The combined efforts of optimising the beamformers for minimum SI leakage and maximum SNR result in optimised sensing and communication metrics.

[0086] The present invention provides a reasonable DL rate at the receiver with minimised SI even at high transmit SNR, which high transmit SNR inevitably increases the SI power at the receiver. Thus, the present invention is applicable notably, but not limited to, FD communication systems, where high transmit SNR is required for proper operation.

[0087] As will be appreciated by one skilled in the art, aspects of the embodiments may be embodied as a system, apparatus, method, or program product. Accordingly, embodiments may take the form of an entirely hardware embodiment, an entirely software-implemented embodiment, including firmware, resident software, microcode, etc., or an embodiment combining software and hardware aspects.

[0088] For example, the disclosed embodiments may be implemented as a hardware circuit comprising custom very-large-scale integration (VLSI) circuits or gate arrays, off-the- shelf semiconductors such as logic chips, transistors, or other discrete components. The disclosed embodiments may also be implemented in programmable hardware devices such as field programmable gate arrays, programmable array logic, programmable logic devices, or the like. As another example, the disclosed embodiments may include one or more physical or logical blocks of executable code which may, for instance, be organised as an object, procedure, or function.

[0089] The method presented hereinbefore may be represented by computer program instructions. Accordingly, in accordance with a further aspect of the invention, a computer program product comprises computer program instructions which, when executed by a microprocessor of a wireless apparatus in accordance with the second aspect of the invention, cause the microprocessor to execute methods in accordance with the first aspect of the present invention, and to accordingly control hardware and / or software blocks or modules of the wireless apparatus.

[0090] Computer program instructions, or code, for carrying out operations for embodiments may be any number of lines and may be written in any combination of one or more programming languages including an object- oriented programming language such as Python, Ruby, Java, Smalltalk, C++, or the like, and conventional procedural programming languages, such as the “C” programming language, or the like, and / or machine languages such as assembly languages. The code may execute entirely on the user’s computer, partly on the user’s computer, as a stand-alone software package, partly on the user’s computer and partly on a remote computer or entirely on the remote computer or server. In the latter scenario, the remote computer may be connected to the user’s computer through any type of network, including a local area network (LAN), wireless LAN (WLAN), or a wide area network (WAN), or the connection may be made to an external computer, for example, through the Internet using an Internet Service Provider (ISP). The computer program instructions may be retrievably stored or transmitted on a computer-readable medium or data carrier. The medium or the data carrier may by tangibly or physically embodied, e.g., in the form of a hard disk, solid state disk, flash memory device or the like. However, the medium or the data carrier may also comprise a modulated electro-magnetic, electrical, or optical signal that is received by the computer by means of a corresponding receiver, and that is transferred to and stored in a memory of the computer.

[0091] The described features, structures, or characteristics of the embodiments may be combined in any suitable manner. In this description, numerous specific details are provided, such as examples of programming, software modules, user selections, network transactions, database queries, database structures, hardware modules, hardware circuits, hardware chips, etc., to provide a thorough understanding of embodiments. One skilled in the relevant art will recognize, however, that embodiments may be practiced without one or more of the specific details, or with other methods, components, materials, and so forth. In other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of an embodiment. Reference throughout this specification to “one embodiment,” “an embodiment,” or similar language means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment. Thus, appearances of the phrases “in one embodiment,” “in an embodiment,” and similar language throughout this specification may, but do not necessarily, all refer to the same embodiment, but mean “one or more but not all embodiments” unless expressly specified otherwise. The terms “including,” “comprising,” “having,” and variations thereof mean “including but not limited to,” unless expressly specified otherwise. An enumerated listing of items does not imply that any or all of the items are mutually exclusive, unless expressly specified otherwise. The terms “a,” “an,” and “the” also refer to “one or more” unless expressly specified otherwise.

[0092] Where aspects of the embodiments are described in this specification with reference to schematic flowchart diagrams and / or schematic block diagrams of methods, apparatuses, systems, and program products according to embodiments it will be understood that each block of the schematic flowchart diagrams and / or schematic block diagrams, and combinations of blocks in the schematic flowchart diagrams and / or schematic block diagrams, can be implemented by code. This code may be provided to a processor of a general-purpose computer, special purpose computer, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, create means for implementing the functions / acts specified in the flowchart diagrams and / or block diagrams.

[0093] It should be noted that, in some implementations or embodiments, the functions noted in the exemplary embodiments shown in the figures may occur out of the order shown in the figures. For example, two blocks shown in succession may, in fact, be executed substantially concurrently, or the blocks may sometimes be executed in the reverse order, depending upon the functionality involved. Other steps and methods may be conceived that are equivalent in function, logic, or effect to one or more blocks, or portions thereof, shown in the figures.

Claims

CLAIMS1. A method (100) of processing communication signals for wireless transmission via multiple antennas of a multiple-input-multiple-output (MIMO) radio equipment (node k), each of the multiple antennas being connected to a transmission (TX) and / or a reception (RX) signal chain, the method comprising:- estimating (110) channel matrices for communication channels (HDL) between the MIMO radio equipment (node k) and one or more target radio equipment (node q, node m), for sensing channels (HRad) between the MIMO radio equipment (node k) and a sector of space coverable by electronic beamforming of wireless signals transmitted by the multiple antennas, and for a self-interference (SI) channel (HS|) between the multiple antennas of the MIMO radio equipment (node k),- determining (120) analogue TX (VtF) and RX (WdF) beamforming parameters for the respective signals transmitted and / or received via the multiple antennas, targeted to balance the properties of the signals transmitted via the respective communication link between a maximum SINR for communication with a respective target radio equipment and a maximum SINR for sensing in accordance with a predetermined or user settable ratio or ratio range, while reducing self-interference (SI) below a predetermined first value,- determining (130) digital TX and / or RX precoding parametersfor the respective signals transmitted and / or received via the multiple antennas targeted to balance properties of the signals transmitted via the respective communication link between a maximum SINR for communication and a maximum SINR for sensing in accordance with the predetermined or user settable ratio or ratio range, while further reducing SI below a predetermined second value.

2. The method of claim 1 , wherein estimating (110) channel matrices, determining (120) analogue TX (VdF) and RX (WdF) beamforming parameters and / or determining (130) digital TX precoding parameters is repeated after a transmission event has begun, and the respective updated parametersare applied for a remainder of signals to be sent during an ongoing transmission event or for signals to be sent in a subsequent transmission event.

3. The method (100) of claim 1 or 2, wherein determining (120) respective parameters for analogue TX and RX beamforming for each of the TX and RX signal chains, respectively, comprises determining respective phase and / or amplitude variation values for each of the TX and RX antennas.

4. The method (100) of claim 1 , 2 or 3, wherein determining (120) respective parameters for analogue TX and RX beamforming for each of the TX and RX signal chains, respectively, is based on known locations of the antennae of the MIMO radio equipment (node k) and the estimated channel matrix for the SI channel (HS|), and / or is targeted to maintain RX signal levels of the SI signal below a predetermined third value.

5. The method (100) of any one or more of claims 1 to 4, wherein determiningR R(130) digital TX precoding parameters (Vb) comprises invoking a numerical solving process.

6. The method (100) of any one or more of claims 1 to 5, wherein determining (130) digital TX precoding parameters (VbB) comprises employing a singular value decomposition of an auxiliary downlink channel.

7. A method (200) of processing communication signals for use in a multiple- input-multiple-output (MIMO) radio equipment (node k) having multiple antennas , each of the multiple antennas being connected to a transmission (TX) and / or a reception (RX) signal chain, the method comprising, for joint communication and sensing (JCAS):- processing communication signals for wireless transmission in accordance with the method (100) of one or more of claims 1 to 6,- parameterising (140 / 210) an analogue SI canceller stage (Cb) that selectively connects each TX RF chain with each RX RF chain targeting to provide a firstlevel of SI cancellation,- parameterising (150 / 220) a digital SI canceller stage (Db) targeting to provide a second level of SI cancellation,- transmitting (160 / 230) the processed wireless signals as one or more directed TX RF beams,- receiving (170 / 240) wireless signals transmitted by one or more remote radio equipment and reflexions of signals transmitted by the multiple-input-multiple- output (MIMO) radio equipment (node k), reflected off of objects in a space covered by the one or more directed TX RF beams,- cancelling (180 / 250) the SI in the analogue (Cb) and digital SI canceller stages (Db),- providing (190 / 260) at least the signals transmitted by one or more remote radio equipment to a digital RX beamformer stage (UbB), for separating the communication signals from the one or more remote radio equipment , and- providing (191 / 270) the separated communication signals from the one or more remote radio equipment and the remaining received signals to a JCAS signal processing stage.

8. Wireless apparatus (300) configured for receiving quadrature-modulated signal components of transmit symbols, comprising two or more antennas (302), circuitry (304) for processing radio frequency signals, one or more microprocessors (306), volatile (308) and non-volatile memory (310), connected via one or more data and / or signal lines or buses (312), wherein the non-volatile memory (310) stores computer program instructions which, when executed by the one or more microprocessors (306), configure components of the wireless apparatus (300) to implement or carry out a method of any one of the preceding claims 1 to 7.

9. The wireless apparatus (300) of claim 8, wherein the circuitry (304) for processing radio frequency signals comprises a low noise amplifier and / or a mixer configured for providing a representation of a received signal at an intermediate frequency.

10. Computer program product comprising computer program instructions which, when executed by a microprocessor of a wireless apparatus (300) according to one or more of claims 8 or 9, cause the wireless apparatus (300) and / or control hardware blocks, modules or components of the wireless apparatus (300), respectively, to execute the method (100, 200) of one or more of claims 1 to 7.

11. Computer readable medium or data carrier retrievably transmitting or storing the computer program product of claim 10.

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