Control device, control program, and control method

Abstract

A control system (120) comprises: a rotary electric machine (40); inverters (20, 30) that connect armature windings (51U-51W) of the rotary electric machine and a DC power supply (10); and a control device (70). The control system (120) is configured such that a zero-axis current flows through the armature windings. The control device comprises: a feedback calculation unit (82) that calculates a feedback operation amount (Vz*) for feedback-controlling a current (Izr) flowing through the armature windings to a current command value (Iz*); and a switch control unit (84) that performs switching control of the inverter on the basis of the calculated feedback operation amount. The feedback calculation unit calculates an adjustment value (ADJ) by multiplying a harmonic component of the zero-axis current flowing through the armature windings by a control gain (k) corresponding to rotation speed (Nr) of the rotary electric machine, and adds or subtracts the calculated adjustment value to or from the feedback operation amount related to the zero-axis current so as to reduce the harmonic component of the zero-axis current.

Description

Control device, control program, and control method CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application is based on Japanese Application No. 2024-145402, filed on August 27, 2024, the contents of which are incorporated herein by reference.

[0002] The present disclosure relates to a control device, a control program, and a control method for a rotating electric machine.

[0003] Conventionally, a system including a rotating electric machine having a multi-phase armature winding and an inverter connecting the armature winding and a DC power supply is known. Among such systems, a system configured to cause a zero-axis current to flow through the armature winding is also known. An example of a system configured to suppress this zero-axis current is disclosed in Patent Document 1.

[0004] JP 2017-169251 A

[0005] However, the system disclosed in the above-mentioned Patent Document 1 does not take into consideration changes in the zero-axis current due to the rotation speed of the rotating electric machine, and it has been found that when the rotation speed increases, the zero-axis current cannot be appropriately suppressed.

[0006] A primary object of the present disclosure is to provide a control device, a control program, and a control method that can suppress the zero-axis current flowing through the armature winding.

[0007] A control device that solves the above problem is a control device for a rotating electric machine that is applied to a control system that includes a rotating electric machine having a multi-phase armature winding and an inverter that connects the armature winding and a DC power source, and is configured so that a zero-axis current flows through the armature winding, and includes: a feedback calculation unit that calculates a feedback manipulation amount for feedback controlling the current flowing through the armature winding to a current command value; and a switch control unit that performs switching control of the inverter based on the calculated feedback manipulation amount, wherein the feedback calculation unit calculates an adjustment value by multiplying a harmonic component of the zero-axis current flowing through the armature winding by a control gain that corresponds to the rotation speed of the rotating electric machine, and adds or subtracts the calculated adjustment value to or from the feedback manipulation amount related to the zero-axis current so as to reduce the harmonic component of the zero-axis current.

[0008] According to the above configuration, even if the rotation speed increases, the harmonic components of the zero-axis current can be appropriately suppressed.

[0009] A control program for solving the above-described problems is a control program executed by a control device of a rotating electric machine that is applied to a control system that includes a rotating electric machine having a multi-phase armature winding and an inverter connecting the armature winding and a DC power source, and is configured so that a zero-axis current flows through the armature winding. The control program includes: a feedback calculation step of calculating a feedback manipulation variable for feedback-controlling the current flowing through the armature winding to a current command value; and a switch control step of performing switching control of the inverter based on the calculated feedback manipulation variable. In the feedback calculation step, an adjustment value is calculated by multiplying a harmonic component of the zero-axis current flowing through the armature winding by a control gain corresponding to the rotation speed of the rotating electric machine, and the calculated adjustment value is added to or subtracted from the feedback manipulation variable related to the zero-axis current so as to reduce the harmonic component of the zero-axis current. A control method for solving the above-described problems is a control method implemented by a control device of a rotating electric machine that is applied to a control system that includes a rotating electric machine having a multi-phase armature winding and an inverter connecting the armature winding and a DC power source, and is configured so that a zero-axis current flows through the armature winding. The control method includes: a feedback calculation step of calculating a feedback manipulation variable for feedback-controlling the current flowing through the armature winding to a current command value; and a switch control step of performing switching control of the inverter based on the calculated feedback manipulation variable. In the feedback calculation step, an adjustment value is calculated by multiplying a harmonic component of the zero-axis current flowing through the armature winding by a control gain corresponding to the rotation speed of the rotating electric machine, and the calculated adjustment value is added to or subtracted from the feedback manipulation variable related to the zero-axis current so as to reduce the harmonic component of the zero-axis current.

[0010] According to the above configuration, even if the rotation speed increases, the harmonic components of the zero-axis current can be appropriately suppressed.

[0011] The above and other objects, features and advantages of the present disclosure will become more apparent from the following detailed description taken in conjunction with the accompanying drawings. 1 is a configuration diagram of a control system, FIG. 2 is a functional block diagram of control processing performed by a control device, FIG. 3 is a diagram showing the relationship between disturbance attenuation characteristics and frequency in a comparative example, FIG. 4 is a diagram showing the relationship between disturbance attenuation characteristics and frequency in this embodiment, FIG. 5 is a diagram showing the relationship between a third harmonic component of the zero-axis current and rotation speed, FIG. 6 is a functional block diagram showing a part of control processing according to a third embodiment, FIG. 7 is a diagram showing the relationship between disturbance attenuation characteristics and frequency, FIG. 8 is a functional block diagram showing a part of control processing according to a fourth embodiment, FIG. 9 is a diagram showing the zero-axis current, FIG. 10 is a diagram showing the zero-axis current, FIG. 11 is a diagram showing the zero-axis current, FIG. 12 is a diagram showing the zero-axis current, FIG. 13 is a functional block diagram showing a part of control processing according to a modified example, FIG. 14 is a diagram showing the relationship between control gain and rotation speed in the modified example, and FIG. 15 is a diagram showing the relationship between control gain and rotation speed in the modified example.

[0012] A first embodiment of a control device and a control program according to the present disclosure will be described below with reference to the drawings. The control device and the control program of the present embodiment are applied to a control system mounted on an electrically powered vehicle such as an electric vehicle or a hybrid vehicle.

[0013] As shown in Fig. 1, the control system 120 includes a battery 10, which is a DC power supply, a first inverter 20, a second inverter 30, and a motor 40, which is a rotating electrical machine. The battery 10 is, for example, a battery pack including a series connection of unit cells. The unit cell is a single battery cell or a series connection of multiple battery cells. The battery cell is, for example, a secondary battery such as a lithium-ion battery.

[0014] The first inverter 20 and the second inverter 30 are power conversion circuits that convert DC power supplied from the battery 10 into three-phase AC power and supply it to the motor 40 .

[0015] The first inverter 20 includes a series connection of a U-phase first upper arm switch SUHa and a U-phase first lower arm switch SULa, a V-phase first upper arm switch SVHa and a V-phase first lower arm switch SVLa, and a W-phase first upper arm switch SWHa and a W-phase first lower arm switch SWLa. Hereinafter, these will be collectively referred to as switches SUHa to SWLa. The upper arm and lower arm may also be collectively referred to as upper and lower arms.

[0016] Similarly, the second inverter 30 includes a series connection of a U-phase second upper arm switch SUHb and a U-phase second lower arm switch SULb, a series connection of a V-phase second upper arm switch SVHb and a V-phase second lower arm switch SVLb, and a series connection of a W-phase second upper arm switch SWHb and a W-phase second lower arm switch SWLb. Hereinafter, these will be collectively referred to as switches SUHb to SWLb.

[0017] In this embodiment, the switches SUHa to SWLa and SUHb to SWLb are voltage-controlled semiconductor switching elements, more specifically, IGBTs. In this case, the high-potential terminal of each switch SUHa to SWLa and SUHb to SWLb is the collector, and the low-potential terminal is the emitter. Freewheeling diodes DUHa, DVHa, DWHa, DULa, DVLa, DWLa, DUHb, DVHb, DWHb, DULb, DVLb, and DWLb are connected in anti-parallel to the switches SUHa, SVHa, SWHa, SULa, SVLa, SWLa, SUHb, SVHb, SWHb, SULb, SVLb, and SWLb, respectively.

[0018] The collectors of the first upper arm switches SUHa, SVHa, SWHa of each phase and the collectors of the second upper arm switches SUHb, SVHb, SWHb of each phase are connected via a positive bus 11, which is an electrical path such as a bus bar. The emitters of the first lower arm switches SULa, SVLa, SWLa of each phase and the emitters of the second lower arm switches SULb, SVLb, SWLb of each phase are connected via a negative bus 12, which is an electrical path such as a bus bar. The emitters of the second lower arm switches SULb, SVLb, SWLb of each phase are connected to the negative bus 12.

[0019] The control system 120 includes a power switch 13. The power switch 13 is, for example, a semiconductor switching element or a mechanical relay. The power switch 13 connects the positive bus 11 and the positive terminal of the battery 10. When the power switch 13 is turned on, the positive terminal of the battery 10 is electrically connected to the collectors of the first upper arm switches SUHa, SVHa, and SWHa of each phase. When the power switch 13 is turned off, the positive terminal of the battery 10 is electrically disconnected from the collectors of the first upper arm switches SUHa, SVHa, and SWHa of each phase.

[0020] The control system 120 includes a capacitor 15. The capacitor 15 functions as a smoothing capacitor. The capacitor 15 is connected in parallel to the series connection of the first upper arm switches SUHa to SWHa of each phase and the first lower arm switches SULa to SWLa of each phase.

[0021] The motor 40 is an on-board main engine that serves as a power source for running the vehicle. The motor 40 includes a rotor 41 and a stator 50. The rotor 41 is capable of transmitting power to the drive wheels of the vehicle. In this embodiment, the motor 40 is a permanent magnet field type synchronous machine. The rotor 41 includes permanent magnets 42 (e.g., neodymium magnets) as field poles.

[0022] The stator 50 has armature windings including a U-phase winding 51U, a V-phase winding 51V, and a W-phase winding 51W. The windings 51U, 51V, and 51W of each phase are arranged at intervals of 120 electrical degrees on the stator core that constitutes the stator 50. The windings 51U, 51V, and 51W of each phase are open windings.

[0023] In each phase, the emitters of first upper switches SUHa, SVHa, SWHa and the collectors of first lower switches SULa, SVLa, SWLa are connected to first ends 51Ua, 51Va, 51Wa of the windings 51U, 51V, 51W, respectively. In each phase, the emitters of second upper switches SUHb, SVHb, SWHb and the collectors of second lower switches SULb, SVLb, SWLb are connected to second ends 51Ub, 51Vb, 51Wb of the windings 51U, 51V, 51W, respectively.

[0024] The control system 120 includes a current sensor 60, a rotation angle sensor 61, and a voltage sensor 62. The current sensor 60 detects the phase current flowing through each of the phase windings 51U, 51V, and 51W. In this embodiment, the current sensor 60 is provided at one of both ends of each of the phase windings 51U, 51V, and 51W closer to the first inverter 20. The current sensor 60 may also be provided at one of both ends of each of the phase windings 51U, 51V, and 51W closer to the second inverter 30. The rotation angle sensor 61 is, for example, a resolver, and detects the electrical angle of the rotor 41. The voltage sensor 62 detects the voltage across the capacitor 15.

[0025] The detection values ​​of the sensors 60 to 62 are input to a control device 70 included in the control system 120. The control device 70 is an electronic control unit (ECU) that performs various controls of the control system 120, and includes a processor 71 as hardware, a storage unit 72, and a communication bus 73 that connects the processor 71 and the storage unit 72. In the control system 120, each on-board device can be controlled by an ECU corresponding to that device. However, for convenience, multiple ECUs are shown as a single control device 70 in FIG. 1.

[0026] The memory unit 72 includes a memory and a storage as hardware. The memory is a storage device for storing data used in the processing of the control device 70. The memory provides the processor 71 with a working area for temporary use when the processor 71 performs processing, for example. The memory includes, for example, a ROM or a RAM. The storage is a storage device for storing various programs and data to be read and executed by the processor 71, and is a non-transitory tangible storage medium. The storage includes, for example, an HDD or a flash memory. The storage stores information about the control program for processing, such as that shown in FIG. 2, which will be described later.

[0027] For example, program information stored on a non-transient physical recording medium is installed in the storage unit 72. The recording medium is, for example, a USB memory, a CD-ROM, or a DVD. Also, for example, program information transmitted via a communication network, such as over the air (OTA), is installed in the storage unit 72.

[0028] The control device 70 controls the on / off of the switches SUHa to SWLa of the first inverter 20 and the switches SUHb to SWLb of the second inverter 30 while the power switch 13 is on, in order to control the control variable of the motor 40 to the command value. More specifically, the control device 70 performs H drive control. The H drive control is a control that switches on and off the switches SUHa to SWLa of the first inverter 20 and the switches SUHb to SWLb of the second inverter 30. In this embodiment, the control variable is torque.

[0029] Next, the control process of the motor 40 executed by the control device 70 will be described with reference to Fig. 2. Fig. 2 is a block diagram of the control process.

[0030] The command value calculation unit 80 calculates the d-axis current command value Id* and the q-axis current command value Iq* in the dq coordinate system, which is a two-phase rotating coordinate system, using a torque-dq map or the like based on the command torque Trq* received from a control device that is higher than the control device 70.

[0031] The current converter 81 calculates the d-axis current value Idr, the q-axis current value Iqr, and the zero-axis current value Izr in the dq0-axis coordinate system, which is a rotating coordinate system, based on the current values ​​Iur, Ivr, and Iwr of each phase detected by the current sensor 60 and the electrical angle θr detected by the rotation angle sensor 61. The zero-axis current is also called a zero-phase current or a z-axis current.

[0032] 2 , a current feedback unit 82 calculates a d-axis voltage command value Vd* as a feedback manipulated variable based on the d-axis current command value Id* and the d-axis current value Idr. Specifically, the current feedback unit 82 has a subtractor 91 that receives the d-axis current command value Id* and the d-axis current value Idr and calculates a d-axis current deviation, which is the difference between the d-axis current command value Id* and the d-axis current value Idr, and a controller 92 that calculates a d-axis voltage command value Vd* as a manipulated variable for feedback controlling the calculated d-axis current deviation to zero. The feedback control is, for example, proportional-integral control (PI control).

[0033] Similarly, the current feedback unit 82 calculates a q-axis voltage command value Vq* as a feedback manipulated variable based on the q-axis current command value Iq* and the q-axis current value Iqr. Specifically, the current feedback unit 82 has a subtractor 93 that receives the q-axis current command value Iq* and the q-axis current value Iqr and calculates a q-axis current deviation, which is the difference between the q-axis current command value Iq* and the q-axis current value Iqr, and a controller 94 that calculates the q-axis voltage command value Vq* as a manipulated variable for feedback controlling the calculated q-axis current deviation to zero. The feedback control is, for example, proportional-plus-integral control.

[0034] Similarly, the current feedback unit 82 calculates a zero-axis voltage command value Vz* as a feedback manipulated variable based on the zero-axis current command value Iz* and the zero-axis current value Izr. Specifically, the current feedback unit 82 includes a subtractor 95 that receives the zero-axis current command value Iz* and the zero-axis current value Izr and calculates a zero-axis current deviation, which is the difference between the zero-axis current command value Iz* and the zero-axis current value Izr, and a controller 96 that calculates a zero-axis voltage command value Vz* as a manipulated variable for feedback control of the calculated zero-axis current deviation to zero. The feedback control may be, for example, proportional-plus-integral control. The zero-axis current command value Iz* is preset to "0" (zero). As will be described in detail later, an adjustment value ADJ is added to the zero-axis voltage command value Vz* calculated by the controller 96, resulting in the final zero-axis voltage command value Vz* output from the current feedback unit 82.

[0035] The drive signal generating unit 84 generates drive signals for the switches SUHa to SWLa and SUHb to SWLb of the first and second inverters 20 and 30 based on the d-axis voltage command value Vd*, the q-axis voltage command value Vq*, and the zero-axis voltage command value Vz* calculated by the current feedback unit 82. The drive signals consist of on commands and off commands for the switches.

[0036] An example of a method for generating drive signals in the drive signal generator 84 will be described. The drive signal generator 84 calculates U-, V-, and W-phase voltage command values ​​Vu*, Vv*, and Vw* in a three-phase fixed coordinate system based on the calculated d-, q-, and zero-axis voltage command values ​​Vd*, Vq*, and Vz* and the electrical angle θr. The voltage command values ​​Vu*, Vv*, and Vw* are shifted in phase by 120° in electrical angle. The drive signal generator 84 generates drive signals for the switches SUHa to SWLb of the first and second inverters 20 and 30 based on a magnitude comparison between the U-, V-, and W-phase voltage command values ​​Vu*, Vv*, and Vw* and a carrier signal (e.g., a triangular wave signal). The drive signal generator 84 in this embodiment corresponds to a switch control unit.

[0037] Based on the generated drive signal, the drive circuit 85 controls the charge / discharge currents of the gates of the switches SUHa-SWLa and SUHb-SWLb of the first and second inverters 20 and 30. As a result, the switching of the switches SUHa-SWLa and SUHb-SWLb of the first and second inverters 20 and 30 is controlled in accordance with the drive signal. The switching patterns of the switches of the inverters 20 and 30, which are switched in accordance with the drive signal, are shifted in phase by 120° electrical angle in each phase.

[0038] The main cause of the zero-axis current is the back electromotive force generated in the motor 40. The harmonic components of the back electromotive force (third harmonic component and ninth harmonic component) become higher in frequency and larger in amplitude as the rotation speed of the motor 40 increases. Accordingly, the harmonic components of the zero-axis current also become higher in frequency and larger in amplitude as the rotation speed increases.

[0039] Here, in FIG. 3 , the disturbance suppression characteristic by the controller 96 (i.e., feedback control (PI control)) is shown by a solid line. The vertical axis represents the disturbance suppression characteristic (unit: dB), and the horizontal axis represents frequency (unit: Hz). Since the third harmonic component of the zero-axis current increases at frequency f1 in FIG. 3 , for example, it is desirable that the disturbance suppression characteristic be configured to be able to attenuate more significantly at frequency f1. However, in reality, this is not the case, and the characteristic is such that the attenuation occurs gradually depending on the frequency. For this reason, the controller 96 alone cannot sufficiently attenuate the harmonic component of the zero-axis current, resulting in an effect due to the zero-axis current.

[0040] Furthermore, the frequency of the third harmonic component of the zero-axis current varies depending on the rotational speed of the motor 40. For example, when the rotational speed of the motor 40 varies such that Nr1 < Nr2 < Nr3 < Nr4 < Nr5, the frequency of the third harmonic component of the zero-axis current also varies such that f1 < f2 < f3 < f4 < f5. For this reason, it is desirable to change the frequency to be significantly attenuated depending on the rotational speed of the motor 40. Furthermore, as described above, the higher the frequency, the larger the amplitude of the zero-axis current. Therefore, it is desirable to attenuate the zero-axis current more significantly as the rotational speed of the motor 40 increases (the rotational speed increases).

[0041] Therefore, in this embodiment, in order to suppress the third harmonic component of the zero-axis current, a resonant regulator 97 as a harmonic controller and a control constant calculation unit 98 as a setting unit are provided in the current feedback unit 82. The resonant regulator 97 and the control constant calculation unit 98 will be described below.

[0042] The resonant regulator 97 is represented by the transfer function shown in equation (eq.1). The resonant regulator 97 calculates the adjustment value ADJ by multiplying the zero-axis current deviation (Iz*-Izr) by equation (eq.1) or the transfer function shown in FIG. 2. Here, "k" is the control gain, "s" is a differential operator, "b" indicates the bandwidth, and "Wc" indicates the center frequency of the resonant regulator 97. Strictly speaking, a transfer function having a first-order characteristic in the denominator is not called a resonant regulator, but in this embodiment, to clarify the aim of the disclosure, equation (eq.1) having a first-order characteristic in the denominator will also be referred to as a resonant regulator.

[0043] As shown in equation (eq.1), the resonant regulator 97 extracts components of a predetermined frequency band from the zero-axis current deviation, multiplies the extracted components by a control gain k, and calculates the adjustment value ADJ, with a second-order transfer characteristic (second-order differential operator) in the denominator. The center frequency of the predetermined frequency band corresponds to the center frequency Wc, and the width of the frequency band corresponds to the bandwidth b. In other words, the resonant regulator 97 uses a band-pass filter with the center frequency Wc and the bandwidth b to extract components of the zero-axis current deviation in a frequency band determined by the center frequency Wc and the bandwidth b, and multiplies the extracted components by the control gain k to calculate the adjustment value ADJ.

[0044] The transfer function (analog filter) shown in equation (eq.1) is digitized and implemented (embedded in a program). In other words, the analog filter shown in equation (eq.1) is converted into a digital filter. When digitizing the transfer function shown in equation (eq.1), it is converted using, for example, a bilinear transformation. At this time, prewarping may be performed.

[0045] The control constant calculation unit 98 calculates the center frequency Wc and the control gain k before the resonant regulator 97 calculates the adjustment value ADJ. The control constant calculation unit 98 sets the center frequency Wc based on the rotational speed of the motor 40, the number of pole pairs, and the order of the harmonic component to be extracted. Specifically, the control constant calculation unit 98 calculates the center frequency Wc using equation (eq.2). In equation (eq.2), "p" is the number of pole pairs of the motor 40, "n" is the order of the harmonic component to be extracted, and "Nr" is the rotational speed (rpm) of the motor 40. Note that in this embodiment, the order "n" is preset to "3" in order to reduce the third-order harmonic component of the zero-axis current. The rotational speed of the motor 40 can be calculated from the angular velocity, etc., determined from the electrical angle θr. For this reason, the control constant calculation unit 98 inputs the electrical angle θr. In addition, in equation (eq.2), the rotation speed is divided by 60 to change the unit from "rpm" to "rps."

[0046] The control constant calculation unit 98 calculates the control gain k using equation (eq.3). In equation (eq.3), "k0" is the electromotive force constant, "n" is the order of the harmonic component to be extracted, and "Nr" is the rotation speed (unit: rpm) of the motor 40. Note that "k0" does not necessarily have to use the numerical value of the motor parameter of the electromotive force constant, and a different numerical value may be used. Also, a map configuration may be used that changes this numerical value depending on the rotation speed.

[0047] Then, the current feedback unit 82 adds the adjustment value ADJ to the zero-axis voltage command value Vz* calculated by the controller 96 to calculate the final zero-axis voltage command value Vz*, and inputs it to the drive signal generation unit 84.

[0048] The following describes the operation and effect of the current control of the control device 70 configured as described above. Figure 4 shows the disturbance suppression characteristics when an adjustment value is added by the resonant regulator 97. That is, Figure 4 shows the disturbance suppression characteristics of the controller 96 and the resonant regulator 97.

[0049] In Fig. 4, the disturbance attenuation characteristics when the rotation speed of the motor 40 is Nr1 are indicated by dashed line R1, the disturbance attenuation characteristics when the rotation speed is Nr2 are indicated by dashed line R2, the disturbance attenuation characteristics when the rotation speed is Nr3 are indicated by dashed line R3, the disturbance attenuation characteristics when the rotation speed is Nr4 are indicated by dashed line R4, and the disturbance attenuation characteristics when the rotation speed is Nr5 are indicated by dashed line R5. Furthermore, the disturbance attenuation characteristics obtained by the controller 96 without using the resonant regulator 97 and without adding the adjustment value ADJ are indicated by solid line R0. Furthermore, Fig. 5 shows the relationship between the third harmonic component of the zero-axis current (unit: A (amperes)) and the rotation speed (unit: rpm). The vertical axis represents the third harmonic component of the zero-axis current, and the horizontal axis represents the rotation speed.

[0050] As shown in FIG. 4 , the disturbance suppression characteristics of the current control of the control device 70 are configured to significantly attenuate disturbances (here, the zero-axis current) in a predetermined frequency band. The predetermined frequency band has a predetermined bandwidth b centered on a center frequency Wc. Here, the center frequency Wc coincides with the frequency f1 at which the amplitude of the third-order harmonic component of the zero-axis current increases when the rotation speed is Nr1. Therefore, when the rotation speed is Nr1, the disturbance suppression characteristics of the current control of the control device 70 significantly attenuate disturbances (here, the third-order harmonic component of the zero-axis current) in the predetermined frequency band at which the amplitude of the third-order harmonic component of the zero-axis current increases. Note that in FIG. 4 , the dashed line swings significantly downward, indicating that the disturbance suppression characteristics significantly attenuate disturbances.

[0051] Furthermore, when the rotation speed is between Nr2 and Nr5, the center frequency Wc coincides with the frequencies f2 to f5 at which the amplitude of the third harmonic component of the zero-axis current becomes large. Therefore, when the rotation speed is between Nr2 and Nr5, the disturbance suppression characteristic by the current control of the control device 70 becomes, as described above, a characteristic that greatly attenuates disturbances (here, the third harmonic component of the zero-axis current) in a predetermined frequency band where the amplitude of the third harmonic component of the zero-axis current becomes large. Therefore, even if the frequency of the third harmonic component of the zero-axis current changes depending on the rotation speed, the current control of the control device 70 can effectively suppress the disturbance.

[0052] Furthermore, the control gain k is set to a larger value as the rotation speed increases. Therefore, even if the amplitude of the third-order harmonic component of the zero-axis current increases with an increase in the rotation speed, the zero-axis current can be maintained at a similar value by increasing the control gain k and the adjustment value ADJ, as shown in FIG. 5 . Note that the dashed line in FIG. 5 indicates the third-order harmonic component of the zero-axis current when the control gain k is constant, and the solid line indicates the third-order harmonic component of the zero-axis current when the control gain k is increased according to the rotation speed. As shown by the dashed line, when the control gain k is constant, the third-order harmonic component of the zero-axis current increases according to the rotation speed. However, as shown by the solid line, when the control gain k is increased according to the rotation speed, the third-order harmonic component of the zero-axis current can be maintained constant even when the rotation speed changes.

[0053] Therefore, even if the amplitude of the third harmonic component of the zero-axis current changes depending on the rotation speed, it can be suitably suppressed by the current control of the control device 70. In this embodiment, in order to clarify the disclosed configuration, the gain characteristic of the transfer function of the controller for a specific frequency is expressed as the control gain.

[0054] Second Embodiment A second embodiment will be described in which the configuration of the control device 70 of the first embodiment is partially modified. Note that the same components as those of the first embodiment are denoted by the same reference numerals, and their description and drawings will be omitted.

[0055] In the second embodiment, upper and lower limit values ​​are predetermined for the center frequency Wc used in the resonant regulator 97. The upper limit value of the center frequency Wc is predetermined according to the maximum rotation speed of the motor 40. The maximum rotation speed is determined by the specifications of the motor 40.

[0056] The lower limit of the center frequency Wc is determined by the cutoff frequency of the controller 96 (PI control). For example, the lower limit may be set to a value of the cutoff frequency x 2π, or may be set to half the cutoff frequency. The lower limit may also be determined according to the motor time constant of the motor 40.

[0057] If the center frequency Wc calculated using equation (eq.2) is greater than the upper limit, the control constant calculation unit 98 sets the upper limit as the center frequency Wc and inputs it to the resonant regulator 97. If the center frequency Wc calculated using equation (eq.2) is less than the upper limit, the control constant calculation unit 98 sets the lower limit as the center frequency Wc and inputs it to the resonant regulator 97. If the center frequency Wc calculated using equation (eq.2) is greater than or equal to the lower limit and less than or equal to the upper limit, the control constant calculation unit 98 inputs the calculated center frequency Wc to the resonant regulator 97.

[0058] According to the second embodiment, it is possible to prevent the value of the center frequency Wc from being too large, causing the adjustment value ADJ to be infinitely close to zero and therefore not being calculated, or the value of the center frequency Wc from being too small, causing the value of the adjustment value ADJ to be unable to be determined.

[0059] Third Embodiment A third embodiment will be described in which the configuration of the control device 70 of the first embodiment is partially modified. Note that the same components as those of the first embodiment are denoted by the same reference numerals, and their description and drawings will be omitted.

[0060] As shown in FIG. 6 , the resonant regulator 197 of the third embodiment is expressed by the transfer function shown in equation (eq.4). The resonant regulator 197 then multiplies the zero-axis current deviation (Iz*-Izr) by the value calculated by equation (eq.4) to calculate the adjustment value ADJ. Here, "k" is a control gain, "s" is a differential operator, and "Wc" is the center frequency of the resonant regulator 197. The other processing and configuration are the same as those of the first embodiment, and therefore will not be described. Note that the control gain k may be a value different from that of the first embodiment.

[0061] In the third embodiment, unlike the first embodiment, the bandwidth b is omitted. Therefore, as shown by the solid line in Fig. 7, the frequency band in which disturbances are significantly attenuated becomes narrower, but on the other hand, the calculation can be simplified and the load can be reduced. The solid line in Fig. 7 shows the characteristics of the transfer function in the third embodiment, and the dashed line in Fig. 7 shows the characteristics of the transfer function in the first embodiment.

[0062] Fourth Embodiment A fourth embodiment will be described in which the configuration of the control device 70 of the first embodiment is partially modified. Note that the same components as those of the first embodiment are denoted by the same reference numerals, and their description and drawings will be omitted.

[0063] In the fourth embodiment, the control delay and the motor delay are identified, and if the sum of the control delay and the motor delay is less than a predetermined value, the control gain k is set to a positive value, and an adjustment value ADJ is added to the zero-axis voltage command value Vz* calculated by the controller 96, as in the first embodiment (as shown in FIG. 2). On the other hand, if the sum of the control delay and the motor delay is equal to or greater than a predetermined value, the control gain k is set to a negative value, and the adjustment value ADJ is subtracted from the zero-axis voltage command value Vz* calculated by the controller 96, as shown in FIG.

[0064] The control delay refers to a delay related to the control of the motor 40, such as the output delay phase Δθa or the phase delay amount Δθb due to current detection. First, the output delay phase Δθa will be described. The output delay phase Δθa (unit: rad) is calculated using equation (eq. 5). "Ts" is the control period (unit: sec) of the processor 71, and "fa" is the frequency (unit: rad / s) of interest using this equation, which is the center frequency in this embodiment. In other words, "fa" = "Wc."

[0065] In other words, it is known that there is a delay of 1.5 control cycles from the calculation of the control command value (voltage command value) to the output (actual voltage application) of the PWM signal (drive signal). The output delay phase Δθa can be calculated by converting the units of the delay of 1.5 control cycles based on the angular velocity (frequency) of the signal being handled (in this embodiment, the high-frequency component of the zero-axis current) and the control cycle of the processor 71. Note that, according to equation (eq. 5), the output delay phase Δθa of the motor 40 increases as the frequency of the third-order harmonic component related to the center frequency increases due to changes in the rotation speed. It is also known that the output delay phase Δθa changes depending on the control cycle of the processor 71 and the angular velocity of the signal being handled.

[0066] Next, the phase delay amount Δθb due to current detection will be described. When the current sensor 60 detects the current in the stator winding 52 of the motor 40, a detection delay may occur. The detection delay time is determined by an RC filter (not shown) provided between the current sensor 60 and the processor 71. Specifically, the phase delay amount Δθb due to current detection is calculated using equation (eq. 6). "R" is the resistance value of the RC filter, "C" is the capacitance of the RC filter, and "fa" is the frequency (unit: rad / s) of interest using this equation, which is the center frequency "Wc" in this embodiment.

[0067] In this embodiment, the control delay is the sum of the output delay phase Δθa and the phase delay amount Δθb, but it may be only one of the output delay phase Δθa and the phase delay amount Δθb due to current detection. Note that the phase delay amount Δθb due to current detection may be so small that it can be ignored.

[0068] Furthermore, motor delay refers to the delay between when a voltage is applied to the motor 40 and when it is output as a current (hereinafter referred to as electrical delay phase φ). The transfer function M(s) (transfer function from voltage to current) of the motor 40 is expressed as a first-order delay system transfer function, and is specifically known to be 1 / (Ls+R). Note that "R" is a resistance value (unit: Ω) indicating the motor resistance of the motor 40, "L" is an inductance value "L" (unit: H) indicating the inductance of the motor 40, and "s" is a differential operator.

[0069] Considering this equation, the electrical delay phase φ of the motor 40 is calculated by equation (eq. 7). Note that "ω" is the angular velocity (unit: rad / s), and when calculating the electrical delay phase φ, an angular velocity equivalent to the center frequency is set.

[0070] The control constant calculation unit 98 sums up the calculated control delay and motor delay, and if the sum is less than a predetermined value of 90 degrees (1 / 4 cycle), the control constant calculation unit 98 sets the calculated control gain k to a positive sign and inputs it to the resonance regulator 97. On the other hand, if the sum is equal to or greater than a predetermined value of 90 degrees (1 / 4 cycle), the control constant calculation unit 98 sets the calculated control gain k to a negative sign and inputs it to the resonance regulator 97.

[0071] The operation and effects of the fourth embodiment will be described.

[0072] When the total value of the control delay and the motor delay is less than 90 degrees (1 / 4 cycle), adding the adjustment value ADJ to the zero-axis voltage command value Vz* makes it possible to suitably suppress the zero-axis current in the motor 40, as shown in Fig. 9. Note that when the total value of the control delay and the motor delay is less than 90 degrees (1 / 4 cycle), subtracting the adjustment value ADJ from the zero-axis voltage command value Vz* will cause the zero-axis current to oscillate, as shown in Fig. 10.

[0073] On the other hand, when the sum of the control delay and the motor delay is equal to or greater than 90 degrees (¼ cycle), if the adjustment value ADJ is subtracted from the zero-axis voltage command value Vz*, the zero-axis current in the motor 40 can be suitably suppressed, as shown in FIG. 12 .

[0074] In addition, when the sum of the control delay and the motor delay is 90 degrees (1 / 4 cycle) or more, adding the adjustment value ADJ to the zero-axis voltage command value Vz* causes the zero-axis current to oscillate, as shown in FIG. 11.

[0075] (Modifications) Modifications in which the control device 70 in each of the above embodiments is changed will be described below.

[0076] In the fourth embodiment, the predetermined value is set to 90 degrees (1 / 4 cycle), but this may be changed. For example, it may be added when the total value is 0 to 60 degrees or less. It may also be subtracted when the total value is 120 to 180 degrees.

[0077] In the above embodiments, the third harmonic component of the zero-axis current is reduced. However, a harmonic component other than the third harmonic component, for example, a ninth harmonic component, may be reduced. In this case, the order [n] may be set to 9.

[0078] Alternatively, the zero-axis current may be configured to reduce both the third-order harmonic component and the ninth-order harmonic component. In this case, as shown in FIG. 13 , the current feedback unit 82 may include a third-order harmonic component resonant regulator 97 and a ninth-order harmonic component resonant regulator 297 connected in parallel. In FIG. 13 , "Wc3" is the center frequency of the third-order harmonic component resonant regulator 97 and corresponds to the frequency of the third-order harmonic component. The control constant calculation unit 98 calculates the center frequency "Wc3" based on equation (eq. 2) with the order "n" set to 3 and inputs it to the third-order harmonic component resonant regulator 97. The control constant calculation unit 98 also calculates a control gain "k" based on equation (eq. 3) with the order "n" set to 3 and inputs it to the third-order harmonic component resonant regulator 97.

[0079] On the other hand, "Wc9" is the center frequency of the 9th harmonic component resonant regulator 297 and corresponds to the frequency of the 9th harmonic component, and the control gain "k'" is the control gain used by the 9th harmonic component resonant regulator 297 and is calculated by the control constant calculation unit 98. Specifically, the control constant calculation unit 98 calculates the center frequency "Wc9" based on equation (eq.2) with the order "n" set to 9 and inputs it to the 9th harmonic component resonant regulator 297. Furthermore, the control constant calculation unit 98 calculates the control gain "k'" based on equation (eq.3) with the order "n" set to 9 and inputs it to the 9th harmonic component resonant regulator 297.

[0080] The resonant regulator 97 for the third harmonic component calculates the adjustment value ADJ in the same manner as in the first embodiment. Moreover, the resonant regulator 297 for the ninth harmonic component calculates the adjustment value ADJ' in the same manner as the resonant regulator 97 for the third harmonic component.

[0081] Then, the current feedback unit 82 adds the adjustment value ADJ and the adjustment value ADJ' to the zero axis voltage command value Vz* calculated by the controller 96 to calculate a final zero axis voltage command value Vz*, and inputs the final zero axis voltage command value Vz* to the drive signal generation unit 84. As described in the fourth embodiment, the adjustment value ADJ or the adjustment value ADJ' may be subtracted in accordance with the total value of the control delay and the motor delay.

[0082] In the above embodiment, the control gain k may be set to "0" (zero) when the rotation speed is equal to or lower than a predetermined rotation speed at which the influence of back electromotive force is small, as shown in Fig. 14. Furthermore, the control gain k may have a large gradient at a specific rotation speed in order to avoid the influence of mechanical resonance, as shown in Fig. 15.

[0083] In the above embodiment, the lower limit of the center frequency may be set so that the sum of the control delay and the motor delay is equal to or greater than a predetermined value.

[0084] In the above embodiment, the sign of the control gain k is changed based on the control delay and the motor delay, but the positive and negative signs of the control gain k may be switched depending on the center frequency, that is, the frequency of the harmonic components of the target zero-axis current. For example, the control gain k may be positive when the center frequency is less than a predetermined frequency, and negative when the center frequency is equal to or greater than the predetermined frequency.

[0085] Furthermore, the control gain k may be set to "0" (zero) when the center frequency is less than a predetermined frequency, and may be set to a negative value when the center frequency is equal to or greater than the predetermined frequency. In other words, the resonant regulator 97, 197, 297 may be controlled only in the high rotation speed region where the zero-axis current (and back electromotive force) becomes large.

[0086] The controller and methods described herein may be implemented by a special-purpose computer configured with a processor and memory programmed to perform one or more functions embodied in a computer program. Alternatively, the controller and methods described herein may be implemented by a special-purpose computer configured with a processor configured with one or more dedicated hardware logic circuits. Alternatively, the controller and methods described herein may be implemented by one or more special-purpose computers configured with a processor and memory programmed to perform one or more functions in combination with a processor configured with one or more hardware logic circuits. Furthermore, the computer program may be stored as instructions executed by a computer on a computer-readable non-transitory storage medium.

[0087] The following additionally describes technical ideas that can be extracted from the above-described embodiments and modified examples. [Configuration 1] A control device (70) for a rotating electric machine applied to a control system (120) including a rotating electric machine (40) having a multi-phase armature winding (51U-51W), and an inverter (20, 30) connecting the armature winding and a DC power supply (10), wherein the control device (70) is configured so that a zero-axis current flows through the armature winding, the control device (70) including: a feedback calculation unit (82) that calculates a feedback manipulated variable (Vz*) for feedback-controlling a current (Izr) flowing through the armature winding to a current command value (Iz*), and a switch control unit (84) that performs switching control of the inverter based on the calculated feedback manipulated variable, and a harmonic controller (97) having a second-order transfer characteristic in a denominator, the harmonic controller extracting a component of a predetermined frequency band including the harmonic component of the zero-axis current from the zero-axis current, and multiplying the extracted component by the set control gain to calculate the adjustment value to be added to or subtracted from the feedback manipulated variable. [Configuration 3] The control device according to Configuration 2, wherein the center frequency (Wc) of the frequency band extracted by the harmonic controller is calculated based on the order (n) of the harmonic component to be extracted, the rotation speed (Nr) of the rotating electric machine, and the number of pole pairs (p) of the rotating electric machine. [Configuration 4] The control device according to Configuration 3, wherein the upper limit of the center frequency is set based on the maximum rotation speed of the rotating electric machine. [Configuration 5] The control device according to Configuration 3 or 4, wherein the lower limit of the center frequency is set based on a cutoff frequency or an electrical time constant of the rotating electric machine.[Configuration 6] The control device according to any one of configurations 3 to 5, wherein the lower limit of the center frequency is set so that a sum of a control delay related to control of the rotating electric machine and a motor delay that is a delay based on the electrical characteristics of the rotating electric machine is equal to or greater than a predetermined value. [Configuration 7] The control device according to any one of configurations 3 to 6, wherein the frequency band extracted by the harmonic controller is set to have a predetermined bandwidth (b). [Configuration 8] The control device according to any one of configurations 1 to 7, wherein when a sum of a control delay related to control of the rotating electric machine and a motor delay that is a delay based on the electrical characteristics of the rotating electric machine is less than a predetermined value, the adjustment value is added, and when the sum is equal to or greater than the predetermined value, the adjustment value is subtracted from the feedback manipulated variable. [Configuration 9] A control program executed by a control device (70) for a rotating electric machine applied to a control system (120) including a rotating electric machine (40) having armature windings (51U-51W) of multiple phases, and an inverter (20, 30) connecting the armature windings and a DC power supply (10), configured so that a zero-axis current flows through the armature winding, includes: a feedback calculation step of calculating a feedback manipulated variable for feedback-controlling a current (Izr) flowing through the armature winding to a current command value (Iz*); and a switch control step of performing switching control of the inverter based on the calculated feedback manipulated variable, the feedback calculation step calculates an adjustment value (ADJ) by multiplying a harmonic component of the zero-axis current flowing through the armature winding by a control gain (k) corresponding to a rotation speed (Nr) of the rotating electric machine, and adds or subtracts the calculated adjustment value to or from the feedback manipulated variable related to the zero-axis current so as to reduce the harmonic component of the zero-axis current.

[0088] Although the present disclosure has been described with reference to the embodiments, it is understood that the present disclosure is not limited to the embodiments or structures. The present disclosure also encompasses various modifications and equivalent modifications. In addition, various combinations and forms, including only one element, more than one element, or less than one element, are also within the scope and spirit of the present disclosure.

Claims

1. A control device (70) for a rotating electric machine applied to a control system (120) including a rotating electric machine (40) having armature windings (51U to 51W) of multiple phases, and an inverter (20, 30) connecting the armature windings and a DC power source (10), configured so that a zero-axis current flows through the armature winding, the control device (70) comprising: a feedback calculation unit (82) that calculates a feedback operation amount (Vz*) for feedback-controlling a current (Izr) flowing through the armature winding to a current command value (Iz*); and a switch control unit (84) that performs switching control of the inverter based on the calculated feedback operation amount, the feedback calculation unit calculates an adjustment value (ADJ) by multiplying a harmonic component of the zero-axis current flowing through the armature winding by a control gain (k) corresponding to a rotation speed (Nr) of the rotary electric machine, and adds or subtracts the calculated adjustment value to or from the feedback manipulated variable related to the zero-axis current so as to reduce the harmonic component of the zero-axis current.

2. The control device according to claim 1, wherein the feedback calculation unit comprises: a setting unit (98) that sets the value of the control gain to a large value according to the rotation speed of the rotating electric machine; and a harmonic controller (97) that extracts components of a predetermined frequency band including harmonic components of the zero-axis current from the zero-axis current, multiplies the extracted components by the set control gain, and calculates the adjustment value to be added to or subtracted from the feedback manipulated variable, the harmonic controller having a second-order transfer characteristic in the denominator.

3. The control device described in claim 2, wherein the center frequency (Wc) of the frequency band extracted by the harmonic controller is calculated based on the order (n) of the harmonic component to be extracted, the rotation speed (Nr) of the rotating electric machine, and the number of pole pairs (p) of the rotating electric machine.

4. The control device according to claim 3, wherein the upper limit of the center frequency is set based on the maximum rotation speed of the rotating electrical machine.

5. The control device according to claim 3, wherein the lower limit of the center frequency is set based on a cutoff frequency or an electrical time constant of the rotating electrical machine.

6. A control device as described in claim 5, wherein the lower limit of the center frequency is set so that the sum of the control delay related to the control of the rotating electric machine and the motor delay, which is a delay based on the electrical characteristics of the rotating electric machine, is greater than or equal to a predetermined value.

7. The control device according to claim 3, wherein the frequency band extracted by the harmonic controller is set to have a predetermined bandwidth (b).

8. A control device according to any one of claims 3 to 7, wherein when the sum of the control delay relating to the control of the rotating electric machine and the motor delay, which is a delay based on the electrical characteristics of the rotating electric machine, is less than a predetermined value, the adjustment value is added, and when the sum is equal to or greater than the predetermined value, the adjustment value is subtracted from the feedback manipulated variable.

9. A control program executed by a control device (70) for a rotating electric machine applied to a control system (120) including a rotating electric machine (40) having armature windings (51U-51W) of multiple phases and an inverter (20, 30) connecting the armature windings and a DC power source (10), the control system being configured so that a zero-axis current flows through the armature winding, includes: a feedback calculation step of calculating a feedback manipulated variable for feedback-controlling a current (Izr) flowing through the armature winding to a current command value (Iz*); and a switch control step of performing switching control of the inverter based on the calculated feedback manipulated variable. the feedback calculation step calculates an adjustment value (ADJ) by multiplying a harmonic component of the zero-axis current flowing through the armature winding by a control gain (k) corresponding to a rotation speed (Nr) of the rotating electric machine, and adds or subtracts the calculated adjustment value to or from the feedback manipulated variable related to the zero-axis current so as to reduce the harmonic component of the zero-axis current.

10. A control method implemented by a control device (70) for a rotating electric machine applied to a control system (120) including a rotating electric machine (40) having a multi-phase armature winding (51U-51W) and an inverter (20, 30) connecting the armature winding and a DC power source (10), the control system being configured so that a zero-axis current flows through the armature winding, the control method comprising: a feedback calculation step of calculating a feedback manipulated variable for feedback-controlling a current (Izr) flowing through the armature winding to a current command value (Iz*); and a switch control step of performing switching control of the inverter based on the calculated feedback manipulated variable. the feedback calculation step multiplies a harmonic component of the zero-axis current flowing through the armature winding by a control gain (k) corresponding to a rotation speed (Nr) of the rotating electric machine to calculate an adjustment value (ADJ), and adds or subtracts the calculated adjustment value to or from the feedback manipulated variable related to the zero-axis current so as to reduce the harmonic component of the zero-axis current.

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