Bidirectional chopper circuit

Abstract

A bidirectional chopper circuit (1) comprises a principal power converter (11), one auxiliary power converter (12), and an inductor (13) that is connected in series to the auxiliary power converter (12). A phase difference that corresponds to the ratio of the frequency of a second carrier signal triA that is used for control of the auxiliary power converter (12) to the frequency of a first carrier signal triM that is used for control of the principal power converter (11) is provided between the first carrier signal triM and the second carrier signal triA.

Description

Bidirectional chopper circuit

[0001] This disclosure relates to a bidirectional chopper circuit that performs bidirectional voltage conversion between a first DC voltage between a pair of first external connection terminals and a second DC voltage between a pair of second external connection terminals.

[0002] In recent years, expectations for electric vehicles (EVs) have been rising from the perspective of improving the global environment. However, electric vehicles suffer from drawbacks such as limited charging infrastructure and short driving range. One method for improving driving range is a known technology that boosts the battery voltage using a bidirectional chopper circuit to achieve high-efficiency, high-output motor drive (see, for example, Non-Patent Document 1).

[0003] Furthermore, in electric vehicles, a technology is known in which the battery voltage (201.6V) is boosted to a high voltage (maximum 650V) using a bidirectional chopper (carrier frequency 10kHz, inductance 329μH), and then the voltage is supplied to the motor using an inverter (see, for example, Non-Patent Document 2). Miniaturizing and lightweighting in-vehicle power converters is an important issue. In particular, the volume and weight of a power converter depend heavily on the inductor, which is the energy storage element. Therefore, miniaturizing and lightweighting the inductor greatly contributes to miniaturizing and lightweighting the power converter. The weight and volume of an inductor can be reduced by reducing its magnetic energy, and thus the above objective can be achieved by reducing its inductance. However, reducing the inductance increases the ripple current contained in the inductor current, which may cause a decrease in power quality and instability in the converter's operation. To reduce the ripple current contained in the inductor current, it is necessary to increase the switching frequency (carrier frequency) of the bidirectional chopper circuit, but from the viewpoint of reducing electromagnetic noise, the limits of miniaturization and weight reduction through high frequency have been pointed out.

[0004] On the other hand, with the aim of miniaturizing and lightening the inductor used in bidirectional choppers, a bidirectional chopper circuit having a main power converter and an auxiliary power converter has been proposed (see, for example, Patent Document 1).

[0005] International Publication No. 2017 / 038122

[0006] Takahiro Kawashima, Shigeyuki Funabiki, Masayoshi Yamamoto, Hideki Asuke, Hiromitsu Terui, and Hideharu Takano, "Characteristic Analysis and Evaluation of Coupled Inductors in Multiphase Translink Boost Chopper Circuits," Journal of the Power Electronics Society, vol. 35, pp. 136-145, 2010 Timothy A. Burress, Steven L. Campbell, Chester Coomer, Curtis William Ayers, Andrew A. Wereszczak, Joseph Philip Cunningham, Laura D. Marino, Larry Eugene Sheibel Eugene, Hua-Tay Lin, "Evaluation of the 2010 Toyota Prius Hybrid Synergy Drive System," Oak Ridge National Laboratory, Oak Ridge Ridge, Tennessee (TN), USA, Technical Report ORNL / TM-2010 / 253, March 2011, G. A. M. Nastion, M. Matsumoto, M. Hagiwara, "Bidirectional Chopper with Single-Cell Auxiliary Full-Bridge Converter for Onboard Battery Energy Storage Systems" "System," IEEE Transactions on Power Electronics, vol. 39, No. 8, pp. 10021-10033, 2024.

[0007] According to the technology described in Patent Document 1, the magnitude of the ripple current contained in the inductor current can be reduced. However, since a large number of single-phase full-bridge power converters are used as auxiliary power converters, the size increases, converter losses increase, and costs increase. Therefore, there is a need for the development of a bidirectional chopper circuit that can reduce the size, converter losses, and costs while reducing the magnitude of the ripple current contained in the inductor current.

[0008] According to one aspect of the present disclosure, a bidirectional chopper circuit that bidirectionally converts voltage between a first DC voltage between a pair of first external connection terminals and a second DC voltage between a pair of second external connection terminals comprises: a main power converter having a first switch section and a second switch section connected in series with each other such that their conduction directions are aligned when on, and when one is on the other the other is off, with the terminals on both sides opposite to the connection side of the first and second switch sections forming a pair of first external connection terminals; a single-phase full-bridge power converter provided on wiring branched from the wiring connecting the first and second switch sections; and an inductor connected in series to the single-phase full-bridge power converter on wiring branched from the wiring connecting the first and second switch sections, wherein the AC input / output side of the single-phase full-bridge power converter is connected to an inductor or a pair of second external connection terminals, and the DC input / output side is connected to a DC capacitor. A pair of second external connection terminals are provided at one of the positions on the wiring that has an inductor and a single-phase full-bridge power converter branched off from the wiring connecting the first switch section and the second switch section, and a phase difference is provided between the first carrier signal used to control the main power converter and the second carrier signal used to control the single-phase full-bridge power converter, corresponding to the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal.

[0009] In this case, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 1, a phase difference of approximately ±90 degrees may be provided between the first carrier signal and the second carrier signal.

[0010] Also, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 1, the value of the DC capacitor voltage of the DC capacitor may be controlled to follow approximately half the magnitude of the value of the first DC voltage.

[0011] Also, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is an integer m of 2 or more, a phase difference in the vicinity of ±45 / m degrees may be provided between the first carrier signal and the second carrier signal.

[0012] Also, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is an integer m of 2 or more and the conduction rate d M of the first switch section is 0 or more and less than 0.5, the conduction rate d M of the first switch section may be controlled so that the DC capacitor voltage of the DC capacitor becomes smaller as d becomes larger.

[0013] Also, when the magnitude of the first DC voltage is V dc1 , the value of the DC capacitor voltage may be controlled to follow a value determined by (1 - d M ) × V dc1 in the vicinity.

[0014] Also, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is an integer m of 2 or more and the conduction rate d M of the first switch section is 0.5 or more and 1 or less, the conduction rate d M of the first switch section may be controlled so that the DC capacitor voltage of the DC capacitor becomes larger as d becomes larger.

[0015] Also, when the magnitude of the first DC voltage is V dc1 , the value of the DC capacitor voltage may be controlled to follow a value determined by d M × V dc1 in the vicinity.

[0016] According to one aspect of the present disclosure, it is possible to realize a bidirectional chopper circuit that can reduce the size, converter loss, and cost while reducing the magnitude of the ripple current included in the inductor current.

[0017] This is a circuit diagram showing a bidirectional chopper circuit according to an embodiment of the present disclosure. This is a circuit diagram (1) showing an example of the arrangement of the auxiliary power converter, inductor, and second external connection terminal. This is a circuit diagram (2) showing an example of the arrangement of the auxiliary power converter, inductor, and second external connection terminal. This is a principle block diagram showing the control of a bidirectional chopper circuit according to an embodiment of the present disclosure. This is a diagram illustrating the relationship between the first carrier signal and the second carrier signal when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 1. This is a block diagram showing the control unit for the main power converter in a bidirectional chopper circuit according to an embodiment of the present disclosure. The frequency f of the first carrier signal SM The frequency f of the second carrier signal relative to SA If the proportion is 1 (f SA / f SM =1) Block diagram of the control unit for the auxiliary power converter. When the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 2 (f SA / f SM This figure illustrates the relationship between the first carrier signal and the second carrier signal in (2). The frequency f of the first carrier signal is... SM The frequency f of the second carrier signal relative to SA If the proportion is an integer m greater than or equal to 2 (f SA / f SM A block diagram of the control unit for the auxiliary power converter in case (f). When the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 2 (f SA / f SM =2) Current conductivity d M This figure illustrates the waveforms when the value is set to 0.7. This figure illustrates the ripple current reduction effect according to the embodiment of the present disclosure. This figure shows the circuit parameters used in the simulation of the bidirectional chopper circuit according to the embodiment of the present disclosure. In the bidirectional chopper circuit according to the embodiment of the present disclosure, the frequency f of the first carrier signal SM The frequency f of the second carrier signal relative to SA If the proportion is 1 (f SA / f SM= 1) The first DC voltage (high-voltage DC voltage) V dc1 From the second DC voltage (low-voltage DC voltage) V dc2 This figure shows the simulated waveform when 110 kW of active power is transmitted. In the bidirectional chopper circuit according to the embodiment of this disclosure, the frequency f of the first carrier signal SM The frequency f of the second carrier signal relative to SA If the ratio is 2 (f SA / f SM =2) The first DC voltage (high-voltage DC voltage) V dc1 From the second DC voltage (low-voltage DC voltage) V dc2 This figure shows the simulated waveform when 110 kW of active power is transmitted.

[0018] The following describes an embodiment of the bidirectional chopper circuit with reference to the drawings. In the following description, components having the same or similar function are denoted by the same reference numerals. Duplication of these components may be omitted. The drawings have been scaled appropriately for ease of understanding.

[0019] Furthermore, in the following description, terms are defined in consideration of the functionality in the embodiments of this disclosure, and may vary depending on the intent or conventions of the user or operator. For example, in the following description, "connected" means "electrically connected." "On" of the switch means that the switch closes and forms an electrical circuit through the switching element. "Off" of the switch means that the switch opens and interrupts the electrical circuit through the switch. Also, the numerical examples given below are just examples, and other numerical values ​​may be used.

[0020] <Configuration of a bidirectional chopper circuit according to an embodiment of the present disclosure> Figure 1 is a circuit diagram showing a bidirectional chopper circuit according to an embodiment of the present disclosure.

[0021] The bidirectional chopper circuit 1 according to the embodiment of this disclosure has a first DC voltage V between a pair of first external connection terminals T1 and G1 dc1and the second DC voltage V between the pair of second external connection terminals T2 and G2 dc2 Voltage conversion is performed bidirectionally between the two. Of the first external connection terminals T1 and G1 and the second external connection terminals T2 and G2, a DC power supply is connected to one and a load or another DC power supply is connected to the other.

[0022] For example, when a DC power supply is connected to the first external connection terminals T1 and G1, and a load is connected to the second external connection terminals T2 and G2, the bidirectional chopper circuit 1 operates as a step-down chopper. In this case, the voltage output by the DC power supply is the first DC voltage V dc1 Therefore, the voltage applied to the load is the second DC voltage V dc2 That is the case.

[0023] Furthermore, for example, if a load is connected to the first external connection terminals T1 and G1, and a DC power supply is connected to the second external connection terminals T2 and G2, the bidirectional chopper circuit 1 operates as a boost chopper. In this case, the voltage applied to the load is the first DC voltage V dc1 Therefore, the voltage output by the DC power supply is the second DC voltage V dc2 That is the case.

[0024] Alternatively, for example, a DC power supply may be connected to the first external connection terminals T1 and G1, and another DC power supply may be connected to the second external connection terminals T2 and G2.

[0025] The bidirectional chopper circuit 1 comprises a main power converter 11, an auxiliary power converter 12, an inductor 13, and a control device 14.

[0026] The main power converter 11 comprises a first switch section S1, a second switch section S2, a first feedback diode D1, and a second feedback diode D2. The first feedback diode D1 is connected in antiparallel to the first switch section S1. The second feedback diode D2 is connected in antiparallel to the second switch section S2. The first switch section S1 and the second switch section S2 are connected in series with each other so that their conduction directions are aligned. The first switch section S1 and the second switch section S2 are controlled to turn on one of the switch sections and turn off the other. Here, the connection point between the first switch section S1 and the second switch section is denoted by P. Also, the voltage appearing across the second switch section S2 (i.e., the potential difference between the ground terminal G1 and the connection point P) is denoted by v. M This is expressed as follows. For example, when a DC power supply is connected to the first external connection terminals T1 and G1, and a load is connected to the second external connection terminals T2 and G2, the voltage output by the main power converter 11 is v M This means that the terminals on both sides opposite to the connection side between the first switch section S1 and the second switch section S2 become the first external connection terminals T1 and G1.

[0027] The auxiliary power converter 12, which consists of a single-phase full-bridge power converter, is installed on wiring branched from a connection point P on the wiring connecting the first switch section S1 and the second switch section S2. The auxiliary power converter 12 includes a third switch section S3, a fourth switch section S4, a fifth switch section S5, a sixth switch section S6, a third feedback diode D3, a fourth feedback diode D4, a fifth feedback diode D5, and a sixth feedback diode D6. The third feedback diode D3 is connected in antiparallel to the third switch section S3. The fourth feedback diode D4 is connected in antiparallel to the fourth switch section S4. The fifth feedback diode D5 is connected in antiparallel to the fifth switch section S5. The sixth feedback diode D6 is connected in antiparallel to the sixth switch section S6. The third switch section S3 and the fourth switch section S4 are connected in series with each other so that their conductive directions are aligned. The fifth switch section S5 and the sixth switch section S6 are connected in series with each other so that their conductive directions are aligned. The third switch section S3 and the fourth switch section S4 connected in series, the fifth switch section S5 and the sixth switch section S6 connected in series, and the DC capacitor 21 are connected in parallel. Here, the DC capacitor voltage of the DC capacitor 21 of the auxiliary power converter 12 is v C The voltage on the AC input / output side of the auxiliary power converter 12 is set to v A Let's assume that.

[0028] Examples of the switch sections S1, S2, S3, S4, S5, and S6, which consist of semiconductor switching elements, include FETs, IGBTs, thyristors, GTOs, and transistors. However, the type of semiconductor switching element itself is not limited to this embodiment, and other semiconductor switching elements may also be used.

[0029] The control device 14 controls the switching operations of the first switch section S1 and the second switch section S2 in the main power converter 11, as well as the switching operations of the third switch section S3, the fourth switch section S4, the fifth switch section S5, and the sixth switch section S6 in the auxiliary power converter 12. The power supply for driving the control device 14 is not shown in the diagram. Details of the control device 14 will be described later.

[0030] The inductor 13 is connected in series with the auxiliary power converter 12 on a wiring branched from connection point P on the wiring connecting the first switch section S1 and the second switch section S2 in the main power converter 11. The voltage applied across the inductor 13 is v L Let i be the inductor current flowing through inductor 13. L In addition, if a DC power supply is connected to the first external connection terminals T1 and G1, and an RL load (inductive load) is connected to the second external connection terminals T2 and G2, the inductance component of the RL load may be used as a substitute for the inductor 13, in which case the inductor 13 can be omitted.

[0031] Furthermore, second external connection terminals T2 and G2 are provided at any position on the wiring that branches off from the connection point P on the wiring connecting the first switch section S1 and the second switch section S2 within the main power converter 11, and on the wiring on which the inductor 13 and the auxiliary power converter 12 are provided.

[0032] Therefore, the auxiliary power converter 12, the inductor 13, and the second external connection terminals T2 and G2 are provided on the same wiring branched from the connection point P on the wiring connecting the first switch section S1 and the second switch section S2 within the main power converter 11. In the example shown in Figure 1, the auxiliary power converter 12 is placed between the main power converter 11 and the inductor 13, and the second external connection terminals T2 and G2 are placed on the side of the inductor 13 opposite to the side to which the auxiliary power converter 12 is connected. However, the arrangement order of the auxiliary power converter 12, the inductor 13, and the second external connection terminals T2 and G2 can be designed arbitrarily. Figures 2A and 2B are circuit diagrams showing examples of the arrangement of the auxiliary power converter, the inductor, and the second external connection terminals. Note that in Figures 2A and 2B, for ease of understanding, the second external connection terminals T2 and G2 are shown as V dc2 As stated above, the low-voltage DC voltage V dc2 This is the voltage of the DC power supply or load.

[0033] In the example shown in Figure 2A, the inductor 13 is positioned on the left side of the auxiliary power converter 12 in the figure, and the low-voltage DC voltage V dc2(i.e., the second external connection terminals T2 and G2) are located on the right side of the auxiliary power converter 12 in the diagram. Also, in the example shown in Figure 2B, the inductor 13 is located on the right side of the auxiliary power converter 12 in the diagram, and the low-voltage DC voltage V dc2 The auxiliary power converter 12 (i.e., the second external connection terminals T2 and G2) is located on the left side in the diagram.

[0034] <Overall configuration of the control device for a bidirectional chopper circuit according to an embodiment of the present disclosure> Figure 3 is a principle block diagram showing the control of a bidirectional chopper circuit according to an embodiment of the present disclosure.

[0035] The bidirectional chopper circuit 1 includes a control device 14 as its control system, which has a main power converter control unit 30 and an auxiliary power converter control unit 40. The control of the bidirectional chopper circuit 1 is controlled by the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12. C Control and the inductor current i flowing through inductor 13 L The control unit 30 for the main power converter controls the DC capacitor voltage of the DC capacitor 21, and the control unit 40 for the auxiliary power converter controls the inductor current i flowing through the inductor 13. L This executes the control of [the system]. More details are as follows:

[0036] The control unit 30 for the main power converter controls the DC capacitor voltage v of the DC capacitor 21 of the auxiliary power converter 12. C The predetermined DC capacitor voltage command value V C * While following this, the voltage appearing across the second switch section S2 in the main power converter 11 (i.e., the potential difference between the ground terminal G1 and the connection point P) v M The energization rates of the first switch section S1 and the second switch section S2 in the main power converter 11 are controlled to output an AC voltage that cancels out the AC voltage component. Based on the energization rates set by the main power converter control unit 30, a switching signal for the main power converter is generated to control the switching operation of each semiconductor switching element in the first switch section S1 and the second switch section S2.

[0037] The auxiliary power converter control unit 40 controls the inductor current i flowing through the inductor 13.L The predetermined inductor current command value i L * A switching signal for the auxiliary power converter is generated to control the power conversion operation of the auxiliary power converter 12 in accordance with this.

[0038] The main power converter control unit 30 generates the switching signal for the main power converter, with a frequency f SM A first carrier signal tri having M The following is used. That is, the control unit 30 for the main power converter receives the voltage command and the first carrier signal tri M The system compares this with the frequency f and generates a switching signal for the main power converter to control the switching operation of the first switch section S1 and the second switch section S2 within the main power converter 11. The auxiliary power converter control unit 40 generates the switching signal for the auxiliary power converter using a frequency f. SA The second carrier signal tri has A The following is used: That is, the auxiliary power converter control unit 40 receives the voltage command and the second carrier signal tri A The first carrier signal tri is compared with the first carrier signal tri to generate a switching signal for the auxiliary power converter to control the switching operation of the third switch section S3, the fourth switch section S4, the fifth switch section S5, and the sixth switch section S6 within the auxiliary power converter 12. M and the second carrier signal tri A For details, see the first carrier signal tri M The second carrier signal tri for the frequency A The following explanation will be divided into two cases: when the frequency ratio is 1 and when it is an integer m greater than or equal to 2.

[0039] <Control by the control device when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 1> First carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If equal to (f SM = f SA ) That is, the first carrier signal tri M frequency f SMThe second carrier signal tri for A frequency f SA is 1 (f SA / f SM = 1), the control by the control device 14 will be described.

[0040] Fig. 4 is a diagram illustrating the relationship between the first carrier signal and the second carrier signal when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 1.

[0041] In the embodiment of the present disclosure, the first carrier signal tri used for controlling the main power converter 11 M frequency f SM to the second carrier signal tri used for controlling the auxiliary power converter 12 A frequency f SA is 1 (f SA / f SM = 1), the main power converter control unit 30 and the auxiliary power converter control unit 40 in the control device 14 advance or delay the phase of the first carrier signal tri M by 90 degrees (π / 2 [rad]) from the phase of the second carrier signal tri A . That is, a phase difference Δθ which is a value in the vicinity of ±90 degrees (±π / 2 [rad]) as shown in Equation 1 is provided between the first carrier signal tri used for controlling the main power converter 11 M and the second carrier signal tri used for controlling the auxiliary power converter 12 A . This phase difference Δθ is, for example, a phase angle based on the frequency of the first carrier signal tri M . When the control device 14 that generates each carrier signal is realized using, for example, an FPGA (Field Programmable Gate Array), it is assumed that a deviation α (that is, an error α) of somewhat from 90 degrees will occur depending on the performance of the FPGA. The error α is assumed to be a value of about 20 degrees, for example, but the numerical values given here are examples and other numerical values may be used.

[0042]

[0043] Also, in the embodiment of the present disclosure, for the frequency f of the first carrier signal tri M with respect to the frequency f of the second carrier signal tri SM A when the ratio is 1 (f SA / f SA / f SM = 1), the main power converter control unit 30 in the control device 14 controls the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12 so as to satisfy Equation 2. V C is the first DC voltage between the pair of first external connection terminals T1 and G1. dc1

[0044]

[0045] Note that an error of about ±10% may occur depending on the performance and accuracy of the voltage sensor and the detection system. Therefore, the auxiliary power converter control unit 40 in the control device 14 controls the DC capacitor voltage v so as to approximately satisfy the relational expression of Equation 2, and follows the size that is approximately half of the first DC voltage V. C dc1

[0046] The voltage appearing across both ends of the second switch unit S2 (that is, the potential difference between the ground terminal G1 and the connection point P) v M is expressed by Equation 3.

[0047]

[0048] In Equation 3, the voltage v M appearing across both ends of the second switch unit S2 includes a DC component v Mdc expressed by Equation 4 and an AC component v Mac expressed by Equation 5. d M indicates the conduction rate (duty ratio) of the first switch unit S1, and d M = V dc2 / V dc1

[0049]

[0050]

[0051] From the auxiliary power converter 12, v MacBy outputting the same amount of AC component, the inductor current i L This reduces the ripple current contained in the voltage. That is, the AC voltage output by the auxiliary power converter 12 is reduced to v Aac Therefore, v Aac = v Mac It is necessary to satisfy this condition. On the other hand, if the conditions of Equation 2 are set, depending on the operating conditions, v Aac = v Mac This cannot be satisfied. Therefore, depending on the operating conditions, v Aac Set it as shown in equation 6 or equation 7. That is, the current flow rate d of the first switch unit S1 M If d is greater than or equal to 0 and less than 0.5 (0 ≤ d M <0.5) is v Aac Set the value represented by Equation 6, and the current flow rate d of the first switch section S1 M If d is between 0.5 and 1 (0.5 < d M ≤1) is v Aac Set it to the value represented by Equation 7.

[0052]

[0053]

[0054] Figure 5 is a block diagram showing the control unit for the main power converter in a bidirectional chopper circuit according to an embodiment of the present disclosure.

[0055] The control unit 30 for the main power converter controls the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12. C The predetermined DC capacitor voltage command value V C * While following this, the voltage v that appears across the second switch section S2 in the main power converter 11 M The energization rates of the first switch section S1 and the second switch section S2 in the main power converter 11 are controlled so that an AC voltage is output that cancels out the AC voltage component. For this purpose, the control unit 30 for the main power converter has a DC capacitor voltage control unit 301, an adder 302, and a divider 303. The DC capacitor voltage control unit 301 controls the DC capacitor voltage command value V C * and the actual DC capacitor voltage v CThe DC capacitor voltage v is controlled by PI control, which takes the deviation between the two as input. C Controls the DC capacitor voltage v C and the first DC voltage V dc1 In order to satisfy the relationship in Equation 2, the DC capacitor voltage command value V C * = V dc1 Set the inductor current i to / 2. L The output of the PI control v depends on the polarity. B * The polarity is changed. Then, in the adder 302, v B * The second DC voltage V dc2 This is added as a feedforward control term, and this is further divided by the divider 303 to obtain the first DC voltage V dc1 By dividing by this, the current conduction rate d of the first switch section S1 is obtained. M Generates.

[0056] Figure 6 shows the frequency f of the first carrier signal. SM The frequency f of the second carrier signal relative to SA If the proportion is 1 (f SA / f SM =1) This is a block diagram showing the control unit for the auxiliary power converter.

[0057] The auxiliary power converter control unit 40 controls the inductor current i flowing through the inductor 13. L The predetermined inductor current command value i L * The energization rate d of each leg of the auxiliary power converter 12 controls the power conversion operation of the auxiliary power converter 12 to follow the power conversion operation of the auxiliary power converter 12. A1 and d A2 The auxiliary power converter control unit 40 includes an inductor current control unit 401, a ripple suppression control unit 402, an adder / subtractor 403, and a divider 404. The inductor current control unit 401 generates the inductor current command value i L * And the actual inductor current i flowing through inductor 13 L The DC component i Lave The inductor current i flowing through the inductor 13 is controlled by PI control, which takes the deviation between the two as input.L Voltage command value v for controlling i * Outputs the inductor current i flowing through inductor 13. L The DC component i Lave The inductor current i flowing through the inductor 13 is measured using a moving average filter or the like. L It can be calculated by detecting only the DC component. The ripple suppression control unit 402 determines the AC voltage v that the auxiliary power converter 12 should output based on the above equation 6 or equation 7. Aac The adder / subtractor 403 calculates the voltage command value v in order to achieve non-interference with the DC capacitor voltage control by the main power converter control unit 30. i * v, which was explained with reference to Figure 6. B * This is added to the AC voltage v that the auxiliary power converter 12 should output. Aac By adding this, the final voltage command value v A * It outputs the DC capacitor voltage v. Then, this is further divided by the divider 303. c By dividing by this, the energization rate d for the leg consisting of the third switch section S3 and the fourth switch section S4 is obtained. A1 Furthermore, the energization rate d for the leg consisting of the fifth switch section S5 and the sixth switch section S6. A2 Calculate.

[0058] <Control by the control device when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is an integer m of 2 or more> First carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If it is 1 / m times (where m is an integer greater than or equal to 2) (f SM = f SA / m) that is, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is an integer m greater than or equal to 2 (f SA / f SMThe control by the control device 14 in (m) will be explained below.

[0059] Figure 7 shows the case where the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 2 (f SA / f SM This diagram illustrates the relationship between the first carrier signal and the second carrier signal in (2).

[0060] For example, the first carrier signal tri used to control the main power converter 11 M If one period is 360 degrees (2π [rad]), the second carrier signal tri used to control the auxiliary power converter 12 A The period of is 360 / m (2π / m [rad]), and the first carrier signal tri M and the second carrier signal tri A The timing of the increase from 0 is shifted by 45 / m degrees (π / (4m) [rad]). The first carrier signal tri M The phase of the second carrier signal tri A By advancing or lagging the phase by 45 / m degrees (π / (4m) [rad]) compared to the phase, the inductor current i flowing through inductor 13 is controlled. L The ripple current can be reduced. In the embodiments of this disclosure, the first carrier signal tri used for controlling the main power converter 11 M frequency f SM The second carrier signal tri used to control the auxiliary power converter 12 for A frequency f SA If the proportion is an integer m greater than or equal to 2 (f SA / f SM =m), the control unit 30 for the main power converter and the control unit 40 for the auxiliary power converter in the control device 14, the first carrier signal tri M The phase of the second carrier signal tri A The phase is advanced or delayed by 45 / m degrees (π / (4m) [rad]) compared to the phase of the first carrier signal tri M and the second carrier signal tri ABetween them, a phase difference Δθ is provided, which is a value in the vicinity of ±45 / m degrees (±π / (4m) [rad]) as shown in Equation 8. This phase difference Δθ is, for example, the first carrier signal tri M This is the phase angle relative to the frequency. For example, when implementing a control device 14 that generates each carrier signal using an FPGA, it is expected that a slight deviation α (i.e., error α) from 45 / m degrees will occur depending on the performance of the FPGA. An error α of approximately 20 degrees is expected, but the value given here is just an example, and other values ​​may also be used.

[0061]

[0062] Furthermore, in the embodiments of this disclosure, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is an integer m greater than or equal to 2 (f SA / f SM = m), and the current conduction rate d of the first switch section S1. M If d is greater than or equal to 0 and less than 0.5 (0 ≤ d M <0.5) The control unit 30 for the main power converter in the control device 14 satisfies equation 9 by controlling the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12. C Control.

[0063]

[0064] In other words, the main power converter control unit 30 satisfies equation 9, and the current rate d of the first switch unit S1 M The larger the value, the DC capacitor voltage v C It is controlled to reduce the value. However, an error of about ±10% may occur depending on the performance and accuracy of the voltage sensor and detection system, so the auxiliary power converter control unit 40 in the control device 14 controls the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12 so that it approximately satisfies the relational equation 9. C However, (1-d M ) × V dc1 It controls the system to follow a value close to the value defined by [the specified parameter].

[0065] Furthermore, in the embodiments of this disclosure, the first carrier signal tri used for controlling the main power converter 11 M frequency f SM The second carrier signal tri used to control the auxiliary power converter 12 for A frequency f SA If the proportion is an integer m greater than or equal to 2 (f SA / f SM = m), and the current conduction rate d of the first switch section S1. M If d is between 0.5 and 1 (0.5 < d M ≤1) The control unit 30 for the main power converter in the control device 14 satisfies equation 10 by setting the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12. C Control.

[0066]

[0067] In other words, the main power converter control unit 30 satisfies equation 10, and the current rate d of the first switch unit S1 M The larger the value, the DC capacitor voltage v C It is controlled to increase the value. However, an error of about ±10% may occur depending on the performance and accuracy of the voltage sensor and detection system, so the auxiliary power converter control unit 40 in the control device 14 controls the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12 so that it approximately satisfies the relation in equation 10. C However, d M ×V dc1 It controls the system to follow a value close to the value defined by [the specified parameter].

[0068] As described above, the voltage v that appears across the second switch section S2 M This is expressed by equation 3. Also, the voltage v appearing across the second switch section S2 is expressed by... M This is the DC component v expressed in Equation 4. Mdc The AC component v expressed by equation 5 Mac This includes the auxiliary power converter 12 to v Mac By outputting the same amount of AC component, the inductor current i L This reduces the ripple current contained in the voltage. That is, the AC voltage output by the auxiliary power converter 12 is reduced to v AacTherefore, v Aac = v Mac The following must be satisfied. When the conditions of equations 9 and 10 are set, regardless of the operating conditions, v Aac = v Mac It can satisfy this condition. Therefore, v Aac Set it to the value represented by equation 11.

[0069]

[0070] As shown in Figure 5, the control unit 30 for the main power converter controls the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12. C While maintaining the voltage at a predetermined level, the voltage v that appears across the second switch section S2 in the main power converter 11 M The energizing ratios of the first switch section S1 and the second switch section S2 in the main power converter 11 are controlled so that an AC voltage is output that cancels out the AC voltage component. At that time, a moving average filter or the like is used to control the DC capacitor voltage v C Only the DC component contained in may be detected. However, the current rate d of the first switch unit S1 M If d is greater than or equal to 0 and less than 0.5 (0 ≤ d M <0.5) is the DC capacitor voltage v C and the first DC voltage V dc1 The DC capacitor voltage command value V is such that the relationship in equation 9 is approximately satisfied. C * Set the current supply rate d of the first switch section S1. M If d is greater than or equal to 0 and less than 0.5 (0 ≤ d M <0.5) is the DC capacitor voltage command value V that can approximately satisfy the relation in Equation 10. C * Set it.

[0071] Figure 8 shows the frequency f of the first carrier signal. SM The frequency f of the second carrier signal relative to SA If the proportion is an integer m greater than or equal to 2 (f SA / f SM A block diagram showing the control unit for the auxiliary power converter in m).

[0072] The auxiliary power converter control unit 40 controls the inductor current i flowing through the inductor 13. L DC component i included Lave The predetermined inductor current command value i L * The energization rate d of each leg of the auxiliary power converter 12 controls the power conversion operation of the auxiliary power converter 12 to follow the power conversion operation of the auxiliary power converter 12. A1 and d A2 The auxiliary power converter control unit 40 includes an inductor current control unit 401, a ripple suppression control unit 402, an adder / subtractor 403, and a divider 404.

[0073] First carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is an integer m greater than or equal to 2 (f SA / f SM The ripple suppression control unit 402 in (m) determines the AC voltage v that the auxiliary power converter 12 should output based on the above equation 11. Aac The following is calculated. The inductor current control unit 401, adder / subtractor 403, and divider 404 are as described with reference to Figure 6.

[0074] <Analysis of Ripple Current> Inductor current i flowing through inductor 13 in the bidirectional chopper circuit 1 according to the embodiment of this disclosure L We will analyze the ripple current contained within it.

[0075] For example, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM =2) The inductor current i flowing through inductor 13 L Ripple current I included Lripple2 This can be theoretically calculated as follows:

[0076] Figure 9 shows the case where the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 2 (f SA / fSM =2) Current conductivity d M This figure illustrates the waveforms when the value is set to 0.7. In Figure 9, the first carrier signal tri used to control the main power converter 11 M , the second carrier signal tri used to control the auxiliary power converter 12 A , the current flow rate d of the first switch section S1 M (=0.7), the voltage v appearing across the second switch section S2. M The current flow rate d of the third switch section S3 and the fourth switch section S4. A1 , the current flow rate d for the fifth switch section S5 and the sixth switch section S6 A2 , the voltage v on the AC input / output side of the auxiliary power converter 12 A The voltage v applied across the inductor 13 L , and the inductor current i flowing through the inductor 13. L This indicates that.

[0077] The voltage appearing across the fourth switch section S4 is v A1 The voltage appearing across the sixth switch section S6 is v A2 In this case, the voltage v on the AC input / output side of the auxiliary power converter 12 A This can be expressed as shown in Equation 12.

[0078]

[0079] DC capacitor voltage v of DC capacitor 21 C This can be expressed as shown in Equation 13.

[0080]

[0081] Current conduction rate d of the first switch section S1 M This can be expressed as shown in Equation 14.

[0082]

[0083] In Figure 9, the voltage v that appears across the second switch section S2. M This can be expressed as shown in equation 15.

[0084]

[0085] In equation 15, θ1 is given by equation 16 and θ7 is given by equation 17.

[0086]

[0087]

[0088] Current conduction rate d of the first switch section S1 M If the value is 0.5 or greater, the AC voltage command value v of the auxiliary power converter 12 A * This is expressed in Equation 18. However, in Equation 18, the voltage term used for voltage control and current control is set to 0 (zero).

[0089]

[0090] At this time, the current flow rate d of the third switch section S3 A1 (Maximum value: 1, Minimum value: -1) is given by equation 19. Here, V C The DC capacitor voltage v C This represents the DC component contained within it.

[0091]

[0092] Furthermore, the current flow rate d of the fifth switch section S5 A2 (Maximum value: 1, Minimum value: -1) is given by equation 20.

[0093]

[0094] From equations 19 and 20, it should be noted that general unipolar modulation is applied to the auxiliary power converter 12. In equations 19 and 20, the energization rate d of the third switch section S3. A1 and the current flow rate d of the fifth switch section S5 A2 Normalizing each of these values ​​from 0 to 1 yields equations 21 and 22.

[0095]

[0096]

[0097] Current conduction rate d of the first switch section S1 MIf is 0.5 or greater, the relationship θ1 ≤ θ2 ≤ θ3 ≤ θ4 always holds. From Figure 9 and Equation 21, the voltage v appearing across the fourth switch section S4 A1 This can be expressed as shown in equation 23.

[0098]

[0099] In equation 23, θ2 is given by equation 24, θ3 is given by equation 25, and θ6 is given by equation 26.

[0100]

[0101]

[0102]

[0103] Similarly, the voltage v that appears across the sixth switch section S6 A2 This can be expressed as shown in equation 27.

[0104]

[0105] In equation 27, θ4 is given by equation 28 and θ5 is given by equation 29.

[0106]

[0107]

[0108] Finally, the voltage v on the AC input / output side of the auxiliary power converter 12 A This can be expressed as shown in equation 30.

[0109]

[0110] According to Kirchhoff's voltage law, the voltage v applied across the inductor 13 L This is given by equation 31.

[0111]

[0112] Therefore, the first carrier signal tri used to control the main power converter 11 M During one period, the inductor current i flows through the inductor 13. L The value increases and decreases multiple times. In Figure 9, the inductor current i flowing through inductor 13L Focusing on the phase interval in which the current increases, from the phase relationship equation above, we get θ3 - θ2 = θ5 - θ4, and θ7 - θ6 > θ3 - θ2. From equation 31, the current increase I at θ3 - θ2 is up1 Equation 32 is obtained by solving for this.

[0113]

[0114] On the other hand, the inductor current i flowing through the inductor 13 L Focusing on the phase interval in which the current decreases, from the phase relationship equation above, we get θ6 - θ5 = θ4 - θ3, and θ6 - θ5 > θ2 - θ1. From equation 31, the current decrease I at θ6 - θ5 is down1 Equation 33 is obtained by finding this value.

[0115]

[0116] From equations 32 and 33, I up1 The absolute value and I down1 The absolute value of is equal to I. up1 The maximum value of the inductor current i flowing through the inductor 13 is L Ripple current I included Lripple2 This is defined as follows. From equation 32, the maximum value of the ripple current I Lripple2 d M When is 1 / √2, it can be expressed as in equation 34.

[0117]

[0118] Note that in Figure 9, the current flow rate d M We focused on the case where the current rate d is 0.7, but M In the range of 0.5 to 1, equations 24 and 25 show that there exist θ2 and θ3 such that θ3 > θ2. In other words, equations 32 and 34 are given by the current-carrying ratio d M This is an equation that holds true in the range of 0.5 to 1. Current conductivity d M Similarly, the ripple current can be analyzed when it is between 0 and 0.5, and the maximum value of the ripple current I Lripple2 This is similarly represented by equation 34.

[0119] Above, the first carrier signal tri M frequency f SMThe second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM =2) The inductor current i flowing through inductor 13 L Ripple current I included Lripple2 The first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA Even if the ratio is a different value, the ripple current can be analyzed in the same way.

[0120] For example, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 3 (f SA / f SM =3) The inductor current i flowing through inductor 13 L The new maximum value of ripple current I Lripple3 This can be expressed by equation 35.

[0121]

[0122] Also, for example, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 1 (f SA / f SM =1) The inductor current i flowing through inductor 13 L The new maximum value of ripple current I Lripple1 This can be expressed by equation 36.

[0123]

[0124] Figure 10 illustrates the ripple current reduction effect according to the embodiment of the present disclosure. In Figure 10, the horizontal axis represents the current conduction rate d of the first switch section. M The vertical axis represents the inductor current i flowing through inductor 13. LThis indicates the maximum value of the ripple current contained within.

[0125] Waveform A is the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 1 (f SA / f SM =1) In the embodiments of this disclosure, the first carrier signal tri M and the second carrier signal tri A When control is performed to create a phase difference Δθ of ±90 degrees between the two, the inductor current i L This shows the maximum value of the ripple current contained in waveform A. For waveform A, the maximum value of the ripple current I Lripple1 This is as shown in Equation 36.

[0126] Waveform B is the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM =2) The inductor current i when DC capacitor control is performed in the embodiment of this disclosure. L This shows the maximum value of the ripple current contained in waveform B. For waveform B, the maximum value of the ripple current I Lripple2 This is as shown in Equation 34.

[0127] Waveform C is the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 3 (f SA / f SM =3) The inductor current i when DC capacitor control is performed in the embodiment of this disclosure. L This shows the maximum value of the ripple current contained in waveform C. For waveform C, the maximum value of the ripple current I is... Lripple3 This is as shown in Equation 35.

[0128] Waveform D represents the inductor current i of a general bidirectional chopper circuit described in Non-Patent Document 2. LThis shows the maximum value of the ripple current contained in waveform C. For waveform C, the maximum value of the ripple current I is... Lripple4 This is represented by equation 37.

[0129]

[0130] Waveform E is the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 1 (f SA / f SM =1) The first carrier signal tri M and the second carrier signal tri A When the control described in Non-Patent Document 3 is performed without providing a phase difference Δθ of ±90 degrees between the two, the inductor current i L This shows the maximum value of the ripple current contained in waveform E. For waveform E, the maximum value of the ripple current I is shown. Lripple5 This is expressed in equation 38.

[0131]

[0132] First carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 1 (f SA / f SM =1) When comparing waveform A and waveform E, the first carrier signal tri M and the second carrier signal tri A By performing control according to the embodiment of this disclosure, which provides a phase difference Δθ of ±90 degrees between the two, it can be seen that the maximum value of the ripple current can be reduced to 1 / 4 compared to a general bidirectional chopper circuit that does not provide a phase difference Δθ. The reason why the maximum value of the ripple current can be reduced to 1 / 4 is that the voltage step of the inductor voltage becomes 1 / 2 and the frequency becomes twice as large compared to a general bidirectional chopper circuit that does not provide a phase difference Δθ.

[0133] Comparing waveform B and waveform E, it can be seen that the DC capacitor control according to the embodiment of this disclosure can reduce the maximum ripple current to 0.17 times that of a typical bidirectional chopper circuit.

[0134] Comparing waveform B and waveform C, in the embodiment of this disclosure, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 3 (f SA / f SM =3) The first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM It can be seen that the maximum value of the ripple current can be reduced to 2 / 3 of that compared to (2).

[0135] Comparing waveform A, waveform B, and waveform C, in the embodiment of this disclosure, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 3 (f SA / f SM =3) When DC capacitor control is performed, and the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM =2) When DC capacitor control is performed, the first carrier signal tri M and the second carrier signal tri A Compared to the case where a phase difference Δθ of ±90 degrees is set between the two, it can be seen that the maximum value of the ripple current can be reduced to 68.6% and 45.8%, respectively.

[0136] <Simulation Results of a Bidirectional Chopper Circuit According to an Embodiment of the Disclosure> Figure 11 shows the circuit parameters used in the simulation of the bidirectional chopper circuit according to an embodiment of the disclosure. "PSCAD / EMTDC" was used for the simulation. The simulation was performed for the purpose of verifying the principle, and therefore assumed an ideal state. That is, an analog control system with zero control delay was assumed, and an ideal switch section with zero dead time was used.

[0137] The rated power of the bidirectional chopper circuit 1 is P = 110 kW, and the first DC voltage is V dc1 = 650V, the second DC voltage is V dc2 = 430V. In this case, if we set the voltage control term to 0 (zero) as shown in Figure 5, then d M =V dc2 / V dc1 = 0.66. Also, the inductor current flowing through inductor 13 is I L = 250A. The first carrier signal tri used to control the main power converter 11 M Regarding the frequency, f SM = 10 kHz. The second carrier signal tri used to control the auxiliary power converter 12 A Regarding the frequency, the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 1 (f SA / f SM =1) is f SA Let = 10 kHz, and the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM =1) is f SA = 20 kHz. The first carrier signal tri M and the second carrier signal tri A The phase relationships with the first carrier signal tri M frequency f SMThe second carrier signal tri A frequency f SA If the proportion is 1 (f SA / f SM DC capacitor voltage V in =1) C V is obtained according to equation 2. C = 0.5V dc1 =325V. The first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM DC capacitor voltage V in (2) C V is obtained according to equation 10. C = d M V dc1 The voltage was set to 430V. The inductance of inductor 13 was set to L = 0.1mH, and the capacitance of DC capacitor 21 was set to C = 0.3mF.

[0138] Figure 12 shows a bidirectional chopper circuit according to an embodiment of the present disclosure, where the frequency f of the first carrier signal SM The frequency f of the second carrier signal relative to SA If the proportion is 1 (f SA / f SM = 1) The first DC voltage (high-voltage DC voltage) V dc1 From the second DC voltage (low-voltage DC voltage) V dc2 This figure shows the simulated waveform when 110 kW of active power is transmitted. In Figure 12, the first carrier signal tri used to control the main power converter 11 is shown. M , the second carrier signal tri used to control the auxiliary power converter 12 A , the current flow rate d of the first switch section S1 M (=0.66), the voltage v appearing across the second switch section S2. M , the voltage v on the AC input / output side of the auxiliary power converter 12 A The voltage v applied across the inductor 13 L , the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12 C , and the inductor current i flowing through the inductor 13.L This is shown. Also, in Figure 12, the first carrier signal tri used for controlling the main power converter 11 is shown. M The waveforms for two periods are shown. In the simulation, the first carrier signal tri M The second carrier signal tri A The phase is delayed by 90 degrees.

[0139] The voltage v that appears across the second switch section S2 M Focusing on this, the current conduction rate d of the first switch section S1 M The first carrier signal tri M If the above is true (d M ≧tri M When the first switch unit S1 is turned on, the voltage that appears across the second switch unit S2 is v M = V dc1 This is the result. On the other hand, the current flow rate d of the first switch section S1 M The first carrier signal tri M If less than (d M <tri M When the first switch section S1 is turned off, the voltage that appears across the second switch section S2 is v M = 0. At this time, the auxiliary power converter 12 receives the voltage v that appears across the second switch section S2. M It outputs an AC voltage that cancels out the AC voltage component of the signal. Specifically, the auxiliary power converter 12 outputs the voltage given by equation 7. The voltage v applied across the inductor 13 L Focusing on this, the equivalent frequency is 2f SM However, this is the first carrier signal tri M and the second carrier signal tri A This is due to a 90-degree phase difference between the first carrier signal tri M and the second carrier signal tri A By introducing a 90-degree phase difference Δθ between the two points, the ripple current can be reduced compared to a typical bidirectional chopper circuit that does not have a 90-degree phase difference Δθ.

[0140] DC capacitor voltage v of DC capacitor 21 in auxiliary power converter 12 CFocusing on this, the DC capacitor voltage v C It can be seen that the signal contains both DC and AC components. Furthermore, it can be seen that the DC component follows the DC capacitor voltage command value of 325V well. In addition, although there is a 10kHz AC component, its magnitude is sufficiently small compared to the DC component.

[0141] Inductor current i flowing through inductor 13 L Focusing on this, the inductor current i L It can be seen that the current contains both DC and AC components. Furthermore, it can be seen that the DC component follows the current command value of 250A well. In addition, there is an AC component of 10kHz, and the ripple current value is 36A. The above value is small compared to the maximum ripple current of 40A given by equation 36, which confirms the validity of the theoretical result described above.

[0142] Figure 13 shows a bidirectional chopper circuit according to an embodiment of the present disclosure, where the frequency f of the first carrier signal SM The frequency f of the second carrier signal relative to SA If the ratio is 2 (f SA / f SM =2) The first DC voltage (high-voltage DC voltage) V dc1 From the second DC voltage (low-voltage DC voltage) V dc2 This figure shows the simulated waveform when 110 kW of active power is transmitted. In Figure 13, the first carrier signal tri used to control the main power converter 11 is shown. M , the second carrier signal tri used to control the auxiliary power converter 12 A , the current flow rate d of the first switch section S1 M (=0.66), the voltage v appearing across the second switch section S2. M , the voltage v on the AC input / output side of the auxiliary power converter 12 A The voltage v applied across the inductor 13 L , the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12 C , and the inductor current i flowing through the inductor 13. LThis is shown. Also, in Figure 13, the first carrier signal tri used to control the main power converter 11 is shown. M The waveforms for two periods are shown. In the simulation, similar to Figure 7, the first carrier signal tri M and the second carrier signal tri A The timing of the increase from 0 is shifted by 22.5 degrees (= 45 / 2 degrees).

[0143] The voltage v that appears across the second switch section S2 M Focusing on this, the current conduction rate d of the first switch section S1 M The first carrier signal tri M If the above is true (d M ≧tri M When the first switch unit S1 is turned on, the voltage that appears across the second switch unit S2 is v M = V dc1 This is the result. On the other hand, the current flow rate d of the first switch section S1 M The first carrier signal tri M If less than (d M <tri M When the first switch section S1 is turned off, the voltage that appears across the second switch section S2 is v M = 0. At this time, the auxiliary power converter 12 receives the voltage v that appears across the second switch section S2. M It outputs an AC voltage that cancels out the AC voltage component of the signal. Specifically, the auxiliary power converter 12 outputs the voltage given by equation 13. The voltage v applied across the inductor 13 L Focusing on this, the zero-voltage section and 4f SM A section occurs where the frequency of the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the ratio is 2 (f SA / f SM =2) is the first carrier signal tri M frequency f SM The second carrier signal tri A frequency f SA If the proportion is 1 (fSA / f SM As compared with (i.e., when / f = 1), it can be seen that the ripple current included in the inductor current i can be further reduced. L

[0144] Focusing on the DC capacitor voltage v of the DC capacitor 21 in the auxiliary power converter 12, it can be seen that the DC capacitor voltage v includes a DC component and an AC component. And it can be seen that the DC component follows well the DC capacitor voltage command value of 430V. Also, regarding the AC component, although there is an AC component of 10 kHz, its magnitude is sufficiently small compared with the DC component. C C

[0145] Focusing on the inductor current i flowing through the inductor 13, it can be seen that the inductor current i includes a DC component and an AC component. And it can be seen that the DC component follows well the current command value of 250A. Also, regarding the AC component, there is an AC component of 10 kHz, and the value of the ripple current is 27A. This is in good agreement with the maximum ripple current given by Equation 36, which is 27.9A, and the validity of the above-described theoretical results can be confirmed. L L

[0146] <Processor and Memory>At least one processor, which is an arithmetic processing unit, is provided in the bidirectional chopper circuit 1. Examples of the arithmetic processing unit include an IC, LSI, CPU, MPU, DSP, etc. The arithmetic processing unit has a control device 14 and other processing units. Each of these units included in the arithmetic processing unit is, for example, a functional module realized by a program executed on the processor. For example, when constructing the control device 14 and other processing units in the form of a program, the functions of each unit can be realized by operating the arithmetic processing unit according to this program. The program for executing each process in the control device 14 and other processing units may be provided in the form of a program product recorded on a computer-readable recording medium (storage medium) such as a semiconductor memory, a magnetic recording medium (magnetic storage medium), or an optical recording medium (optical storage medium). Alternatively, the control device 14 and other processing units may be realized as a semiconductor integrated circuit in which a program for realizing the functions of each unit is written.

[0147] In addition, at least one memory, which is a storage device, is provided in the bidirectional chopper circuit 1. The memory also includes various storage units within the control device 14 and other processing units. Examples of the memory include an electrically erasable and recordable non-volatile memory such as EEPROM (registered trademark), or a random access memory such as DRAM, SRAM, etc. that can be read and written at high speed. Also, the storage device may have a configuration such as an HDD (hard disk drive) or SSD (solid state drive). A program for operating the control device 14 and other processing units is stored in the memory. In the memory, programs, command values, detection values, and circuit parameters related to the main power converter control unit 30 and the auxiliary power converter control unit 40 within the control device 14 are stored. Various programs and various data related to the main power converter 11 are stored in the memory. Various programs and various data related to the auxiliary power converter 12 are stored in the memory.

[0148] Although the present disclosure has been described in detail above, it is not limited to the individual embodiments and modifications described above. These embodiments and modifications can be added, replaced, modified, partially deleted, etc., in any way that does not depart from the gist of the present disclosure or from the spirit of the present disclosure derived from the claims and their equivalents. Furthermore, these embodiments and modifications can be implemented in combination. For example, the order of operations and processes in the embodiments and modifications described above are shown as examples only and are not limited thereto. The same applies when numerical values ​​or mathematical formulas are used in the description of the embodiments and modifications described above.

[0149] 1 Bidirectional chopper circuit 11 Main power converter 12 Auxiliary power converter 13 Inductor 14 Control device 21 DC capacitor 30 Control unit for main power converter 40 Control unit for auxiliary power converter 301 DC capacitor voltage control unit 301 302 Adder 303 Divider 401 Inductor current control unit 402 Ripple suppression control unit 403 Adder / subtractor 404 Divider D1 First feedback diode D2 Second feedback diode D3 Third feedback diode D4 Fourth feedback diode D5 Fifth feedback diode D5 Fifth feedback diode P Connection point Q1, Q2 Input / output terminals T1, G1 First external connection terminals T2, G2 Second external connection terminals T3, G3 Third external connection terminals T4, G4 Fourth external connection terminals S1 First switch section S2 Second switch section S3 Third switch section S4 Fourth switch section S5 Fifth switch section S6 Sixth switch section

Claims

1. A bidirectional chopper circuit that performs bidirectional voltage conversion between a first DC voltage between a pair of first external connection terminals and a second DC voltage between a pair of second external connection terminals, comprising: a main power converter having a first switch section and a second switch section connected in series with each other so that their conduction directions are aligned when ON, and when one is ON the other is OFF, the terminals on both sides opposite to the connection side of the first switch section and the second switch section are the pair of first external connection terminals; a single-phase full-bridge power converter provided on wiring branched from the wiring connecting the first switch section and the second switch section; and an inductor connected in series to the single-phase full-bridge power converter on wiring branched from the wiring connecting the first switch section and the second switch section, wherein the inductor or the pair of second external connection terminals are connected to the AC input / output side of the single-phase full-bridge power converter, and a DC capacitor is connected to the DC input / output side. A bidirectional chopper circuit comprising: a pair of second external connection terminals provided at any position on the wiring on which the inductor and the single-phase full-bridge power converter are provided, which are branched from the wiring connecting the first switch section and the second switch section; and a phase difference corresponding to the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is provided between the first carrier signal used to control the main power converter and the second carrier signal used to control the single-phase full-bridge power converter.

2. The bidirectional chopper circuit according to claim 1, wherein, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 1, a phase difference of approximately ±90 degrees is provided between the first carrier signal and the second carrier signal.

3. The bidirectional chopper circuit according to claim 1 or 2, wherein, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is 1, the value of the DC capacitor voltage of the DC capacitor is controlled to follow approximately half the magnitude of the value of the first DC voltage.

4. The bidirectional chopper circuit according to claim 1, wherein, when the ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is an integer m of 2 or more, a phase difference of approximately ±45 / m degrees is provided between the first carrier signal and the second carrier signal.

5. The ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is an integer m of 2 or more, and the current flow rate d of the first switch section is... M When the current supply rate d of the first switch section is 0 or greater and less than 0.5, M The bidirectional chopper circuit according to claim 1 or 4, wherein the DC capacitor voltage of the DC capacitor is controlled to decrease as the value of the DC capacitor increases.

6. The magnitude of the first DC voltage is V dc1 In that case, the value of the DC capacitor voltage is (1-d M ) × V dc1 A bidirectional chopper circuit according to claim 5, which is controlled to follow a value in the vicinity of a value determined by [a specific parameter].

7. The ratio of the frequency of the second carrier signal to the frequency of the first carrier signal is an integer m of 2 or more, and the current flow rate d of the first switch section is... M When the current conduction rate d of the first switch section is 0.5 or more and 1 or less, M The bidirectional chopper circuit according to claim 1 or 4, wherein the DC capacitor voltage of the DC capacitor increases as the value of the DC capacitor increases.

8. When the magnitude of the first DC voltage is V dc1 , the value of the DC capacitor voltage is controlled to follow the vicinity of the value determined by d M ×V dc1 The bidirectional chopper circuit according to claim 7

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