Resonance compensation control method for three-phase inverter

Abstract

The invention belongs to the technical field of inverters, and particularly relates to a three-phase inverter resonance compensation control method, which comprises a power module and a resonance compensation module, and is characterized in that the power module comprises a power conversion circuit, a CL output filter and a protection circuit of a three-phase inverter; the resonance compensation module is mainly a proportional resonance controller (PR controller) and is used for generating harmonic feedback compensation. The control architecture of the whole inverter is double closed-loop control, the voltage control is an outer loop, the current control is an inner loop, and the PR controller is feedback control. Wherein a voltage outer loop adopts proportion and integral control, and a current inner loop only adopts proportion control, so that phase delay introduced by PI of the inner loop and PI of the outer loop is avoided. By setting the control parameters of the voltage control loop and the current control loop, multiple harmonics can be effectively suppressed.

Description

Technical Field

[0001] The invention belongs to the technical field of inverters, and in particular relates to a resonance compensation control method for a three-phase inverter. Background Art

[0002] With the shortage of traditional energy and the increasing severity of environmental problems, distributed generation technologies such as photovoltaics, wind power and fuel cells have developed rapidly. Microgrids, as a system that integrates distributed generation, energy storage and local loads, are becoming more and more widely used. In the isolated operation of these microgrids, nonlinear loads cause output current distortion, which leads to output voltage distortion and reduces the quality of voltage output. According to national standards and IEEE standards, the total harmonic distortion (THD) of the grid voltage should be controlled within 5%. When the PCC (point of common connection) voltage harmonic problem of the microgrid is serious, it may interfere with the normal operation of electrical equipment in the microgrid. Therefore, measures must be taken to suppress the PCC voltage harmonics caused by nonlinear loads to ensure the power quality of the microgrid and reduce interference to the grid when it is connected to the grid.

[0003] The traditional control strategy often uses a voltage and current double closed loop plus a PR controller. The current inner loop uses the capacitor current i on the inverter side. C Feedback control often involves multiple capacitors or complex filtering circuits, which requires more sensors to monitor the capacitor currents at different nodes separately. There are multiple capacitors or multiple current paths in the circuit, and the current of each capacitor needs to be monitored separately. The system has a large number of sensors and high costs. When the PR controller is on the main line, it assumes the task of leading the dynamic response of the system and directly affects the control output of the system. In this case, the PR controller must process all frequency components in the system. However, since the PR controller responds slowly to frequency components other than the base frequency, its dynamic response becomes slower. Especially in the face of rapid changes or disturbances, the PR controller on the main line is in the outer voltage loop, and the bandwidth of the outer voltage loop is about 1 / 10 of the inner current loop. The response speed is slow, the speed of adjusting the system output is slow, and the dynamic performance is weak. Summary of the invention

[0004] The present invention aims to provide a three-phase inverter resonance compensation control method to reduce the impact of harmonics on the stability of power grid operation, enhance system robustness and dynamic response characteristics, and reduce costs; adjust the control parameters of the voltage loop and the current loop to effectively suppress harmonics and improve system stability.

[0005] In order to achieve the above object, the present invention provides a three-phase inverter resonance compensation control method, comprising the following steps:

[0006] Step 1: Detect the inductor current i on the inverter side Labc , detect output voltage voabc , detect the LC filter output current i oabc ;

[0007] Step 2: The frequency-locked loop (FLL) controls the system frequency at the reference frequency and converts the inverter frequency into an angle using the formula ph = 2π × ω.

[0008] Step 3: Set the current loop control parameter K, where the transfer function of the current loop control is:

[0009]

[0010] Step 4: Set the voltage loop control parameters KP and KI, where the transfer function of the voltage loop control is:

[0011]

[0012] Step 5: Tune the PR controller parameter K h , the transfer function of the proportional resonant controller is:

[0013]

[0014] Step 6: v oabc The coordinate transformation is transformed from the abc coordinate system to the dq axis coordinate system, and then the control parameters of the voltage outer loop are optimized and the quasi-proportional resonant controller is used to eliminate the influence of low-order harmonics to obtain the inverter port reference current i ref abc ;

[0015] Step 7: Inverter port reference current i ref abc , inverter side inductor current i Labc , output voltage v oabc Transform from the abc coordinate system to the dq axis coordinate system, input it into the current loop, and then obtain the inverter port voltage u in the αβ coordinate system through coordinate transformation αβ ;

[0016] Step 8: Inverter port voltage u αβ The PWM drive signal of the inverter switch device is output through SVPWM modulation to drive the inverter switch to generate stable three-phase alternating current.

[0017] The working principle and beneficial effect of this scheme are: by detecting the inductor current i Labc , output voltage v oabc , LC filter output current i oabc , and then convert the abc coordinate system into the dq coordinate system to obtain the i required for control Ldq 、v odq After passing through the voltage outer loop and the proportional resonant controller, the inverter port current reference value i is obtained. ref abc, the current reference value and i Ldq 、v odq Input to the current inner loop controller, and the inverter port voltage u in the αβ coordinate system is obtained through the current inner loop controller αβ Finally, the PWM drive signal of the inverter switch device is output through SVPWM modulation. Through the action of PWM drive signal, the inverter outputs three-phase AC power. This reduces the impact of harmonics on the stability of power grid operation, enhances system robustness and dynamic response characteristics, and reduces costs; adjusts the control parameters of the voltage loop and current loop, effectively suppresses harmonics, and improves system stability.

[0018] Optionally, a filter inductor current sensor is used to detect the inductor current i Labc , use the filter capacitor voltage sensor to detect the output voltage v oabc , use the filter output current sensor to detect the LC filter output current i oabc .

[0019] Optionally, in step 1, the inverter side inductor current i is detected Labc . Reduce the number of sensors.

[0020] Optionally, in step 4, a multi-resonant harmonic compensator is added, and the transfer function is:

[0021]

[0022] The harmonic voltage distortion is further reduced, the resonance peak is further effectively suppressed, and the attenuation effect of high-amplitude low-order harmonics is enhanced.

[0023] Optionally, in step 4, an FLL frequency-locked loop is added to keep the reference frequency ω of the inverter always at ω ref . Ensure that the power output by the inverter matches the voltage and frequency of the grid and achieves synchronization with the grid voltage.

[0024] Optionally, in step 5, a proportional resonant controller is added to the voltage outer loop for feedback control.

[0025] Optionally, voltage feedforward is used in the voltage outer loop in step 6 to enhance the dynamic performance of the system. BRIEF DESCRIPTION OF THE DRAWINGS

[0026] Figure 1 is a schematic diagram of a three-phase inverter resonance compensation control method in an embodiment of the present invention;

[0027] Figure 2 This is a voltage inner loop control diagram in an embodiment of the present invention;

[0028] Figure 3 is a Bode diagram of the system in the embodiment of the present invention under different current inner loop gains;

[0029] Figure 4 This is a block diagram of the voltage outer loop control in an embodiment of the present invention;

[0030] Figure 5 This is a simplified voltage outer loop control block diagram in an embodiment of the present invention;

[0031] Figure 6 1 is the Bode diagram of the system in the embodiment of the present invention when K=15.4, KP=0.15 and the loads Z=10, Z=50, and Z=100;

[0032] Figure 7 1 is the Bode diagram of the system in the embodiment of the present invention when K=15.4, KP=0.143 and loads Z=10, Z=50, and Z=100;

[0033] Figure 8 is an open-loop Bode diagram of the system in an embodiment of the present invention when KI is 1, 5, 15, 20, 30, and 40 respectively;

[0034] Fig. 9 The Bode diagram of the system in the embodiment of the present invention at KP=0.01, 0.05, 0.143;

[0035] Fig.10 A system control block diagram of a three-phase inverter resonance compensation control method with a multi-resonance harmonic compensator added in an embodiment of the present invention;

[0036] Fig.11 A Bode diagram of a system in an embodiment of the present invention with a multi-resonant harmonic compensator added;

[0037] Fig.12 The bode diagram of the system in the embodiment of the present invention with the resonant harmonic compensator adding the resonant compensation at 50 Hz;

[0038] Fig.13 It is a control block diagram of the entire system after adding quasi-proportional resonance control in an embodiment of the present invention;

[0039] Fig.14 1 is the open-loop Bode diagram of the system at loads Z=10, 50, and 10000 in an embodiment of the present invention;

[0040] Fig.15 : is the closed-loop Bode diagram of the system at load Z=10, 50, 10000 in the embodiment of the present invention;

[0041] Fig.16 1 and 2 are voltage and current waveforms and THD after the system control method is applied in the embodiment of the present invention. DETAILED DESCRIPTION

[0042] The following is further described in detail with reference to the accompanying drawings through specific implementation methods:

[0043] Example

[0044] A three-phase inverter resonance compensation control method, the principle diagram is as shown in the attached Figure 1 As shown, the following simulation experiments prove that the control parameters of the voltage and current dual closed-loop control system are adjusted while optimizing the refining control strategy. The specific derivation and result proof are combined with the attached drawings of the specification as follows:

[0045] In this simulation verification experiment, the parameters are as follows:

[0046] f s =20KHz; L=500μH; r=0.2; ω=2π×60rad / s; C=22μF; V dc =300V.

[0047] Step 1: Detect the inverter side inductor current i through the sensor Labc , detect output voltage v oabc , detect the LC filter output current i oabc ;

[0048] Step 2: The frequency-locked loop (FLL) controls the system frequency at the reference frequency and converts the inverter frequency into an angle using the formula ph = 2π × ω.

[0049] Step 3: Set the current inner loop parameter K. The specific process is as follows:

[0050] like Figure 1 Current inner loop control block diagram, first set the current inner loop control parameter K, set the load impedance to Z, then Then the current loop open-loop transfer function is:

[0051]

[0052] The closed-loop transfer function can also be obtained:

[0053]

[0054] Assume the rated load Z = 10Ω, then:

[0055]

[0056] The current bandwidth is the switching frequency. Pick The switching frequency is the bandwidth, that is: ω bi =2π×(0.2×20)KHz≈25Krad / s. The goal is to obtain a -3dB amplitude gain in the current bandwidth. According to the current loop gain formula:

[0057]

[0058] There are When G dB =20×log 10 G(jω bi )=20×log 10 (0.707)≈-3dB. The above formula shows that the system gain at the ω frequency is 0.707, which is -3dB; it ensures that the system response is attenuated to half of the original amplitude near the ω frequency, thereby reaching a balance point that can respond to frequency components without causing excessive oscillation of the system. Substituting all the calculated parameters into the current loop gain formula, the gain K = 15.4 is obtained. Substituting the above derived parameters into the simulation model, the bode diagram of the system under different current inner loop gains is obtained as follows: Figure 2 It can be guaranteed that the bandwidth will not be less than ω under all loading conditions. bi .

[0059] Step 4: Set the control parameters KP and KI of the voltage loop. The specific process is as follows:

[0060] like Figure 3 is the control block diagram of the voltage outer loop. According to the control block diagram, the control parameters KP and KI of the voltage loop are adjusted by the transfer function. Figure 3 It can be seen that the voltage outer loop includes the control of the current inner loop, so the control block diagram of the voltage control can be simplified as Figure 4 As shown, we have:

[0061]

[0062] The voltage control open loop transfer function is:

[0063]

[0064] In the current loop. Known Let KI = 0;

[0065]

[0066] Given that K = 15.4, let KP = 0.15. Figure 5 The Bode diagrams for loads Z = 10, Z = 50, and Z = 100 are shown. It can be seen intuitively from the figure that the current inner loop can actively damp the LC resonance and the system has sufficient phase margin. Even under light loads, it can be assumed that z → ∞ is simplified as follows:

[0067]

[0068] but:

[0069]

[0070] Bandwidth selection is a compromise between transient response and anti-interference. The bandwidth of the voltage outer loop is generally designed to be greater than 10 times the base frequency and less than 1 / 10 of the switching frequency. For the system involved in the simulation of the present invention, it is greater than 500HZ and less than 2kHZ. The simulation verification of the present invention selects 1.3kHZ, then ω bv =2π×1.3kHZ≈8krad / s.

[0071] Order |G v (jω bv )| 2 =0.5, we can get KP=0.143.

[0072] Before simplification:

[0073]

[0074] Then there are:

[0075]

[0076] Where KP = 0.143, K = 15.4, KI = 0;

[0077] Draw as Figure 6 From the Bode diagrams of loads Z=10, Z=50, and Z=100 shown, it can be seen that the impact of the load on the system bandwidth is very small and can be ignored, thus ensuring the system bandwidth.

[0078] Applying the Routh-Hurwitz stability criterion to the voltage closed-loop transfer function G v (s) The stable boundary can be obtained:

[0079] Draw the open-loop Bode diagrams at KI of 10, 20, 30, and 40, as shown in Figure 7 As shown. The larger KI is, the more it can eliminate the steady-state error; the smaller KI is, the more it can ensure that the integral term does not affect other frequencies. After comprehensive consideration, KI=30 is selected. Therefore, the system control parameters after tuning are: K=15.4, KP=0.143, KI=30.

[0080] Step 5: Tune the PR controller parameter K h , the specific process is as follows:

[0081] The same by Figure 3 It can be seen that the control of the voltage outer loop can also write the transfer function:

[0082]

[0083] Draw out Z O (s) The Bode plots at KP = 0.01, 0.05, and 0.143 are shown in Figure 2. Figure 8 As shown in the figure, with the increase of KP, the harmonic voltage distortion is reduced, and the resonance peak is effectively suppressed. However, the attenuation of high-amplitude low-order harmonics by this control system is not enough, so a multi-harmonic compensator needs to be added to achieve better control effect.

[0084] The principle of resonant harmonic compensation is: The control block diagram of adding multi-resonance harmonic compensator is as follows Fig. 9 As shown, the transfer function based on this is:

[0085]

[0086] The transfer function and control parameters can be plotted as follows: Fig.10 From the Bode diagram, we can see that the high-amplitude low-order harmonics are effectively suppressed, but the amplitude at the 50HZ fundamental wave is still high. On this basis, adding a resonance compensation at 50HZ can give Fig.11 Bode diagram of the voltage shown.

[0087] The added resonant compensator only affects the frequency near the resonance, so its effect on the dynamic performance of the inverter is negligible. In order to prevent the offset of the frequency w from causing the poor effect of the resonant compensator, the following method can be used:

[0088]

[0089] By adding quasi-proportional resonant control, the control block diagram of the optimized entire system can be obtained as follows: Fig.12 As shown, the closed-loop transfer function of the entire system can be obtained:

[0090]

[0091] The open-loop transfer function can be obtained from the control block diagram:

[0092]

[0093] So the closed-loop transfer function is:

[0094]

[0095] The control parameters of the simulation verification experiment are K = 15.4, KP = 0.143, KI = 30, K h1 =2, K h5 =20, K h7 =10, K h11 =40, K h13=50.

[0096] Step 6: v oabc The coordinate transformation is transformed from the abc coordinate system to the dq axis coordinate system, and then the control parameters of the voltage outer loop are optimized and the quasi-proportional resonant controller is used to eliminate the influence of low-order harmonics to obtain the inverter port reference current i ref abc ;

[0097] Step 7: Inverter port reference current i ref abc , inverter side inductor current i Labc , output voltage v oabc Transform from the abc coordinate system to the dq axis coordinate system, input it into the current loop, and then obtain the inverter port voltage u in the αβ coordinate system through coordinate transformation αβ .

[0098] Step 8: Inverter port voltage u αβ The PWM drive signal of the inverter switch device is output through SVPWM modulation, driving the inverter switch to generate stable three-phase AC power.

[0099] At load Z = 10, 50, 10000, the following is drawn: Fig.13 The system open loop Bode diagram shown in Fig.14 The closed-loop Bode diagram of the system is shown. From the Bode diagram, it can be seen that no matter how the load changes, it will not affect the effect and stability of the system. Fig.15 The voltage and current waveforms and system THD after using the method of the present invention. It can be seen that the dynamic response of the inverter output under different load conditions changes, the voltage waveform remains relatively stable under these different load conditions, and the three-phase voltage waveform output by the inverter basically maintains a good sine waveform throughout the test process, indicating that the inverter has strong voltage stability. Frequency domain analysis shows that the harmonic distortion of the inverter is small, and the THD is 0.10%, which is far lower than the 5% specified by the national standard and IEEE standard. This is a relatively ideal level, indicating that the output quality of the inverter is high.

[0100] The above are only embodiments of the present invention. The invention is not limited to the fields involved in this implementation case. The common sense such as the known specific structures and characteristics in the scheme is not described in detail here. The ordinary technicians in the relevant field know all the common technical knowledge in the technical field to which the invention belongs before the application date or the priority date, can obtain all the existing technologies in the field, and have the ability to apply the conventional experimental means before that date. The ordinary technicians in the relevant field can improve and implement this scheme in combination with their own abilities under the enlightenment given by this application. Some typical known structures or known methods should not become obstacles for ordinary technicians in the relevant field to implement this application. It should be pointed out that for those skilled in the art, without departing from the concept of the present invention, several deformations and improvements can be made, which should also be regarded as the scope of protection of the present invention, which will not affect the effect of the implementation of the present invention and the practicality of the patent. The scope of protection required by this application shall be based on the content of its claims, and the specific implementation methods and other records in the specification can be used to explain the content of the claims.

Claims

1. A three-phase inverter resonance compensation control method, characterized in that: The following steps are involved: Step 1: Detect the inductor current i Labc , detect output voltage v oabc , detect the LC filter output current i oabc ; Step 2: The frequency-locked loop (FLL) controls the system frequency at the reference frequency and converts the inverter frequency into an angle using the formula ph = 2π × ω. Step 3: Set the current loop control parameter K, where the transfer function of the current loop control is: Step 4: Set the voltage loop control parameters KP and KI, where the transfer function of the voltage loop control is: Step 5: Tune the PR controller parameter K h , the transfer function of the proportional resonant controller is: Step 6: v oabc The coordinate transformation is transformed from the abc coordinate system to the dq axis coordinate system, and then the control parameters of the voltage outer loop are optimized and the quasi-proportional resonant controller is used to eliminate the influence of low-order harmonics to obtain the inverter port reference current i ref abc ; Step 7: Inverter port reference current i ref abc , inverter side inductor current i Labc 、Output voltage v oabc Transform from the abc coordinate system to the dq axis coordinate system, input it into the current loop, and then obtain the inverter port voltage u in the αβ coordinate system through coordinate transformation αβ ; Step 8: Inverter port voltage u αβ The PWM drive signal of the inverter switch device is output through SVPWM modulation to drive the inverter switch to generate stable three-phase alternating current.

2. A three-phase inverter resonance compensation control method according to claim 1, characterized in that: Use filter inductor current sensor to detect inductor current i Labc , use the filter capacitor voltage sensor to detect the output voltage v oabc , use the filter output current sensor to detect the LC filter output current i oabc .

3. A three-phase inverter resonance compensation control method according to claim 1, characterized in that: In step 1, the inductor current i on the inverter side is detected. Labc .

4. The three-phase inverter resonance compensation control method according to claim 1, characterized in that: In step 5, a multi-resonant harmonic compensator is added, and the transfer function is:

5. A three-phase inverter resonance compensation control method according to claim 1, characterized in that: In step 2, an FLL frequency-locked loop is added to keep the inverter reference frequency ω at all times. ref .

6. A three-phase inverter resonance compensation control method according to claim 1, characterized in that: In step 5, a proportional resonant controller is added to the voltage outer loop.

7. A three-phase inverter resonance compensation control method according to claim 1, characterized in that: The voltage outer loop in step 6 adopts voltage feed-forward.

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