An electromagnetic flow converter

Abstract

The present invention relates to an electromagnetic flow converter, including a power supply circuit, a controllable constant current excitation circuit, a controllable constant current source, an excitation detection circuit, a controllable amplifier circuit, an AD conversion circuit, an empty pipe detection circuit, a keyboard and display circuit, and a communication interface circuit. The controllable current source and the controllable gain amplifier are used in combination to automatically adjust the amplification factor of the operational amplifier circuit and the excitation current of the sensor respectively; according to the current calculated flow value, the appropriate amplification factor and the appropriate excitation current are automatically selected, so that the measurement range of the flow meter becomes wider, the range ratio is larger, and the scope of application is wider. Different amplification factors and different excitation currents constitute a combination of multiple traditional flow meters. In any combination, it is equivalent to an independent traditional electromagnetic flow meter working, and can be seamlessly switched, using only the same set of hardware circuits and the same set of software algorithms, which greatly saves costs and improves the efficiency of resource use.

Description

Technical Field

[0001] The invention relates to an electromagnetic flowmeter, in particular to an electromagnetic flow converter. Background Art

[0002] The traditional electromagnetic flow converter is the core component of the electromagnetic flowmeter. The instrument coefficient is set by setting the converter coefficient and the sensor coefficient. The converter coefficient is mainly used to calibrate the converter performance so that the same tooling can work on the converter and achieve the same flow output, and then obtain the corresponding converter coefficient. The main purpose is to make the converter coefficient reach a consistent level between the converters and ultimately achieve the purpose of direct replacement. The sensor coefficient is usually obtained by calibration through a water flow calibration device. In order to improve the accuracy of on-site use, the sensor and converter are usually used in combination. Although the principle and goal are to be replaceable, in actual applications, in order to ensure measurement accuracy, they are still used in combination.

[0003] Traditional electromagnetic flow converters have only one set of calibration parameters. Once calibrated at the factory, their excitation current and instrument factor are fixed. Their accuracy is only valid within the calibration range, and generally only one accuracy index is given. For applications with a wide range ratio and certain requirements for accuracy, this traditional electromagnetic flow converter is pale and powerless.

[0004] In order to achieve normal use at low flow rates on site, the operational amplifier circuit is usually designed as a multi-speed adjustable circuit. By manually setting the amplification gear, the voltage is amplified, thereby realizing the measurement of low flow rate signals. However, this solution can only be used in situations where the flow rate is always relatively low. When the flow rate increases, the signal output by the sensor will also increase. After amplification by the amplifier circuit, it is very likely that the operational amplifier circuit will reach saturation and lose its linear amplification function, ultimately causing measurement errors. This solution requires manual intervention in the magnification setting, and cannot take into account the measurement of large and small flow rates at the same time. In addition, there is only one set of calibration parameters in the design of traditional electromagnetic flowmeters, resulting in large differences in accuracy at low and high flow rates.

[0005] In order to avoid the problem of accuracy difference between large and small flow rates, multi-stage correction is usually used to improve accuracy. However, this method is only a temporary solution and does not address the root cause. When the flow rate is too small, it is because the effective signal output by the sensor is too small, which makes it easy to be interfered. At the same time, because the voltage after amplification by the operational amplifier is too low, the noise of the operational amplifier circuit itself is likely to contribute to the flow rate, which ultimately leads to deviations in the measurement results.

[0006] The excitation current is set manually to match the sensor, and the excitation current is usually changed by disassembling the sampling resistor. In this way, the sensor matching is achieved completely manually, and of course, automated assembly can be achieved by using robot assembly. However, once an irregular installation occurs on site, for example, for convenience, the sensor is taken out and installed on the pipe first, and then the converter is installed. In the end, the problem is that the delivery is a matching set, but during installation, it is not installed in a matching set, resulting in mixed installation and mixed use. For sensors and converters with the same excitation current, it may be possible to use them by simply changing the caliber, but if the excitation current is different, it will directly lead to measurement deviation problems. Summary of the invention

[0007] Traditional electromagnetic flowmeters have only one set of instrument coefficients, which greatly restricts the development of intelligent flowmeters, resulting in the inability to achieve a larger range ratio and the inability to take into account both large and small flow rates at the same time.

[0008] Therefore, the range ratio of the flow meter is improved by automatically adjusting the magnification of the operational amplifier and the excitation current. The measurement accuracy is ensured by multi-level parameter storage and automatic call.

[0009] In view of the problem that traditional electromagnetic flowmeters cannot achieve a larger range ratio during use, that is, taking into account both large and small flow rates at the same time, the amplification factor of the operational amplifier is automatically adjusted to avoid manual settings, and automatic multi-speed signal amplification is achieved, thereby achieving a wide range ratio. Since it is impossible for ideal electronic components to exist, the amplification factor is corrected using the calibration parameters switched between gears. When switching gears, the corresponding calibration parameters are automatically multiplied to achieve flow calculation to achieve the goal of a wide range ratio. In order to achieve higher-precision measurements, flow calibration can be performed at different gears, and the calibration parameters of the corresponding gears are used to calculate the flow when switching gears. While achieving a wide range ratio, the accuracy of flow conversion is improved.

[0010] The acquisition circuit uses an amplifier circuit with variable amplification factor and designs multiple amplification gears. Each gear corresponds to a measurement range. When the calculated flow rate exceeds the range, the gear can be automatically switched. There are overlapping measurement intervals between gears. When it is higher than the upper limit of the current gear or lower than the lower limit of the current gear, it will automatically switch. Since there are overlapping measurement intervals between gears, it can effectively avoid misjudgment and frequent switching due to interference and other factors. The advantage of this treatment is that it increases the amplitude of the analog amplifier circuit output, which is conducive to subsequent data acquisition.

[0011] In order to further improve the range ratio, the excitation circuit is adjusted using a controllable constant current source. In order to avoid mutual interference, a constant current control circuit that can implement photoelectric isolation technology is preferably used. Based on the constant current control circuit controlled by PWM, the current is regulated by changing the duty cycle. In order to improve the output accuracy of the excitation current, the excitation current is corrected and the correction result is saved. There are also many ways to implement the isolated constant current control circuit. Among them, the PWM method can accurately control the current size due to the small number of pins used, so this method is preferred in the present invention. Of course, the use of a voltage reference combined with a controllable gain amplifier can also achieve multi-speed current regulation.

[0012] Under different excitation currents, the signal amplitude of the sensor output is different. Therefore, the same instrument coefficient cannot be used. In order to ensure the normal use of the flow meter, multiple excitation currents are calibrated to obtain the relationship between the excitation current and the sensor output, thereby realizing the correction of the instrument coefficient.

[0013] The technical solution of the present invention is as follows:

[0014] An electromagnetic flow converter includes: a power supply circuit, a controllable constant current excitation circuit, a controllable constant current source, an excitation detection circuit, a controllable amplifier circuit, an AD conversion circuit (i.e., an ADC chip, used to convert analog quantities into digital quantities), an empty pipe detection circuit, a keyboard and display circuit, and a communication interface circuit. Among them, the power supply circuit uses an isolated switching power supply to provide multiple groups of different voltages to provide a stable isolated power supply for the electromagnetic flow converter; the controllable constant current excitation circuit is used to provide a direction-controllable current for the excitation coil of the sensor; the controllable constant current source is used to control the current flowing through the excitation coil of the controllable constant current excitation circuit; the excitation detection circuit is used to determine whether the controllable constant current excitation circuit is working normally; the controllable amplifier circuit is used to perform differential acquisition, bandpass filtering, level shifting and other processing on the analog voltage signal, so that the output voltage signal is convenient for AD conversion; the AD conversion circuit is used to process the voltage signal output by the controllable amplifier circuit, convert the analog quantity into a digital quantity, and facilitate MCU processing; the empty pipe detection circuit is used to determine whether the liquid is full; the keyboard and display circuit are used to realize local human-computer interaction; and the communication interface circuit is used to realize remote data interaction.

[0015] In circuit design, RC filters are used for the power supply of operational amplifiers to suppress power supply ripple. The core of the controllable amplifier circuit is to dynamically adjust the feedback network.

[0016] The controllable amplifier circuit includes: CD4053BPWR chip, operational amplifier U10A, operational amplifier U10B, resistors R30, R31, R32, R33, R34, R35, R36, R37, R38, R39, R40, R41 and capacitors C12, C13, C14, C15, C16, C17, C18;

[0017] Pin 16 of the CD4053BPWR chip is connected to a +5V power supply; Pins 6 and 8 of the CD4053BPWR chip are connected to ground; Pins 9 and 10 of the CD4053BPWR chip are connected to the pins of the microcontroller (the microcontroller controls the switch state of the CD4053BPWR chip through the high and low levels of the pins); Pins 2 and 4 of the CD4053BPWR chip are connected; Pin 7 of the CD4053BPWR chip is connected to a -5V power supply and is grounded through capacitor C18; Resistors R38, R39, R40, and R41 are connected in series, and resistor R The end of 41 not connected to R40 is grounded; Pin 1 of the CD4053BPWR chip is connected to the node between resistors R38 and R39, Pin 3 of the chip is connected to the node between resistors R39 and R40, and Pin 5 of the chip is connected to the node between resistors R40 and R41; The end of resistor R38 not connected to resistor R39 is connected to one end of resistor R35 and the output end of operational amplifier U10A respectively; The other end of resistor R35 is connected to one end of resistor R36 and the inverting input end of operational amplifier U10B respectively, The output end is connected to the other end of the resistor R36 and one end of the resistor R34 respectively; the other end of the resistor R34 is connected to one end of the capacitor C15 and the ADC input end respectively, and the other end of the capacitor C15 is grounded; the non-inverting input end of the operational amplifier U10B is connected to the capacitor C13, the resistor R30, and one end of the resistor R31 respectively, and the other end of the resistor R31 is connected to the other end of the capacitor C13 and one end of the capacitor C12 and is grounded; the other end of the capacitor C12 is connected to the other end of the resistor R30 and is connected to the +5V power supply; the inverting input end of the operational amplifier U10A is connected to the CD The pin 15 of the 4053BPWR chip is connected; the non-inverting input terminal of the operational amplifier U10A is connected to one end of the resistor R33, and the other end of the resistor R33 is connected to the analog voltage signal; the positive power supply terminal of the operational amplifier U10A is connected to one end of the resistor R32 and one end of the capacitor C14 respectively, the other end of the resistor R32 is connected to the +5V power supply, and the other end of the capacitor C14 is grounded; the negative power supply terminal of the operational amplifier U10A is connected to one end of the resistor R37 and one end of the capacitor C16 respectively, the other end of the resistor R37 is connected to the -5V power supply, and the other end of the capacitor C16 is grounded.

[0018] The models of operational amplifier U10A and operational amplifier U10B are TL072IDR or TLC2272AI.

[0019] The controllable constant current excitation circuit includes: a photocoupler, an inverter U5, an operational amplifier U6B, an operational amplifier U6A, a metal-oxide semiconductor field effect transistor, an H-bridge circuit, resistors R12, R13, R14, R15, R16, R17, R18, R19, R20, R21, R22, R23, R24 and capacitors C4, C5, C6, C7, C8, C9, C10;

[0020] The model of the inverter U5 is SN74AHC1G04; the optocoupler uses a high-speed optocoupler, such as 6N137; the model of the operational amplifier U6B and the operational amplifier U6A is TLC27L2IDR; the metal-oxide semiconductor field effect transistor is a MOS tube, model ZVN2120.

[0021] Pin 2 of the photocoupler is connected to the power supply 3.3V through resistor R13; Pin 3 of the photocoupler is connected to the PWM signal; Pin 5 of the photocoupler is grounded; Pin 6 of the photocoupler is connected to one end of resistor R14 and one end of R15 respectively, and Pin 8 of the photocoupler is connected to one end of resistor R11; The other end of resistor R14, the other end of resistor R11 and one end of capacitor C5 are connected and connected to V1 (V1 is the power supply of the photocoupler. The inverter U5 behind will shape the output waveform of the photocoupler, and the waveform voltage is determined by the voltage V2 of the inverter U5, so , V1 voltage range is very wide, the value is relatively arbitrary); the other end of capacitor C5 is grounded; the other end of resistor R15 is connected to the input end of inverter U5, the output end of inverter U5 is connected to one end of resistor R16, the other end of resistor R16 is connected to one end of resistor R17 and one end of capacitor C8 respectively, the other end of capacitor C8 is grounded; the other end of resistor R17 is connected to one end of resistor R18 and one end of capacitor C7 respectively, the other end of resistor R18 is connected to the same end of operational amplifier U6B (the first stage operational amplifier, using a 3rd order low pass filter to effectively avoid high frequency leakage) The inverting input terminal of the operational amplifier U6B is connected to one end of the resistor R21, the other end of the resistor R21, the other end of the capacitor C7, and the output terminal of the operational amplifier U6B are all connected to the non-inverting input terminal of the operational amplifier U6A, the negative power supply terminal of the operational amplifier U6A is grounded, and the positive power supply terminal of the operational amplifier U6A uses a low-power filter composed of a resistor R12 and a capacitor C6 to suppress power supply noise; the inverting input terminal of the operational amplifier U6A is connected to one end of the resistor R20 and the capacitor C10 The other end of the resistor R20 is connected to one end of the resistors R22, R23, and R24 respectively, and the other ends of the resistors R22, R23, and R24 are connected and grounded; the other end of the capacitor C10 is connected to the output end of the operational amplifier U6A; the output end of the operational amplifier U6A is connected to one end of the resistor R19, and the other end of the resistor R19 is connected to the gate of the metal-oxide semiconductor field effect transistor, the drain of the metal-oxide semiconductor field effect transistor is connected to the H-bridge circuit, and the source of the H-bridge circuit is connected to one end of the resistors R22, R23, and R24 respectively.

[0022] The power supply voltage V2 of the inverter U5 is provided by a high-precision voltage reference chip.

[0023] Alternatively, the power supply voltage V2 of the inverter U5 is provided by a reference voltage generating circuit, which uses a reference voltage divider plus an op amp follower (a 1.25v or 2.5v voltage reference chip is used to obtain a reference voltage, which is then divided by a resistor to obtain a reference voltage V2ref, and V2ref is followed by an op amp to obtain a power supply voltage V2).

[0024] The communication interface circuit includes RS485 module, 4-20mA module, hart module, bluetooth module, wifi module and sub-g module.

[0025] The electromagnetic flow converter has the function of storing and calling multi-speed correction parameters, and automatically adjusts the controllable amplification circuit and controllable excitation current according to the current calculated flow rate; different amplification factors and different excitation currents form a variety of measurement combinations, and each combination sets a measurement range, so that there are overlapping measurement intervals between the combinations. When the calculated flow rate exceeds the measurement range of this gear, the gear is switched to avoid frequent gear switching; in order to improve the measurement accuracy, each combination has a calibrated instrument coefficient; in order to reduce the calibration complexity, the influence factors of the amplification factor and the excitation current are decomposed, and the influence factors of changing the amplification factor and the excitation current separately are calibrated, and then the instrument coefficients of all combinations are obtained.

[0026] This method can also reduce the number of calibrations. The number of calibrations saved is m*n-(m+n-1), where m and n are the number of levels of excitation current and amplification factor, respectively.

[0027] The beneficial effects of the present invention are as follows: the controllable current source and the controllable gain amplifier are used in combination to automatically control the amplification factor of the operational amplifier circuit and the excitation current of the sensor; the controllable current source controls the excitation current, and the controllable gain amplifier controls the amplification factor. According to the current calculated flow value, the appropriate amplification factor and the appropriate excitation current are automatically selected, so that the measurement range of the flow meter becomes wider, the range ratio is larger, and the scope of application is wider. When the flow rate becomes lower or higher, by adjusting the amplification factor, it is ensured that the final output voltage of the operational amplifier circuit is not distorted and is as large as possible, and the output voltage must meet the voltage requirements of the ADC acquisition. In the process of amplification factor conversion, there is an overlapping area of ​​the factor, so that the amplification factor (gear) is not switched too frequently, ensuring stability; when the flow rate is very low or very high, by adjusting the excitation current, the output voltage of the sensor is changed, so that the amplified voltage after passing through the amplifier circuit meets the voltage requirements of the ADC acquisition. The combination of these two methods can greatly widen the range ratio.

[0028] By using a controllable gain amplifier, a smaller flow device can be used to calibrate the flow meter, solving the problem in traditional calibration methods where the flow range of the device cannot meet the measurement range.

[0029] By combining the controllable current source and the controllable gain amplifier, the measuring range of the flow meter becomes wider, the range ratio becomes larger, and the application range becomes wider.

[0030] Multiple gear parameter calibration, storage and automatic call to ensure measurement accuracy.

[0031] There is overlap in the measurement ranges between gears, and the hysteresis comparator model is used to avoid frequent gear switching.

[0032] The operating mode of the controllable circuit can be set to automatic and manual. In manual mode, parameter correction is convenient.

[0033] Different amplification factors and different excitation currents constitute a combination of multiple traditional flow meters. In any combination, it is equivalent to an independent traditional electromagnetic flow meter working and can be switched seamlessly, using only the same set of hardware circuits and the same set of software algorithms, which greatly saves costs and improves the efficiency of resource utilization.

[0034] In order to improve the measurement accuracy, each combination is calibrated in the traditional way to obtain multiple instrument coefficients. When the electromagnetic flowmeter works in a certain combination, the corresponding instrument coefficient is automatically called for flow calculation, which ensures the output accuracy of the flowmeter while expanding the measurement range of the electromagnetic flowmeter.

[0035] In order to reduce the complexity of calibration, the two dimensions of the influence of the amplification factor (gain) and the influence of the excitation current are decomposed to form a matrix flowmeter model. At this time, by fixing the amplification factor (gain) or the excitation current, the influence factors of the excitation current and gain are calculated respectively, and then the coefficients of all combinations are derived, which reduces the calibration requirements and saves manpower and material resources.

[0036] In general, the present invention uses a controllable current source and a controllable gain amplifier to make the signal easier to collect, and then uses the influence factor decomposition method to reduce the complexity of calibration, so that the electromagnetic flow converter with this structure is more practical. BRIEF DESCRIPTION OF THE DRAWINGS

[0037] Figure 1 Electromagnetic flowmeter system block diagram;

[0038] Figure 2 Basic schematic diagram of controllable gain amplifier circuit;

[0039] Figure 3 A circuit diagram of an amplifier with controllable magnification according to an embodiment of the present invention;

[0040] Figure 4 Traditional constant current excitation circuit diagram (constant current part);

[0041] Figure 5 A controllable constant current excitation circuit diagram of an embodiment of the present invention;

[0042] Figure 6 Figure 5 A partial enlarged diagram of

[0043] Figure 7A reference voltage generating circuit;

[0044] Figure 8 Basic processing mode of traditional electromagnetic flowmeter acquisition system. DETAILED DESCRIPTION

[0045] The following is combined with Figure 1-8 The present invention is described in further detail.

[0046] The basic processing mode of the traditional electromagnetic flowmeter acquisition system is as follows: Figure 8 As shown, the voltage output by the sensor is extremely weak, so the preamplifier circuit usually uses an instrument amplifier to process the signal, and then uses a low-pass filter to filter the signal, and then removes the DC component through DC blocking capacitor coupling to obtain a low-frequency AC signal, which is then amplified by the amplifier circuit and finally, after level shifting, a positive signal that is convenient for ADC processing is obtained, and then enters the ADC.

[0047] In the process of using traditional electromagnetic flowmeters, since their magnification and excitation current are fixed values, they cannot achieve a larger range ratio, that is, the problem of taking into account both large and small flow rates. Since the excitation current is fixed, when the flow rate is low, according to the law of electromagnetic induction, the signal output by the sensor will be very low. Affected by noise interference, the signal output by the amplifier circuit at this time has a relatively large noise contribution value, resulting in poor accuracy of the calculated flow value, or even exceeding the tolerance. When the flow rate is high, the output signal of the sensor is large. When the magnification is set to a large value, it is possible that the output of the op amp will be saturated, resulting in signal distortion.

[0048] In order to solve the above problems, a controllable gain amplifier and a controllable current source are used together to automatically adjust the amplification factor of the operational amplifier circuit and the excitation current of the sensor, so that the final output voltage of the operational amplifier circuit is not distorted and is as large as possible, and meets the acquisition conditions of the ADC, thereby realizing a flow meter with a wide range ratio.

[0049] After the hardware circuit is fixed, the relationship between different amplification (gain) levels is determined, that is, each amplifier circuit can correspond to a coefficient, through which the output voltage value of the amplifier circuit can be converted into an equivalent output voltage value of a fixed amplification factor.

[0050] The excitation current affects the output voltage of the sensor. Through the relationship between the excitation current and the magnetic field strength, it can be known that the excitation current is proportional to the magnetic field strength. According to the law of electromagnetic induction, the magnetic field strength is proportional to the induced voltage, and the induced voltage is proportional to the flow rate of the fluid. Therefore, the excitation current is proportional to the flow rate of the fluid. That is, there is a proportional relationship between different excitation currents, which can be equivalently converted through coefficients.

[0051] The present invention achieves automatic multi-level signal amplification by automatically adjusting the amplification ratio of the operational amplifier to avoid manual setting, thereby achieving a wide range ratio. The basic principle of the controllable gain amplifier is as follows Figure 2 As shown in the figure, by controlling the switches of S1 to S3, different magnifications can be obtained. In order to realize intelligent control, a multiplexed analog switch is usually used to replace the switches shown in S1 to S3. It can be directly controlled by a single-chip microcomputer, that is, the magnification is set by the single-chip microcomputer independently, which can determine which magnification position the current op amp circuit is in, facilitating the calculation of flow. There are many types of analog switches, including single-channel multi-channel and multi-channel multi-channel, such as the commonly used 74HC series analog switches and CD405x series switches.

[0052] The acquisition circuit has multiple amplification gears, each gear corresponds to a measurement range, which is limited by the acquisition voltage of the ADC. When the calculated flow rate (ADC input voltage corresponds one to one) exceeds the range, the gear is automatically switched. Overlapping measurement intervals are set between the gears. When it is higher than the upper limit of the current gear or lower than the lower limit of the current gear, it is automatically switched. Since there are overlapping measurement intervals between the gears, the working model is similar to a hysteresis comparator, which can effectively avoid misjudgment and frequent switching due to interference and other factors. The advantage of this treatment is that the amplitude of the analog amplifier circuit output is increased, which is conducive to subsequent data acquisition.

[0053] In order to reduce the influence of power supply ripple, an RC filter is applied to the power supply end of the op amp. The basic principle of the multiplier controllable amplifier is to change the feedback network, such as Figure 3 As shown in (circuit diagram of controllable amplifier circuit of one embodiment of the present invention), the feedback network is composed of R38, R39, R40, and R41. Pins 9 and 10 of the CD4053BPWR chip are connected to the pins of the microcontroller. The microcontroller can obtain different amplification factors by controlling the control levels of UFB and UFC.

[0054] In the traditional constant current excitation circuit, the constant current circuit commonly used is as follows Figure 4 As shown. In order to change the excitation current, it is usually achieved by changing the resistance. However, once this method is implemented, the excitation current becomes a fixed value during use, which limits the output range of the sensor. There are two ways to achieve this: 1. Change the voltage divider resistor, 2. Change the sampling resistor. The essence of changing the voltage divider resistor (R3, R6, R7) is to change the reference voltage of the op amp's non-inverting input terminal, and the essence of changing the sampling resistor (R8, R9, R10) is to change the feedback voltage of the circuit (the voltage at the inverting input terminal). When the voltages at the non-inverting and inverting input terminals of the op amp are equal, the voltage reaches a balanced state. The excitation current is calculated by the following formula:

[0055]

[0056] In order to further improve the range ratio, the excitation circuit uses a controllable constant current source for adjustment. In order to avoid mutual interference, a constant current control circuit based on photoelectric isolation technology is implemented, that is, a constant current control circuit based on PWM control is used to adjust the current by changing the PWM duty cycle. In order to improve the output accuracy of the excitation current, the excitation current is corrected and the corrected result is saved. Compared with the traditional excitation constant current circuit ( Figure 4 ) is different in that Vref is generated in different ways. The traditional method directly obtains it through resistor voltage division, while the present invention uses a reference voltage Vref obtained by optical coupling isolation, shaping, and filtering of a PWM pulse. Figure 5 (Controllable constant current excitation circuit diagram of one embodiment of the present invention) is shown.

[0057] The PWM waveform sent by the single-chip microcomputer passes through the high-speed photocoupler U4, and is shaped by the inverter U5 to obtain a PWM waveform with controllable amplitude. The amplitude of the waveform is determined by V2. The role of the resistor R15 is to provide current limiting protection for the input port of the inverter. The PWM waveform output by the inverter U5 passes through a third-order low-pass filter composed of resistors R16, R17, R18, R21, capacitors C7, C8, C9, and operational amplifier U6B (where R16 and C8 are of significance in preventing AC signal leakage to the filter) to obtain a reference voltage Vref, which is proportional to the duty cycle of the PWM.

[0058] The output current of the H-bridge can be calculated by formula (1). In order to reduce the interference of power supply noise on the operational amplifier U6A, an RC filter (R12, C6) is added to the positive power supply port of the operational amplifier U6A to improve the ripple of the positive power supply terminal of the operational amplifier U6A, improve the stability of the output of the operational amplifier U6A, and ensure the accuracy of the output current. According to the following characteristics of the operational amplifier U6A, the voltage at the inverting input terminal is consistent with the non-inverting input terminal, that is, the voltage across the sampling resistors R22, R23, and R24 is consistent with the voltage (Vref) obtained by the pre-stage filter. According to Ohm's law, it can be seen that the current on the sampling resistor is proportional to the duty cycle of the PWM. Adjusting the PWM duty cycle controls the output current of the circuit.

[0059] The generation of V2 can be provided by a high-precision voltage reference chip, or by using a reference voltage divider plus an op amp follower. In order to more conveniently obtain the corresponding reference voltage, the latter is adopted, that is, a reference voltage is generated by a reference chip, and then the target voltage is obtained by a resistor divider, and the final reference voltage is obtained after passing through an op amp follower. The use of a follower can improve the driving capability. Figure 7 (reference voltage generating circuit) is Figure 5The inverter U5 in the circuit provides the power supply voltage V2. The reference voltage is obtained by using the 1.25v or 2.5v voltage reference chip U9 (TL431 or TL432), and the reference voltage V2ref is obtained by resistor voltage division. After V2ref is followed by the op amp, the output voltage V2 is obtained. The purpose of using the voltage follower is to reduce the output impedance, improve the driving ability of V2, avoid voltage fluctuations caused by changes in the load carried by V2, and affect the high-level voltage value of the PWM waveform. It is also an important measure to ensure the accuracy of current output.

[0060] According to the law of electromagnetic induction, under different excitation currents, the magnetic field strength generated on the excitation coil is different, and the signal amplitude of the sensor output is also different. The magnetic induction intensity is proportional to the excitation current, and a proportional coefficient can be used to characterize the influence of the excitation current on the sensor output signal. In order to ensure the accuracy of flow calculation, multiple excitation currents are calibrated to obtain the relationship between the excitation current and the sensor output, thereby realizing the correction of the instrument coefficient.

[0061] When the fluid flow rate is relatively low, the output voltage of the sensor is particularly small. At this time, affected by the interference noise, the signal-to-noise ratio is relatively low. Simply increasing the amplification factor of the operational amplifier will inevitably lead to a decrease in the accuracy of the calculated flow rate. For this reason, when the fluid flow rate is relatively low, the signal voltage output by the sensor is increased by increasing the excitation current to achieve the purpose of improving the signal-to-noise ratio, thereby ensuring the measurement accuracy. When the fluid flow rate is relatively high, the output voltage of the sensor may reach the limit voltage of the ADC after passing through the amplifier circuit (under the minimum amplification factor). For this reason, the output voltage of the sensor is reduced by reducing the excitation current to obtain a larger range ratio.

[0062] Specifically:

[0063] In order to accurately control the output current, the PWM current control circuit correction parameters are stored in the flowmeter. Through the correction parameters, the duty cycle of the target current can be calculated to control the excitation current. In order to ensure the consistency of the output flow when the amplifier ratio is switched, the coefficients under each magnification are calibrated to obtain coefficients under multiple magnifications, so that when the magnification is switched, the corresponding coefficients are called for flow calculation, which are referred to as the multiplier coefficients in the following text. The multiplier coefficients are obtained by changing the amplifier ratio under a fixed flow; in order to ensure the accuracy of flow calculation under different excitation currents, calibration is performed under each excitation current to obtain the proportional coefficients under multiple currents, referred to as the current coefficients.

[0064] Correction method of excitation current: replace the excitation coil with a standard resistor, control the excitation excitation so that the H bridge is no longer alternately turned on, set the PWM wave with a duty cycle by the microcontroller, record the voltage value across the standard resistor (divide by the resistance to get the current), and thus get the relationship between the excitation current and the PWM duty cycle. Use the two-point method to get a linear relationship, that is, first set two duty cycles, measure the corresponding voltage value to get the current value, assuming that the two duty cycles are a and b, and the measured voltages are V a 、V b , the resistance of the standard resistor is R.

[0065] Duty Cycle Voltage Calculate current a <![CDATA[V a ]]> <![CDATA[V a / R]]> b <![CDATA[V b ]]> <![CDATA[V b / R]]>

[0066] Then the current at any duty cycle x is:

[0067]

[0068] The duty cycle x corresponding to any current I is:

[0069]

[0070] The calibration parameters are saved in (a, V a )(b,V b ), calculate the duty cycle at the specified current according to the above formula.

[0071] Method for calibration of the multiplier coefficient: perform calibration at a fixed flow rate (i.e., a fixed sensor output signal, which can be simulated with a signal generator), and obtain and record the corresponding ADC value (ADC value is the digital value output by the ADC chip) by adjusting the amplification factor. Note that during the calibration process, it is necessary to ensure that the ADC voltage value meets the input conditions and that the operational amplifier does not have saturation distortion. For the sake of convenience, assume that the amplification factors of each gear are M0, M1, and M2, respectively. Under the same sensor input, the corresponding ADC values ​​are ADC0, ADC1, and ADC2. Since the sensor input is the same, and the ADC value has a one-to-one correspondence with the output value of the amplifier circuit, there is the following relationship between the ADC value and the operational amplifier amplification factor:

[0072]

[0073] Let K0 = 1 but

[0074]

[0075] Since the ADC value is directly proportional to the flow rate when the magnification is fixed, when the gear is changed, each ADC value is normalized by formula (5), that is, after the formula transformation, the ADC value used to calculate the flow rate is consistent.

[0076] Calibration method of current coefficient: When switching between multiple current levels, the output voltage of the same sensor at the same flow rate will be different. In theory, the excitation current is proportional to the sensor output signal. In order to improve the accuracy, the calibration method is used to obtain the proportional relationship, which can better characterize the actual flow rate. Under the same flow rate and the same magnification, change the excitation current and record the ADC value. Suppose the excitation currents are Ia, Ib, and Ic respectively, and the ADC values ​​are ADC_a, ADC_b, and ADC_c respectively. Since the flow rate is fixed, there must be a set of coefficients K, so that the following formula holds:

[0077] ADC_a*Ka=ADC_b*Kb=ADC_c*Kc

[0078]

[0079] The current can be switched between multiple gears. The smaller the excitation current, the smaller the output signal of the sensor. The amplification factor can be switched between multiple gears. The higher the amplification factor, the larger the ADC value. Assume that a flow meter has 3 (increasable) excitation current gears and 3 (increasable) amplification factor gears. Assuming that it is calibrated at the amplification factor M0 and the excitation current gear Ia, the coefficients are (1, K1, K2) (1, Kb, Kc), and other coefficients can be expanded as shown in the table.

[0080] (Ia->Ib->Ic M0->M1->M2 increase in sequence)

[0081]

[0082] When switching between gears, the ADC value can be normalized according to the coefficients in the table, and then the flow rate can be calculated according to the flow rate relationship of the calibration point M0.

[0083] When powered on, the flow meter starts with the default excitation current and magnification, detects whether it is in automatic magnification mode, and if so, automatically switches the magnification, and detects whether it is in automatic current control mode, and if so, automatically switches the excitation current.

[0084] When the excitation current and the amplification factor are at their maximum, that is, (Ic M2), the lower limit of the detectable flow rate reaches its minimum; when the excitation current and the amplification factor are at their minimum, that is, (Ia M0), the upper limit of the detectable flow rate reaches its maximum.

[0085] When the detected ADC value (its corresponding flow rate) is small, it is adjusted by increasing the amplification factor and the excitation current, that is, adjusting toward the lower right side of the graph; when the detected ADC value (its corresponding ADC flow rate) is large, it is adjusted by reducing the amplification factor and the excitation current, that is, adjusting toward the upper left side of the graph.

[0086] The measurable range of each gear is mainly limited by the operational amplifier output without distortion and the ADC input voltage meeting the input requirements (usually less than the power supply voltage). The range of guaranteed accuracy is limited by the signal-to-noise ratio of the sensor output signal. The smaller the sensor output signal, the higher the degree of noise contribution and the worse the accuracy.

[0087] Due to the existence of multiple gears, the upper and lower limits of measurement within the gear can be flexibly allocated reasonably. When the measured value exceeds the upper or lower limit, switching is performed. If there is an overlapping area in the measurement range, the current gear is kept unchanged in the overlapping area, and the gear is changed only when it exceeds the range.

[0088] The above implementation modes are only used to illustrate the present invention, but not to limit the present invention. Ordinary technicians in the relevant technical field can make various changes and modifications without departing from the essence and scope of the present invention. Therefore, all equivalent technical solutions also belong to the protection scope of the present invention.

[0089] The contents not described in detail in this specification belong to the prior art known to professional and technical personnel in this field.

Claims

1. An electromagnetic flow converter, characterized in that: include: Power supply circuit, controllable constant current excitation circuit, controllable constant current source, excitation detection circuit, controllable amplifier circuit, AD conversion circuit, empty pipe detection circuit, keyboard and display circuit and communication interface circuit; wherein, the power supply circuit uses an isolated switching power supply to provide multiple sets of different voltages to provide a stable isolated power supply for the electromagnetic flow converter; the controllable constant current excitation circuit is used to provide a direction-controllable current for the excitation coil of the sensor; the controllable constant current source is used to control the current flowing through the excitation coil of the controllable constant current excitation circuit; the excitation detection circuit is used to determine whether the controllable constant current excitation circuit is working normally; the controllable amplifier circuit is used to perform differential acquisition, bandpass filtering and level shifting processing on the analog voltage signal, so that the output voltage signal is convenient for AD conversion; the AD conversion circuit is used to process the voltage signal output by the controllable amplifier circuit, convert the analog quantity into a digital quantity, which is convenient for MCU to process; the empty pipe detection circuit is used to determine whether the liquid is full; the keyboard and display circuit are used to realize local human-computer interaction; the communication interface circuit is used to realize remote data interaction; The controllable amplifier circuit includes: a CD4053BPWR chip, an operational amplifier U10A, an operational amplifier U10B, resistors R30, R31, R32, R33, R34, R35, R36, R37, R38, R39, R40, R41 and capacitors C12, C13, C14, C15, C16, C17, C18; Pin 16 of CD4053BPWR chip is connected to +5V power supply; Pin 6 and Pin 8 of CD4053BPWR chip are grounded; Pin 9 and Pin 10 of CD4053BPWR chip are connected to the pins of microcontroller; Pin 2 and Pin 4 of CD4053BPWR chip are connected; Pin 7 of CD4053BPWR chip is connected to -5V power supply and grounded through capacitor C18; Resistors R38, R39, R40 and R41 are connected in series, and the end of resistor R41 not connected to R40 is grounded; Pin 1 of the chip is connected to the node between resistors R38 and R39, pin 3 of the chip is connected to the node between resistors R39 and R40, and pin 5 of the chip is connected to the node between resistors R40 and R41; the end of resistor R38 not connected to resistor R39 is connected to one end of resistor R35 and the output end of operational amplifier U10A respectively; the other end of resistor R35 is connected to one end of resistor R36 and the inverting input end of operational amplifier U10B respectively, and the output end of operational amplifier U10B is connected to the other end of resistor R36 and the inverting input end of resistor R39. The other end of resistor R34 is connected to one end of capacitor C15 and the ADC input terminal respectively, and the other end of capacitor C15 is grounded; the non-inverting input terminal of operational amplifier U10B is connected to capacitor C13, resistor R30, and one end of resistor R31 respectively, and the other end of resistor R31 is connected to the other end of capacitor C13 and one end of capacitor C12 and grounded; the other end of capacitor C12 is connected to the other end of resistor R30 and connected to the +5V power supply; the inverting input terminal of operational amplifier U10A is connected to the CD4053BPWR chip The pin 15 of the chip is connected; the non-inverting input terminal of the operational amplifier U10A is connected to one end of the resistor R33, and the other end of the resistor R33 is connected to the analog voltage signal; the positive power supply terminal of the operational amplifier U10A is connected to one end of the resistor R32 and one end of the capacitor C14 respectively, the other end of the resistor R32 is connected to the +5V power supply, and the other end of the capacitor C14 is grounded; the negative power supply terminal of the operational amplifier U10A is connected to one end of the resistor R37 and one end of the capacitor C16 respectively, the other end of the resistor R37 is connected to the -5V power supply, and the other end of the capacitor C16 is grounded; The controllable constant current excitation circuit includes: a photoelectric coupler, an inverter U5, an operational amplifier U6B, an operational amplifier U6A, a metal-oxide semiconductor field effect transistor, an H-bridge circuit, resistors R12, R13, R14, R15, R16, R17, R18, R19, R20, R21, R22, R23, R24 and capacitors C4, C5, C6, C7, C8, C9, C10; Pin 2 of the photocoupler is connected to the power supply 3.3V through resistor R13; Pin 3 of the photocoupler is connected to the PWM signal; Pin 5 of the photocoupler is grounded; Pin 6 of the photocoupler is connected to one end of resistor R14 and one end of R15 respectively, and Pin 8 of the photocoupler is connected to one end of resistor R11; The other end of resistor R14, the other end of resistor R11 and one end of capacitor C5 are connected and connected to V1; The other end of capacitor C5 is grounded; The other end of resistor R15 is connected to the input end of inverter U5, and the output end of inverter U5 is connected to one end of resistor R16, and the other end of resistor R16 is connected to one end of resistor R17 and one end of capacitor C8 respectively, and the other end of capacitor C8 is grounded; The other end of resistor R17 is connected to one end of resistor R18 and one end of capacitor C7 respectively, and the other end of resistor R18 is connected to the in-phase input end of operational amplifier U6B and one end of capacitor C9 respectively, and the other end of capacitor C9 is grounded; The inverting input end of operational amplifier U6B is connected to resistor R21 The negative power supply terminal of the operational amplifier U6A is grounded, and the positive power supply terminal of the operational amplifier U6A uses a low-power filter composed of a resistor R12 and a capacitor C6 to suppress the power supply noise; the inverting input terminal of the operational amplifier U6A is connected to one end of the resistor R20 and one end of the capacitor C10 respectively; the other end of the resistor R20 is connected to one end of the resistors R22, R23, and R24 respectively, and the other ends of the resistors R22, R23, and R24 are connected and grounded; the other end of the capacitor C10 is connected to the output terminal of the operational amplifier U6A; the output terminal of the operational amplifier U6A is connected to one end of the resistor R19, and the other end of the resistor R19 is connected to the gate of the metal-oxide semiconductor field effect transistor, the drain of the metal-oxide semiconductor field effect transistor is connected to the H-bridge circuit, and the source of the H-bridge circuit is connected to one end of the resistors R22, R23, and R24 respectively.

2. The electromagnetic flow converter according to claim 1, characterized in that: The operational amplifier U10A and the operational amplifier U10B are of the type TL072IDR or TLC2272AI.

3. The electromagnetic flow converter according to claim 1, characterized in that: The model of the inverter U5 is SN74AHC1G04; the photocoupler uses a high-speed photocoupler, model 6N137; the model of the operational amplifier U6B and the operational amplifier U6A is TLC27L2IDR; the metal-oxide semiconductor field effect transistor is a MOS tube, model ZVN2120.

4. The electromagnetic flow converter according to claim 1, characterized in that: The power supply voltage V2 of the inverter U5 is provided by a high-precision voltage reference chip.

5. The electromagnetic flow converter according to claim 1, characterized in that: The power supply voltage V2 of the inverter U5 is provided by a reference voltage generating circuit, which uses a reference voltage divider plus an op amp follower; A 1.25v or 2.5v voltage reference chip is used to obtain a reference voltage, and a reference voltage V2ref is obtained through resistor voltage division. After V2ref is followed by an operational amplifier, a power supply voltage V2 is obtained.

6. The electromagnetic flow converter according to claim 1, characterized in that: The communication interface circuit includes RS485 module, 4-20mA module, hart module, bluetooth module, wifi module and sub-g module.

7. The electromagnetic flow converter according to claim 1, characterized in that: The electromagnetic flow converter has the function of storing and calling multi-speed correction parameters, and automatically adjusts the controllable amplification circuit and controllable excitation current according to the current calculated flow rate; different amplification factors and different excitation currents form a variety of measurement combinations, and each combination sets a measurement range, so that there are overlapping measurement intervals between the combinations. When the calculated flow rate exceeds the measurement range of this gear, the gear is switched to avoid frequent gear switching; in order to improve the measurement accuracy, each combination has a calibrated instrument coefficient; in order to reduce the calibration complexity, the influence factors of the amplification factor and the excitation current are decomposed, and the influence factors of changing the amplification factor and the excitation current separately are calibrated, and then the instrument coefficients of all combinations are obtained.

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