A composite frequency modulation method for isolated AC-DC converter with zero vector

Abstract

The present invention proposes a composite frequency modulation method for an isolated AC-DC converter containing a zero vector. The present invention improves the DAB-ACDC converter topology, introduces an LC series resonant network, and draws on an extended phase-shift modulation scheme to design a high-low frequency PWM composite frequency modulation strategy containing a zero vector. While reducing switching losses, the synchronous rectification characteristics of MOSFET are used to reduce conduction losses and achieve natural commutation without any lossy buffer circuit or forced commutation strategy. Further, soft switching is achieved in combination with the LC resonance characteristics, reducing the harmonic content of the transformer. In addition, it also has an automatic power factor correction function, and the modulation parameters do not need to be calculated in real time, which effectively reduces the design difficulty of the closed-loop control scheme.

Description

Technical Field

[0001] The invention belongs to the technical field of converter frequency modulation, and in particular relates to a composite frequency modulation method of an isolated AC-DC converter containing a zero vector. Background Art

[0002] With the rapid development of distributed power generation and the rise of the electric vehicle industry, bidirectional AC-DC converters, as the core interface converters in interconnected microgrid systems and electric vehicle charging piles, have become the focus of current research. Depending on whether the AC and DC sides are isolated, bidirectional AC-DC converters can be divided into isolated and non-isolated converters; according to the number of power conversion stages, they can be divided into single-stage structures and two-stage structures.

[0003] The non-isolated bidirectional AC-DC converter has a limited transformation ratio and serious common-mode interference on the AC and DC sides. When used in high-power applications, a modular parallel structure is required, which can easily cause circulating current problems. If an industrial frequency transformer is used for isolation, it has disadvantages such as large weight and volume, high cost and noise, and large electromagnetic interference and energy loss. Therefore, high-frequency isolated bidirectional AC-DC converters have become the mainstream solution. Existing high-frequency isolation solutions mostly adopt a two-stage structure, including a front-stage AC-DC circuit to achieve power factor correction and provide a stable DC bus voltage, and a rear-stage DC-DC circuit to change the DC voltage gain and achieve electrical isolation, while further controlling the rectified DC voltage. The rear stage usually adopts a dual active full bridge (DAB) topology. The relevant research on DAB converters has been quite mature. The single phase shift modulation, extended phase shift modulation, double phase shift modulation and triple phase shift modulation schemes proposed based on the DAB topology have been widely used.

[0004] Although the DAB topology has advantages such as easy soft switching and high control freedom, the two-stage converter has low energy transmission efficiency and poor reliability. At the same time, a large number of sensors are required to achieve closed-loop control of voltage and current, and an independent control loop needs to be set for each stage. In order to solve the above problems, some scholars have proposed a single-stage isolated DAB-ACDC bidirectional converter with higher integration and power density based on the DAB topology. It has no DC link and can achieve single-stage energy conversion.

[0005] However, for this new type of DAB-ACDC converter, existing solutions often rely on a large number of iterative calculations to optimize control parameters. At the same time, variable frequency control is required during the modulation process to achieve important features such as power factor correction, soft switching, and stress optimization. This makes it difficult to match the converter filter parameters, and the controller computing power requirements are not low. In some solutions, the AC grid current even has obvious distortion. At the same time, they generally have a common problem: the transformer leakage inductance and filter inductance can be regarded as "current sources" on both sides of the cycloconverter. The cycloconverter working as a "current breaker" will generate voltage spikes caused by current cutoff. In this regard, related research at home and abroad usually adopts the solution of adding an absorption circuit, but this will produce undesirable harmonic distortion and additional losses when the system is in a steady state. All of the above problems are not conducive to the use of converters in engineering projects. Therefore, it is necessary to improve the existing modulation schemes. Summary of the invention

[0006] The purpose of the present invention is to solve the problems of the insufficiency of the existing topological structures and modulation schemes, and to provide a composite frequency modulation method for an isolated AC-DC converter containing a zero vector.

[0007] The present invention is implemented by the following technical scheme. The present invention proposes a single-stage isolated bidirectional AC-DC converter topology structure, the topology structure comprising: a first filtering unit, an AC full-bridge unit, a resonant converter, a DC full-bridge unit and a second filtering unit; the first filtering unit, the AC full-bridge unit, the resonant converter, the DC full-bridge unit and the second filtering unit are connected in sequence;

[0008] The resonant converter comprises a transformer and a resonant network, wherein the resonant network is an LC resonant network and is connected in series with the transformer.

[0009] Furthermore, the resonant network is equivalent to a power transmission inductor L eq The voltage between the midpoints of the AC and DC full bridge arms is expressed as v p1 (t) and v s (t), v p2 (t) is the voltage v calculated to the secondary side of the transformer p1 (t), equivalent inductance L eq The relationship with the resonant network is expressed by equation (1):

[0010]

[0011] In the formula, ω r =2πf r , f r is the resonant frequency, ω s =2πf s , f sis the switching frequency.

[0012] The present invention proposes a composite frequency modulation method for an isolated AC-DC converter containing a zero vector, wherein the modulation method utilizes the topological structure to realize modulation;

[0013] Define the switching function S ac (t) and S dc (t), acting on the AC full bridge and DC full bridge respectively, so that the voltage waveforms on both sides of the transformer and the resonant network are modulated into high-frequency quasi-square waves v p1 (t) and v s (t), and their pulse duty ratios are D ac (t) / 2 and D dc (t) / 2, that is:

[0014]

[0015] v p1 (t)(or v p2 (t)) and v s (t) The pulse center interval is the phase shift angle φ, whose magnitude and sign directly affect the transmission power; when φ>0, the converter works in the power forward transmission mode, that is, AC-DC mode, and vice versa when φ<0, it is DC-AC mode; the zero vector is compared with v ac (t) The compared power frequency square wave signal acts on the carrier u 1 On, realize in v ac (t) When in the positive half cycle, the Q of the AC full-bridge unit 11 ~Q 41 Always on, Q 12 ~Q 42 At the switching frequency f s For high frequency switching, v ac (t) The same applies when it is in the negative half cycle; the upper and lower switching devices of the same bridge arm of the DC full bridge are complementary turned on, i Leq is the current flowing through the equivalent inductor, and the turn-on and turn-off processes of the AC full-bridge switch tube are both ZCS.

[0016] Furthermore, the current relationship between the two sides of the AC / DC full bridge can also be expressed by a switching function as follows:

[0017]

[0018] Furthermore, the working state of the converter is controlled by the switching function, which is the mathematical expression of the converter when it operates according to the established modulation target. Its essence is still the result of the combined action of the input voltage and the phase shift angle φ. Therefore, at the converter working level, the circuit variables are used to replace the switching function. Therefore, under the high-frequency switching cycle time scale, equation (2) can be rewritten as equation (4):

[0019]

[0020] Using the fundamental component method, taking m = 1, the equivalent inductor current phasor expression (5) can be derived. Combining equations (4) and (5), according to equation (6), the complex power transmitted from the primary side to the secondary side of the converter during the high-frequency cycle can be obtained, where P T is the active power, Q T is the reactive power; T = 0, the unity power factor operation on the AC side can be guaranteed, that is, only active power is allowed to be transmitted from the grid side. At this time, the phase shift angle φ and the duty cycle must satisfy the constraint relationship (7);

[0021]

[0022]

[0023]

[0024] Under the condition that equation (7) holds true, by comparing equations (4) and (5), we can know that the resonant current i Leq With v p1 Keep the same frequency and phase, which means that all the switching devices of the AC full bridge can be turned on and off with zero current. ac is constant at 1, so we can get:

[0025]

[0026]

[0027] Where K = nV ac / V dc Defined as the voltage transfer ratio, its maximum value K max =nV L / V dc ; Formula (8) shows that D ac Follow v ac (t) changes, for a certain phase shift angle φ, D dc Can be obtained by v ac (t) The instantaneous value is uniquely determined; Formula (9) shows that within a high-frequency cycle, when the circuit parameters remain unchanged, the transmission power is only determined by the single variable of the phase shift angle φ, and the sign of the phase shift angle φ controls the direction of power flow; Note that ω L <<ω s , and then the time domain expression of the high-frequency resonant current can be derived:

[0028]

[0029] From formula (3), we can know that i iThe AC current after filtering is rectified by the resonant current. ac Can be obtained by n|i Leq |Averaging over the high-frequency switching cycle yields:

[0030]

[0031] The above formula shows that i ac With v ac In phase, automatic power factor correction is achieved.

[0032] Furthermore, since the entire modulation process is synchronized with the grid voltage, no additional phase-locked loop is required, and the corresponding average power is:

[0033]

[0034] Formula (8) itself contains the constraint condition of K / cosφ≤1. Substituting this condition into formula (12), the maximum transmission power of the converter can be obtained as:

[0035]

[0036] Same as the AC side, i o Also by i Leq After rectification by the switching function:

[0037]

[0038] Furthermore, during the freewheeling process, when the current flows through the anti-parallel diode, a trigger pulse is given to the corresponding MOSFET to achieve zero voltage turn-on of the switch device. Leq (t 0 )>0, Q 5 Always ZVS.

[0039] Furthermore, when the DC full bridge satisfies the minimum duty cycle constraint (15), Q 6 and Q 8 Capable of full range soft switching;

[0040]

[0041] Substituting equation (8) into equation (15), the minimum duty cycle condition for achieving soft switching can be expressed by inequality (16). If the power frequency cycle is taken as the reference, Q 6 and Q 8 The soft switching interval is normalized and M zvs If M zvs With K max The functional relationship can be expressed as formula (17):

[0042]

[0043]

[0044] In the formula,

[0045] Furthermore, for the control of the modulation variable phase shift angle, a voltage and current double closed-loop control strategy is adopted. The control strategy is specifically as follows: the DC current i dc It contains a DC component and a double frequency AC component, so when determining i dc The average value will inevitably bring a delay of half the power frequency cycle. In order to speed up the system response, the Smith predictor plus PI control method is used for the current inner loop to eliminate the time lag caused by the averaging process. po (s) Letter of communication to the actual accused, G q (s) is the undelayed state transfer function of the controlled object model.

[0046] The present invention improves the DAB-ACDC converter topology, introduces an LC series resonant network, and draws on the extended phase-shift modulation scheme to design a high-low frequency PWM composite frequency modulation strategy with a zero vector. While reducing the switching loss, the MOSFET synchronous rectification characteristics are used to reduce the conduction loss, achieve natural commutation, and do not require any lossy buffer circuit or forced commutation strategy. Further combined with the LC resonance characteristics, soft switching is achieved, reducing the harmonic content of the transformer. In addition, it also has an automatic power factor correction function, and the modulation parameters do not need to be calculated in real time, which effectively reduces the design difficulty of the closed-loop control scheme. BRIEF DESCRIPTION OF THE DRAWINGS

[0047] Figure 1 This is a topological diagram of a single-stage isolated bidirectional AC-DC converter.

[0048] Figure 2 Simplified equivalent circuit diagram for the converter.

[0049] Figure 3 It is the principle diagram of the implementation of the zero-vector PWM modulation strategy; (a) is the waveform of the key parameters of the circuit under the industrial frequency cycle and (b) is the waveform of the key parameters of the circuit under the switching cycle.

[0050] Figure 4 This is a schematic diagram of the soft switching range.

[0051] Figure 5 This is a schematic diagram of the soft switching ratio curve.

[0052] Figure 6 This is the block diagram of phase shift angle double closed-loop control.

[0053] Figure 7This is the steady-state experimental waveform of the forward working mode; (a) is Q 11 , Q 12 , Q 5 Drive signal and AC voltage v ac Waveform, (b) is the bridge arm midpoint voltage v p1 、v s And AC voltage and current v ac 、i ac Waveform.

[0054] Figure 8 The following are the waveform diagrams of the soft switching principle experiment; (a) is the waveform of the Main window, and (b) is the waveform of the Zoom1 and Zoom2 windows.

[0055] Fig. 9 Soft switching waveforms of DC side switch tubes Q5~Q8; (a) is the drive signal and drain-source voltage waveform of Q5 and Q7, and (b) is the drive signal and drain-source voltage waveform of Q6 and Q8.

[0056] Fig.10 This is the waveform diagram of the load power sudden increase and decrease experiment.

[0057] Fig.11 This is the experimental waveform of DC side current and resonant current.

[0058] Fig.12 This is the converter efficiency curve.

[0059] Fig.13 It is a block diagram of the high-low frequency PWM composite frequency modulation method containing a zero vector. DETAILED DESCRIPTION

[0060] The technical solutions in the embodiments of the present invention will be described clearly and completely below in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0061] The present invention proposes a high-low frequency PWM composite frequency modulation method containing a zero vector, which has an automatic power factor correction function and realizes single-stage AC / DC energy conversion. And this characteristic is realized by the modulation method itself, is not affected by the load characteristics, and does not rely on the control system. At the same time, single variable control reduces the performance requirements for the controller, does not require real-time calculation, and is conducive to engineering applications. And the converter soft switching range is expanded, and there is no need to make too many restrictions on the phase shift angle and duty cycle, thereby improving the converter power transmission capacity, reducing power reflux and device current stress, and improving the converter power density and efficiency. In addition, the AC side full-bridge switch loss is reduced by half, and half of the MOSFETs are kept in the on state during the freewheeling period, avoiding the commutation problem, and no clamping absorption circuit is required, so the system is simpler and more reliable.

[0062] Combination Figure 1-Figure 13 The present invention first improves the existing DAB-ACDC converter and introduces an LC series resonant network, which not only facilitates soft switching, but also filters high-order harmonics at the switching frequency level and reduces transformer harmonic losses. The improved topology is as follows Figure 1 shown.

[0063] The resonant network is equivalent to a power transmission inductor L eq , thus the simplified equivalent model of the converter is obtained as Figure 2 As shown. p1 (t) and v s (t) represents the voltage between the midpoints of the AC and DC full bridge arms, v p2 (t) is the voltage v calculated to the secondary side of the transformer p1 (t). Equivalent inductance L eq The relationship with the resonant network can be expressed by equation (1).

[0064]

[0065] In the formula, ω r =2πf r , f r is the resonant frequency, ω s =2πf s , f s is the switching frequency.

[0066] The implementation principle of the proposed zero-vector PWM modulation strategy can be obtained by Figure 3 Define the switch function S ac (t) and S dc (t), acting on the AC full bridge and DC full bridge respectively, so that the voltage waveforms on both sides of the transformer and the resonant network are modulated as Figure 3 (a) shows the high-frequency quasi-square wave v p1 (t) and v s(t), and their pulse duty ratios are D ac (t) / 2 and D dc (t) / 2. That is:

[0067]

[0068] Combination Figure 3 From the time scale in (b), we can see that the converter has 6 operating modes in one switching cycle, and the values ​​of the switching function in each mode are listed in Table 1. p1 (t)(or v p2 (t)) and v s (t) The pulse center interval is the phase shift angle φ, whose magnitude and sign directly affect the transmission power. When φ>0, the converter works in the power forward transmission mode, that is, AC-DC mode, and vice versa when φ<0, it is DC-AC mode. Figure 3 (a) and (b) respectively show the power frequency period T in AC-DC mode. L =1 / f L Time scale and switching period T s =1 / f s The waveform of the key parameters of the circuit in the time scale is similar to the DC-AC mode. The positive direction of the circuit variable is Figure 1 To avoid repetition, the theoretical analysis of the present invention is only carried out for the AC-DC mode.

[0069] Table 1 Switching function values ​​under different modes

[0070]

[0071] The modulation wave and carrier wave used to generate the switching signal are also Figure 3 As shown in (a), the zero vector is ac (t) The compared power frequency square wave signal acts on the carrier u 1 On v ac When (t) is in the positive half cycle, Q 11 ~Q 41 Always on, Q 12 ~Q 42 At the switching frequency f s For high frequency switching, v ac This high- and low-frequency combined modulation method can effectively reduce the switching loss of the circuit, and with the help of the synchronous rectification characteristics of MOSFET, it provides a path for the current during the primary side freewheeling period, eliminating the design of the clamping absorption circuit and achieving reduced conduction loss.

[0072] The upper and lower switching devices of the same bridge arm of the DC full bridge are complementary turned on, i Leq is the current flowing through the equivalent inductance, from Figure 3 It is not difficult to find in (b) that under the proposed zero vector modulation strategy, the turn-on and turn-off processes of the AC full-bridge switch are both ZCS.

[0073] The open-loop power factor correction characteristics of the modulation strategy proposed in the present invention will be verified below. The input and output voltage relationship of the two full bridges is expressed by equation (2). Similarly, the current relationship on both sides of the full bridge can also be expressed by the switching function as follows:

[0074]

[0075] The working state of the converter is controlled by the switching function, which is the mathematical expression of the converter when it operates according to the established modulation target. Its essence is still the result of the combined action of the input voltage and the phase shift angle φ. Therefore, at the converter working level, the circuit variables are used to replace the switching function. Therefore, under the high-frequency switching cycle time scale, equation (2) can be rewritten as equation (4):

[0076]

[0077] Using the fundamental component method, taking m = 1, the equivalent inductor current phasor expression (5) can be derived. Combining equations (4) and (5), the complex power transmitted from the primary side to the secondary side of the converter during the high-frequency cycle can be obtained according to equation (6). Where P T is the active power, Q T is the reactive power. T = 0, the unity power factor operation on the AC side can be guaranteed, that is, only active power is allowed to be transmitted from the grid side. At this time, the phase shift angle φ and the duty cycle must satisfy the constraint relationship (7).

[0078]

[0079]

[0080]

[0081] Under the condition that equation (7) holds true, by comparing equations (4) and (5), we can know that the resonant current i Leq With v p1 Keeping the same frequency and phase, this means that all switching devices in the AC full bridge can be turned on and off with zero current. Figure 3 It can be seen that D ac is constant at 1, so we can get:

[0082]

[0083]

[0084] Where K = nV ac / Vdc Defined as the voltage transfer ratio, its maximum value K max =nV L / V dc Formula (8) shows that D ac Follow v ac (t) changes, for a certain phase shift angle φ, D dc Can be obtained by v ac (t) The instantaneous value is uniquely determined; Formula (9) shows that within a high-frequency cycle, when the circuit parameters remain unchanged, the transmission power is only determined by the single variable of the phase shift angle φ, and the sign of the phase shift angle φ controls the direction of power flow. Note that ω L <<ω s , and then the time domain expression of the high-frequency resonant current can be derived:

[0085]

[0086] From formula (3), we can know that i i The AC current after filtering is rectified by the resonant current. ac Can be obtained by n|i Leq |Averaging over the high-frequency switching cycle yields:

[0087]

[0088] The above formula shows that i ac With v ac In phase, automatic power factor correction is achieved. In addition, since the entire modulation process is synchronized with the grid voltage, the present invention does not require an additional phase-locked loop. The corresponding average power is:

[0089]

[0090] It should be noted that equation (8) itself contains the constraint condition of K / cosφ≤1. Substituting this condition into equation (12), the maximum transmission power of the converter can be obtained as:

[0091]

[0092] Same as the AC side, i o Also by i Leq After rectification by the switching function:

[0093]

[0094] The soft switching performance of the proposed modulation strategy will be analyzed in detail below.

[0095] The modulation strategy proposed in the present invention can achieve ZCS of all MOSFETs on the AC side, regardless of the load. Since the resonant network is inductive, it provides conditions for soft switching of the DC full bridge. The implementation principle is as follows: During the freewheeling process, when the current flows through the anti-parallel diode, a trigger pulse is given to the corresponding MOSFET, and the zero voltage turn-on of the switch device can be achieved. For example, when i Leq (t 0 )>0, Q 5 In summary, when the DC full bridge meets the minimum duty cycle constraint (15), Q 6 and Q 8 Full range soft switching is also possible.

[0096]

[0097] From formula (8), we can see that D dc Following the grid voltage v ac (t), so Q will only appear when the K value is small within the power frequency cycle. 6 and Q 8 Hard switching. Substituting equation (8) into equation (15), the minimum duty cycle condition for achieving soft switching can be expressed by inequality (16). Figure 4 For further intuitive explanation, the green shaded area in the figure indicates that the minimum duty cycle constraint is met, which can ensure Q 6 and Q 8 If the power frequency cycle is used as the reference, Q 6 and Q 8 The soft switching interval is normalized and M zvs If M zvs With K max The functional relationship can be expressed as formula (17).

[0098]

[0099]

[0100] In the formula, according to Figure 5 It can be seen that Q 6 and Q 8 The soft switching ratio of K max The decrease of K and the increase of transmission power make it larger. At the maximum power, when K is taken max =0.2, Q 6 and Q 8 The soft switching accounts for up to 90%.

[0101] For the reverse working mode, the analysis is similar. Table 2 summarizes the soft switching of all switches in all working modes of the converter. It can be seen from the table that the method can achieve soft switching of all switches: 5 / 6 of the switches are global soft switches.

[0102] Table 2 Soft switching of the converter under the method described

[0103]

[0104] For the control of the modulation variable phase shift angle, a voltage and current double closed-loop control scheme is adopted. The block diagram is as follows: Figure 6 shown.

[0105] From formula (14), we can see that the DC current i dc It contains a DC component and a double frequency AC component, so when determining i dc The average value will inevitably bring a delay of half the power frequency cycle. In order to speed up the system response, the Smith predictor plus PI control scheme is used for the current inner loop to eliminate the time lag caused by the averaging process. po (s) Letter of communication to the actual accused, G q (s) is the undelayed state transfer function of the controlled object model.

[0106] In order to verify the correctness of the theoretical derivation of the present invention and the excellent performance of the proposed zero-vector modulation strategy, the proposed zero-vector PWM modulation strategy and phase-shift angle dual closed-loop control strategy were verified by steady-state, soft switching and dynamic experiments based on a rated power 450W bidirectional converter experimental platform. The hardware parameters are shown in Table 3.

[0107] Table 3 DAB-ACDC converter system parameters

[0108]

[0109] The steady-state operation results of the converter are as follows: Figure 7 shown. Figure 7 (a) and (b) show the switch drive signal, AC voltage and current, and AC / DC full-bridge midpoint voltage waveforms in the forward working mode of the converter. The controller sets a phase shift angle of φ = 1 rad, so that the converter transmits 300W of active power in the forward direction. Figure 7 (a) It can be seen that the AC side switch tube driving signal g 11 , g 12 Following AC voltage v ac The polarity changes the switching frequency or grid frequency; and the DC side switch tube drive signal g 5 Pulse center relative g 12 , g 12The pulse center is switched at high frequency with a constant phase shift angle of φ. The relative phase shift of the DC full-bridge pulse makes the transformer primary side voltage v p1 The pulse center always leads v s The pulse center ensures the unidirectional transmission of power, such as Figure 7 (b). And it can be seen that v s The pulse width follows v p1 pulse amplitude, which makes the AC voltage v ac and current i ac The same frequency and phase are completely maintained, that is, automatic power factor correction is achieved. The measured PF = 0.996. This characteristic is the result of the modulation scheme proposed by the present invention and is not affected by the load. In the experiment, when the load is adjusted to resistive inductive properties, Figure 7 The waveform has not changed.

[0110] In order to verify the advantages of soft switching described above, the converter DC-AC working mode is taken as an example to verify the soft switching implementation principle. Figure 8 The voltage and current waveforms of the transformer and resonant network are given. Figure 8 (b) shows the transformer primary voltage v p1 With the resonant current i Leq At the same time, Q 6 and Q 8 For global ZVS, Q 5 , Q 7 The ZVS operation must satisfy the minimum duty cycle condition (15). Figure 8 In (b), the Zoom1 and Zoom2 windows respectively show the ZVS operation experimental results of the DC full-bridge switch tube when the critical case just satisfies equation (15) and when equation (15) is fully satisfied.

[0111] Fig. 9 Given Figure 8 (b) The drive signal and drain-source voltage waveforms of the DC side switch tubes Q5~Q8 in Zoom2. It can be seen that the drive signal g x When the MOSFET threshold voltage is reached, its drain-source voltage v DSx It has basically dropped to 0V, where x=5, 6, 7, 8, that is, Q5~Q8 have all achieved ZVS.

[0112] In order to verify the limit performance and dynamic performance of the method, Fig.10 Taking the AC-DC operation mode of the converter as an example, the load is adjusted to make the output power of the converter change from Increased to Falling back to The experimental results show that the alternating current i acRapid response, as the load power increases and decreases, the input power increases and decreases accordingly, and the resonant current i Leq and the DC output current i o 、i dc It changes with the change and the response cycle is short. This shows that the proposed PI control plus Smith predictive control phase shift angle double closed-loop control system has good dynamic characteristics.

[0113] Fig.11 Verify the resonant current i Leq and the DC output current i o 、i dc The relationship between o and i Leq The envelopes of are coincident. Note that as the current value decreases, i o The waveform gradually becomes smaller than i Leq , the envelopes of the two deviate, this is because at this time Q 5 and Q 7 The on-time of the switch is getting shorter and shorter, and only a small part of i Leq The waveform is rectified to the DC side. The red waveform in the figure is i o The filtered DC current i dc , the current is the superposition of the double power frequency sinusoidal signal and the DC signal, which is completely consistent with equation (14). dc The second harmonic components in the circuit have no adverse effects in many AC-DC applications including communication power supplies, energy storage systems and electric vehicle chargers. o and i dc It can be seen from the waveform that the converter generates almost no reflux power.

[0114] The efficiency curve of the converter when operating at different transmission powers is as follows: Fig.12 As shown, at a rated transmission power of 450W, the efficiency of the converter in forward and reverse working modes reaches 97.1% and 96.7% respectively.

[0115] The block diagram of the high-low frequency PWM composite frequency modulation method with zero vector proposed by the present invention is as follows: Fig.13As shown. The present invention is mainly composed of AC full-bridge modulation, DC full-bridge modulation and control links. Based on the proposed AC / DC converter topology, this method invents a zero-vector PWM modulation strategy based on extended phase shifting, which reduces switching losses and has an open-loop power factor correction function. It also utilizes the synchronous rectification characteristics of MOSFET and does not require an additional buffer absorption circuit, effectively improving the efficiency of the converter. The AC full-bridge part can achieve zero-current switching of all switching devices in the full range, and the DC full-bridge part selects the voltage conversion ratio K value according to the transmission power size, which can increase the zero-voltage switching range as much as possible. The modulation scheme does not require real-time iterative calculation, has low current stress, a large transmission power range, and almost no power reflux, and the realization of soft switching will not react on the phase shift angle. The control link eliminates the half-power frequency cycle lag effect generated in the process of solving the DC average current, and improves the dynamic response speed of the control loop.

[0116] The above is a detailed introduction to the composite frequency modulation method of an isolated AC-DC converter containing a zero vector proposed in the present invention. Specific examples are used in this article to illustrate the principles and implementation methods of the present invention. The description of the above embodiments is only used to help understand the method of the present invention and its core idea. At the same time, for those skilled in the art, according to the idea of ​​the present invention, there will be changes in the specific implementation method and application scope. In summary, the content of this specification should not be understood as limiting the present invention.

Claims

1. A composite frequency modulation method for an isolated AC-DC converter containing a zero vector, characterized in that: The topological structure includes: a first filtering unit, an AC full-bridge unit, a resonant converter, a DC full-bridge unit, and a second filtering unit; the first filtering unit, the AC full-bridge unit, the resonant converter, the DC full-bridge unit, and the second filtering unit are connected in sequence; The resonant converter comprises a transformer and a resonant network, wherein the resonant network is an LC resonant network and is connected in series with the transformer; The current relationship on both sides of the AC / DC full bridge can also be expressed by the switching function: The input voltage and the phase shift angle φ act together on the high-frequency switching cycle time scale to obtain equation (4): Using the fundamental component method, taking m = 1, the equivalent inductor current phasor expression (5) can be derived. Combining equations (4) and (5), according to equation (6), the complex power transmitted from the primary side to the secondary side of the converter during the high-frequency cycle can be obtained, where P T is the active power, Q T is the reactive power; T = 0, the unity power factor operation on the AC side can be guaranteed, that is, only active power is allowed to be transmitted from the grid side. At this time, the phase shift angle φ and the duty cycle must satisfy the constraint relationship (7); Under the condition that equation (7) holds true, by comparing equations (4) and (5), we can know that the resonant current i Leq With v p1 Keep the same frequency and phase, which means that all the switching devices of the AC full bridge can be turned on and off with zero current. ac is constant at 1, so we can get: Where K = nV ac / V dc Defined as the voltage transfer ratio, its maximum value K max =nV L / V dc ; Formula (8) shows that D ac Follow v ac (t) changes, for a certain phase shift angle φ, D dc Can be obtained by v ac (t) The instantaneous value is uniquely determined; Formula (9) shows that within a high-frequency cycle, when the circuit parameters remain unchanged, the transmission power is only determined by the single variable of the phase shift angle φ, and the sign of the phase shift angle φ controls the direction of power flow; Note that ω L <<ω s , and then the time domain expression of the high-frequency resonant current can be derived: From formula (3), we can know that i i The AC current after filtering is rectified by the resonant current. ac Can be obtained by n|i Leq |Averaging over the high-frequency switching cycle yields: The above formula shows that i ac With v ac In phase, automatic power factor correction is achieved; The voltage waveforms on both sides of the transformer and the resonant network are modulated into high-frequency quasi-square waves v p1 (t) and v s (t), and their pulse duty ratios are D ac (t) / 2 and D dc (t) / 2.

2. The method according to claim 1, characterized in that The resonant network is equivalent to a power transmission inductor L eq The voltage between the midpoints of the AC and DC full bridge arms is expressed as v p1 (t) and v s (t), v p2 (t) is the voltage v calculated to the secondary side of the transformer p1 (t), equivalent inductance L eq The relationship with the resonant network is expressed by equation (1): In the formula, ω r =2πf r , f r is the resonant frequency, ω s =2πf s , f s is the switching frequency.

3. The method according to claim 1, characterized in that Since the entire modulation process is synchronized with the grid voltage, no additional phase-locked loop is required, and the corresponding average power is: Formula (8) itself contains the constraint condition of K / cosφ≤1. Substituting this condition into formula (12), the maximum transmission power of the converter can be obtained as: Same as the AC side, i o Also by i Leq After rectification by the switching function:

4. The method according to claim 3, characterized in that In the freewheeling process, when the current flows through the anti-parallel diode, a trigger pulse is given to the corresponding MOSFET to achieve zero voltage turn-on of the switch device. Leq When (t0)>0, Q5 is always ZVS.

5. The method according to claim 4, characterized in that When the DC full bridge meets the minimum duty cycle constraint (15), Q6 and Q8 in the DC full bridge can achieve full range soft switching; Substituting equation (8) into equation (15), the minimum duty cycle condition for achieving soft switching can be expressed by inequality (16). If the power frequency cycle is used as the benchmark, the soft switching intervals of Q6 and Q8 are normalized and expressed by M zvs If M zvs With K max The functional relationship can be expressed as formula (17): In the formula, 6. The method according to claim 5, characterized in that For the control of the modulation variable phase shift angle, a voltage and current double closed-loop control strategy is adopted. The control strategy is specifically as follows: the DC current i dc It contains a DC component and a double frequency AC component, so when determining i dc The average value will inevitably bring a delay of half the power frequency cycle. In order to speed up the system response, the Smith predictor plus PI control method is used in the current inner loop to eliminate the time lag caused by the averaging process.

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