Signal transmitting link and method, transceiving link, integrated circuit and electromagnetic wave device

Abstract

The invention discloses a signal transmitting link and method, a receiving and transmitting link, an integrated circuit and an electromagnetic wave device. The signal transmitting link comprises a signal transmitting main path and a signal calibration link which are integrated in the same integrated circuit; wherein the signal calibration link is configured to be used for calibrating the signal emission main path to obtain compensation information; and the signal emission main path is configured to perform compensation operation according to the compensation information and then generate a radio frequency emission signal so as to realize target detection and / or communication.

Description

[0001] This application claims priority to the Chinese patent application filed with the China Patent Office on June 14, 2023, with application number 202310702586.5 and invention name “Signal transmission, calibration, compensation and transceiver link, IQ mixer, integrated circuit, sensor and equipment”, the content of which should be understood as incorporated into this application by reference. Technical Field

[0002] The embodiments of the present disclosure relate to, but are not limited to, the technical field of electromagnetic wave devices, and in particular to a signal transmission link and method, a transceiver link, an integrated circuit, and an electromagnetic wave device. Background Art

[0003] In a frequency modulated continuous wave (FMCW) radar, the frequency difference between the transmitted signal and the echo signal can be used to calculate the distance of the target. The farther the target is, the greater the frequency of the corresponding intermediate frequency signal; and the closer the target is, the smaller the frequency of the corresponding intermediate frequency signal. After receiving multiple continuous pulse signals, the distance information and radial velocity information of the target can be obtained by performing Fourier transform in the fast time and slow time dimensions. In addition, the direction of arrival (DOA) of the echo signal reflected by the target can also be measured through multi-antenna technology.

[0004] In order to improve the parameter estimation performance (such as distance, radial velocity and DoA) of the FMCW radar system, the signal transmission link needs to be calibrated and compensated. For the calibration of the signal transmission link, additional equipment can be used to measure the performance of the RF device and make certain compensation. However, this calibration method cannot calibrate the RF device in real time because a large number of peripherals are required, and the RF device parameter indicators will change with changes in the environment. Summary of the invention

[0005] The following is a summary of the subject matter described in detail herein. This summary is not intended to limit the scope of the claims.

[0006] The signal transmission link and method, the transceiver link, the integrated circuit and the electromagnetic wave device of the embodiments of the present disclosure.

[0007] A signal transmission link comprises a signal transmission main path and a signal calibration link integrated in the same integrated circuit; wherein: the signal calibration link is configured to calibrate the signal transmission main path to obtain compensation information; and the signal transmission main path is configured to generate a radio frequency transmission signal after performing a compensation operation according to the compensation information, so as to achieve target detection and / or communication.

[0008] A signal transceiver link comprises the signal transmission link described above, and a signal receiving link; the signal receiving link comprises a receiving end local oscillator, a receiving end mixer, an analog-to-digital converter, and a receiving end baseband digital module; wherein the receiving end local oscillator is configured to provide a receiving end local oscillator signal; the receiving end mixer is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal; the echo signal is a signal formed by the signal transmitted by the signal transmission link being reflected and / or scattered by a target object; the analog-to-digital converter is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; the receiving end baseband digital module is configured to process the receiving end digital baseband signal to achieve target detection and / or wireless communication.

[0009] An integrated circuit comprises the signal transmission link described above, or the signal transceiver link described above.

[0010] An electromagnetic wave device comprises: a carrier; the integrated circuit mentioned above, which is arranged on the carrier; an antenna, which is arranged on the carrier, or the antenna and the integrated circuit are integrated into a single device and arranged on the carrier; the antenna comprises a transmitting antenna and a receiving antenna; wherein the integrated circuit is connected to the antenna for transmitting electromagnetic wave signals and / or receiving electromagnetic wave signals.

[0011] A user terminal device comprises: a device body; and the electromagnetic wave device described above arranged on the device body; wherein the electromagnetic wave device is used for target detection and / or wireless communication to provide reference information for the operation of the device body.

[0012] A signal transmission method, comprising: determining the phase of a radio frequency transmission signal of each signal transmission link; determining the initial phase of a baseband signal in each signal transmission link according to the phase of the radio frequency transmission signal; generating an initial baseband signal according to the determined initial phase; compensating the initial baseband signal using pre-acquired compensation information to obtain the baseband signal; and performing a phase shift operation on a transmitting end local oscillator signal based on the baseband signal to obtain the radio frequency transmission signal.

[0013] A non-transitory computer-readable storage medium stores computer-readable instructions. When the instructions are executed by a processor, the processor performs the signal transmission method described above.

[0014] The signal transmission link, transceiver link, integrated circuit, electromagnetic wave device and equipment of the embodiments of the present disclosure, since the signal calibration link is integrated into the integrated circuit including the main transmission path, the signal calibration link can perform calibration operations on the main transmission path in real time, and the calibration operation of the calibration link can be unaffected by changes in the operating environment of the main transmission path, so that the main transmission path can obtain more accurate calibration information, thereby improving the signal processing performance of the main transmission path.

[0015] Furthermore, the compensation information generated by the signal calibration link can solve the problems of orthogonal imbalance, LO leakage and harmonic distortion, thereby effectively improving the signal quality in the main transmission path of the digital phase shifter architecture.

[0016] Other aspects will become apparent upon reading and understanding the drawings and detailed description. BRIEF DESCRIPTION OF THE DRAWINGS

[0017] The accompanying drawings are used to provide a further understanding of the technical solution of the present disclosure and constitute a part of the specification. Together with the embodiments of the present disclosure, they are used to explain the technical solution of the present disclosure and do not constitute a limitation on the technical solution of the present disclosure. The shapes and sizes of the components in the accompanying drawings do not reflect the actual proportions and are only intended to illustrate the contents of the present disclosure.

[0018] Figure 1A A simplified schematic diagram of a signal transmission link simulating a phase shifter architecture;

[0019] Figure 1B for Figure 1A A simplified schematic diagram of an analog phase shifter in a signal transmission link is shown;

[0020] Figure 2 A schematic diagram of the structure of a signal transmission link provided for an exemplary embodiment of the present disclosure;

[0021] Figure 3 A waveform diagram of an FMCW transmission signal and an echo signal using sawtooth wave modulation;

[0022] Figure 4 A schematic diagram of the structure of another signal transmission link provided for an exemplary embodiment of the present disclosure;

[0023] Figure 5A Schematic diagram of the signal model of the transmitter's orthogonal modulation when the orthogonality is unbalanced;

[0024] Figure 5B Schematic diagram of the signals generated by a real mixer;

[0025] Figure 5C It is a schematic diagram of third-order harmonic distortion;

[0026] Fig. 6AA structural schematic diagram of a signal transmission link is provided for an embodiment of the present disclosure;

[0027] Figure 6B A schematic diagram of quadrature imbalance compensation provided by an embodiment of the present disclosure;

[0028] Figure 6C A first schematic diagram of harmonic distortion compensation provided by an embodiment of the present disclosure;

[0029] Fig.6D A second schematic diagram of harmonic distortion compensation provided by an embodiment of the present disclosure;

[0030] Fig. 7A A schematic diagram of a digital phase shifter architecture in a signal transmission link according to an embodiment of the present application;

[0031] Figure 7B is a schematic diagram of a signal transmission link including a compensation unit in an embodiment of the present disclosure;

[0032] Figure 8 A schematic diagram of the structure of a signal transceiver link provided for an exemplary embodiment of the present disclosure;

[0033] Fig. 9 A schematic diagram of the structure of another signal transceiver link provided for an exemplary embodiment of the present disclosure;

[0034] Fig.10 A schematic diagram of another transceiver link in an embodiment of the present application;

[0035] Fig.11 A schematic diagram of a transceiver link including TX IQ Mod, RX IQ De-Mod and LO Freq Diff in an embodiment of the present application;

[0036] Fig.12 In the embodiment of the present application, a Fig.11 The structure shown is a schematic diagram of the transceiver link combined with BIST;

[0037] Fig.13 It is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present application;

[0038] Fig.14 A schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application;

[0039] Fig.15 It is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application;

[0040] Fig.16 This is a schematic diagram of the structure of a mixer in an embodiment of the present application;

[0041] Fig.17 This is a schematic diagram of the structure of a compensation unit in a transmitter in an embodiment of the present application;

[0042] Fig.18 A schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module in an embodiment of the present application;

[0043] Fig.19 A schematic diagram of a digital pre-compensation HD3 architecture based on a frequency doubling waveform generator module in an embodiment of the present application;

[0044] Fig. 20 A schematic diagram of calibration compensation of a transmission link based on a digital phase shifter architecture in an embodiment of the present application;

[0045] Fig.21 A flowchart of a signal transmission method provided by an exemplary embodiment of the present disclosure. DETAILED DESCRIPTION

[0046] In order to make the purpose, technical solution and advantages of the present disclosure more clear, the embodiments of the present disclosure will be described in detail below with reference to the accompanying drawings. It should be noted that the embodiments and features in the embodiments of the present disclosure can be combined with each other arbitrarily without conflict.

[0047] Unless otherwise defined, the technical terms or scientific terms used in the embodiments of the present disclosure should be understood by people with ordinary skills in the field to which the present disclosure belongs. The "first", "second" and similar words used in the embodiments of the present disclosure do not indicate any order, quantity or importance, but are only used to distinguish different components. "Include" or "comprising" and similar words mean that the elements or objects before the word cover the elements or objects listed after the word and their equivalents, without excluding other elements or objects.

[0048] Radar is an electronic device that uses electromagnetic waves to detect targets. The radar chip transmits a beam through a signal transmission link. When the transmitted beam encounters an obstacle, the echo reflected by the obstacle is received by the receiving antenna and transmitted to the radar chip. The radar chip determines the position, distance, speed and other information of the target relative to the electromagnetic wave emission point. With the development of microelectronics and other technologies, radar has gradually been widely used, especially millimeter wave radar (such as automotive radar). Due to its small antenna size, it has been widely used in autonomous driving, smart home devices and industrial automation devices. At present, the miniaturization and integration of radar has become the current development trend.

[0049] Figure 1AThis is a simplified schematic diagram of the signal transmission link that simulates the phase shifter architecture. Figure 1B for Figure 1A The following is a simplified schematic diagram of an analog phase shifter in a signal transmission link. Figure 1A As shown, for a transmission link, when the sensor transmits a signal, a signal generator 11 such as a phase locked loop (PLL) generates a local oscillator (LO) signal (such as a frequency sweep signal in the 77 GHz band), which may be an FMCW signal; an analog phase shifter (Analog PS) 12 performs a phase shift operation on the received LO signal, and then radiates the received LO signal to a predetermined spatial region through a transmitting antenna 13, so as to perform operations such as target detection and measurement. Optionally, Figure 1A In the transmission chain structure shown in FIG. 1 , the corresponding analog phase shifter architecture can be shown as Figure 1B As shown, the specific phase shift principle can be shown as follows:

[0050]

[0051] in

[0052] That is, two signals (i.e., LOI and LOQ) are obtained by performing 0° and 90° phase shifts on the received LO signal (i.e., LOIN), and are fitted into an output signal LOOUT after being amplified by a power amplifier (PA). That is, the output signal LOOUT is phase shifted by θ compared to the LOIN input signal, wherein the value of the phase shift θ is determined by the amplitude A of the two signals LOI and the amplitude B of LOQ.

[0053] In addition, the above analog phase shifter architecture can also be implemented by means of a delay line unit, that is, by using the narrow-band assumption of the signal to perform phase shifting by means of time delay. The principle is shown in the following formula:

[0054]

[0055] Where τ is the delay time of the delay line.

[0056] Since the above-mentioned analog phase shifters have low phase modulation precision (Resolution) and phase modulation accuracy (Accuracy), they cannot meet the current sensor requirements. At the same time, although the phase modulation precision and phase modulation accuracy can be improved through calibration, the analog architecture phase shifter needs to be calibrated off-line, which will greatly increase the difficulty and complexity of engineering implementation and product mass production. At the same time, analog phase shifters also have serious problems such as large area, high loss, stability and channel coupling.

[0057] Since the signal transmission link adopts the analog phase shifter architecture, it has problems such as low phase modulation precision and accuracy. Therefore, it cannot meet the high performance requirements of the vehicle radar system.

[0058] like Figure 2 As shown, the embodiment of the present disclosure provides a signal transmission link, which is applied to a radar system. The signal transmission link includes: a baseband digital module 201, a digital-to-analog converter (Dto-Analog The digital-to-analog converter module 202 comprises two identical digital-to-analog converters, wherein: the transmitting baseband digital module 201 is configured to generate two orthogonal transmitting digital baseband signals, and send the two orthogonal transmitting digital baseband signals to a digital-to-analog converter module 202 respectively; the digital-to-analog converter module 202 is configured to convert the two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals; the transmitting local oscillator 203 is configured to provide a transmitting local oscillator signal TX_LO; the transmitting orthogonal modulator 204 is configured to perform frequency shifting and phase shifting operation on the transmitting local oscillator signal TX_LO based on the two transmitting analog baseband signals, so as to form a FMCW RF transmitting signal after a predetermined phase shift.

[0059] In the disclosed embodiment, the transmitting digital baseband signal provided by the transmitting baseband digital module 201 includes preset phase information; the digital-to-analog conversion module 202 performs digital-to-analog conversion on the received transmitting digital baseband signal to convert the transmitting digital baseband signal into a transmitting analog baseband signal (without changing the phase information); the transmitting orthogonal modulator 204 mixes the received transmitting analog baseband signal with the transmitting local oscillator signal TX_LO generated by the transmitting local oscillator 203 to achieve frequency shifting of the transmitting local oscillator signal based on the transmitting analog baseband signal while performing a preset phase shift operation to form a FMCW RF transmission signal after a predetermined phase shift.

[0060] The signal transmission link of the embodiment of the present disclosure forms a digital phase shifter architecture by using a transmitting baseband digital module 201, a digital-to-analog conversion module 202 and a transmitting orthogonal modulator 204. Since the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower side lobes, so its phase shift phase can be generated very accurately, making the phase modulation accuracy higher, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of small link loss, low cost, and no need for offline calibration, and can support more flexible wave transmission schemes, such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0061] In the disclosed embodiment, since the transmitting baseband digital module 201 provides a digital signal, in order to further adapt the signal characteristics, the transmitting modulator is set as an orthogonal modulator (IQ Modulator), and the digital-to-analog conversion module 202 is set as an orthogonal digital-to-analog converter (IQ DAC).

[0062] In the disclosed embodiment, the transmitting local oscillator 203 may include a phase-locked loop (PLL) structure, and may provide an electromagnetic wave (eg, laser, microwave, etc.) signal.

[0063] In some exemplary embodiments, the signal transmission link further includes a power amplifier (PA) 205, wherein the power amplifier 205 is configured to amplify the power of the phase-shifted RF signal and output the amplified signal to the transmitting antenna.

[0064] In some exemplary embodiments, the signal transmission link further includes: a transmitting antenna 206, wherein: the transmitting antenna 206 is configured to radiate the amplified signal to a preset spatial area.

[0065] In the disclosed embodiment, the signal amplified by the power amplifier 205 can be radiated to a preset spatial region through the packaged integrated or external transmitting antenna 206. That is, the transmitting local oscillator 203, the digital phase shifter and the transmitting antenna 206 can be integrated into a device or can be separate components; for example, the transmitting local oscillator 203 and the digital phase shifter can be integrated into a package to form a SoC chip, while the transmitting antenna 206 can be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the transmitting antenna 206 can also be integrated into the chip package to form AiP or AoP, with a chip structure with a packaged antenna.

[0066] In some exemplary embodiments, the frequency bandwidth of the swept frequency signal is above 2 GHz.

[0067] The electromagnetic wave of the transmission signal emitted by the transmitting antenna of the frequency modulated continuous wave radar system is a high-frequency frequency modulated continuous wave, and the echo signal received by the receiving antenna of the frequency modulated continuous wave radar system is the electromagnetic wave reflected / scattered back by the target object. Figure 3 FIG. 1 shows a waveform diagram of an exemplary FMCW transmission signal and an echo signal. Figure 3As shown, the frequencies of the transmitted signal and the echo signal change regularly with time. The frequency modulated continuous wave is generally sawtooth, triangle, etc. The present disclosure takes the sawtooth shape as an example for explanation. The electromagnetic wave in each frequency modulation period T is called a chirp. The frequency of each chirp signal increases linearly with time. In the embodiment of the present disclosure, the bandwidth range B of a chirp is greater than or equal to 2GHz.

[0068] In some exemplary embodiments, the transmitting end digital baseband signal is a single-tone signal, and the transmitting end local oscillator signal is a swept frequency signal.

[0069] In the disclosed embodiment, the transmitting end local oscillator 203 may be configured to provide an FMCW signal in a centimeter wave band or a millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves, and the transmitting end baseband digital module 201 may be configured to provide a single-tone transmitting end digital baseband signal at the MHz level (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz level single-tone transmitting end digital baseband signal to obtain a single-tone transmitting end analog baseband signal in the corresponding frequency range, and the transmitting end orthogonal modulator 204 may be configured to perform up-mixing or down-mixing operations on the received millimeter wave frequency band FMCW signal based on the received single-tone transmitting end analog baseband signal, so as to realize a preset phase shift operation on the FMCW signal.

[0070] Exemplarily, the FMCW signal in the 3.1 GHz frequency band may include a swept frequency signal between 3.1 GHz and 10.6 GHz, such as 7.163-8.812 GHz; the FMCW signal in the 77 GHz frequency band may include a swept frequency signal between 76 GHz and 81 GHz, or swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, or 79 GHz to 81 GHz.

[0071] In some other exemplary embodiments, the digital baseband signal at the transmitting end is a swept frequency signal, and the local oscillator signal at the transmitting end is a single tone signal.

[0072] In the disclosed embodiment, the transmitting end local oscillator 203 may be configured to provide a single-tone transmitting end local oscillator signal in a centimeter wave band or a millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves, and the transmitting end baseband digital module 201 may be configured to provide a MHz (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.) level transmitting end digital baseband FMCW signal, that is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz level transmitting end digital baseband FMCW signal to obtain a transmitting end analog baseband FMCW signal in a corresponding frequency range, and the transmitting end orthogonal modulator 204 may be configured to perform up-mixing or down-mixing operations on the received single-tone transmitting end local oscillator signal in the centimeter wave band or the millimeter wave band based on the received transmitting end analog baseband FMCW signal, so as to realize the preset phase shift and frequency sweep operations on the single-tone transmitting end local oscillator signal.

[0073] Exemplarily, the single-tone transmitting local oscillator signal in the 3.1 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 3.1 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc.; the single-tone transmitting local oscillator signal in the 77 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.

[0074] In some exemplary embodiments, the signal transmission link also includes: a low pass filter (LPF) 207, which is arranged between the digital-to-analog conversion module 202 and the transmitting end orthogonal modulator 204, and is configured to low pass filter the transmitting end analog baseband signal output by the digital-to-analog conversion module 202 and output it to the transmitting end orthogonal modulator 204.

[0075] like Figure 2 As shown, the transmitting end baseband digital module 201 generates two orthogonal digital baseband signals, namely, an I-channel digital baseband signal and a Q-channel digital baseband signal, and sends the generated digital baseband signals to the digital-to-analog conversion module 202 (including two identical DACs, namely, IQ DACs) to obtain two analog baseband signals. Then, the two analog baseband signals are input into the low-pass filter 207 to filter out the out-of-band noise signals, and are orthogonally modulated by the transmitting end orthogonal modulator 204 to obtain modulated RF signals, and then the modulated RF signals are radiated out through the power amplifier 205 and the transmitting antenna 206.

[0076] In some exemplary embodiments, the signal transmission link may further include a direct digital frequency synthesizer (DDFS) ( Figure 2The direct digital frequency synthesizer is not shown in the figure and is arranged between the transmitting baseband digital module 201 and the digital-to-analog conversion module 202. The direct digital frequency synthesizer can be configured to implement at least one of a variety of signal waveforms and transmission modes such as CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), Digital IF (Digital Intermediate Frequency) based on the received source signal to achieve flexible configuration of signal transmission form and transmission waveform.

[0077] like Figure 4 As shown, the embodiment of the present disclosure further provides a signal transmission link, which is applied to an electromagnetic wave transmitting device. The signal transmission link includes a first signal source 41 and a digital phase shift module 42; wherein,

[0078] The first signal source 41 is configured to generate a first analog signal; and

[0079] The digital phase shift module 42 is configured to perform frequency shift and / or phase shift on the first analog signal by using a digital quadrature modulation method to form an FMCW radio frequency transmission signal.

[0080] The signal transmission link provided by the embodiment of the present disclosure includes a first signal source 41 and a digital phase shift module 42. The first signal source 41 can be configured to provide a first analog signal, and the digital phase shift module 42 can be configured to generate a phase shift signal in the digital domain. The digital phase shift module 42 can also phase-shift the first analog signal based on the generated phase shift signal to perform a preset phase shift operation on the first analog signal.

[0081] In some exemplary embodiments, the first signal source 41 may be a transmitting local oscillator, and the first analog signal may be a transmitting LO signal.

[0082] In some exemplary embodiments, the signal transmission chain further includes a power amplifier (not shown in the figure) configured to amplify the FMCW RF transmission signal.

[0083] In some exemplary embodiments, the signal transmission link further includes a transmission antenna 43 configured to radiate the power-amplified FMCW RF transmission signal to a predetermined area.

[0084] like Figure 4As shown, in some optional embodiments, the signal transmission link may include a first signal source 41, a digital phase shift module (Digital PS) 42 and a transmitting antenna 43, that is, the first signal source 41 may be configured to provide an LO signal, and the digital phase shift module 42 may be configured to perform a preset phase shift operation on the received LO signal, so that the phase-shifted LO signal is radiated to a preset spatial area through the transmitting antenna 43. Among them, the first signal source 41 may also be an architecture including a phase-locked loop PLL, which can provide electromagnetic wave (such as laser, microwave, etc.) signals. Among them, the first signal source 41, the digital phase shift module 42 and the transmitting antenna 43 may be integrated into an integrated device, or may be separate components; for example, the first signal source 41 and the digital phase shift module 42 may be integrated into a package to form a SoC chip, and the transmitting antenna 43 may be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the transmitting antenna 43 may also be integrated on the chip package to form AiP or AoP, with a chip structure of a packaged antenna.

[0085] In some exemplary embodiments, the digital phase shift module 42 includes a second signal source 423, a digital-to-analog conversion module 422, and a mixer 421 connected in sequence; wherein,

[0086] The second signal source 423 is configured to generate a first digital signal;

[0087] The digital-to-analog conversion module 422 is configured to convert the first digital signal into a second analog signal;

[0088] The mixer 421 is configured to perform frequency shift and / or phase shift on the first analog signal based on the second analog signal to form an FMCW RF transmission signal.

[0089] like Figure 4As shown, the digital phase shift module 42 in the embodiment of the present disclosure may include a mixer (Mixer) 421, a digital-to-analog conversion module (i.e., DAC) 422, and a second signal source (e.g., a digital baseband signal source Baseband) 423, etc., that is, the second signal source 423 may be configured to provide a first digital signal; the digital-to-analog conversion module 422 may be configured to perform digital-to-analog conversion on the received first digital signal to convert the first digital signal into a second analog signal; the mixer 421 may be configured to perform a mixing operation on the received second analog signal with the received first analog signal from the first signal source 41, so as to achieve a phase shift operation of setting the above-mentioned first analog signal using the first digital signal. Optionally, when the above-mentioned signal transmission link provides a swept frequency signal, such as providing an FMCW laser signal or an FMCW microwave signal, a swept frequency transmission signal may be provided based on the first signal source 41, and / or a swept frequency first digital signal may be provided based on the second signal source 423, so that after mixing by the mixer 421, a swept frequency continuous wave signal is output.

[0090] In some optional embodiments, based on Figure 4 In the structure shown, the first signal source 41 can be configured to provide an FMCW signal (i.e., a first analog signal) in a centimeter wave band or a millimeter wave band (e.g., 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves, and the second signal source 423 can be configured to provide a first digital signal at a MHz level (e.g., 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog conversion module 422 performs digital-to-analog conversion on the first digital signal at the MHz level to obtain a second analog signal in a corresponding frequency range, and the mixer 421 can be configured to perform up-mixing or down-mixing operations on the received millimeter wave band FMCW signal based on the received second analog signal of a fixed frequency band, so as to realize a preset phase shift operation on the FMCW signal.

[0091] In some optional embodiments, the centimeter wave signal in the 3.1 GHz frequency band may include 3.1 GHz to 10.6 GHz, for example, 3.1 GHz, 5 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc.; the millimeter wave signal in the 77 GHz frequency band may include 76 GHz to 81 GHz signals, for example, swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, 79 GHz to 81 GHz, or fixed frequency band signals such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.

[0092] In some exemplary embodiments, the first digital signal includes two orthogonal transmitting digital baseband signals;

[0093] The second signal source 423 is a baseband digital module at the transmitting end, and the digital-to-analog conversion module 422 includes two identical digital-to-analog converters;

[0094] The transmitting baseband digital module is configured to generate two orthogonal transmitting digital baseband signals and send the two orthogonal transmitting digital baseband signals to a digital-to-analog converter respectively;

[0095] The digital-to-analog conversion module 422 is configured to convert two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals.

[0096] based on Figure 4 In the structure shown, since the second signal source 423 provides the first digital signal, in order to further adapt the signal characteristics, the mixer 421 can be set to an IQ Mixer, and the digital-to-analog conversion module 422 is an IQ DAC. At the same time, the second signal source 423 can be configured to provide a digital baseband signal source (DDFS) for phase shifting and / or provide a corresponding source signal as a waveform controller (Waveform Control). Among them, DDFS is a phase-adjustable digital baseband signal source that generates a digital baseband signal.

[0097] The following is an analysis and explanation of the problems existing in the transmitter of the digital phase shifter:

[0098] Figure 5A Schematic diagram of the signal model of the transmitter's orthogonal modulation when the orthogonality is unbalanced. Figure 5A As shown, DSP is a digital signal processor, DAC is a digital-to-analog converter, LPF is a low-pass filter, and LO is a local oscillator signal source. I (f) is the frequency domain expression of the LPF output signal of channel I, E I (f) is the baseband frequency response of channel I, A′ I (f) is the frequency domain expression of the baseband output signal of channel I; S I (f) is the frequency domain expression of the output signal of the I channel modulator, G I (f) is the RF frequency response of channel I, S′ I (f) is the frequency domain expression of the RF output signal of channel I; where α I is the amplitude gain of the I channel mixer, is the phase response of the I channel mixer; A Q (f) is the frequency domain expression of the Q channel LPF output signal, E Q (f) is the Q channel baseband frequency response, A′ Q (f) is the frequency domain expression of the Q channel baseband output signal; S Q (f) is the frequency domain expression of the Q channel modulator output signal, G Q(f) is the Q channel RF frequency response, S′ Q (f) is the frequency domain expression of the Q channel RF output signal; where α Q is the amplitude gain of the Q channel mixer, is the phase response of the Q channel mixer; RF (f) is the frequency domain expression of the transmitted modulated signal. In the disclosed embodiment, the radio frequency includes all circuits from the mixer to the radio frequency signal, and the baseband includes all circuits from the DAC to the mixer.

[0099] according to Figure 5A The signal model shown, the output of the orthogonal demodulator U RF (f) can be expressed as

[0100]

[0101] Among them, A(ff m )V1(ff m ) is the frequency f m nearby desired signals;

[0102] For the frequency -f m nearby desired signals;

[0103] A * (-f+f m )V2(ff m ) is the frequency f m nearby image signals;

[0104] For the frequency -f m nearby image signals;

[0105]

[0106] use Indicates U RF (f) The complex baseband form of the signal, i.e.

[0107]

[0108] Among them, when f<0, u(f)=0; when f≥0, u(f)=1.

[0109] Therefore, the baseband equivalent model of the quadrature regulator is

[0110]

[0111] Where A(f)V1(f) is the frequency f m Nearby desired signal; A *(-f)V2(f) is the frequency at -f m Nearby image signals

[0112] The relative image ratio (RIR) is used to measure the ability of a quadrature mixer to suppress unwanted image frequency components. Therefore, the RIR is defined as the power ratio of the desired signal and the image signal components, which can be written as

[0113]

[0114] The above equations show that the cascaded gain and phase response of the I and Q channels need to be equal (the cascaded gain and phase response of each channel should be considered), otherwise the channels are unbalanced and the image components are not completely suppressed.

[0115] Figure 5B This is a schematic diagram of the signals generated by a real mixer. Figure 5B As shown, FIN is the input signal of the mixer, FOUT is the output signal of the mixer, and FLO is the local oscillator signal used by the mixer, wherein FIN is the baseband signal with a frequency of, FLO is the high-frequency signal with a frequency of FLO, and FLO is the signal with a frequency of FLO±FBB.

[0116] An ideal mixer will produce an output that is the product of its two inputs. In frequency, the outputs should be FLO+FBB and FLO+FBB, with nothing else. If either input is not driven, there will be no output. However, a real mixer will also produce some energy at FBB and FLO. The energy produced at FBB can be ignored because it is far from the desired output and will be filtered out by the RF devices after the mixer output. Regardless of the energy produced at FBB, the energy produced at FLO can be a problem. It is very close to or within the desired output signal and is difficult or impossible to remove by filtering, which will also filter out the desired signal. There are other ways that the LO can leak to the system output, such as through the power supply or across the silicon itself. Regardless of how the local oscillator leaks, its leakage can be referred to as LO Leakage (LOL).

[0117] Although analog (RF) circuits can be approximated by linear models with small signal representation, the nonlinearity of the device often produces interesting and important phenomena that cannot be predicted. The input and output characteristics of nonlinear memoryless systems can be approximated as:

[0118] y(t)=α1x(t)+α2x 2 (t)+α3x 3 (t)+…≈α1x(t)+α2x 2 (t)+α3x 3 (t)

[0119] Many of the nonlinear effects of these devices are caused by third-order terms. And xq(t)=Asin(ωt+φ), then

[0120]

[0121] in:

[0122]

[0123] If the system has odd symmetry, that is, the system is completely differential, then the even harmonic terms can cancel each other out, and what remains are the odd harmonic terms, which is also called harmonic distortion.

[0124] Figure 5C This is a schematic diagram of third-order harmonic distortion. Figure 5C As shown, the baseband signal generated by the baseband processor is a baseband signal with a frequency of BB. After being processed by the IQ modulator, a desired signal Wanted, an image signal IMG and a third-order harmonic HD3 can be obtained. The signal output by the power amplifier PA includes the desired signal Wanted, the image signal IMG and the third-order harmonic HD3.

[0125] The main source of HD3 is the third-order nonlinearity of the baseband analog device. As can be seen from the above derivation, the third-order harmonic component HD3 of the baseband BB signal is located at -3BB. After passing through the mixer, the baseband HD3 is up-converted to LO-3BB, that is, C-IM3. After being amplified by the PA, the signal generated by the modulation is difficult to distinguish at high frequencies.

[0126] Based on the above description, it can be known that for the digital orthogonal modulation signal transmission link, it is necessary to face problems such as IQImbalance, harmonic distortion, and LO Leakage. After the ideal single-tone baseband signal passes through the orthogonal up-mixer, multiple non-ideal single-tone signals with different frequencies will be generated at high frequencies. Therefore, from the perspective of system architecture, a digital pre-compensation module is added to the transmitter to solve the problems introduced by the non-ideal characteristics of analog devices in the digital orthogonal modulation radar TX. When designing the system architecture, it not only includes a pre-compensation module, but also a calibration loop to achieve real-time calibration and compensation.

[0127] Fig. 6A The present invention provides a schematic diagram of the structure of a signal transmission link. Fig. 6AAs shown, the signal transmission link includes a signal transmission main path and a signal calibration link integrated in the same integrated circuit; wherein: the signal calibration link is configured to calibrate the signal transmission main path to obtain compensation information; and the signal transmission main path is configured to generate a radio frequency transmission signal after performing a compensation operation according to the compensation information, so as to achieve target detection and / or communication.

[0128] Since the signal calibration link is integrated into the integrated circuit including the main transmission path, the signal calibration link can perform calibration operations on the main transmission path in real time, and the calibration operation of the calibration link can be unaffected by changes in the operating environment of the main transmission path, so that the main transmission path can obtain more accurate calibration information, thereby improving the signal processing performance of the main transmission path.

[0129] Furthermore, the compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter. Since the compensation information generated by the signal calibration link can solve the orthogonal imbalance, LO leakage, and harmonic distortion problems, the signal quality in the main transmission path of the digital phase shifter architecture is effectively improved.

[0130] In one implementation, the integrated circuit is a millimeter wave radar chip, and / or the radio frequency transmission signal is a FMCW signal.

[0131] In another implementation, the integrated circuit is an ultra wide band (UWB) chip.

[0132] The main signal transmission path includes a first signal source and a phase shifter; wherein the first signal source is configured to generate a first analog signal; and the phase shifter is configured to perform frequency shifting and / or phase shifting on the first analog signal to form a radio frequency transmission signal.

[0133] When the phase shifter is a non-orthogonal architecture, the phase shifter includes a second signal source and a transmitting mixer, wherein the second signal source is configured to generate a second analog signal, and the transmitting mixer is configured to perform mixing processing on the first analog signal and the second analog signal to form the radio frequency transmission signal;

[0134] When the phase shifter is an orthogonal architecture, the phase shifter includes: a second signal source, a digital-to-analog conversion module and a transmitting-end mixer; wherein the second signal source is configured to generate a first digital signal; the digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and the transmitting-end mixer is configured to perform frequency shifting and / or phase shifting on the first analog signal based on the second analog signal to form the RF transmission signal.

[0135] The main transmitting path also includes a compensation circuit, wherein the signal input end of the compensation circuit is connected to the second signal source, and the signal input end is connected to the phase shifter, and the compensation circuit is used to combine the compensation signal and the signal output by the second signal source before outputting it.

[0136] When the signal output by the signal transmission main path is two-way orthogonal signals, the compensation signal used by the compensation circuit is two-way orthogonal signals;

[0137] When the signal output by the signal transmission main path is not two orthogonal signals, the compensation signal used by the compensation circuit is a signal of the same type as the signal output by the second signal source, wherein the signal type is a digital signal or an analog signal.

[0138] Wherein, the compensation circuit includes a compensation signal generator and an adder, wherein:

[0139] The compensation signal generator is configured to generate the compensation signal;

[0140] The adder is connected to the compensation signal generator and the second signal source, and is used to perform a signal superposition operation on the signal output by the second signal source and the compensation signal output by the compensation signal generator.

[0141] Specifically, when the signal output by the signal transmission main path is two orthogonal signals, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal imbalance compensation signal unit and a local oscillator leakage compensation signal unit;

[0142] When the signal output by the signal transmission main path is not a two-path orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leakage compensation signal unit;

[0143] Among them, the compensation signal generated by the harmonic compensation signal unit is used to eliminate the harmonic signal of the main frequency signal in the main signal transmission path; the compensation signal of the local oscillator leakage compensation signal unit is used to compensate for the leakage signal generated by the transmitting end local oscillator signal in the main signal transmission path; the compensation signal generated by the orthogonal imbalance compensation signal unit is used to compensate for the image signal of the main frequency signal in the main signal transmission path.

[0144] The compensation signal used by the orthogonal imbalance compensation circuit has the same frequency, the same amplitude, and the opposite phase between the image signal corresponding to the signal transmission main path and the image signal corresponding to the desired signal generated by the signal transmission main path.

[0145] In an exemplary embodiment, the compensation signal used by the quadrature imbalance compensation circuit is determined according to a signal output by a second signal source and a frequency-inverted complex conjugate signal of the signal output by the second signal source.

[0146] Among them, the method for obtaining the compensation signal generated by the orthogonal imbalance compensation circuit includes: obtaining the product of a preset pre-compensation coefficient and the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal; calculating the difference between the signal output by the second signal source and the adjustment signal to obtain the compensation signal generated by the orthogonal imbalance compensation circuit.

[0147] The pre-compensation coefficient is determined based on a ratio of an amplitude of a desired signal to an amplitude of an image signal corresponding to the desired signal.

[0148] Figure 6B Schematic diagram of quadrature imbalance compensation provided by an embodiment of the present disclosure. Figure 6B As shown, in order to achieve compensation of the TX orthogonal modulator, that is, to suppress the existence of the unwanted image signal, a pre-compensation method can be used. Let hIQ be the pre-compensation coefficient, which acts on the complex conjugate signal A*(-f) of the baseband signal to be transmitted. The frequency domain expression of the pre-compensated baseband signal can be written as

[0149] A pc (f) = A(f) - h IQ ·A * (-f)→a pc (n) = a(n) - h IQ ·a * (n)

[0150] When this pre-compensated signal is now used as the input to the quadrature modulator, the frequency domain form of the modulator output can be expressed as:

[0151]

[0152] From the above formula, we can see that we only need to ensure that V2(-f t )-h IQ V1(f t )=0, the image component can be suppressed by pre-compensation. Therefore, the pre-compensation coefficient can be expressed as

[0153]

[0154] Where the precompensation coefficients represent the frequency response of the optimal precompensation filter, which will be used in the digital domain of the modulator to predistort (precompensate) the I and Q signals before transmission. t ) and V1(f t ) is the ratio between them.

[0155] In an exemplary embodiment, the compensation signal generated by the local oscillator leakage compensation signal unit is generated based on a leakage signal corresponding to the first analog signal used by the phase shifter.

[0156] The compensation signal generated by the local oscillator leakage compensation signal unit has the same frequency and amplitude as the leakage signal but an opposite phase.

[0157] Specifically, LO Leakage Correction (LOC) is achieved by generating a signal with equal amplitude to LOL but opposite phase to cancel it. After obtaining the exact amplitude and phase of LOL, the cancellation signal can be generated by applying a DC offset to the input of the transmitter. The orthogonal mixer structure enables it to generate cancellation signals well. Since there are orthogonal signals of LO in the mixer, signals of any phase and amplitude can be generated at the frequency of LO. It can be interpreted geometrically in the complex plane. The first line is the result of the synthesis of IQ two-way time domain, and the second line is a geometric interpretation from the perspective of the IQ two-way signal spectrum. After applying a DC bias to the transmit signal, the output of the mixer will contain the desired transmit signal as well as the desired LOL cancellation signal. The deliberately generated cancellation signal LOC will cancel the useless LOL, leaving only the desired transmit signal. Ideally,

[0158] T× LOC =h DC,I ×LO I -h DC,Q ×LO Q =-T× LOL .

[0159] Therefore, the compensation method of LO leakage can be expressed as: pc (n) = a(n) - h DC .

[0160] In practice, LOL can be modeled as follows:

[0161] The ideal IQ is α I cos(w m t) and α Q sin(w m t);

[0162] The actual IQ is α I cos(w m t+φ I ) and α Q sin(w m t+φ Q );

[0163] LO leakage is

[0164] Therefore, LO Leakage needs to be eliminated in the following ways:

[0165]

[0166]

[0167] Therefore, LOL can be calibrated by adding different DC offsets to the I and Q signals.

[0168] In an exemplary embodiment, the compensation signal generated by the harmonic compensation signal unit has the same frequency and amplitude as the harmonic signal in the signal transmission main path, and has an opposite phase.

[0169] Specifically, the harmonic compensation signal unit includes an n-th power module or an n-fold frequency signal generator; wherein:

[0170] An n-th power module, wherein the signal input end of the n-th power module is connected to the signal output end of the second signal source, and is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source using the signal output by the second signal source, to obtain the compensation signal;

[0171] The n-fold frequency signal generator is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source to obtain the compensation signal, wherein the value of n is a positive integer.

[0172] Here, the value of n is an odd number. Furthermore, the value of n is 3.

[0173] Figure 6C and Fig.6D Schematic diagram of harmonic distortion compensation provided by the embodiment of the present disclosure. Figure 6C and Fig.6D As shown, similar to the principle of eliminating LOL, there are two compensation methods for third-order harmonic distortion, namely, a compensation architecture based on an n-th power module and a compensation architecture based on an n-fold frequency waveform generator.

[0174] a pc (n) = a(n) - h HD3 ·(a * (n)) 3

[0175] a pc (n) = a(n) - h HD3 ·a3(n)

[0176] like Fig. 7AAs shown, the signal transmission link (TX digital phase shifter architecture) of the digital phase shifter architecture may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (Direct Digital Frequency Synthesizer, DDFS for short), an IQ digital to analog converter (Digital to Analog Convertor, DAC for short), a low-pass filter (Low-Pass Filter, LPF for short), an IQ modulator (IQ modulator / IQ Mixer), a power amplifier (Power Amplifier, PA), etc., that is, the baseband signal source is configured to provide a digital phase-shifted source signal (that is, the aforementioned first digital signal), and the direct digital frequency synthesizer may be configured to implement CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), Digital IF (Digital Intermediate Multiplexing) based on the received source signal. Frequency, digital intermediate frequency) and other signal waveforms and wave transmission methods to achieve flexible configuration of signal transmission form and transmission waveform. Among them, the signal amplified by the power amplifier can be radiated to a preset space area through a transmitting antenna packaged as an integral unit or externally.

[0177] With respect to the signal transmission link of the digital phase shifter architecture in the embodiment of the present disclosure, since the digital phase shifter architecture is configured to generate a baseband signal sequence in the digital domain, and can generate an analog baseband signal (i.e., a second analog signal) through a DAC, and then modulate the transmission signal to a high frequency through an orthogonal mixer, that is, because the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower side lobes, and therefore its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy.

[0178] In some optional embodiments, for the signal transmission link of the digital phase shifter architecture, when the RF LO frequency sweep is used to realize the transmitted signal as an FMCW signal, a compensation unit may be added to the signal transmission link to solve the problems of TX IQ imbalance (Imbalance), signal leakage (such as TX LO Leakage) and harmonic distortion (HD) caused by IQ mismatch (Mismatch). Fig. 7A As shown, a TX compensation unit can be set between the TX DDFS and the IQ DAC to perform calibration and compensation operations on the signal transmission link of the digital phase shifter architecture, so as to achieve an operation to solve at least one of the above problems. Among them, the HD caused by the third-order nonlinearity of the baseband can be referred to as HD3.

[0179] In some optional embodiments, such as Figure 7B As shown, the TX compensation unit (TX compensation) may include at least one of a TXLO leakage compensation unit (TX LO leakage compensation), a TX IQ imbalance compensation unit (TX IQ Imbalance compensation) and a TX HD3 compensation unit (TX HD3 compensation), wherein the TX LO leakage compensation unit may be configured to perform compensation operations for signal leakage, the TX IQ imbalance compensation unit may be configured to perform compensation operations for IQ imbalance, and the TX HD3 compensation unit may be configured to perform compensation operations for the above-mentioned HD3. The TXIQ imbalance compensation unit is configured to be used to perform compensation operations for at least one of an IQ modulator imbalance (TX IQ Modulator imbalance) and an IQ channel imbalance (IQ channel imbalance). In addition, when the compensation unit includes at least two of the TX LO leakage compensation unit, the TX IQ imbalance compensation unit and the TX HD3 compensation unit, compensation may be performed synchronously (such as in parallel) or sequentially (such as in series) according to actual needs and signal characteristics. Figure 7B As shown, LO leakage compensation can be performed first, then IQ compensation, and finally HD3 compensation.

[0180] In some optional embodiments, for the signal transmission link of the digital phase shifter architecture, an error correction module (TX DAC Board Error Correction) for DAC and an AWGN (additive white gaussian noise) module for Gaussian white noise may also be included, which are not shown in the figure and may be added or deleted according to actual needs. Among them, I in IQ mentioned in the embodiments of the present disclosure can be expressed as the abbreviation of In-Phase (i.e., in-phase), Q can be expressed as the abbreviation of Quadrature (i.e., orthogonal), and RF can be expressed as the abbreviation of Radio Frequency (i.e., radio frequency).

[0181] In an optional embodiment, the compensation operation for IQ Imbalance can be achieved by compensating the conjugate signal of the BB (baseband) signal to reversely cancel the image component, and this compensation operation method is not affected by the calibration method of IQ Imbalance. The compensation for LO Leakage can be achieved by adjusting the DC component (i.e., DC bias) of the two IQ paths, and the calibration method of LO Leakage has no effect on its compensation scheme. For HD3, since the third-order harmonic distortion of the orthogonal mixer V / IConverter is the main source of HD3, and the harmonic distortion will be affected by the DC bias, when both the LO Leakage and HD3 of the transmission link need to be calibrated, it is necessary to calibrate HD3 after calibrating LO Leakage to ensure the accurate performance of HD3 calibration.

[0182] In addition, the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency doubling waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process. Specifically:

[0183] In an optional embodiment, for a digital pre-compensation architecture based on a digital cubic module, LO Leakage may be calibrated and compensated first, and then, under a stable DC bias, when both LO Leakage and HD3 of the transmission link need to be calibrated, it is necessary to calibrate HD3 after calibrating LO Leakage to ensure accurate performance of HD3 calibration; when LO Leakage, IQ imbalance and HD3 of the transmission link need to be calibrated, IQ imbalance calibration is performed after calibrating LO Leakage, and finally HD3 calibration is performed to ensure accurate performance.

[0184] In an optional embodiment, for a digital pre-compensation architecture based on a frequency doubling waveform generator module, after calibrating and compensating for LO Leakage, the compensation coefficient of HD3 can be calibrated, and IQImbalance can be calibrated and IQ Imbalance can be compensated under a stable DC bias; subsequently, the actual waveforms of the IQ and Q signals and the compensation coefficient of HD3 are calculated respectively through the compensation results, and the waveform information of 3x and 5x that need to be pre-compensated is calculated in reverse.

[0185] The embodiment of the present disclosure also provides a signal transceiver link, including a signal transmission link and a signal reception link, such as Figure 8 or Fig. 9 As shown, the signal transmission link may include: a transmitting baseband digital module 201, a digital-to-analog conversion module 202, a transmitting local oscillator 203 and a transmitting orthogonal modulator 204, wherein: the transmitting baseband digital module 201 is configured to generate two orthogonal transmitting digital baseband signals, and send the generated transmitting digital baseband signals to the digital-to-analog conversion module 202; the digital-to-analog conversion module 202 is configured to convert the transmitting digital baseband signal into a transmitting analog baseband signal; the transmitting local oscillator 203 is configured to provide a transmitting local oscillator signal TX_LO; the transmitting orthogonal modulator 204 is configured to perform a phase shift operation on the transmitting local oscillator signal TX_LO based on the transmitting analog baseband signal to obtain a phase-shifted RF signal.

[0186] The signal receiving link may include a receiving end local oscillator 302, a receiving end mixer 303, an analog-to-digital converter (ADC) 304 and a receiving end baseband digital module 305; wherein the receiving end local oscillator 302 is configured to provide a receiving end local oscillator signal; the receiving end mixer 303 is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal; the analog-to-digital converter 304 is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; the receiving end baseband digital module 305 is configured to process the receiving end digital baseband signal to achieve target detection and / or wireless communication, for example, to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.

[0187] In the embodiment of the present disclosure, the two ideal I-channel digital baseband signals and Q-channel digital baseband signals generated by the transmitting end baseband digital module 201 can be converted into very ideal complex signals after passing through the digital-to-analog conversion module 202, and the phase of the complex signals can be accurately controlled by the transmitting end baseband digital module 201. Figure 8 or Fig. 9 The receiver structure in the signal transceiver link shown can effectively obtain the phase information of the radio frequency signal of the signal transmission link, thereby realizing phase modulation of multiple antennas.

[0188] In some exemplary embodiments, the signal transmission link may further include: a power amplifier 205, wherein the power amplifier 205 is configured to amplify the power of the phase-shifted radio frequency signal and output the amplified signal to the transmission antenna.

[0189] In some exemplary embodiments, the signal transmission link may further include: a transmitting antenna 206, wherein the transmitting antenna 206 is configured to radiate the amplified signal to a preset spatial region.

[0190] In some exemplary embodiments, the signal receiving link may further include a receiving antenna 301, wherein the receiving antenna 301 is configured to receive an echo signal, where the echo signal is a signal formed when the signal transmitted by the signal transmitting link is reflected and / or scattered by a target object.

[0191] In some exemplary embodiments, the local oscillator signal at the receiving end may be a swept frequency signal, or the local oscillator signal at the receiving end may be a single tone signal.

[0192] In the embodiment of the present disclosure, in the above signal transceiver link, the frequency of the TX-LO signal received by the transmitting end orthogonal modulator 204 in the signal transmission link and the frequency of the RX-LO signal received by the receiving end mixer 303 in the signal receiving link may be the same. For example, assuming that the signal output by the transmitting end baseband digital module 201 is a sine wave of x MHz, the TX-LO signal and the RX-LO signal may both be sine waves of z GHz, wherein x and z are both positive numbers, generally between 0 and 1000.

[0193] Below through Figure 8 The signal transceiver link shown introduces the principle of the present disclosure. As mentioned above, for the FMCW radar system, the signal transmission link can have two wave transmission schemes: 1) the transmitting end local oscillator signal is swept, and the transmitting end digital baseband signal is single-tone; 2) the transmitting end local oscillator signal is single-tone, and the transmitting end digital baseband signal is swept. Assuming that the transmitting end local oscillator signal TX_LO, the transmitting end digital baseband signal, and the modulated transmission signal are represented by TLO(t), BB(t) and TX(t), respectively, and the subscripts I and q are used to represent the I-path signal and the Q-path signal, and the superscript a is used to represent its complex signal form, then under the two wave transmission schemes, the signals at each stage of the signal transmission link can be represented as follows:

[0194] 1) The local oscillator signal at the transmitting end is swept, and the digital baseband signal at the transmitting end is single tone

[0195]

[0196] 2) Single tone of local oscillator signal at the transmitting end, frequency sweep of digital baseband signal at the transmitting end

[0197]

[0198] in, is the sweep slope of the FMCW signal, f bb is the starting frequency of the transmitting digital baseband signal, f tlo is the starting frequency of the transmitting local oscillator signal, is the initial phase of the transmitting digital baseband signal, is the initial phase of the local oscillator signal at the transmitting end.

[0199] Assumptions Figure 8 The receiving end local oscillator signal RX_LO in the signal receiving link shown is represented by RLO(t). In this embodiment, assuming that the receiving end local oscillator signal RX_LO is a swept frequency signal, RLO(t) can be expressed as:

[0200]

[0201] Among them, f rlo is the starting frequency of the local oscillator signal at the receiving end, and Φ0 is the initial phase of the local oscillator signal at the receiving end. Then, the RF signal transmitted by the signal transmission link generates an echo signal after being reflected / scattered by the target. After passing through the receiving antenna 301, the receiving mixer 303 and the receiving low-pass filter 307, the following is obtained:

[0202]

[0203] Wherein, τ represents the time delay of the RF signal transmitted by the signal transmission link and returned to the signal receiving link after being reflected / scattered by the target.

[0204] The above formula shows that both of the above two frequency sweeping methods can be used to control the initial phase of the digital baseband signal at the transmitting end. To perform precise phase modulation, thereby achieving high-precision and high-accuracy phase shifting, avoiding direct phase shifting at high frequencies.

[0205] In some optional embodiments, the receiving antenna 301 can be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in other optional embodiments, the receiving antenna can also be integrated on the chip package to form AiP or AoP, that is, a chip structure with a packaged antenna.

[0206] In some exemplary embodiments, the signal receiving link may further include a low noise amplifier (LNA) 306 disposed between the receiving antenna 301 and the receiving end mixer 303 , which performs low noise amplification on the echo signal received by the receiving antenna 301 and then sends it to the receiving end mixer 303 .

[0207] In some exemplary embodiments, the signal receiving link may further include a low-pass filter (LPF) 307 and a high-pass filter (HPF) 308 connected in series, which are arranged between the receiving mixer 303 and the analog-to-digital converter 304. The low-pass filter 307 and the high-pass filter 308 form a bandpass filter for filtering out-of-band noise.

[0208] In some exemplary embodiments, Figure 8 As shown, in the signal receiving link, the receiving end mixer 303 may be a real mixer, and the analog-to-digital converter 304 may be a real analog-to-digital converter.

[0209] In the disclosed embodiment, although the signal transmission link adopts a digital phase-shifting architecture, the signal receiving link may include a receiver of an orthogonal receiving architecture or a non-orthogonal receiving architecture. Therefore, it can effectively be compatible with sensors of receiving links of various architectures, effectively reducing the development cost of the entire transceiver link system.

[0210] In other exemplary embodiments, Fig. 9 As shown, in the signal receiving link, the receiving end mixer 303 may be an orthogonal mixer, and the analog-to-digital converter 304 may be an orthogonal analog-to-digital converter.

[0211] In order to match the signal transmission link of the digital phase shifter architecture, the embodiment of the present disclosure adjusts the receiving mixer 303 in the signal receiving link to an IQ demodulator (IQ Demodulator), and adjusts the analog-to-digital converter 304 to an IQ ADC. The echo signal received by the receiving antenna is processed by the above-mentioned low-noise amplifier 306, the receiving mixer 303, the low-pass filter 307, the high-pass filter 308 and the analog-to-digital converter 304 in sequence, and then converted into an IQ digital baseband signal. The subsequent receiving baseband digital module 305 processes the IQ digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics (i.e., micro-Doppler).

[0212] When the receiving end mixer 303 is an orthogonal mixer and the receiving end local oscillator signal is a swept frequency signal, the receiving end local oscillator signal RX_LO can be expressed as:

[0213]

[0214] In other exemplary embodiments, the local oscillator signal at the receiving end of the signal receiving link may be a single tone signal as shown in formula (12) or formula (13) in addition to being a frequency sweep signal as shown in formula (9) or formula (11).

[0215] The receiving mixer 303 is a real mixer: RLO(t)=cos(2πf rlot+Φ0)(12).

[0216] The receiving end mixer 303 is an orthogonal mixer:

[0217] When the local oscillator signal at the receiving end is a single-tone signal, it is only necessary to add a digital domain signal processing flow (including a digital domain mixing operation) in the receiving end baseband digital module 305 to obtain the required digital baseband signal.

[0218] In summary, the embodiments of the present disclosure can expand various system-level technical solutions according to the combination of different transmission schemes and receiving schemes (for example, whether the transmitting end uses a digital baseband signal single tone, a local oscillator signal sweep frequency or a digital baseband signal sweep frequency, a local oscillator signal single tone; whether the receiving end uses a real mixer, a real analog-to-digital converter or an orthogonal mixer, an orthogonal analog-to-digital converter; whether the receiving end uses a single-tone local oscillator signal or a swept frequency local oscillator signal).

[0219] Fig.10 FIG. 1 is a schematic diagram of another transceiver link in an embodiment of the present application. Fig.10 As shown, a transceiver link may include a transmitting link and a receiving link, etc. The transmitting link (i.e., a transmitter) may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (TXDDFS), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), a power amplifier (PA), etc. connected in sequence, and the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna. The receiving link may include a low noise amplifier (LNA), a real mixer (Real Mixer), a trans-impedance amplifier (TIA), a low-pass filter (LPF), a high-pass filter (HPF), a real digital-to-analog converter (Real ADC), etc. connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, Real Mixer, TIA, LPF, HPF and Real ADC in sequence and converted into a real digital baseband signal. The subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.

[0220] In the above transceiver link, the frequency of the TX-LO signal received by the IQ modulator in the transmitting link and the frequency of the RX-LO signal received by the Real Mixer in the receiving link may be the same. Fig.10As shown, if the signal output by the Baseband is a sine wave of x MHz, then the TX-LO signal and the RX-LO signal can both be sine waves of z GHz.

[0221] exist Fig.10 In the embodiment shown, the transmitting link adopts a digital phase-shifting architecture, while the receiving link can adopt components of an analog architecture, that is, no IQ components are used, so it can effectively be compatible with sensors of the receiving link of the analog architecture, effectively reducing the development cost of the entire transceiver link system.

[0222] Optionally, in an embodiment of the present application, the receiving link may include a receiving antenna, that is, the receiving antenna may be connected through a peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the receiving antenna may also be integrated on the chip package to form AiP or AoP, that is, a chip structure with a packaged antenna.

[0223] In an exemplary embodiment, the signal calibration link includes a signal receiving link corresponding to the signal transmission main path, wherein the signal receiving link is used to receive and process the echo signal corresponding to the FMCW RF transmission signal; wherein:

[0224] The signal receiving link is configured to obtain a sampling signal from the signal transmitting main path and obtain configuration information of a compensation signal according to the sampling signal; wherein the frequencies of the transmitting end local oscillator signal used in the signal transmitting main path and the receiving end local oscillator signal used in the signal receiving link are different.

[0225] In some optional embodiments, in order to match the transmission link of the digital phase shifter architecture, relevant adjustments may be made to the receiving link, such as Fig.11 As shown, Fig.11 Schematic diagram of a transceiver link including TX IQ Mod, RX IQ De-Mod and LO Freq Diff in an embodiment of the present application. Fig.11 The transceiver link shown may include Fig.10 The similar transmission link architecture and receiving link in the example (in order to avoid redundancy, the same parts will not be elaborated here), Fig.10The Real Mixer in the receiving link is adjusted to an IQ demodulator (IQ Demodulator), and the Real ADC is adjusted to an IQ ADC. That is, the receiving link may include a low noise amplifier (LNA), an IQ demodulator (IQ Demodulator), a transimpedance amplifier (TIA), a low pass filter (LPF), a high pass filter (HPF), an IQ digital-to-analog converter (IQ ADC), etc., which are connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, IQ Demodulator, TIA, LPF, HPF and IQ ADC in sequence and converted into an IQ digital baseband signal. The subsequent digital signal processing module processes the IQ digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics (i.e., micro-Doppler).

[0226] At the same time, based on Fig.11 When the transceiver link shown in the figure is self-calibrated, it is only necessary to directly connect the signal output port of the transmission link with the signal input port of the receiving link through a transmission line, that is, the transmission link directly sends the transmission signal to the receiving link through the transmission line, so as to realize the self-calibration operation of the receiving and / or transmission link without passing through the transmitting antenna and the receiving antenna. At this time, there is a certain frequency deviation between the TX-LO signal received by the IQ modulator in the transmission link and the RX-LO signal received by the IQ Demodulator in the receiving link. For example, Fig.11 As shown, if the signal output by the Baseband is a sine wave of x MHz, the TX-LO signal can be a sine wave of z GHz. At this time, the RX-LO signal is converted into a digital signal after being processed by a down-mixer (i.e., an IQ Demodulator in the receiving link), a low-pass filter, and a high-pass filter, so as to perform TX IQ imbalance calibration.

[0227] In some optional embodiments, such as Fig.11 As shown, the transmitting link (the Transmitter, TX shown in the figure) can be calibrated by adding a receiving link (the Receiver, RX shown in the figure), and the TXIQ imbalance compensation unit in the transmitting link performs compensation based on the calibrated data. At the same time, the transmitting link (the Transmitter, TX shown in the figure) can also be calibrated by multiplexing the receiving link (the Receiver, RX shown in the figure) actually used for signal transmission and reception, and the TX IQ imbalance compensation unit in the transmitting link and / or receiving link performs compensation based on the calibrated data. Among them, Fig.11This is the signal chain for TX IQ imbalance. Since the signal chain for TX harmonic signal problem is similar to the signal chain for TX IQ imbalance, Fig.11 The TX IQ imbalance compensation unit in the embodiment is replaced by a TX HD3 compensation unit to solve the problem of TX HD3. Similar implementations can also be performed in other embodiments, which will not be described in detail in the following for the sake of simplicity.

[0228] When the frequency of the local oscillator signal at the transmitting end is the same as the frequency of the local oscillator signal at the receiving end, a frequency adjustment circuit can be used to perform frequency adjustment, wherein the signal calibration link also includes a frequency adjustment circuit configured to adjust the frequency of at least one of the sampling signal and the receiving end local oscillator signal.

[0229] Furthermore, a logic built-in self test (BIST) in the integrated circuit is used as a frequency adjustment circuit, thereby reducing hardware costs.

[0230] In one implementation, the frequency adjustment circuit is connected between the receiving end mixer and the receiving end local oscillator in the signal receiving link, and is configured to adjust the frequency of the receiving end local oscillator signal, wherein the receiving end local oscillator is used to generate the receiving end local oscillator signal, and the receiving end mixer is used to demodulate the received signal using the receiving end local oscillator signal; the signal receiving link is connected between the signal transmission main path and the transmitting antenna, and is configured to obtain a sampling signal from the signal transmission main path, and use the signal output by the frequency adjustment circuit to process the sampling signal to obtain the configuration information of the compensation signal, which can be specifically referred to Fig.12 The circuit shown.

[0231] based on Fig.11 Based on the structure shown, in order to achieve accurate calibration of the transceiver link, Fig.11 The RX-LO port of the IQ Demodulator of the receiving link shown in the figure is set with an internal self-test module (Built-in Self-Test, referred to as BIST) module, that is, Fig.12 As shown, Fig.12 In the embodiment of the present application, a Fig.11 The structure shown is combined with the schematic diagram of the BIST transceiver link. Fig.11On the basis of the transceiver link structure shown, an IQ BIST architecture is set at the RX-LO port of the IQ Demodulator of the receiving link to input an LO signal with a preset frequency deviation at the RX-LO port of the IQ Demodulator of the receiving link. For example, an IQ BIST composed of a phase angle converter and an IQ modulator (IQ Modulator) is used to form a frequency-shifted signal based on the frequency deviation signal of another input signal BIST-LO of the IQ modulator, such as a TX-LO signal, through a phase angle converter through an IQ modulator, and then input to the RX-LO port of the IQ Demodulator. For example, if the TX-LO signal is a zGHz sine wave, and the BIST-LO signal is a y MHz sine wave, the frequency-shifted signal input to the RX-LO port of the IQ Demodulator is (zGHz-yMHz). It should be noted that, between different embodiments, x, y, and z are all schematic values, and the specific values ​​may be the same or different.

[0232] In some optional embodiments, based on Fig.12 The IQ BIST architecture of the transceiver link structure shown in the figure can also calibrate the transmit link of the digital phase shifter architecture by multiplexing the receive link in the transceiver link; wherein, in other embodiments, the calibration operation of the transmit link using the receive link, and the calibration operation of the receive link using the transmit link, can be implemented by multiplexing the corresponding receive link or transmit link in the link that actually performs signal transmission and reception, and can also be implemented by adding a corresponding calibration receive link or calibration transmit link to implement the calibration operation of the corresponding transmit link or receive link in the link that actually performs signal transmission and reception.

[0233] Optionally, the IQBIST may include a phase angle converter and an IQ modulator (IQ Modulator), the phase angle converter is used to realize the calibration of the I path and Q path in the transmission link of the digital architecture respectively, and the other input signal BIST-LO of the IQ modulator may be a y MHz sine wave (sine wave), which is used to simulate the characteristics related to the echo signal formed by the reflection of the transmission signal by the target, wherein Fig.12 Where x, y, and z are all positive numbers, and x≠y≠z, and can generally be between 0 and 1000.

[0234] Optional, in Fig.12In the transceiver link shown, a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may be provided in the transmitting link (e.g., between the TXDDFS and the IQ DAC), and / or a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may be provided in the receiving link (e.g., after the Real ADC), so that the transmitted and / or received signal may be supplemented based on the calibration parameters (or coefficients) obtained by the above self-calibration operation to solve problems such as IQ imbalance.

[0235] In another implementation, the signal input end of the frequency adjustment circuit is connected between the signal transmission main path and the transmission antenna, and the signal output end of the frequency adjustment circuit is connected between the signal receiving link and the receiving antenna, wherein the frequency adjustment circuit is configured to adjust the frequency of the RF transmission signal output by the signal transmission main path; correspondingly, the signal receiving link is configured to obtain a sampling signal from the frequency adjustment circuit, and obtain configuration information of a compensation signal according to the sampling signal, for details, see Fig.13 The circuit shown.

[0236] Fig.13 Schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present application. Fig.13 As shown, based on Fig.12 The structure shown can set the above-mentioned IQ BIST module between the signal output port of the transmitting link and the signal input port of the receiving link, that is, the transmitting link directly sends the transmitting signal to the receiving link through the IQ BIST module, so as to realize the self-calibration operation of the receiving link and / or the transmitting link without passing through the transmitting antenna and the receiving antenna.

[0237] It should be noted that in Figures 11 to 13 In the transmission chain structure shown, only an IQ compensation unit (TXIQ Imbalance compensation) is illustrated. In actual applications, an LO compensation unit (TXLO leakage compensation) and an HD3 compensation unit (TXHD3compensation) may be added to the transmission chain based on actual needs to form a compensation unit (TXcompensation) including an LO compensation unit (TX LO leakage compensation), an IQ compensation unit (TX IQ Imbalancecompensation) and / or an HD3 compensation unit (TX HD3compensation).

[0238] In an exemplary embodiment, when the RF transmission signal transmitted by the signal transmission main path is two orthogonal signals, the signal calibration link is provided with an orthogonal processing circuit and two signal receiving links, wherein each signal receiving link does not support processing of orthogonal echo signals, wherein the orthogonal processing circuit is configured to perform RF processing on the received orthogonal signal to obtain two signals, and send the two signals to the two signal receiving links respectively.

[0239] Fig.14 Schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application, Fig.15 Schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present application.

[0240] like Fig.14 As shown, a transceiver link, combined with Fig. 9 and Fig.13 The structure and related description shown in the figure, the transceiver link may include a transmission link, a receiving link and a calibration link. The transmission link may include a TX digital baseband signal source (TX Baseband), a direct digital frequency synthesizer (TX DDFS), a compensation unit (Compensation), an IQ digital-to-analog converter (IQDAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator) and a power amplifier (PA) connected in sequence, and the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna. The receiving link may include a low noise amplifier (LNA), a real mixer (RealMixer), a trans-impedance amplifier (TIA), a high-pass filter (HPF), a variable gain amplifier (VGA), a real digital-to-analog converter (RealADC) and an RX Baseband for TXRF calibration, etc., which are connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, Real Mixer, TIA, HPF, VGA and Real ADC in sequence and converted into a real digital baseband signal. The subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.

[0241] For the transmit link, the compensation unit (TXcompensation) disposed between the TX DDFS and the IQ DAC may include an LO compensation unit (TX LO leakage compensation), an IQ compensation unit (TX IQImbalance compensation) and an HD3 compensation unit (TX HD3 compensation) to implement compensation operations corresponding to LO leakage, IQ Imbalance and HD3 in the transmit link of the digital phase shifter architecture.

[0242] In some optional embodiments, a calibration module may be provided between the transmitting link and the receiving link, and the calibration compensation unit may be configured to multiplex the receiving link to perform operations such as calibration on the transmitting link of the above-mentioned digital phase shifter architecture. At the same time, the compensation unit may be based on the parameters or coefficients obtained by the calibration operation of the calibration module to implement compensation operations on the transmitting signal at the transmitting link end. In other embodiments, a corresponding receiving compensation unit may be provided in the receiving link simultaneously or separately, that is, at this time, the receiving compensation unit may be based on the parameters or coefficients obtained by the above-mentioned calibration operation to implement compensation for the echo signal at the receiving link end.

[0243] like Fig.14 As shown, the above-mentioned calibration module may include a BIST unit and an auxiliary circuit unit, that is, the output port of the transmitting link is connected to any node between the RealMixer and the Real ADC in the receiving link through the BIST unit and the auxiliary circuit unit. For example, the IQ Modulator in the transmitting link generates a radio frequency signal of (z GHz±x MHz) based on the digital phase-shifted baseband signal of x MHz and the LO signal of z GHz, and then outputs it to the BIST unit through the output port. The BIST unit performs a frequency deviation operation of y MHz on the received radio frequency signal to obtain an analog echo signal of (z GHz±x MHz±y MHz), and then uses the IQ De-Modulator in the auxiliary unit to down-convert the frequency to obtain a preset intermediate frequency signal (z GHz±x MHz±y MHz-z GHz=±x MHz±y MHz). The intermediate frequency signal is then input to a preset node in the receiving link to implement the calibration operation in the transmitting link.

[0244] Optionally, the auxiliary circuit unit may be an orthogonal demodulator circuit, and the output end of the auxiliary circuit unit may be connected to any node among the node between the TIA and the HPF, the node between the HPF and the VGA, and the node between the VGA and the Real ADC in the receiving link. In addition, in order to maximize the reuse of the receiving link structure, after the output port of a transmitting link passes through the BIST unit and the auxiliary circuit unit, the I and Q branches may be respectively connected to different transmitting links, that is, Fig.14 As shown, two receiving links are multiplexed to calibrate one transmitting link. After the calibration of the transmitting link is completed, the compensation units such as the LO compensation unit (TX LO leakage compensation), the IQ compensation unit (TX IQ Imbalance compensation) and / or the HD3 compensation unit (TX HD3 compensation) in the compensation module (TX compensation) can be used to implement compensation operations corresponding to the LO leakage, IQ Imbalance and HD3 problems in the transmitting link of the digital phase shifter architecture based on the parameters obtained by the calibration.

[0245] In some optional embodiments, the BIST unit may include a phase angle converter and an IQ modulator (IQ Modulator) connected in sequence, and the auxiliary circuit unit may include an LNA, an IQ De-Modulator and a TIA connected in sequence, that is, the phase angle converter receives the RF signal output from the transmission link, and one input end of the IQ Modulator is connected to the output end of the phase angle converter, and the other input end receives the BIST-LO signal of y MHz to generate a preset echo signal. The LNA sends the received echo signal to an input end of the IQ De-Modulator after amplification, and the other input end of the IQ De-Modulator is used to receive the RX-LO signal of z GHz. The two output branches (i.e., the I branch and the Q branch) of the IQ De-Modulator are respectively connected to the corresponding nodes in the corresponding receiving links after the TIA, so as to output the generated preset intermediate frequency signal to the two receiving links, thereby realizing a more efficient multiplexing receiving link design while realizing the calibration operation.

[0246] It should be noted that, for the calibration operation in the embodiment of the present application, if the transmitting link transmits a swept frequency signal, in the actual calibration operation, the TX LO signal can be used as a single tone signal for point-by-point calibration; at the same time, the TXLO signal can also be used as a swept frequency signal for large bandwidth calibration operations, and even the swept frequency bandwidth calibration can be used once to implement the calibration operation for the swept frequency signal of the entire frequency band.

[0247] based on Fig.14 The structure shown in the figure can be realized by cascading at least two BIST units in order to further suppress the preset degree of HD3, LO Leakage, IQ Imbalance, etc. in the transmission link of the digital phase shifter architecture; Fig.15 As shown, by two BIST units connected in series, the noise caused by the above defects can be suppressed to a level of -50 dB, thereby effectively reducing the difficulty of development and design of related link analog devices.

[0248] In some optional embodiments, based on the transmission link of the digital phase shifter architecture described in the embodiments of the present application, when calibrating and compensating for IQ Imbalance, the compensation coefficient of IQ Imbalance can be obtained in the time domain (Time-Domain) based on spectrum analysis, and the compensation coefficient of IQ Imbalance can also be obtained in the frequency domain (Frequency-Domain) based on the spectrum peak ratio.

[0249] In some optional embodiments, in order to further improve the accuracy of the IQ Imbalance compensation coefficient, the ideal compensation coefficient may be approached by iterative calibration and compensation, or the ideal compensation coefficient may be obtained by multi-observation calibration and compensation.

[0250] For example, for the iterative calibration and compensation method, it can be determined whether to stop the iterative operation based on the size relationship between the compensation coefficients of the two calibration compensations, or whether the difference between the compensation coefficients of the two calibration compensations meets the preset iterative conditions, and the compensation coefficient obtained when the iterative operation is stopped is used as the final compensation coefficient in the current scene for subsequent operations. For the multi-observation calibration and compensation method, after multiple (for example, three) calibration and compensation operations, the measurement data obtained from each operation can be subjected to FFT (fast Fourier transform) and the corresponding amplitude and phase information can be obtained. The measurement can be subtracted and normalized to obtain relevant data, and an observation matrix can be constructed; the corresponding compensation coefficient can be reversely solved based on the data obtained by inverting the observation matrix.

[0251] In some optional embodiments, based on the similar idea of ​​obtaining the IQ Imbalance compensation coefficient, the compensation coefficient of LO leakage and / or HD3 may be obtained by adopting a method such as iterative calibration and compensation, or a method of multiple observation calibration and compensation.

[0252] In the above examples of transmitters with compensation units, the problem of solving harmonic distortion in the transmitter is mentioned. It is found through research that some harmonic distortion may be caused by components including nonlinear characteristics such as mixers in the transmitter.

[0253] by Fig.16 For example, Fig.16 Schematic diagram of the structure of the mixer 421 in the embodiment of the present application. Fig.16 As shown, the mixer 421 includes a voltage-current converter (V / I Converter), a current switch (Current Switch) and a current-voltage converter (I / V Converter). The voltage-current converter converts the received voltage signal into a current signal; the current switch is connected to the voltage-current converter and the second signal generator, and is used to process the current signal output by the voltage-current converter using the local oscillator signal; the current-voltage converter is connected to the current switch, and is used to convert the current signal output by the current switch into a voltage signal.

[0254] In the above structure, since the voltage-current converter is provided with a transistor amplifier, based on the nonlinear characteristics of the transistor amplifier and the low frequency characteristics of the baseband signal, the current signal output by the voltage-current converter contains a harmonic signal corresponding to the baseband signal. For example, the HD caused by the third-order nonlinearity of the baseband can be referred to as HD3. Similarly, the harmonic caused by the fifth-order nonlinearity is called HD5. When the current switch processes the current signal output by the voltage-current converter, the frequency of the harmonic is converted to the RF band after up-conversion. Since the operation complexity of suppressing the harmonic signal in the RF band is high, the hardware cost is high. If the harmonic signal in the RF band is not removed, it will affect the signal quality of the radar reception and transmission, and thus affect the accuracy of the radar measurement.

[0255] The compensation unit is used to input the generated cancellation signal into the signal transmission link to cancel the harmonic signal in the radio frequency signal. The compensation unit is independent of the first signal generator.

[0256] To this end, the compensation unit may include a cancellation signal generator. The cancellation signal output by the compensation unit can suppress the harmonic signal in the radio frequency signal, reduce the harmonic component in the radio frequency signal, and thus improve the signal quality of the radio frequency signal output by the transmitter.

[0257] In the embodiment of the present application, for the harmonic signal in the signal transmission link, the compensation unit uses feedback or according to the characteristics of the transmission wave to input the generated cancellation signal into the signal transmission link to cancel the harmonic signal in the RF signal output by the signal transmission link. The cancellation signal has the characteristics of opposite phase and similar amplitude to the harmonic signal transmitted in the RF transmission circuit, so as to achieve the purpose of suppressing the harmonic signal.

[0258] In some examples, the compensation unit generates a compensation signal including a cancellation effect according to parameters such as the phase, frequency, or amplitude of the baseband signal generated by the first signal generator, or even the path length combined with the LO signal.

[0259] For example, Fig. 7A An example of a transmitter in which a compensation unit is connected to a signal transmission link is shown in FIG. Fig. 7A In the structure shown, the compensation unit is a TX compensation unit. The TX compensation unit includes a generator (not shown) that can generate a cancellation signal according to the characteristics of the transmission wave. The cancellation signal generator can be, for example, Fig. 7A The TX HD3 compensation unit shown in Figure 2.

[0260] Among them, the baseband processor ( Fig. 7A The baseband frame (hereinafter referred to as the baseband frame) controls the orthogonal digital baseband signal generated by the TX DDFS. The TX compensation unit generates an orthogonal compensation signal according to the parameters of the orthogonal digital signal, combines the orthogonal compensation signal and the orthogonal digital signal, and sends them to the IQ DAC to be converted into an analog baseband signal. After LPF filtering, it is sent to the mixer (i.e., Fig. 7A The IQ modulator in the RF transmitter is mixed to obtain a radio frequency signal based on the mixing of the TX LO signal and the analog baseband signal. The PA amplifies the mixed signal and outputs it through the transmitting antenna. The compensation signal cancels at least part of the harmonic signal in the RF transmitting circuit, such as the HD3 harmonic signal. Therefore, the clutter in the transmitted RF signal will be greatly reduced. The RF signal can be an FMCW signal.

[0261] In some other examples, the compensation unit generates a compensation signal according to harmonic information obtained through feedback from the radio frequency transmission circuit. Fig.17 , Fig.17 for Fig. 7A The schematic diagram of the structure of the compensation unit in the transmitter is shown in FIG. Fig.17 As shown, the compensation unit includes an acquisition circuit and a cancellation signal generator.

[0262] The acquisition circuit is coupled to the RF transmission circuit, and is used to acquire the signal in the RF transmission circuit to obtain an acquisition signal. The acquisition signal (or sampling signal) can reflect the waveform information (also called harmonic parameters) in the harmonic signal, such as the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, the power of the main frequency signal, etc.

[0263] It should be noted that the harmonic parameters reflected by the acquisition signal are related to the information carried by the signal that can be acquired by the acquisition circuit. For example, if the acquisition circuit is a power acquisition circuit, the corresponding acquisition signal includes the power of the main frequency. For another example, if the acquisition circuit uses at least part of the circuit of the receiver, the acquisition signal reflects the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, the power of the main frequency signal, etc.

[0264] At least one of the above harmonic parameters can be extracted by analog circuits. For example, the power of the main frequency signal is output through a coupler and a power detector. Or the advantage of the digital circuit in the radar chip in frequency domain calculation is used to extract harmonic parameters. For example, by coupling the RF transmitting circuit, a signal identical to the signal transmitted at the coupling point is obtained as a collection signal. The collection signal carries the main frequency signal and the harmonic signal. The collection signal is converted into a digital signal by an ADC and handed over to the digital circuit for calculation in the frequency domain to obtain more harmonic parameters.

[0265] In one implementation, the input end of the acquisition circuit is connected to the output end or the signal detection end of the mixer. This method can detect the harmonic signal generated by the voltage-current converter and has a simplified acquisition circuit.

[0266] In an exemplary embodiment, the signal calibration link has a first input end connected between a voltage-to-current converter and a current switch in a mixer at a transmitting end in the main signal transmission path, a second input end between the main signal transmission path and a transmitting antenna, and a signal output end connected to a compensation circuit in the main signal transmission path, and is used to obtain a signal in the main signal transmission path from at least one of the first input end and the second input end, and determine compensation information based on the obtained signal. Fig.18 , 19 In one connection mode of the structure shown, the input end of the signal calibration link is connected to the detection end between the voltage-to-current converter and the current switch, and is coupled to the ADC.

[0267] See also Fig. 20 As shown, the signal calibration link includes a calibration demodulator, a multiplexer and a calibration module; wherein:

[0268] The calibration demodulator is configured to obtain the signal in the signal transmission main path from the second input terminal and perform demodulation processing;

[0269] The multiplexer has two signal input terminals and one signal input terminal, one of which is connected between the voltage-to-current converter and the current switch in the transmitting mixer, and the other is connected to the signal input terminal of the calibration demodulator, and is used to output a signal corresponding to one of the two signal input terminals;

[0270] The calibration module is configured to determine the compensation information according to the signal output by the multiplexer.

[0271] In another embodiment, the input end of the acquisition circuit is connected to the RF output end or RF detection end of the RF transmission circuit. The RF output end is exemplified as the output end of the RF transmission circuit. The RF detection end is exemplified as the input end or output end of at least one PA in the RF transmission circuit. This method can acquire more accurate harmonic parameters in the RF transmission circuit, but has a more complex circuit structure.

[0272] In some chips including a BIST module, the acquisition circuit can obtain the acquisition signal through part or all of the circuits in the BIST module. Fig.19 As shown, the input end of the acquisition circuit is coupled to the RF output end, which includes a down converter, a filter, etc. in sequence, and is connected to the IQ ADC to output a digital acquisition signal. Among them, the down converter, the filter, etc. can reuse the BIST module or the receiver.

[0273] The collected signal is input to a cancellation signal generator, which is at least one circuit in the compensation unit. The cancellation signal generator is connected to the first signal generator, so that the signal received by the radio frequency transmitting circuit includes both the baseband signal and the cancellation signal.

[0274] For example, the cancellation signal generator includes the cancellation signal generator mentioned above and a digital circuit for extracting harmonic information, wherein the digital circuit for extracting harmonic information can be configured independently, or at least partially shared with the digital circuit in the radar chip.

[0275] The digital circuit for extracting harmonic information uses, for example, a digital circuit for processing difference frequency baseband signals in a radar chip to extract harmonic information such as harmonic frequency, main frequency, and main frequency power, and provides the information to a cancellation signal generator. The cancellation signal generator generates a cancellation signal according to the received parameters.

[0276] For another example, the digital circuit for extracting harmonic information extracts the main frequency amplitude in the collected signal, and calculates the harmonic amplitude based on the difference between the preset main frequency amplitude and the harmonic amplitude. The cancellation signal generator generates a harmonic signal compensation signal based on the calculated harmonic amplitude and other pre-configured harmonic parameters. Among them, the pre-configured harmonic parameters can be calculated based on the sweep frequency range, phase, etc. of the main frequency signal to be transmitted by the radar chip.

[0277] The cancellation signal generator in the cancellation signal generator can be configured independently with the first signal generator, or at least partially shared. For example, the cancellation signal generated by the cancellation signal generator is input into the first signal generator, so that the baseband signal output by the first signal generator includes the cancellation signal. The cancellation signal generator may include a third harmonic generator and a fifth harmonic generator.

[0278] For another example, the compensation unit further includes an adder, coupled to the cancellation signal generator and the first signal generator, to combine the baseband signal generated by the first signal generator and the cancellation signal generated by the cancellation signal generator. In this embodiment, the cancellation signal includes a cancellation signal Signal_HD3 generated by the third harmonic generator to cancel the third harmonic, and a cancellation signal Signal_HD5 generated by the fifth harmonic generator to cancel the fifth harmonic. The cancellation signals Signal_HD3 and Signal_HD5 and the baseband signal generated by the first signal generator are combined by the adder and output to the RF transmission circuit.

[0279] In summary, the transmitter circuit examples provided in the present application that pre-input a cancellation signal into a radio frequency transmitting circuit in a feedback manner can ensure that the radio frequency signal transmitted by the chip contains sufficiently low harmonic signals under different environments.

[0280] In order to effectively suppress harmonic signals according to the actual operating environment of the chip during use, for example, considering the impact of ambient temperature on semiconductor devices, the present application also provides a method for canceling harmonic signals in the above transmitter using a feedback mechanism, including:

[0281] Step 10: performing a collection operation on the signal in the signal transmission link to obtain a collection signal; wherein the signal transmission link is used to generate a radio frequency signal for radar detection, wherein the radio frequency signal includes a harmonic signal.

[0282] Step 20: Detect the collected signal, generate a cancellation signal for canceling the harmonic signal, and output the cancellation signal to the signal transmission link.

[0283] The method provided in the embodiment of the present application performs a collection operation on the signal in the signal transmission link to obtain a collection signal, and uses the collection signal to generate a cancellation signal, and outputs it to the signal transmission link, so as to use the cancellation signal to suppress the harmonic signal in the radio frequency signal and reduce the harmonic component in the radio frequency signal, thereby improving the signal quality of the radio frequency signal output by the transmitter, and further improving the receiving performance of the receiver for the radio frequency signal.

[0284] The following is combined with Figures 11 to 20 , take the transmitter and its working process as an example:

[0285] For example, Fig.11 An example of using a feedback mechanism to extract harmonic information from a transmitter so that a compensation unit generates a corresponding cancellation signal is shown in FIG. Fig.11 In the structure shown, the compensation unit includes a TX HD3 compensation unit as an example. The TXHD3 compensation unit generates a compensation signal according to the waveform characteristics of the received signal. The feedback mechanism can be executed in the calibration mode of the radar chip to prevent the signal transmission power of the radar chip from being weakened during normal detection.

[0286] The baseband processor in the transmitter (such as Fig.11 The baseband frame in the TX DDFS controls the orthogonal digital baseband signal generated by the TX compensation unit, and then sends it to the IQ DAC to be converted into an analog baseband signal. The analog baseband signal is mixed with an analog cancellation signal for canceling the harmonic signal in the transmission link. The analog baseband signal is filtered by the LPF and then enters the first mixer (i.e., Fig.11 The first mixer uses the TX LO to mix the received filtered signal to obtain an RF signal. The RF signal is coupled through the receiver and output to the TX HD3 calibration circuit (such as Fig.11 The TX HD3 calibration box in FIG. 1 ). The TX HD3 calibration circuit can be regarded as a digital circuit for extracting harmonic information.

[0287] In the receiver, the LNA amplifies the signal output by the transmitter and outputs it to the second mixer (i.e., Fig.11The IQ demodulator in the transmitter is used to obtain a demodulated signal, and the demodulated signal is sent to a transimpedance amplifier for amplification. The demodulated signal is then sent to the TX HD3 calibration circuit after passing through the LPF and HPF in sequence, and then through the IQ ADC for analog-to-digital conversion. The TX HD3 calibration circuit extracts the harmonic information in the transmitter from the feedback signal, and converts it into the parameters required to generate the cancellation signal through the upper controller, and provides it to the TX HD3 compensation unit. Among them, the harmonic information obtained by the TX HD3 calibration circuit includes one or more of the following parameters: the initial phase, starting frequency, cutoff frequency, frequency change duration, center frequency, etc. of the harmonic signal (or main frequency signal). The TX HD3 calibration unit or the upper controller determines the parameters used to generate the cancellation signal in the compensation unit according to the harmonic information, such as the initial phase, the frequency of the cancellation signal, the delay, etc.

[0288] It should be noted that the cancellation operations of the third harmonic and / or the fifth harmonic in the above examples may be determined according to the requirements of the transmitter.

[0289] In some optional embodiments, based on the transmitting link of the digital phase shifter architecture described in the embodiment of the present application, when calibrating and compensating the HD3, the main source of the generation of HD3 in the active mixer is the third harmonic in the nonlinearity of V / IConverter, so the HD3 can be calibrated and compensated by the following method: Fig.18 The cubic module-based compensation architecture shown in Fig.19 The compensation structure shown is based on a triple frequency waveform generator to achieve this.

[0290] Fig. 20 FIG. 1 is a schematic diagram of calibration compensation of a transmission link based on a digital phase shifter architecture in an embodiment of the present application. Fig. 20 As shown, based on the relevant technical contents of the calibration and compensation operation for IQ Imbalance, LO leakage and HD3 in the embodiment of the present application: the compensation operation for IQ Imbalance can be achieved by compensating the conjugate signal of the BB (baseband) signal to reversely cancel the image component, and this compensation operation mode is not affected by the calibration mode of IQ Imbalance. The compensation for LO Leakage can be achieved by adjusting the DC component (i.e., DC bias) of the IQ two-way, and the calibration mode of LO Leakage has no effect on its compensation scheme. For HD3, since the third-order harmonic distortion of the orthogonal mixer V / I Converter is the main source of HD3, and the harmonic distortion will be affected by the DC bias, when both LO Leakage and HD3 of the transmission link need to be calibrated, it is necessary to calibrate HD3 after calibrating LO Leakage to ensure the accurate performance of HD3 calibration.

[0291] In addition, the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency doubling waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process. Specifically:

[0292] In an optional embodiment, for a digital pre-compensation architecture based on a digital cubic module, LO Leakage may be calibrated and compensated first, and then the root cause of the HD3 problem, i.e., the compensation coefficient of HD3, may be calibrated under a stable DC bias; then, IQ Imbalance may be calibrated, and after IQ Imbalance is further compensated, the third harmonic distortion may be compensated for the two IQ paths based on the results of IQ Imbalance pre-compensation.

[0293] In an optional embodiment, for a digital pre-compensation architecture based on a frequency doubling waveform generator module, after calibrating and compensating for LO Leakage, the compensation coefficient of HD3 can be calibrated, and IQImbalance can be calibrated and IQ Imbalance can be compensated under a stable DC bias; subsequently, the third harmonic distortion is compensated for the IQ paths based on the results of IQImbalance pre-compensation according to the results after compensation.

[0294] In another optional embodiment, for a digital pre-compensation architecture based on a frequency doubling waveform generator module, after calibrating and compensating for LO Leakage, the compensation coefficient of HD3 can be calibrated, and IQImbalance can be calibrated and compensated for IQ Imbalance under a stable DC bias; subsequently, the actual waveforms of the IQ and Q signals and the compensation coefficient of HD3 are calculated respectively through the compensation results, and the waveform information of 3x and 5x that need to be pre-compensated is calculated in reverse.

[0295] In another optional embodiment, for the digital pre-compensation architecture based on the frequency doubling waveform generator module, LO Leakage can be calibrated and compensated first, and then the pre-compensation coefficients of HD3 and IQ Imbalance can be calibrated simultaneously through multiple (such as three) observations, and then the pre-compensation coefficients at the HD3 mirror position can be calibrated again through observations (such as two times); finally, the 3-fold and 5-fold frequency coefficients that need to be pre-compensated are calculated back through the pre-compensation coefficients at the HD3 and HD3 mirror positions. It should be noted that the observations in the embodiments of the present application are used to represent operations such as testing and comparative analysis of different test results.

[0296] See also Fig. 20It can be seen from the structure shown that the acquisition circuit has two acquisition branches, and the two acquisition branches can be switched dynamically; wherein, the input end of one acquisition branch is connected between the voltage-current converter and the current switch; the input end of the other acquisition branch is connected to the output end of the power amplifier, and the acquisition branch is provided with an IQ demodulator; in addition, Fig. 20 In the structure shown, the acquisition circuit is further provided with a multiplexer, wherein the input end of the multiplexer is respectively connected to the output ends of the two acquisition branches, and the output end is used to output the acquisition signal.

[0297] exist Figures 18 to 20 In the structure shown, the acquisition signal may be an analog signal, that is, the output end of the acquisition circuit is connected to the IQ ADC; or, the acquisition signal may be a digital signal, and the acquisition circuit at least includes the IQ ADC.

[0298] It should be noted that when the RF signal sent by the signal transmission link is not an orthogonal signal, Figures 18 to 20 The acquisition circuit can be completed by using non-orthogonal components. For example, a single-ended down-conversion mixer can be used to replace the IQ demodulator, and a single-ended analog-to-digital converter can be used to replace the IQ ADC.

[0299] like Fig.21 As shown, the embodiment of the present disclosure further provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link, and the signal transmission method includes:

[0300] Step 2101: determine the phase of the radio frequency transmission signal of each signal transmission link;

[0301] Step 2102: determining the initial phase of the baseband signal in each of the signal transmission links according to the phase of the radio frequency transmission signal;

[0302] Step 2103: Generate an initial baseband signal according to the determined initial phase;

[0303] Step 2104: Compensate the initial baseband signal using the pre-acquired compensation information to obtain the baseband signal;

[0304] Step 2105: Perform a phase shift operation on the transmitting end local oscillator signal based on the baseband signal to obtain the radio frequency transmission signal.

[0305] The signal transmission method of the embodiment of the present disclosure generates a digital baseband signal in the digital domain by using a digital phase shifter architecture, which has better orthogonality and lower sidelobes. The phase shift phase can be generated very accurately, so that the phase modulation accuracy is higher, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of small link loss, low cost, and no need for offline calibration. It can also support more flexible wave transmission schemes, such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.

[0306] In some exemplary embodiments, the digital baseband signal at the transmitting end is a single-tone signal, and the local oscillator signal at the transmitting end is a swept-frequency signal; or, the digital baseband signal at the transmitting end is a swept-frequency signal, and the local oscillator signal at the transmitting end is a single-tone signal.

[0307] In some exemplary embodiments, the frequency bandwidth of the swept frequency signal is above 2 GHz.

[0308] The embodiment of the present disclosure also provides an integrated circuit, which may include any of the above signal transceiver links. Optionally, the integrated circuit may be a millimeter wave radar chip (chip or die).

[0309] In some optional embodiments, the integrated circuit may be an AiP (Antenna-In-Package) chip structure, an AoP (Antenna-On-Package) chip structure, or an AoC (Antenna-On-Chip) chip structure.

[0310] According to some other embodiments of the present disclosure, an electromagnetic wave device is further proposed in the embodiments of the present disclosure. The electromagnetic wave device may include an antenna, and an integrated circuit as described above. The integrated circuit is electrically connected to the antenna for receiving and transmitting electromagnetic wave signals. For example, the electromagnetic wave device may include: a carrier, an integrated circuit and an antenna as described in any of the above embodiments, etc., wherein the integrated circuit may be arranged on the carrier; the antenna may be arranged on the carrier, or the integrated circuit may be integrated with the integrated circuit and arranged on the carrier (that is, the antenna may be an antenna provided in an AiP, AoP or AoC structure at this time); wherein the integrated circuit is connected to the antenna (that is, the sensor chip or the integrated circuit is not integrated with an antenna at this time, such as a conventional SoC, etc.), for receiving and transmitting electromagnetic wave signals. The carrier may be a printed circuit board PCB.

[0311] An embodiment of the present disclosure provides a device, which may include: a device body; and an electromagnetic wave device as described above, which is arranged on the device body; wherein the electromagnetic wave device is used for target detection and / or communication to provide reference information for the operation of the device body.

[0312] The present disclosure also provides an electronic device, which can be expressed in the form of a general-purpose computing device. The components of the electronic device may include, but are not limited to: at least one processing unit, at least one storage unit, a bus connecting different system components (including a storage unit and a processing unit), a display unit, etc. Among them, the storage unit stores a program code, and the program code can be executed by the processing unit so that the processing unit executes the method described in this specification according to various exemplary embodiments of the present disclosure. The storage unit may include a readable medium in the form of a volatile storage unit, such as a random access memory unit (RAM) and / or a cache memory unit, and may further include a read-only memory unit (ROM).

[0313] The storage unit may also include a program / utility having a set (at least one) of program modules, such program modules including but not limited to: an operating system, one or more application programs, other program modules, and program data, each of which or some combination may include an implementation of a network environment.

[0314] The bus may represent one or more of several types of bus structures, including a memory unit bus or memory unit controller, a peripheral bus, an accelerated graphics port, a processing unit, or a local bus using any of a variety of bus architectures.

[0315] The electronic device may also communicate with one or more external devices (e.g., keyboards, pointing devices, Bluetooth devices, etc.), may communicate with one or more devices that enable a user to interact with the electronic device, and / or may communicate with any device that enables the electronic device to communicate with one or more other computing devices (e.g., routers, modems, etc.). This communication may be performed through an input / output (I / O) interface. In addition, the electronic device may also communicate with one or more networks (e.g., local area networks (LANs), wide area networks (WANs), and / or public networks, such as the Internet) through a network adapter. The network adapter may communicate with other modules of the electronic device through a bus. It should be understood that, although not shown in the figure, other hardware and / or software modules may be used in conjunction with the electronic device, including but not limited to: microcode, device drivers, redundant processing units, external disk drive arrays, RAID systems, tape drives, and data backup storage systems, etc.

[0316] For example, the electronic device in the embodiments of the present disclosure may also include: a device body; and an electromagnetic wave device as described in any of the above embodiments arranged on the device body; wherein the electromagnetic wave device can be used to realize functions such as target detection and / or wireless communication.

[0317] Specifically, based on the above embodiments, in an optional embodiment of the present disclosure, the electromagnetic wave device can be arranged outside the device body, or arranged inside the device body, and in other optional embodiments of the present disclosure, the electromagnetic wave device can also be arranged partially inside the device body and partially outside the device body. The embodiments of the present disclosure are not limited to this, and the specific situation may be determined.

[0318] In an optional embodiment, the device body can be a component or product used in fields such as smart cities, smart homes, transportation, smart homes, consumer electronics, security monitoring, industrial automation, in-cabin detection (such as smart cockpits), medical devices, and health care. For example, the device body can be intelligent transportation equipment (such as cars, bicycles, motorcycles, ships, subways, trains, etc.), security equipment (such as cameras), liquid level / flow rate detection equipment, smart wearable devices (such as bracelets, glasses, etc.), smart home devices (such as sweeping robots, door locks, televisions, air conditioners, smart lights, etc.), various communication devices (such as mobile phones, tablet computers, etc.), and gates, smart traffic lights, smart signs, traffic cameras, and various industrialized robotic arms (or robots), etc. It can also be various instruments for detecting life characteristic parameters and various devices equipped with the instruments, such as life characteristic detection in car cabins, indoor personnel monitoring, smart medical equipment, consumer electronic equipment, etc.

[0319] The embodiment of the present disclosure also provides a non-transitory computer-readable storage medium on which computer-readable instructions are stored. When the instructions are executed by a processor, the processor executes the signal transmission method as described above.

[0320] Through the description of the above implementation methods, it is easy for those skilled in the art to understand that the example implementation methods described here can be implemented by software, or by combining software with necessary hardware. The technical solution according to the implementation methods of the present disclosure can be embodied in the form of a software product, which can be stored in a non-volatile storage medium (which can be a CD-ROM, a USB flash drive, a mobile hard disk, etc.) or on a network, including several instructions to enable a computing device (which can be a personal computer, a server, or a network device, etc.) to execute the above method according to the implementation methods of the present disclosure.

[0321] The software product may use any combination of one or more readable media. The readable medium may be a readable signal medium or a readable storage medium. The readable storage medium may be, for example, but not limited to, a system, device or device of electricity, magnetism, light, electromagnetic, infrared, or semiconductor, or any combination thereof. More specific examples (non-exhaustive list) of readable storage media include: an electrical connection with one or more wires, a portable disk, a hard disk, a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or flash memory), an optical fiber, a portable compact disk read-only memory (CD-ROM), an optical storage device, a magnetic storage device, or any suitable combination thereof.

[0322] Computer readable storage media may include data signals propagated in baseband or as part of a carrier wave, wherein readable program codes are carried. Such propagated data signals may take various forms, including but not limited to electromagnetic signals, optical signals, or any suitable combination thereof. The readable storage medium may also be any readable medium other than a readable storage medium, which may send, propagate, or transmit a program for use by or in conjunction with an instruction execution system, device, or device. The program codes contained on the readable storage medium may be transmitted using any suitable medium, including but not limited to wireless, wired, optical cable, RF, etc., or any suitable combination thereof.

[0323] Program code for performing the operations of the present disclosure may be written in any combination of one or more programming languages, including object-oriented programming languages ​​such as Java, C++, etc., and conventional procedural programming languages ​​such as "C" or similar programming languages. The program code may be executed entirely on the user computing device, partially on the user device, as a separate software package, partially on the user computing device and partially on a remote computing device, or entirely on a remote computing device or server. In cases involving a remote computing device, the remote computing device may be connected to the user computing device through any type of network, including a local area network (LAN) or a wide area network (WAN), or may be connected to an external computing device (e.g., through the Internet using an Internet service provider).

[0324] The computer-readable medium carries one or more programs. When the one or more programs are executed by a device, the computer-readable medium implements the aforementioned functions.

[0325] Those skilled in the art will appreciate that the above modules can be distributed in the device according to the description of the embodiment, or can be changed accordingly and only used in one or more devices different from the embodiment. The modules of the above embodiments can be combined into one module, or further divided into multiple sub-modules.

[0326] According to an embodiment of the present disclosure, a computer program is proposed, including a computer program or an instruction, which, when executed by a processor, can execute the method described above. In an optional embodiment, the above-mentioned integrated circuit can be a millimeter wave radar chip. The type of digital functional module in the integrated circuit can be determined according to actual needs. For example, in a millimeter wave radar chip, the receiving end baseband digital module can be used for functions such as distance-dimensional Doppler transform, velocity-dimensional Doppler transform, constant false alarm detection, wave direction detection, point cloud processing, etc., to obtain information such as the distance, horizontal angle, pitch angle, speed, height, micro-Doppler motion characteristics, shape, size, surface roughness and dielectric properties of the target.

[0327] It should be noted that radio devices can achieve functions such as target detection and / or communication by transmitting and receiving radio signals to provide the device body with detection target information and / or communication information, thereby assisting or even controlling the operation of the device body.

[0328] For example, when the above-mentioned device body is applied to an advanced driver assistance system (i.e., ADAS), the radio device (such as millimeter-wave radar) as a vehicle-mounted sensor can assist the ADAS system to realize application scenarios such as adaptive cruise control, automatic braking assistance (i.e., AEB), blind spot detection warning (i.e., BSD), assisted lane change warning (i.e., LCA), reversing assistance warning (i.e., RCTA), parking assistance, rear vehicle warning, collision avoidance, pedestrian detection, etc.

[0329] The technical features of the above-described embodiments may be arbitrarily combined. To make the description concise, not all possible combinations of the technical features in the above-described embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0330] The above-described embodiments only express the preferred embodiments of the present disclosure and the technical principles used. The description is relatively specific and detailed, but it cannot be understood as limiting the scope of the invention patent. It is possible for those skilled in the art to make various obvious changes, readjustments and substitutions without departing from the protection scope of the present disclosure. Therefore, although the present disclosure is described in more detail through the above embodiments, the present disclosure is not limited to the above embodiments, and may also include more other equivalent embodiments without departing from the concept of the present disclosure, and the protection scope of the patent of the present disclosure is determined by the scope of the attached claims.

Claims

1. A signal transmission link, It is characterized in that It includes a signal transmission main path and a signal calibration link integrated in the same integrated circuit; wherein: The signal calibration link is configured to calibrate the signal transmission main path to obtain compensation information; and the signal transmission main path is configured to generate a radio frequency transmission signal after performing a compensation operation according to the compensation information, so as to achieve target detection and / or communication.

2. The signal transmission link according to claim 1, It is characterized in that The compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.

3. The signal transmission link according to claim 1, It is characterized in that The integrated circuit is a millimeter wave radar chip, and / or the radio frequency transmission signal is a FMCW signal.

4. The signal transmission link according to claim 3, Features: The main signal transmission path includes a first signal source and a phase shifter; wherein the first signal source is configured to generate a first analog signal; and the phase shifter is configured to perform frequency shifting and / or phase shifting on the first analog signal to form a radio frequency transmission signal.

5. The signal transmission link according to claim 4, Features: When the phase shifter is a non-orthogonal architecture, the phase shifter includes a second signal source and a transmitting mixer, wherein the second signal source is configured to generate a second analog signal, and the transmitting mixer is configured to perform mixing processing on the first analog signal and the second analog signal to form the radio frequency transmission signal; When the phase shifter is an orthogonal architecture, the phase shifter includes: a second signal source, a digital-to-analog conversion module and a transmitting-end mixer; wherein the second signal source is configured to generate a first digital signal; the digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and the transmitting-end mixer is configured to perform frequency shifting and / or phase shifting on the first analog signal based on the second analog signal to form the RF transmission signal.

6. The signal transmission link according to claim 5, It is characterized in that The main transmission path also includes a compensation circuit, wherein the signal input end of the compensation circuit is connected to the second signal source, and the signal input end is connected to the phase shifter, and the compensation circuit is used to combine the compensation signal and the signal output by the second signal source before outputting it.

7. The signal transmission link according to claim 6, It is characterized in that When the signal output by the signal transmission main path is two-way orthogonal signals, the compensation signal used by the compensation circuit is two-way orthogonal signals; When the signal output by the signal transmission main path is not two orthogonal signals, the compensation signal used by the compensation circuit is a signal of the same type as the signal output by the second signal source, wherein the signal type is a digital signal or an analog signal.

8. The signal transmission link according to claim 6, It is characterized in that The compensation circuit comprises a compensation signal generator and an adder, wherein: The compensation signal generator is configured to generate the compensation signal; The adder is connected to the compensation signal generator and the second signal source, and is used to perform a signal superposition operation on the signal output by the second signal source and the compensation signal output by the compensation signal generator.

9. The signal transmission link according to claim 8, Features: When the signal output by the signal transmission main path is two orthogonal signals, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal imbalance compensation signal unit and a local oscillator leakage compensation signal unit; When the signal output by the signal transmission main path is not a two-path orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leakage compensation signal unit; Among them, the compensation signal generated by the harmonic compensation signal unit is used to eliminate the harmonic signal of the main frequency signal in the main signal transmission path; the compensation signal of the local oscillator leakage compensation signal unit is used to compensate for the leakage signal generated by the transmitting end local oscillator signal in the main signal transmission path; the compensation signal generated by the orthogonal imbalance compensation signal unit is used to compensate for the image signal of the main frequency signal in the main signal transmission path.

10. The signal transmission link according to claim 9, Features: The compensation signal generated by the harmonic compensation signal unit has the same frequency and amplitude as the harmonic signal in the signal transmission main path, and has an opposite phase.

11. The signal transmission link according to claim 10, It is characterized in that The harmonic compensation signal unit includes an n-th power module or an n-fold frequency signal generator; An n-th power module, wherein the signal input end of the n-th power module is connected to the signal output end of the second signal source, and is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source using the signal output by the second signal source, to obtain the compensation signal; The n-fold frequency signal generator is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source to obtain the compensation signal, wherein the value of n is a positive integer.

12. The signal transmission link according to claim 11, It is characterized in that The value of n is an odd number.

13. The signal transmission link according to claim 12, It is characterized in that The value of n is 3.

14. The signal transmission link according to claim 9, Features: The compensation signal generated by the local oscillator leakage compensation signal unit is generated based on a leakage signal corresponding to the first analog signal used by the phase shifter.

15. The signal transmission link according to claim 14, Features: The compensation signal generated by the local oscillator leakage compensation signal unit has the same frequency and amplitude as the leakage signal but an opposite phase.

16. The signal transmission link according to claim 9, Features: The compensation signal used by the orthogonal imbalance compensation circuit has the same frequency, the same amplitude, and the opposite phase between the image signal corresponding to the signal transmission main path and the image signal corresponding to the desired signal generated by the signal transmission main path.

17. The signal transmission link according to claim 16, Features: The compensation signal used by the orthogonal imbalance compensation circuit is determined according to the signal output by the second signal source and a complex conjugate signal of a frequency inversion of the signal output by the second signal source.

18. The signal transmission link according to claim 17, Features: The method for obtaining the compensation signal generated by the orthogonal imbalance compensation circuit includes: Obtaining a product of a preset pre-compensation coefficient and the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal; The difference between the signal output by the second signal source and the adjustment signal is calculated to obtain a compensation signal generated by the orthogonal imbalance compensation circuit.

19. The signal transmission link according to claim 18, It is characterized in that The pre-compensation coefficient is determined based on a ratio of an amplitude of a desired signal to an amplitude of an image signal corresponding to the desired signal.

20. The signal transmission link according to any one of claims 1 to 19, Features: The signal calibration link includes a signal receiving link corresponding to the signal transmission main path, wherein the signal receiving link is used to receive and process the echo signal corresponding to the FMCW radio frequency transmission signal; wherein: The signal receiving link is configured to obtain a sampling signal from the signal transmitting main path, and obtain configuration information of a compensation signal according to the sampling signal; The frequencies of the transmitting end local oscillator signal used in the signal transmission main path and the receiving end local oscillator signal used in the signal receiving link are different.

21. The signal transmission link according to claim 20, It is characterized in that The signal calibration link also includes: The frequency adjustment circuit is configured to adjust the frequency of at least one of the sampling signal and the receiving end local oscillator signal.

22. The signal transmission link according to claim 21, Features: The signal input end of the frequency adjustment circuit is connected between the signal transmission main path and the transmission antenna, and the signal output end of the frequency adjustment circuit is connected between the signal receiving link and the receiving antenna, wherein the frequency adjustment circuit is configured to adjust the frequency of the radio frequency transmission signal output by the signal transmission main path; correspondingly, the signal receiving link is configured to obtain a sampling signal from the frequency adjustment circuit, and obtain configuration information of a compensation signal according to the sampling signal; or, The frequency adjustment circuit is connected between the receiving end mixer and the receiving end local oscillator in the signal receiving link, and is configured to adjust the frequency of the receiving end local oscillator signal, wherein the receiving end local oscillator is used to generate the receiving end local oscillator signal, and the receiving end mixer is used to demodulate the received signal using the receiving end local oscillator signal; the signal receiving link is connected between the signal transmission main path and the transmitting antenna, and is configured to obtain a sampling signal from the signal transmission main path, and process the sampling signal using the signal output by the frequency adjustment circuit to obtain configuration information of the compensation signal.

23. The signal transmission link according to any one of claims 20 to 22, Features: When the radio frequency transmission signal transmitted by the signal transmission main path is two orthogonal signals, the signal calibration link is a signal receiving link that supports processing of echo signals including the two orthogonal signals.

24. The signal transmission link according to any one of claims 20 to 22, Features: When the radio frequency transmission signal transmitted by the signal transmission main path is two orthogonal signals, the signal calibration link is provided with an orthogonal processing circuit and two signal receiving links, wherein each signal receiving link does not support processing of orthogonal echo signals; The orthogonal processing circuit is configured to perform radio frequency processing on the received orthogonal signal to obtain two signals, and send the two signals to two signal receiving links respectively.

25. The signal transmission link according to any one of claims 1 to 19, Features: The signal calibration link, the first input end of the signal calibration link is connected between the voltage-to-current converter and the current switch in the transmitting end mixer in the signal transmission main path, the second input end is between the signal transmission main path and the transmitting antenna, and the signal output end is connected to the compensation circuit in the signal transmission main path, and is used to obtain the signal in the signal transmission main path from at least one of the first input end and the second input end, and determine the compensation information based on the obtained signal.

26. The signal transmission link according to claim 25, It is characterized in that The signal calibration link includes a calibration demodulator, a multiplexer and a calibration module; wherein: The calibration demodulator is configured to obtain the signal in the signal transmission main path from the second input terminal and perform demodulation processing; The multiplexer has two signal input terminals and one signal input terminal, one of which is connected between the voltage-to-current converter and the current switch in the transmitting mixer, and the other is connected to the signal input terminal of the calibration demodulator, and is used to output a signal corresponding to one of the two signal input terminals; The calibration module is configured to determine the compensation information according to the signal output by the multiplexer.

27. The signal transmission link according to claim 26, Features: When the radio frequency transmission signal transmitted by the signal transmission main path is two orthogonal signals, the acquisition signal transmitted by the signal calibration link is two orthogonal signals.

28. The signal transmission link according to claim 27, It is characterized in that The acquisition signal transmitted by the signal calibration link is two orthogonal signals, and the signal calibration link also includes: The analog-to-digital converter is connected between the signal acquisition circuit and the calibration module, and is configured to perform analog-to-digital conversion processing on the acquisition signal output by the signal acquisition circuit.

29. A signal transceiver link, It is characterized in that A signal transmission link and a signal receiving link as claimed in any one of claims 1 to 28; The signal receiving link includes a receiving end local oscillator, a receiving end mixer, an analog-to-digital converter, and a receiving end baseband digital module; wherein, The receiving end local oscillator is configured to provide a receiving end local oscillator signal; The receiving end mixer is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal; the echo signal is a signal formed by the signal transmitted by the signal transmission link being reflected and / or scattered by the target object; The analog-to-digital converter is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; The receiving end baseband digital module is configured to process the receiving end digital baseband signal to achieve target detection and / or wireless communication.

30. The signal transceiver link according to claim 29, It is characterized in that The receiving mixer is a real mixer, and the analog-to-digital converter is a real analog-to-digital converter; or, The receiving-end mixer is an orthogonal mixer, and the analog-to-digital converter is an orthogonal analog-to-digital converter.

31. The signal transceiver link according to claim 30, It is characterized in that The receiving end local oscillator signal is a swept frequency signal; or, the receiving end local oscillator signal is a single tone signal.

32. An integrated circuit, It is characterized in that It comprises the signal transmission link as claimed in any one of claims 1 to 28, or the signal transceiver link as claimed in any one of claims 29 to 31.

33. The integrated circuit according to claim 32, wherein the integrated circuit is a millimeter wave radar chip.

34. An electromagnetic wave device, It is characterized in that include: Carrier; An integrated circuit as claimed in claim 32 or 33, arranged on a carrier; An antenna is arranged on the carrier, or the antenna and the integrated circuit are integrated into a single device and arranged on the carrier; the antenna includes a transmitting antenna and a receiving antenna; Wherein, the integrated circuit is connected to the antenna and is used for transmitting electromagnetic wave signals and / or receiving electromagnetic wave signals.

35. A user terminal device, It is characterized in that include: Equipment body; as well as The electromagnetic wave device as claimed in claim 34 is disposed on the device body; Wherein, the electromagnetic wave device is used for target detection and / or wireless communication to provide reference information for the operation of the device body.

36. A signal transmission method, It is characterized in that include: Determining the phase of the radio frequency transmission signal of each of the signal transmission links; Determining the initial phase of the baseband signal in each of the signal transmission links according to the phase of the radio frequency transmission signal; generating an initial baseband signal according to the determined initial phase; After compensating the initial baseband signal using the pre-acquired compensation information, the baseband signal is obtained; A phase shift operation is performed on a transmitting end local oscillator signal based on the baseband signal to obtain the radio frequency transmission signal.

37. A non-transitory computer-readable storage medium having computer-readable instructions stored thereon, which, when executed by a processor, causes the processor to execute the signal transmission method according to claim 36.

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