All-condition DAB inductance parameter optimization selection method based on three-phase shift modulation

Abstract

The invention relates to the technical field of power electronic transformers, and discloses an all-condition DAB inductance parameter optimization selection method based on three-phase-shift modulation, which aims at DAB wide-voltage and full-power-range operation conditions, adopts a three-phase-shift modulation strategy, establishes a stress model and a loss model of each part of the DAB, performs per-unit processing on evaluation indexes under different working conditions, and obtains a DAB inductance parameter optimization selection result. And obtaining an evaluation function for inductance parameter optimization selection, determining evaluation function weight coefficients under different working conditions according to a DAB working scene, and obtaining an optimal inductance parameter value in all working conditions after obtaining an all-working-condition comprehensive evaluation function through weighting. According to the full-working-condition DAB inductance parameter optimization selection method based on three-phase-shift modulation, the current stress and the loss index of the DAB can be optimized within the wide output voltage and output power range, comprehensive high-efficiency operation of the DAB within the wide voltage range and the full working condition is achieved, and device stress is effectively reduced.

Description

Technical Field

[0001] The invention relates to the technical field of power electronic transformers, and in particular to a method for optimizing and selecting DAB inductance parameters under all working conditions based on three-phase shift modulation. Background Art

[0002] Power electronic transformer (PET) usually refers to a new type of power electronic device composed of power electronic current and high-frequency transformer with power conversion and electrical isolation functions. In PET, high-frequency transformer (HFT) is used to replace the industrial frequency transformer, which has the advantages of small size and weight, low noise, low loss, etc. At the same time, it has higher flexibility and can realize the functions of reactive power compensation and harmonic control on the AC side, access to renewable energy and energy storage equipment, fault isolation, etc. Due to the above characteristics, PET has become a key device for realizing power conversion and multi-directional energy transmission in occasions such as smart grids and energy Internet. Isolated DC-DC converter is the core functional link for realizing electrical isolation and voltage level conversion in medium-high voltage and large-capacity modular PET. Among them, dual active bridge (DAB) converter has become the most commonly used topology in this type of converter due to its advantages such as high-frequency isolation, symmetrical structure, and easy soft switching.

[0003] When the energy storage load is connected to the output end of PET, the output voltage and power of DAB vary in a wide range, and the current stress and device loss also increase accordingly; at the same time, with the widespread application of third-generation semiconductor power devices in power electronic transformers, the operating frequency and power of DAB are further improved, and the inductance loss also increases accordingly. High-frequency transformers and additional inductors often occupy the main part of the volume and weight of DAB. From the perspective of optimizing the power density and efficiency of PET, it is necessary to optimize the inductance parameters of DAB in a wide voltage and full power range, so that DAB has relatively low loss and current stress in all working conditions.

[0004] Aiming at the application scenario of DAB wide-range operation, many scholars have optimized its current stress, return power and soft switching characteristics from the perspective of innovative modulation strategies, and proposed a series of optimized modulation methods based on the traditional single phase shift (SPS) modulation, such as dual phase shift (DPS) modulation, three phase shift (TPS) modulation, pulse width modulation (PWM) plus TPS hybrid modulation, etc. However, the current optimization of inductance parameters for DAB is mostly only carried out for the optimization design of the transformer body structure. DAB adopts SPS modulation, which makes it difficult to combine the advantages of optimized modulation strategy with optimized inductance parameters, or does not consider the wide range of voltage and load changes in the actual converter, so the optimal design solution cannot be obtained. Summary of the invention

[0005] The purpose of the present invention is to address the deficiencies of the above-mentioned technology and provide a method for optimizing the selection of DAB inductor parameters under all operating conditions based on three-phase shift modulation, which can optimize the current stress and loss indicators of DAB within a wider output voltage and output power range, achieve comprehensive high-efficiency operation of DAB within a wide voltage range and all operating conditions, and effectively reduce device stress.

[0006] To achieve the above-mentioned purpose, the present invention designs a full-operating-condition DAB inductor parameter optimization selection method based on three-phase-shift modulation. Aiming at the DAB wide voltage and full-power range operating conditions, a three-phase-shift modulation strategy is adopted, and stress models and loss models of various parts of the DAB are established. The evaluation indicators under different operating conditions are normalized to obtain the evaluation function for the optimal selection of inductor parameters. The weight coefficient of the evaluation function under different operating conditions is determined according to the DAB working scenario. After the comprehensive evaluation function of all operating conditions is obtained by weighting, the optimal inductor parameter value in all operating conditions is obtained.

[0007] Preferably, the method comprises the following steps:

[0008] A) According to the operating conditions of DAB, determine the range of its output voltage and output power, specify the voltage transformation ratio of DAB, and calculate its rated current, rated power and rated inductance;

[0009] B) determining constraints based on the DAB operating range determined in step A);

[0010] C) Adopt TPS modulation strategy for DAB and obtain the expression of modulation strategy in all working conditions;

[0011] D) Based on the TPS modulation strategy, the expression of DAB characteristic current under different switching modes in all working conditions can be obtained;

[0012] E) Based on the expressions of the DAB inductor current and the current at the switching moment obtained in step C), the normalized stress model and loss model of each DAB component in the full working condition are obtained by combining the iGSE model and the Dowell model;

[0013] F) Taking the current stress and the losses of each part of the DAB device as the optimization target, the per unit value of the inductance of the DAB as the parameter to be optimized, and establishing an evaluation function;

[0014] G) Selecting the corresponding current or duty cycle index for all optimization targets for simplified representation, and designing weights for different optimization targets according to the loss distribution in the DAB obtained in step E), and finally obtaining a dimensionless evaluation function;

[0015] H) According to the actual use scenario of DAB, weights are set for the proportions of different powers in the full working condition, and the evaluation function is weighted according to the weights to obtain the evaluation function under the current voltage transformation ratio:

[0016] I) optimizing the comprehensive evaluation function according to the constraint conditions obtained in step B), and obtaining the local optimal inductance value under the current voltage transformation ratio;

[0017] J) According to the distribution law of the DAB output voltage determined in step B), weights can be set for different voltage transformation ratios in all operating conditions, and the local optimal inductance values ​​can be weighted averaged to obtain the global optimal inductance value that minimizes electrical stress and loss in all operating conditions.

[0018] Preferably, in step A), the rated current I n , Rated power P n and rated inductance L n as follows:

[0019] I n =mU1 / (4f s L n ), P n =mU1 2 / (8f s L n )

[0020] m=nu o / U1∈[0,2],P=P o / P n ∈[0,1]

[0021] In the formula, u o is the output voltage, P o is the output power, m is the voltage transformation ratio, the range of which is determined by the output capacity of the converter, n is the transformer turns ratio, U1 is the DC input voltage, and f s is the operating frequency of DAB, P is the output power Po per unit value.

[0022] Preferably, in step B), the constraint condition is: the maximum value of the inductor current I max Less than the maximum collector repetitive peak current I of the switching device CRM , the effective value of the inductor current I rms Less than the continuous collector DC current I of the switching device C-nom , control accuracy e control Greater than the minimum accuracy requirement η, the actual output voltage u o 、Current i o and power P o Within the allowable operating range of the converter, that is:

[0023] I max CRM , I rms C-nom , e control <η,i o ≤I n ,u o ≤U max , P o ≤P n

[0024] Where U max The maximum voltage that the converter can output.

[0025] Preferably, in step C), a TPS modulation strategy is adopted for DAB: when the DAB transformation ratio m≠1, TPS modulation is adopted, including two switching modes mode1 and mode2; when m=1, TPS degenerates into SPS modulation, and the expression of the modulation strategy in all working conditions is as follows:

[0026] m=1:D1=D2=1,D f ∈[0,0.5], P = 4D f (1-D f ),(SPS)

[0027]

[0028] Where D1 is the output voltage pulse width of the primary H-bridge of DAB in half a cycle T hs The duty cycle inside, D2 is the duty cycle of the secondary H-bridge output voltage, D f It is the proportion of the primary-secondary voltage phase difference within half a cycle of π.

[0029] Preferably, in step D), the expression of the DAB characteristic current under different switching modes in all working conditions includes the maximum value of the inductor current I max and the effective value of the inductor current I​​rms , the sum of the currents of all devices at the switching moment is I sw , all expressions for the inductance L n The reference value is normalized to per unit, and the expressions of inductor current and switching current are as follows:

[0030]

[0031]

[0032] Preferably, in step E), the conduction loss of all switching devices is:

[0033] P cod_loss (m,P,L k )=R on I rms_sw 2 =R on 2I rms 2 ∝I rms 2 (m,P,L k )

[0034] Where, L k The current inductance is proportional to the rated inductance L n The per unit value, R on is the on-resistance of the switching device or its anti-parallel diode, I rms-sw 2 is the square sum of the effective current values ​​of all devices in one switching cycle. From the inductor current model in step C), we know that I rms-sw 2 =2·I rms 2 , using the normalized inductor current effective value I rms 2 (m,P,L k ) characterizes conduction loss;

[0035] Switching loss model for all switching devices:

[0036] P sw_loss (m,P,L k ) = f s [E on (I sw )+E off (I sw )]U1 / U nom ∝I sw (m,P,L k )

[0037] In the formula, E on is the turn-on energy of the switching device, Eoff is the shut-off energy, U nom is the nominal voltage of the test condition in the device data sheet. By fitting the curve in the manual, the turn-off energy corresponding to a certain turn-off current can be obtained. The secondary switch current is converted to the primary side, and the sum of the total switch currents of all switch devices I sw (m,P,L k ) characterizes the switching loss P of all devices sw_loss ;

[0038] Since the DAB device loss model includes the device current effective value and the switch current, the DAB inductor current maximum value I max Characterizing the electrical stress F of switching devices I :

[0039] F I (m,P,L k )∝I max (m,P,L k )

[0040] Winding loss model of inductor and transformer:

[0041]

[0042] In the formula, I nth is the nth harmonic of the inductor current, F Rnth is the AC resistance coefficient corresponding to the nth harmonic, R DC is the DC resistance of the winding;

[0043] Transformer core loss model:

[0044] P s-HFT (m,P,L k )=P s-HFT (D1,U1)∝D1 1+β1-α1

[0045] Core loss model of inductor:

[0046] P s-La (m,P,L k )=P s-La (D f ,U L )∝(m+1) β2 D f 1+β2-α2

[0047] Where α1 and β1 are the Steinmetz coefficients of the core material used in the high-frequency transformer HFT, and α2 and β2 are the additional inductance L a Steinmetz coefficient of the core material used, U LThe voltage on the additional inductor is DAB.

[0048] Preferably, in step F), the current stress and the losses of each part of the DAB device are optimized, and the L k is the parameter to be optimized, and according to the constraints obtained in DAB step B), the evaluation function f is obtained:

[0049] f=k1F I.pu +k2P con_loss.pu +k3P sw_loss.pu +k4P s.pu +k5P Cu.pu

[0050] k1~k5 are weight coefficients of different optimization objectives, F I.pu is the per unit value of converter current stress, P con_loss.pu P is the per-unit value of device conduction loss, sw_loss.pu is the per-unit switching loss value of the device, P s.pu P is the per unit value of the core loss of magnetic components (including transformers and inductors), Cu.pu is the per unit value of magnetic component winding loss.

[0051] Preferably, in step G), the dimensionless evaluation function is finally obtained as follows:

[0052] f(m,P,L k )=k1I max.pu +k2I rms.pu 2 +k3I sw.pu +k4[D1 1+β1-α1 +(1+m) β2 D f 1+β2-α2 ]

[0053] +k5I rms.pu 2

[0054] In the formula, I max.pu is the maximum per unit value of the inductor current, I rms.pu is the per-unit value of the inductor current, I sw.pu is the per unit value of the switching current of the switching device.

[0055] Preferably, in step H), weights p are set for the proportions of different powers in the full operating condition. i , according to p i The evaluation function is weighted to obtain the evaluation function under the current voltage transformation ratio:

[0056]

[0057] In step I), the comprehensive evaluation function is optimized according to the constraints obtained in step B), and the local optimal inductance value under the current voltage transformation ratio can be obtained: L opt-loc (m).st min{f opt (m,L k )};

[0058] In step J), according to the distribution law of the DAB output voltage determined in step B), weights u can be set for different voltage transformation ratios in the full working condition. i , the local optimal inductance value is weighted averaged to obtain the global optimal inductance value that minimizes electrical stress and loss in all working conditions

[0059] Compared with the prior art, the present invention has the following advantages:

[0060] 1. Combine the TPS optimization modulation strategy with the algorithm for optimizing inductance, and include the electrical stress and loss in the converter into the calculation of the evaluation function, so that the evaluation index is more reasonable and the consideration is more comprehensive;

[0061] 2. Considering the application of DAB in power electronic transformers and other occasions with wide voltage and output power range, the optimization results have a wider range of applicability;

[0062] 3. Under the inductance value taken by this optimization algorithm, efficient operation of DAB can be achieved in all working conditions, effectively reducing the electrical stress and loss of the converter. BRIEF DESCRIPTION OF THE DRAWINGS

[0063] Figure 1 It is a flow chart of the method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to the present invention;

[0064] Figure 2 A topological diagram of DAB in the embodiment;

[0065] Figure 3 Schematic diagram of the operating range of DAB in the embodiment;

[0066] Figure 4 Schematic diagram of typical waveforms of two switching modes of the TPS modulation strategy adopted by the present invention under buck conditions;

[0067] Figure 5 is the loss proportion of each part of DAB calculated by the loss model in the embodiment;

[0068] Figure 6 The evaluation function f calculated by the present invention for the voltage wide range ratio opt Curve, and the local optimal inductance value L opt-loc . DETAILED DESCRIPTION

[0069] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0070] like Figure 2 and Figure 3 As shown in FIG. 1 , the topology and operating range of the dual active bridge converter (DAB) of this embodiment are schematically shown. The DAB is composed of two H-bridges and a high-frequency transformer and its additional inductor. The switch devices T1 to T4 and their anti-parallel diodes D1 to D4 form the primary H-bridge, and the devices S1 to S4 and their anti-parallel diodes VD1 to VD4 form the secondary H-bridge. The DAB generates a time-varying voltage u across the transformer and the additional inductor through the H-bridge circuits on both sides. p and u s To achieve power transmission, the size and direction of DAB power flow can be adjusted by controlling the phase difference between the two.

[0071] Based on the above topology, such as Figure 1 As shown in the figure, a method for optimizing the selection of DAB inductor parameters under all working conditions based on three-phase shift modulation is proposed. Aiming at the DAB wide voltage and full power range operating conditions, a three-phase shift modulation strategy is adopted, and the stress model and loss model of each part of the DAB are established. The evaluation indicators under different working conditions are normalized to obtain the evaluation function for optimizing the selection of inductor parameters. The weight coefficients of the evaluation function under different working conditions are determined according to the DAB working scenario. After the comprehensive evaluation function of all working conditions is obtained by weighting, the optimal inductor parameter value in all working conditions is obtained.

[0072] The specific steps include:

[0073] A) From the operating range of DAB, it can be known that DAB can work in constant current output, constant voltage output, and constant power output. According to the operating conditions of DAB, the range of its output voltage and output power is determined, the voltage transformation ratio of DAB is specified, and its rated current I is calculated. n , Rated power P n and rated inductance L n as follows:

[0074] I n =mU1 / (4f s L n ), P n =mU1 2 / (8f s L n )

[0075] m=nu o / U1∈[0,2],P=P o / P n ∈[0,1]

[0076] In the formula, u o is the output voltage, P o is the output power, m is the voltage transformation ratio, the range of which is determined by the output capacity of the converter, n is the transformer turns ratio, U1 is the DC input voltage, and f s is the operating frequency of DAB, P is the output power P o The per unit value of

[0077] B) Based on the DAB operating range determined in step A), determine the constraints: the maximum inductor current I max Less than the maximum collector repetitive peak current I of the switching device CRM , the effective value of the inductor current I rms Less than the continuous collector DC current I of the switching device C-nom , control accuracy e control Greater than the minimum accuracy requirement η, the actual output voltage u o 、Current i o and power P o Within the allowable operating range of the converter, that is:

[0078] I max CRM , I rms C-nom , e control <η,i o ≤I n ,u o ≤U max , P o ≤P n

[0079] Where U max is the maximum voltage that the converter can output;

[0080] C) Using TPS modulation strategy for DAB can achieve soft switching in all working conditions and effectively reduce current stress and loss, such as Figure 4 As shown in the figure, when the DAB ratio m≠1, TPS modulation is adopted, including two switching modes mode1 and mode2; when m=1, TPS degenerates into SPS modulation, and the three degrees of freedom of TPS modulation are: the pulse width of the DAB primary H-bridge output voltage in half a period T hs The duty cycle D1 within the period, the duty cycle D2 of the secondary H-bridge output voltage, and the proportion D of the original secondary voltage phase difference within half a period of π f ​​The switching mode of the converter under a certain working condition is determined by the output power per unit value P and the DAB ratio m through the expression of the modulation strategy in the full working condition, as follows:

[0081]

[0082] D) Based on the TPS modulation strategy, the expression of DAB characteristic current under different switching modes in all working conditions can be obtained, including the maximum inductor current I max and the effective value of the inductor current I rms , the sum of the currents of all devices at the switching moment is I sw , all expressions for the inductance L n The reference value is normalized to per unit, and the expressions of inductor current and switching current are as follows:

[0083]

[0084]

[0085] E) Based on the expressions of the DAB inductor current and the current at the switching moment obtained in step C), combined with the iGSE model and the Dowell model, the normalized stress model and loss model of each DAB component in the full working condition are obtained:

[0086] Conduction losses of all switching devices:

[0087] P cod_loss (m,P,L k )=R on I rms_sw 2 =R on 2I rms 2 ∝I rms 2 (m,P,L k )

[0088] Where, L k The current inductance is proportional to the rated inductance L n The per unit value, R on is the on-resistance of the switching device or its anti-parallel diode, which is mainly related to the current amplitude and temperature. When the model of the switching device is determined and the losses of the converter under different working conditions are compared, it can be assumed that the on-resistance remains unchanged, I rms-sw 2 is the square sum of the effective current values ​​of all devices in one switching cycle. From the inductor current model in step C), we know that I rms-sw 2 =2·I rms 2 , using the normalized effective value of the inductor current I rms2 (m,P,L k ) characterizes conduction loss;

[0089] The switching loss of a switching device is related to its operating voltage, switching frequency, and current at the switching time. The switching loss model of all switching devices is:

[0090] P sw_loss (m,P,L k ) = f s [E on (I sw )+E off (I sw )]U1 / U nom ∝I sw (m,P,L k )

[0091] In the formula, E on is the turn-on energy of the switching device, E off is the shut-off energy, U nom is the nominal voltage of the test condition in the device data sheet. By fitting the curve in the manual, the turn-off energy corresponding to a certain turn-off current can be obtained. The secondary switch current is converted to the primary side, and the sum of the total switch currents of all switch devices I sw (m,P,L k ) characterizes the switching loss P of all devices sw_loss , the secondary switching current is converted to the primary side;

[0092] Since the DAB device loss model includes the device current effective value and the switch current, the DAB inductor current maximum value I max Characterizing the electrical stress F of switching devices I :

[0093] F I (m,P,L k )∝I max (m,P,L k )

[0094] In order to calculate the winding loss of the inductor under high-frequency non-sinusoidal excitation, the inductor current is usually decomposed by Fourier, and the total winding loss is calculated according to the effective value of each harmonic component and the corresponding AC resistance coefficient. The winding loss model of the inductor and transformer is:

[0095]

[0096] In the formula, I nth is the nth harmonic of the inductor current, F Rnth is the AC resistance coefficient corresponding to the nth harmonic, R DC is the DC resistance of the winding;

[0097] Transformer core loss model:

[0098] P s-HFT (m,P,L k )=P s-HFT (D1,U1)∝D1 1+β1-α1

[0099] Core loss model of inductor:

[0100] P s-La (m,P,L k )=P s-La (D f ,U L )∝(m+1) β2 D f 1+β2-α2

[0101] Where α1 and β1 are the Steinmetz coefficients of the core material used in the high-frequency transformer HFT, and α2 and β2 are the additional inductance L a Steinmetz coefficient of the core material used, U L The voltage on the DAB additional inductor;

[0102] F) Taking the current stress and loss of each part of the DAB device as the optimization target, the inductance per unit value of DAB is L k is the parameter to be optimized, and according to the constraints obtained in DAB step B), the evaluation function f is obtained:

[0103] f=k1F I.pu +k2P con_loss.pu +k3P sw_loss.pu +k4P s.pu +k5P Cu.pu

[0104] k1~k5 are weight coefficients of different optimization objectives, F I.pu is the per unit value of converter current stress, P con_loss.pu P is the per-unit value of device conduction loss, sw_loss.pu is the per-unit switching loss value of the device, P s.pu P is the per unit value of the core loss of magnetic components (including transformers and inductors), Cu.pu is the per unit value of magnetic component winding loss;

[0105] G) All optimization targets are simplified by selecting their corresponding current or duty cycle indicators, and according to the loss distribution in the DAB obtained in step E) and the weight coefficients of different optimization targets, a dimensionless evaluation function is finally obtained:

[0106] f(m,P,L k )=k1Imax.pu +k2I rms.pu 2 +k3I sw.pu +k4[D1 1+β1-α1 +(1+m) β2 D f 1+β2-α2 ]+k5I rms.pu 2

[0107] In the formula, I max.pu is the maximum per unit value of the inductor current, I rms.pu is the per-unit value of the inductor current, I sw.pu is the switching current per unit value of the switching device, Figure 5 is the loss ratio of each part. In this example, the electrical stress weight k1 is designed to be 0.2. According to the loss ratio, the corresponding weights of the losses of other parts can be calculated as follows:

[0108] k1=0.2, k2=0.8×0.1466=0.4238, k3=0.8×0.3109=0.2487

[0109] k4=0.8×0.1593=0.1275, k5=0.8×0.3832=0.3066;

[0110] H) According to the actual use scenario of DAB, set the weight p for the proportion of different powers in the whole working condition. i , weight the evaluation function according to the weight, and obtain the evaluation function under the current voltage transformation ratio

[0111] I) According to the constraints obtained in step B), the comprehensive evaluation function is optimized to obtain the local optimal inductance value under the current voltage transformation ratio: L opt-loc (m).st min{f opt (m,L k )},like Figure 6 As shown in the figure, it is the evaluation function under different voltage ratios in all working conditions and its corresponding optimal inductance value;

[0112] J) According to the distribution law of the DAB output voltage determined in step B), the weights u can be set for different voltage ratios in the full working condition. i , the local optimal inductance value is weighted averaged to obtain the global optimal inductance value that minimizes electrical stress and loss in all working conditions

[0113]

[0114] In addition, in step E), when calculating the core loss model of the inductor and the core loss model of the transformer, according to the iGSE model, the transformer unit iron loss calculation formula under two-level rectangular wave voltage excitation is as follows:

[0115] p s =2 α+β k i f s α B m β D 1-α

[0116] In the formula, α, β and k i is the Steinmetz coefficient of the core material of the magnetic component, f s is the operating frequency of the magnetic component, B m is the peak value of the magnetic density within one working cycle of the magnetic element, and D is the duty cycle of the excitation voltage.

[0117] Since the iron loss of the magnetic components in the DAB under two-level voltage excitation is related to the peak-to-peak value of the magnetic density in one fundamental wave cycle, the peak-to-peak value of the magnetic density in one fundamental wave cycle can be expressed by the duty cycle D and the input voltage U:

[0118] ΔB=2B m =DU / (2f s N p A c )

[0119] Among them, N p is the number of turns on the primary side, A c is the cross-sectional area of ​​the core, and B m Substituting the iron loss calculation formula into the DAB medium and high frequency transformer and the additional inductor core loss expression can be obtained:

[0120] P s-HFT (m,P,L k )=P s-HFT (D1,U1)∝D1 1+β1-α1

[0121] P s-La (m,P,L k )=P s-La (D f ,U L )∝(m+1) β2 D f 1+β2-α2 .

[0122] The present invention is based on the full-operating-condition DAB inductor parameter optimization selection method of three-phase-shift modulation, combines the TPS optimization modulation strategy with the algorithm for optimizing inductance, and simultaneously incorporates the electrical stress and loss in the converter into the calculation of the evaluation function, so that the evaluation index is more reasonable and the consideration is more comprehensive; the application of DAB in a wide voltage and wide output power range in power electronic transformers and other occasions is taken into consideration, and the application scope of the optimization result is wider; under the inductance value taken by the optimization algorithm, the efficient operation of DAB in all operating conditions can be achieved, and the electrical stress and loss of the converter are effectively reduced.

[0123] Finally, it should be pointed out that the above content is a further detailed description of the invention in combination with specific implementation methods. It cannot be considered that the specific implementation of the present invention is limited to these descriptions. For ordinary technicians in the technical field to which the present invention belongs, simple replacements made without departing from the concept of the present invention should be regarded as belonging to the protection scope of the present invention. The above embodiments are only more representative examples of the present invention. Obviously, the present invention is not limited to the above embodiments, and there can be many variations. Any simple modifications, equivalent changes and modifications made to the above embodiments based on the technical essence of the present invention should be considered to belong to the protection scope of the present invention.

[0124] At the same time, it should be noted that the description of the above technical solutions is exemplary, and this specification can be embodied in different forms and should not be construed as being limited to the technical solutions set forth herein. On the contrary, providing these descriptions will make the disclosure of the present invention thorough and complete, and will fully convey the scope disclosed in this specification to those skilled in the art. In addition, the technical solutions of the present invention are limited only by the scope of the claims. The features of the various embodiments of the present invention may be partially or completely combined or spliced ​​with each other, and may be performed in various different configurations as those skilled in the art may fully understand. The embodiments of the present invention may be performed independently of each other, or may be performed together in a mutually dependent relationship.

[0125] For ordinary technicians in the technical field to which the present invention belongs, several simple deductions or substitutions can be made without departing from the concept of the present invention, and the above structures should be regarded as belonging to the protection scope of the present invention.

Claims

1. A method for optimizing DAB inductor parameters under all working conditions based on three-phase shift modulation, characterized in that: Aiming at the DAB wide voltage and full power range operating conditions, a three-phase shift modulation strategy is adopted, the stress model and loss model of each part of the DAB are established, the evaluation indicators under different working conditions are normalized, and the evaluation function for the optimal selection of inductance parameters is obtained. The weight coefficient of the evaluation function under different working conditions is determined according to the DAB working scenario, and the optimal inductance parameter value in all working conditions is obtained after the comprehensive evaluation function of all working conditions is obtained by weighted method.

2. The method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to claim 1 is characterized in that: The steps include: A) According to the operating conditions of DAB, determine the range of its output voltage and output power, specify the voltage transformation ratio of DAB, and calculate its rated current, rated power and rated inductance; B) determining constraints based on the DAB operating range determined in step A); C) Adopt TPS modulation strategy for DAB and obtain the expression of modulation strategy in all working conditions; D) Based on the TPS modulation strategy, the expression of DAB characteristic current under different switching modes in all working conditions can be obtained; E) Based on the expressions of the DAB inductor current and the current at the switching moment obtained in step C), the normalized stress model and loss model of each DAB component in the full working condition are obtained by combining the iGSE model and the Dowell model; F) Taking the current stress and the losses of each part of the DAB device as the optimization target, the per unit value of the inductance of the DAB as the parameter to be optimized, and establishing an evaluation function; G) Selecting the corresponding current or duty cycle index for all optimization targets for simplified representation, and designing weights for different optimization targets according to the loss distribution in the DAB obtained in step E), and finally obtaining a dimensionless evaluation function; H) According to the actual use scenario of DAB, weights are set for the proportions of different powers in the full working condition, and the evaluation function is weighted according to the weights to obtain the evaluation function under the current voltage transformation ratio: I) optimizing the comprehensive evaluation function according to the constraint conditions obtained in step B), and obtaining the local optimal inductance value under the current voltage transformation ratio; J) According to the distribution law of the DAB output voltage determined in step B), weights can be set for different voltage transformation ratios in all operating conditions, and the local optimal inductance values ​​can be weighted averaged to obtain the global optimal inductance value that minimizes electrical stress and loss in all operating conditions.

3. The method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to claim 2 is characterized in that: In step A), the rated current I n , Rated power P n and rated inductance L n as follows: I n =mU1 / (4f s L n ),P n =mU1 2 / (8f s L n ) m=no o / U1∈[0,2],P=P o / P n ∈[0,1] In the formula, u o is the output voltage, P o is the output power, m is the voltage ratio, n is the transformer turns ratio, U1 is the DC input voltage, f s is the operating frequency of DAB, P is the output power P o per unit value.

4. The method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to claim 3 is characterized in that: In step B), the constraint condition is: the maximum value of the inductor current I max Less than the maximum collector repetitive peak current I of the switching device CRM , the effective value of the inductor current I rms Less than the continuous collector DC current I of the switching device C-nom , control accuracy e control Greater than the minimum accuracy requirement η, the actual output voltage u o 、Current i o and power P o Within the allowable operating range of the converter, that is: I max <I CRM ,I rms <I C-nom ,e control <η,i o ≤I n ,u o ≤U max ,P o ≤P n Where U max The maximum voltage that the converter can output.

5. The method for optimizing and selecting DAB inductor parameters under all-operating conditions based on three-phase shift modulation according to claim 4 is characterized in that: In the step C), the TPS modulation strategy is adopted for DAB: when the DAB transformation ratio m≠1, TPS modulation is adopted, including two switching modes mode1 and mode2; when m=1, TPS degenerates into SPS modulation, and the expression of the modulation strategy in all working conditions is as follows: m=1:D1=D2=1,D f ∈[0,0.5],P=4D f (1-D f ),(SPS) Where D1 is the output voltage pulse width of the primary H-bridge of DAB in half a cycle T hs The duty cycle inside, D2 is the duty cycle of the secondary H-bridge output voltage, D f It is the proportion of the primary-secondary voltage phase difference within half a cycle of π.

6. The method for optimizing and selecting DAB inductor parameters under all-operating conditions based on three-phase shift modulation according to claim 5 is characterized in that: In step D), the expression of DAB characteristic current under different switching modes in all working conditions includes the maximum value of inductor current I max and the effective value of the inductor current I rms , the sum of the currents of all devices at the switching moment is I sw , all expressions for the inductance L n The reference value is normalized to per unit, and the expressions of inductor current and switching current are as follows:

7. The method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to claim 6 is characterized in that: In step E), the conduction loss of all switching devices is: P cod_loss (m,P,L k )=R on and rms_sw 2 =R on 2I rms 2 ∝I rms 2 (m,P,L k ) Where, L k The current inductance is proportional to the rated inductance L n The per unit value, R on is the on-resistance of the switching device or its anti-parallel diode, I rms-sw 2 is the square sum of the effective current values ​​of all devices in one switching cycle. From the inductor current model in step C), we know that I rms-sw 2 =2·I rms 2 , using the normalized inductor current effective value I rms 2 (m,P,L k ) characterizes conduction loss; Switching loss model for all switching devices: P sw_loss (m,P,L k )=f s [E on (I sw )+E off (I sw )]U1 / U nom ∝I sw (m,P,L k ) In the formula, E on is the turn-on energy of the switching device, E off is the shut-off energy, U nom is the nominal voltage of the test condition in the device data sheet. By fitting the curve in the manual, the turn-off energy corresponding to a certain turn-off current can be obtained. The secondary switch current is converted to the primary side, and the sum of the total switch currents of all switch devices I sw (m,P,L k ) characterizes the switching loss P of all devices sw_loss ; Since the DAB device loss model includes the device current effective value and the switch current, the DAB inductor current maximum value I max Characterizing the electrical stress F of switching devices I : F I (m,P,L k )∝I max (m,P,L k ) Winding loss model of inductor and transformer: In the formula, I nth is the nth harmonic of the inductor current, F Rnth is the AC resistance coefficient corresponding to the nth harmonic, R DC is the DC resistance of the winding; Transformer core loss model: P s-HFT (m,P,L k )=P s-HFT (D1,U1)∝D1 1+β1-α1 Core loss model of inductor: P s-La (m,P,L k )=P s-La (D f ,U L )∝(m+1) β2 D f 1+β2-α2 Where α1 and β1 are the Steinmetz coefficients of the core material used in the high-frequency transformer HFT, and α2 and β2 are the additional inductance L a Steinmetz coefficient of the core material used, U L The voltage on the added inductor is DAB.

8. The method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to claim 7 is characterized in that: In the step F), the current stress and the loss of each part of the DAB device are optimized. k is the parameter to be optimized, and according to the constraints obtained in DAB step B), the evaluation function f is obtained: f=k1F I.pu +k2P con_loss.pu +k3P sw_loss.pu +k4P s.pu +k5P Cu.pu k1~k5 are weight coefficients of different optimization objectives, F I.pu is the per unit value of converter current stress, P con_loss.pu is the per-unit value of device conduction loss, P sw_loss.pu is the per-unit switching loss value of the device, P s.pu P is the per unit value of the core loss of magnetic components (including transformers and inductors), Cu.pu is the per unit value of magnetic component winding loss.

9. The method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to claim 8 is characterized in that: In the step G), the dimensionless evaluation function is finally obtained as follows: f(m,P,L k )=k1I max.pu +k2I rms.pu 2 +k3I sw.pu +k4[D1 1+β1-α1 +(1+m) β2 D f 1+β2-α2 ]+k5I rms.pu 2 In the formula, I max.pu is the maximum per unit value of the inductor current, I rms.pu is the per-unit value of the inductor current, I sw.pu is the per unit value of the switching current of the switching device.

10. The method for optimizing and selecting DAB inductor parameters under all working conditions based on three-phase shift modulation according to claim 9 is characterized in that: In the step H), weights p are set for the proportions of different powers in the full operating condition. i , according to p i The evaluation function is weighted to obtain the evaluation function under the current voltage transformation ratio: In step I), the comprehensive evaluation function is optimized according to the constraints obtained in step B), and the local optimal inductance value under the current voltage transformation ratio can be obtained: L opt-loc (m).st min{f opt (m,L k )}; In the step J), according to the distribution law of the DAB output voltage determined in step B), the weights u can be set for different voltage transformation ratios in the full working condition. i , the local optimal inductance value is weighted averaged to obtain the global optimal inductance value that minimizes electrical stress and loss in all working conditions The present invention relates to the technical field of power electronic transformers, and discloses a method for optimizing the selection of DAB inductance parameters under all working conditions based on three-phase shift modulation. For the DAB wide voltage and full power range operating conditions, a three-phase shift modulation strategy is adopted, a stress model and a loss model of each part of the DAB are established, and the evaluation indicators under different working conditions are normalized to obtain an evaluation function for optimizing the selection of inductance parameters. The weight coefficient of the evaluation function under different working conditions is determined according to the DAB working scene, and the optimal inductance parameter value in the full working condition is obtained after the comprehensive evaluation function of the full working condition is obtained by weighting. The method for optimizing the selection of DAB inductance parameters under all working conditions based on three-phase shift modulation can optimize the current stress and loss indicators of the DAB within a wider output voltage and output power range, realize the comprehensive high-efficiency operation of the DAB within a wide voltage range and full working conditions, and effectively reduce the device stress.

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