Phase shift frequency modulation control method and device for bidirectional AC-DC converter with wide voltage range and low stress

Abstract

The invention discloses a phase-shift frequency-modulation control method and device for a bidirectional AC-DC converter with a wide voltage range and low stress. According to the phase-shift frequency-modulation control method and device for the bidirectional AC-DC converter with the wide voltage range and the low stress, a variable, namely the alternating current side full-bridge duty ratio D1, is added in traditional expanded phase-shift control to divide the converter into a boost state and a buck state. The method comprises the following steps: respectively solving inductive current energy transmission models in two different working states and mathematical constraint model constraint conditions of an alternating current side duty ratio D1, a direct current side duty ratio D2, a switching frequency f and an inter-bridge phase shift angle delta; therefore, whether the converter works in a boost state or a buck state, a switch tube reset condition, a power factor correction condition and a full AC range soft switching condition can be satisfied. According to the method, while the advantages of single-phase-shift frequency conversion control are guaranteed, the input and output range of the direct-current side voltage is expanded through mutual cooperation of control variables, and the alternating-current side voltage 220V AC and the direct-current side voltage 48-192V are achieved on the basis of not depending on the adjustment range of a transformer. Current stress of a switch tube is reduced, loss of the converter is reduced, and overall efficiency is improved.

Description

Technical Field

[0001] The present invention belongs to the technical field of converter control, and in particular relates to a phase-shift frequency modulation control method and device for a bidirectional AC-DC converter with a wide voltage range and low stress. Background Art

[0002] Traditional DAB type bidirectional AC-DC converters usually work in a boost state, and use the transformer ratio to adjust the input voltage of the secondary side voltage so that the amplitude of the AC side matrix bridge output voltage is smaller than the amplitude of the DC side matrix bridge output voltage converted to the primary side. However, for occasions where the transformer ratio is fixed or the adjustment range is small, it will not be possible to flexibly control the bidirectional AC-DC converter in a boost conversion state. In order to reduce the loss in the energy transfer process, the existing technology generally has a certain amount of reflux power in the modulation of the DAB type bidirectional AC-DC converter to ensure the soft switching characteristics of the converter switch tube, and compensating the reflux power will cause the switch tube to bear greater current stress.

[0003] Phase shift modulation is a common control scheme for DAB converters, but single phase shift frequency conversion control can only work in the boost state. Once the amplitude of the AC side matrix bridge output voltage is greater than the amplitude of the DC side matrix bridge output voltage converted to the primary side, the control strategy will not be able to output normally. In addition, the maximum and effective values ​​of the inductor current of this strategy are relatively large, and the switch stress of the switch tube is relatively high, resulting in large losses in the converter and reducing the energy transmission efficiency of the converter. Summary of the invention

[0004] The present invention proposes a wide voltage range and low stress bidirectional AC-DC converter phase shift frequency modulation control method and device, which are used to solve the problem that the single phase shift frequency conversion control has a small adjustment range and cannot use the transformer for flexible voltage regulation.

[0005] To achieve the above objectives, the embodiments of the present invention provide the following technical solutions:

[0006] In a first aspect, the present invention provides a phase-shift frequency modulation control method for a bidirectional AC-DC converter with a wide voltage range and low stress, the method comprising:

[0007] The converter working state is divided into two states: boost and buck. The AC side full-bridge duty cycle D1 is combined with single frequency conversion phase shift modulation as a variable to solve the instantaneous value, average value and maximum value of the inductor current at time t0 in different control modes under different states.

[0008] According to the switch reset conditions and AC side power factor correction conditions required for each working mode, the phase shift modulation in the boost or buck state, the external phase shift angle δ of the AC side matrix bridge and the DC side full bridge and the duty cycle D2 of the DC side full bridge under each working mode control are calculated respectively, and two working modes that can simultaneously meet the required soft switching characteristics and power factor correction in the boost or buck state are selected;

[0009] Calculate the instantaneous value, maximum value, effective value and AC side current of the inductor current in two working modes of boost or buck;

[0010] Through the instantaneous value, maximum value, effective value of the inductor current and the AC side current of the two working modes of the two working modes of the two states of boost or buck, as well as the external phase shift angle δ and the DC side switch duty cycle D2, the working range, mode switching conditions and AC side matrix bridge duty cycle D1 range that meet the two working modes of the two working modes of the two states of boost or buck are obtained respectively;

[0011] According to the value range of the AC side matrix bridge duty cycle D1 in the two working modes of the boost or buck states, mathematical models of the maximum value of the inductor current, the AC side full bridge duty cycle D1 and the switching frequency f are established respectively, and the appropriate AC side full bridge duty cycle D1 and the switching frequency f are selected through the obtained mathematical model to optimize the maximum value of the inductor current;

[0012] The converter state is selected according to the voltage relationship on both sides of the transformer, and the converter phase-shift frequency modulation control is realized by optimizing the working range of each working mode in each state, the AC side full-bridge duty cycle D1 and the switching frequency f for the maximum value of the inductor current.

[0013] In one implementation, when the converter is in the boost state, the AC side matrix bridge duty cycle D1 and the DC side switch duty cycle D2 satisfy the following relationship:

[0014] 0≤D1≤1&0≤D2≤1&D1≥D2,

[0015] When the converter is in the step-down state, the AC side matrix bridge duty cycle D1 and the DC side switch duty cycle D2 satisfy the following relationship:

[0016] 0≤D1≤1&0≤D2≤1&D1≤D2.

[0017] In one embodiment, two operating modes of boost or buck are selected that can simultaneously meet the required soft switching characteristics and power factor correction, wherein:

[0018] The external phase shift angle δ and the DC side switch duty cycle D2 in working mode 1 are:

[0019]

[0020] The external phase shift angle δ and the DC side switch duty cycle D2 in working mode 2 are:

[0021]

[0022] Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

[0023] In one implementation, the operating ranges, mode switching conditions, and required soft switching ranges of the two operating modes are:

[0024] Working mode range of working mode 1:

[0025]

[0026] Soft switching range in operating mode 1:

[0027]

[0028] Working mode range of working mode 2:

[0029]

[0030] Soft switching range of working mode 2:

[0031]

[0032] Switching conditions between working mode 1 and working mode 2:

[0033] When δ-D2≤1, the converter operates in mode 1; when δ-D2>1, the converter operates in mode 2.

[0034] In one implementation, the boundary value range of the AC side matrix bridge duty cycle D1 of working mode 1 is as follows:

[0035] Boundary 1:

[0036]

[0037] Boundary 2:

[0038]

[0039] Boundary 3:

[0040]

[0041] Boundary 4:

[0042]

[0043] Boundary 5:

[0044] D1=1

[0045] Mode 1 realizes full AC range soft switching. The AC side matrix bridge duty cycle D1 is as follows:

[0046] Boundary 6:

[0047]

[0048] Boundary 7:

[0049]

[0050] The intersection of the boundary range of the AC side matrix bridge duty cycle D1 of mode 1 and the AC side matrix bridge duty cycle D1 of mode 1 realizing full AC range soft switching is taken as the final value range of the AC side matrix bridge duty cycle D1 of mode 1;

[0051] The boundary value range of the AC side matrix bridge duty cycle D1 in working mode 2 is as follows:

[0052] Boundary 1:

[0053]

[0054] Boundary 2:

[0055]

[0056] Boundary 3:

[0057]

[0058] Boundary 4:

[0059] D1=1

[0060] Working mode 2 realizes full AC range soft switching. The AC side matrix bridge duty cycle D1 is as follows:

[0061] Boundary 5:

[0062]

[0063] The intersection of the boundary range of the AC side matrix bridge duty cycle D1 of working mode 2 and the AC side matrix bridge duty cycle D1 of working mode 2 to achieve full AC range soft switching is taken as the final value range of the AC side matrix bridge duty cycle D1 of working mode 2.

[0064] In one implementation, according to the value range of the AC side matrix bridge duty cycle D1 of each working mode in each state, respectively establish a mathematical model of the maximum value of the inductor current, the AC side full bridge duty cycle D1, and the switching frequency f, and select a suitable AC side full bridge duty cycle D1 and the switching frequency f through the obtained mathematical model to optimize the maximum value of the inductor current, including:

[0065] According to the obtained range of the AC side matrix bridge duty cycle D1, the maximum value of the inductor current in each mode can be simplified and a mathematical model of the maximum value of the inductor current and other control variables can be established;

[0066] The maximum value expression of the inductor current in working mode 1 is:

[0067]

[0068] Substitute the external phase shift angle δ of working mode 1 and the DC side switch duty cycle D2 into the above formula and simplify it:

[0069]

[0070] The maximum value expression of the inductor current in working mode 2 is:

[0071]

[0072] Substitute the external phase shift angle δ of working mode 2 and the DC side switch duty cycle D2 into the above formula and simplify it:

[0073] i max =I B

[0074] The maximum value of working mode 2 is obtained by mathematical model as a constant value I B , working mode 1 is a ternary function that is only related to the values ​​of D1, f and ωt;

[0075] According to the obtained mathematical model calculation, the appropriate AC side matrix bridge duty cycle D1 and switching frequency f are selected to optimize the maximum value of the inductor current.

[0076] In one implementation, the converter state selection is performed according to the voltage magnitude relationship between the two sides of the transformer, specifically:

[0077] When v i ≤nV o , that is, 0≤D2≤D1≤1, the converter works in the boost state;

[0078] When v i ≥nV o , that is, 0≤D1≤D2≤1, the converter works in the step-down state;

[0079] Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, and n is the ratio of the primary side to the secondary side of the high-frequency transformer.

[0080] In one implementation, the AC side duty cycle D1 and the switching frequency f of the working mode 1 and the working mode 2 in the optimized boost state should be:

[0081] D1=1

[0082]

[0083] The optimized AC side duty cycle D1 and switching frequency f of mode 1 and mode 2 of the buck state are: Buck state working mode 1:

[0084]

[0085] Buck working state mode 2:

[0086] D1=1

[0087]

[0088] Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

[0089] In a second aspect, the present invention provides a bidirectional AC-DC converter phase-shift frequency modulation control device with a wide voltage range and low stress, the device comprising:

[0090] Calculation module 1 is used to divide the converter working state into two states: boost and buck, and combine the AC side full-bridge duty cycle D1 as a variable with single-frequency phase-shift modulation to solve the instantaneous value, average value and maximum value of the inductor current at time t0 in different control modes under different states;

[0091] The mode screening module is used to calculate the external phase shift angle δ of the AC side matrix bridge and the DC side full bridge and the duty cycle D2 of the DC side full bridge under the control of each working mode of the phase shift modulation in the boost or buck state according to the switch reset condition and AC side power factor correction condition required for each working mode, and screen out two working modes in the boost or buck state that can simultaneously meet the required soft switching characteristics and power factor correction;

[0092] The calculation module 2 is used to calculate the instantaneous value, maximum value, effective value and AC side current of the inductor current in two working modes of the boost or buck state;

[0093] The calculation module three is used to obtain the working range, mode switching condition and AC side matrix bridge duty cycle D1 range of the two working modes ...

[0094] A selection and optimization module is used to establish mathematical models of the maximum value of the inductor current, the AC side full-bridge duty cycle D1, and the switching frequency f according to the value range of the AC side matrix bridge duty cycle D1 of the two working modes of the boost or buck states, and select appropriate AC side full-bridge duty cycle D1 and switching frequency f through the obtained mathematical model to optimize the maximum value of the inductor current;

[0095] The PWM modulation module is used to select the converter state according to the voltage relationship on both sides of the transformer, and realize the phase-shift frequency modulation control of the converter by optimizing the working range of each working mode in each state, the AC side full-bridge duty cycle D1 and the switching frequency f for the maximum value of the inductor current.

[0096] In a third aspect, the present invention provides a storage medium storing computer-readable instructions, which, when executed by one or more processors, enables the one or more processors to perform the steps of the above-mentioned wide voltage range and low stress bidirectional AC-DC converter phase-shift frequency modulation control method.

[0097] Compared with the prior art, the present invention has the following advantages:

[0098] The present invention divides the single-phase frequency conversion control into a boost state and a buck state by using new variables, and respectively establishes the energy transmission model of the inductor current in two different states and the mathematical constraint model of the control variable, and uses a wide voltage range constraint to perform a frequency conversion phase shift method for optimizing the maximum value of the inductor current. This control strategy will ensure the advantages of single-phase frequency conversion control while using the mutual coordination of the control variables to expand the input and output range of the DC side voltage on the basis of not relying on the transformer adjustment range. On this basis, the inductor current maximum value optimization model is used to reduce the effective value of the inductor current, reduce the current stress of the switch tube, reduce the loss of the converter, and improve the overall efficiency. BRIEF DESCRIPTION OF THE DRAWINGS

[0099] The accompanying drawings are part of the present invention and are used to provide a further understanding of the present invention. The exemplary embodiments of the present invention and their descriptions are used to explain the present invention, but do not constitute an improper limitation of the present invention. Obviously, the accompanying drawings described below are only some embodiments. For ordinary technicians in this field, other accompanying drawings can be obtained based on these accompanying drawings without creative work.

[0100] Figure 1 It is a DAB type bidirectional AC-DC converter topology;

[0101] Figure 2 A flow chart of a phase-shift frequency modulation control method for a bidirectional AC-DC converter with wide voltage range and low stress provided by one embodiment of the present invention;

[0102] Figure 3 This is a waveform diagram of an optimized phase-shift frequency conversion mode working mode in one embodiment of the present invention; Figure 3 (a) is the working waveform of working mode 1 in boost state, Figure 3 (b) is the working waveform of working mode 2 in boost state, Figure 3 (c) is the working waveform of working mode 1 in the step-down state, Figure 3 (d) is the working waveform of working mode 2 in the buck state;

[0103] Figure 4 In one embodiment of the present invention, the phase shift frequency conversion mode division is optimized; wherein Figure 4 (a) is the value range of the switching frequency f that needs to be satisfied by the different AC side matrix bridge duty cycles D1 in boost state working mode 1. Figure 4 (b) is the value range of the switching frequency f that needs to be satisfied by the different AC side matrix bridge duty cycles D1 in boost state working mode 2. Figure 4 (c) is the value range of the switching frequency f that needs to be satisfied by the different AC side matrix bridge duty cycles D1 in the buck state working mode 1 and working mode 2;

[0104] Figure 5A flow chart of switching the phase-shift frequency conversion control mode of a bidirectional AC-DC converter in one embodiment of the present invention;

[0105] Figure 6 This is the working state of the phase-shift frequency conversion mode transformer in one embodiment of the present invention; wherein, Figure 6 (a) is the change of voltage and current on both sides of the inductor current in the boost state in working mode 1. Figure 6 (b) is the change of voltage and current on both sides of the inductor current in the boost state in working mode 2; Figure 6 (c) is the change of voltage and current on both sides of the inductor current in the buck state in working mode 1. Figure 6 (d) is the change of voltage and current on both sides of the voltage and current when the voltage is in the step-down state in working mode 2;

[0106] Figure 7 The AC side voltage and current, the DC side voltage and the transformer operating current in the phase-shift frequency conversion mode in one embodiment of the present invention are: Figure 7 (a) is the AC side voltage and current, DC side voltage and transformer operating current in the boost state. Figure 7 (b) AC side voltage and current, DC side voltage and transformer operating current in the step-down state;

[0107] Figure 8 This is the FFT analysis of the AC side current when the phase-shift frequency conversion mode is working in the forward direction in one embodiment of the present invention, where Figure 8 (a) is the waveform of the current FFT analysis with Fundamental (50Hz) = 4.053 and THD = 2.44%. Figure 8 (b) is the waveform of the current FFT analysis with Fundamental (50 Hz) = 4.349 and THD = 3.27%;

[0108] Fig. 9 It is a comparison between the conventional control and the optimized phase-shifted variable frequency modulation inductor current stress optimization in the boost state in one embodiment of the present invention;

[0109] Fig.10 This is a diagram showing the effect of optimizing the phase-shift frequency conversion in one embodiment of the present invention, in which the output voltage can be continuously boosted from 48V to 192V when the input voltage is 110VAC. DETAILED DESCRIPTION

[0110] In order to make the purpose, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. The following embodiments are used to illustrate the present invention but are not used to limit the scope of the present invention.

[0111] In order to deepen the recognition and understanding of the present invention, the technical solution of the present invention is further introduced below in conjunction with the accompanying drawings and specific implementation methods.

[0112] A bidirectional AC-DC converter is a power electronic converter with bidirectional transmission capability that can directly connect the AC grid and DC components. Figure 1 As shown, it consists of eight switch tubes S 1a , S 1b , S 2a , S 2b , S 3a , S 3b , S 4a , S 4b The AC side bidirectional switch matrix is ​​composed of four groups of bidirectional switches, S5~S8 constitutes a DC full bridge, L i , C i The filter on the AC side, C o The DC side voltage stabilizing capacitor, high frequency transformer T and equivalent inductor L are formed, and the DC side voltage source is v i , the DC side voltage source is V o , the voltage on the left side of the equivalent leakage inductance v p , the voltage on the left side of the equivalent leakage inductance v q The transformer ratio is n:1. The bidirectional AC-DC main circuit has the advantages of high power density and low cost.

[0113] Reference Figure 2 As shown, the embodiment of the present invention provides a phase-shift frequency modulation control method for a bidirectional AC-DC converter with a wide voltage range and low stress, and the method steps include:

[0114] S100: The converter working state is divided into two states: boost and buck, and the AC side full-bridge duty cycle D1 is combined with the single-frequency phase-shift modulation as a variable to respectively solve the instantaneous value, average value and maximum value of the inductor current at time t0 of different control modes under different states.

[0115] It should be noted that in this example, since the AC side voltage peak is less than the DC side voltage, the AC side full bridge duty cycle D1 is greater than the DC side full bridge duty cycle D2. The AC side full bridge duty cycle D1 is used as a variable and combined with the external mode under the extended phase shift modulation. At this time, there are 4 possible working modes of triple phase shift. In order to retain the advantages of extended phase shift modulation, a specific working mode will be selected for subsequent calculations.

[0116] When the converter working state is divided into boost state, there are 0≤D1≤1&0≤D2≤1&D1≥D2, the AC side full bridge duty cycle D1 is used as a variable and combined with the external mode under the extended phase shift modulation. There are 4 possible working modes, but only two working modes meet the subsequent constraints. Recorded as boost working mode 1 Figure 3 (a) shows, boost operation mode 2 as Figure 3 Table 1 shows the working range of boost working mode 1 and boost working mode 2, the instantaneous value, average value and maximum value of the inductor current at time t0.

[0117] Table 1

[0118]

[0119] When the converter working state is divided into buck state, there are 0≤D1≤1&0≤D2≤1&D1≤D2, the AC side full bridge duty cycle D1 is used as a variable and combined with the external mode under the extended phase shift modulation. There are 4 possible working modes, but only two working modes meet the subsequent constraints. It is recorded as buck working mode 1. Figure 3 (c) as shown, buck working mode 2 is as follows Figure 3 Table 2 shows the operating range of buck working mode 1 and buck working mode 2, the instantaneous value, average value and maximum value of the inductor current at time t0.

[0120] Table 2

[0121]

[0122] S200: According to the switch reset conditions and AC side power factor correction conditions required for each working mode, the phase shift modulation in the boost or buck state, the external phase shift angle δ of the AC side matrix bridge and the DC side full bridge and the duty cycle D2 of the DC side full bridge under each working mode control are calculated respectively, and two working modes that can simultaneously meet the required soft switching characteristics and power factor correction in the boost or buck state are selected.

[0123] Furthermore, the switch reset conditions and AC side power factor correction conditions required for each mode are obtained through the equivalent circuit model of the converter and the inductor current working waveform.

[0124] Furthermore, at this time, the switch reset conditions required for each mode are:

[0125]

[0126] AC side power factor correction conditions for each mode:

[0127] i i =I ref

[0128] Among them, I B Represents the reset initial value of the inductor current in each switching cycle, I ref =Isinωt, I represents the peak value of the AC side current.

[0129] It should be noted that, by comparing the boost state and the buck state, it is found that the expressions of the converter's working modes are the same but the constraints are different. At this time, the expressions of the external phase shift angle δ and the DC side switch duty cycle D2 for the converter to achieve soft switching and satisfy the power factor correction on the AC side and the working modes of different states are also the same. Therefore, in order to simplify the calculation, the mode expression solution process of the boost state and the buck state will be combined.

[0130] In order to ensure that all switches of the converter can achieve soft switching and meet the power factor correction on the AC side, the expressions of the external phase shift angle δ and the DC side switch duty cycle D2 of the phase-shift frequency conversion control working mode are optimized as follows:

[0131] Working Mode 1:

[0132]

[0133] Working Mode 2:

[0134]

[0135] Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

[0136] S300: Calculate the instantaneous value, maximum value, effective value and AC side current of the inductor current in two working modes of the boost or buck state.

[0137] Table 3 shows the instantaneous value, maximum value, effective value and AC side current of the inductor current in the two working modes of optimized phase-shift frequency conversion.

[0138] Table 3

[0139]

[0140]

[0141] S400: By using the instantaneous value, maximum value, effective value of the inductor current and the AC side current of the two working modes in the two working states of boost or buck, as well as the external phase shift angle δ and the DC side switch duty cycle D2, the working range, mode switching conditions and AC side matrix bridge duty cycle D1 range that meet the two working modes of boost or buck are respectively obtained.

[0142] In order to ensure that the optimized D2 and δ work in different mode ranges of the optimized phase-shift frequency conversion and meet the soft switching condition range of the converter, the working ranges of working mode 1 and working mode 2 in the boost state and the soft switching conditions that need to be met are shown in Table 4.

[0143] Table 4

[0144]

[0145] Working mode switching process reference Figure 5 As shown. The mode switching conditions between working mode 1 and working mode 2 are:

[0146] When δ-D2≤1, the converter operates in operating mode 1, and when δ-D2>1, the converter operates in operating mode 2.

[0147] Furthermore, the boundary value range of D1 in working mode 1 is as follows:

[0148] Boundary 1:

[0149]

[0150] Boundary 2:

[0151]

[0152] Boundary 3:

[0153]

[0154] Boundary 4:

[0155]

[0156] Boundary 5:

[0157] D1=1 (9)

[0158] Working mode 1 realizes full AC range soft switching. The value of D1 is as follows Figure 4 (b), wherein:

[0159] Boundary 6:

[0160]

[0161] Boundary 7:

[0162]

[0163] Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

[0164] The intersection of the boundary range of the AC side matrix bridge duty cycle D1 of working mode 1 and the AC side matrix bridge duty cycle D1 of working mode 1 to achieve full AC range soft switching is taken as the final value range of the AC side matrix bridge duty cycle D1 of working mode 1.

[0165] At this time, the boundary value range of D1 in working mode 2 is as follows Figure 4 (c), where:

[0166] Boundary 1:

[0167]

[0168] Boundary 2:

[0169]

[0170] Boundary 3:

[0171]

[0172] Boundary 4:

[0173] D1=1 (15)

[0174] Working mode 2 realizes full AC range soft switching. The value of D1 is as follows Figure 5 (d), where:

[0175] Boundary 5:

[0176]

[0177] Among them, v i is the AC bridge input voltage value, V ois the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

[0178] The intersection of the boundary range of the AC side matrix bridge duty cycle D1 of working mode 2 and the AC side matrix bridge duty cycle D1 of working mode 2 to achieve full AC range soft switching is taken as the final value range of the AC side matrix bridge duty cycle D1 of working mode 2.

[0179] It should be noted that since the value range of D1 is a quadratic function, it is necessary to pay attention to the boundary condition that Δ<0 always holds. For example, when Δ>0, the soft switch D1 working in mode 1 takes values ​​of boundary 6 and boundary 7, and when Δ<0, the value range of D1 is [0, 1].

[0180] Through the instantaneous value, maximum value, effective value of the inductor current and the AC side current of the two working modes, as well as the external phase shift angle δ and the DC side switch duty cycle D2 of each working mode, the AC side duty cycle D1 range that meets the working range of working mode 1 and working mode 2 and the required soft switching conditions is obtained. At this time, the selection of D1 is related to f.

[0181] S500: According to the value range of the AC side matrix bridge duty cycle D1 of the two working modes of the boost or buck states, mathematical models of the maximum value of the inductor current, the AC side full-bridge duty cycle D1 and the switching frequency f are established respectively, and the appropriate AC side full-bridge duty cycle D1 and the switching frequency f are selected through the obtained mathematical model to optimize the maximum value of the inductor current.

[0182] In the embodiment of the present application, the step S500 specifically includes the following steps:

[0183] According to the obtained range of the AC side matrix bridge duty cycle D1, the maximum value of the inductor current in each mode can be simplified and a mathematical model of the maximum value of the inductor current and other control variables can be established;

[0184] The maximum value expression of the inductor current in working mode 1 is:

[0185]

[0186] Substitute the external phase shift angle δ of working mode 1 and the DC side switch duty cycle D2 into the above formula and simplify it:

[0187]

[0188] The maximum value expression of the inductor current in working mode 2 is:

[0189]

[0190] Substitute the external phase shift angle δ of working mode 2 and the DC side switch duty cycle D2 into the above formula and simplify it:

[0191] i max =I B (20)

[0192] The maximum value of working mode 2 is obtained by mathematical model as a constant value I B , Mode 1 is a ternary function that is only related to the values ​​of D1, f and ωt;

[0193] According to the obtained mathematical model calculation, the appropriate AC side matrix bridge duty cycle D1 and switching frequency f are selected to optimize the maximum value of the inductor current.

[0194] Specifically, according to the obtained mathematical model of the maximum inductor current in working modes 1 and 2 under the buck-boost state, and the value range of the AC side duty cycle D1 when working in modes 1 and 2 under different buck-boost states obtained in step 400, the variation law of the inductor current with D1 and f is obtained and the appropriate AC side matrix bridge duty cycle D1 and switching frequency f are selected within the range to optimize the maximum inductor current.

[0195] Furthermore, when t is constant, the AC side matrix bridge duty cycle D1 of working mode 1 is close to 1, and the switching frequency should be within a range that takes into account the reduction of inductor current stress.

[0196] Specifically, D1 and f are selected and appropriately used to optimize the maximum value of the inductor current. The maximum value of the inductor current in each mode can be simplified by combining the value range of D1, and a mathematical model of the maximum value of the inductor current and other control variables is established. Through the mathematical model, the maximum value of working mode 2 can be obtained as a constant I B , while working mode 1 is a ternary function that is only related to the values ​​of D1, f and ωt. When t is constant, the larger the AC side duty cycle D1 of working mode 1, the smaller the inductor current, and the larger f, the smaller the inductor current. In order to reduce the inductor current stress, the closer the AC side matrix bridge duty cycle D1 (0≤D1≤1) of working mode 1 is to 1, the better, and the larger the switching frequency f, the better. However, too high a switching frequency f will increase the switching loss of the converter. The switching frequency should be reasonably selected within the range of considering reducing the inductor current stress. At this time, it is necessary to select the switching frequency that needs to be met by the different AC side matrix bridge duty cycles D1 in different modes, and reduce the switching frequency as much as possible to reduce the loss while reducing the current stress.

[0197] Through the instantaneous value, maximum value, effective value of the inductor current and the AC side current of the two working modes, as well as the external phase shift angle δ and the DC side switch duty cycle D2 of each working mode, the AC side duty cycle D1 range that meets the working range of working mode 1 and working mode 2 and the required soft switching conditions is obtained. At this time, the selection of D1 is related to f.

[0198] At this time, the value range of the switching frequency f in boost working mode 1 is as follows: Figure 4 (a), where:

[0199] Boundary 1:

[0200]

[0201] Boundary 2:

[0202]

[0203] Boundary 3:

[0204]

[0205] The value range of switching frequency f in boost mode 2 is as follows: Figure 4 (b), wherein:

[0206]

[0207] The switching frequency f in the buck state has a range of values ​​such as Figure 4 (c), where:

[0208] Boundary 1:

[0209]

[0210] Boundary 2:

[0211]

[0212] Boundary 3:

[0213]

[0214] The switching of the value of f between boundary 1 and boundary 2 is also the process of switching the converter between working mode 1 and working mode 2 in the buck state.

[0215] After comprehensive consideration, the AC side duty cycle D1 and switching frequency f of working mode 1 and working mode 2 in the boost state should be:

[0216] D1=1 (28)

[0217]

[0218] Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

[0219] The AC side duty cycle D1 and switching frequency f of working mode 1 and working mode 2 in the buck state should be:

[0220] Buck state working mode 1:

[0221]

[0222] Buck state working mode 2:

[0223] D1=1

[0224]

[0225] S600: The converter state is selected according to the voltage magnitude relationship between the two sides of the transformer, and the converter phase shift frequency modulation control is realized by optimizing the working range of each working mode in each state, the AC side full bridge duty cycle D1 and the switching frequency f for the maximum value of the inductor current.

[0226] Continue to refer to Figure 5 The converter state is selected according to the voltage relationship on both sides of the transformer, specifically:

[0227] When v i ≤nV o , that is, 0≤D2≤D1≤1, the converter works in the boost state;

[0228] When v i ≥nV o , that is, 0≤D1≤D2≤1, the converter works in the step-down state;

[0229] Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, and n is the ratio of the primary side to the secondary side of the high-frequency transformer.

[0230] Specifically, the duty cycle D2 of the DC side full bridge and the inter-bridge phase shift angle δ obtained in the above step S200 and the duty cycle D1 and the switching frequency f of the AC side matrix bridge obtained in step S500 are substituted into the converter for phase-shift frequency conversion modulation.

[0231] The present invention analyzes the traditional single-phase variable frequency control. This scheme adds the variable of AC side full-bridge duty cycle D1 to the traditional extended phase shift control to divide the converter into a boost state and a buck state. The inductor current energy transmission model of two different working states and the mathematical constraint model constraints of AC side duty cycle D1, DC side duty cycle D2, switching frequency f and bridge phase shift angle δ are solved respectively, so that the converter can meet the switch tube reset condition, power factor correction condition and full AC range soft switch whether it is working in the boost state or the buck state. The solved control variable constraint condition is established and used to optimize the inductor current maximum value, and a variable frequency phase shift method for optimizing the inductor current maximum value that meets the wide voltage range constraint range is established. This control strategy will ensure the advantages of single-phase variable frequency control while using the mutual cooperation of control variables to expand the input and output range of DC side voltage, and achieve an AC side voltage of 220VAC and a DC side voltage of 48~192V without relying on the transformer adjustment range. On this basis, the inductor current maximum optimization model is used to reduce the effective value of the inductor current, reduce the current stress of the switch tube, reduce the loss of the converter, and improve the overall efficiency.

[0232] In order to further illustrate the technical solution of the present invention, a Matlab simulation model of a bidirectional AC-DC converter with optimized phase-shift frequency conversion control and traditional single-phase-shift frequency conversion control is built below. The simulation platform parameters are shown in Table 5.

[0233] Table 5

[0234] <![CDATA[Input voltage V i > <![CDATA[110V ac ]]> <![CDATA[Output voltage V o > 48～192V <![CDATA[Reset current I B > 5A Transformer ratio n 1:1 <![CDATA[Equivalent leakage inductance L f > 100μH <![CDATA[AC side reference current I ref > 4A

[0235] Taking forward operation as an example, the converter transfers energy from the AC side to the DC side under the control of optimized phase-shift frequency conversion. Figure 6 (a) shows the change of voltage and current on both sides of the inductor current when the working boost state is in mode 1. Figure 6 (b) shows the change of voltage and current on both sides of the inductor current when working in mode 2. Figure 6 (c) shows the change of voltage and current on both sides of the inductor current when the working buck state is in mode 1. Figure 6 (d) shows the change of voltage and current on both sides of the inductor current when working in mode 2. When the DC side is connected to the rated load, the AC side transfers energy to the DC side. At this time, the AC side voltage and current and the DC side voltage waveforms are shown in the figure. Combined with the simulation results in Table 5, it can be seen that the modulation method of the present invention greatly broadens the voltage output range of the DC side of the converter compared with the single phase shift control. Figure 7(a) Phase shift frequency conversion boost state and Figure 7 (b) The AC voltage and current waveforms in the phase-shifted frequency conversion step-down state. It can be clearly observed that under the control of the optimized phase-shifted frequency conversion, the converter AC voltage and the AC output current are in phase and the DC output voltage is the rated voltage. At the same time, the current waveform distortion rate of the AC side current FFT analysis is less than 5%. The current FFT analysis waveform is as follows: Figure 8 (a) Figure 8 The simulation results are shown in Table 6.

[0236] Table 6

[0237]

[0238]

[0239] According to the simulation results in Table 6, when the converter works in the boost state, the effective value of the inductor current, the inductor current stress and the switching frequency of the switch tube can be effectively reduced within the full power range. Fig. 9 By comparing the inductor current stress and switching frequency simulation waveforms of the present invention with those of the traditional modulation method, it can be found that the modulation method of the present invention has significantly lower inductor current stress than the traditional single-phase-shift modulation; the modulation method of the present invention also has significantly lower switching frequency than the traditional single-phase-shift modulation. At this point, it can be concluded that the modulation method of the present invention does significantly reduce the switching frequency and inductor current of the switch tube, reduces the switching loss of the switch tube, reduces the current loss of the converter, and improves the conversion efficiency of the converter compared to the single-phase-shift frequency conversion. In addition, Fig.10 It means that after pre-charging the converter to 48V, the converter can operate in buck mode and boost mode, continuously adjusting the output level from 48 to 192V.

[0240] It should be noted that the maximum value of the inductor current under this control is also the point where the DC side full-bridge switch tube is subjected to the maximum stress. While controlling the transformer inductor current in the critical current mode to solve the full-range voltage soft switching of the bidirectional AC-DC converter, the transformer inductor current stress and the DC side full-bridge switch off current are reduced by optimization, thereby improving the efficiency of the converter's bidirectional energy transfer and reducing the loss during energy flow.

[0241] The following is an embodiment of a bidirectional AC-DC converter phase-shift frequency modulation control device with a wide voltage range and low stress of the present invention, which can be used to execute an embodiment of a bidirectional AC-DC converter phase-shift frequency modulation control method with a wide voltage range and low stress of the present invention. For details not disclosed in the embodiment of the bidirectional AC-DC converter phase-shift frequency modulation control device with a wide voltage range and low stress of the present invention, please refer to the embodiment of the bidirectional AC-DC converter phase-shift frequency modulation control method with a wide voltage range and low stress of the present invention.

[0242] In one embodiment, a bidirectional AC-DC converter phase-shift frequency modulation control device with wide voltage range and low stress is proposed, the device comprising:

[0243] Calculation module 1 is used to divide the converter working state into two states: boost and buck, and combine the AC side full-bridge duty cycle D1 as a variable with single-frequency phase-shift modulation to solve the instantaneous value, average value and maximum value of the inductor current at time t0 in different control modes under different states;

[0244] The mode screening module is used to calculate the external phase shift angle δ of the AC side matrix bridge and the DC side full bridge and the duty cycle D2 of the DC side full bridge under the control of each working mode of the phase shift modulation in the boost or buck state according to the switch reset condition and AC side power factor correction condition required for each working mode, and screen out two working modes in the boost or buck state that can simultaneously meet the required soft switching characteristics and power factor correction;

[0245] The calculation module 2 is used to calculate the instantaneous value, maximum value, effective value and AC side current of the inductor current in two working modes of the boost or buck state;

[0246] The calculation module three is used to obtain the working range, mode switching condition and AC side matrix bridge duty cycle D1 range of the two working modes ...

[0247] A selection and optimization module is used to establish mathematical models of the maximum value of the inductor current, the AC side full-bridge duty cycle D1, and the switching frequency f according to the value range of the AC side matrix bridge duty cycle D1 of the two working modes of the boost or buck states, and select appropriate AC side full-bridge duty cycle D1 and switching frequency f through the obtained mathematical model to optimize the maximum value of the inductor current;

[0248] The PWM modulation module is used to select the converter state according to the voltage relationship on both sides of the transformer, and realize the phase-shift frequency modulation control of the converter by optimizing the working range of each working mode in each state, the AC side full-bridge duty cycle D1 and the switching frequency f for the maximum value of the inductor current.

[0249] In one embodiment, a storage medium storing computer-readable instructions is proposed. When the computer-readable instructions are executed by one or more processors, the one or more processors execute the steps of the above-mentioned wide voltage range and low stress bidirectional AC-DC converter phase-shift frequency modulation control method.

[0250] Those skilled in the art can understand that all or part of the processes in the above-mentioned embodiments can be implemented by instructing the relevant hardware through a computer program, and the computer program can be stored in a computer-readable storage medium. When the program is executed, it can include the processes of the embodiments of the above-mentioned methods. Among them, the aforementioned storage medium can be a non-volatile storage medium such as a disk, an optical disk, a read-only memory (ROM), or a random access memory (RAM).

[0251] The technical features of the above embodiments may be combined arbitrarily. To make the description concise, not all possible combinations of the technical features in the above embodiments are described. However, as long as there is no contradiction in the combination of these technical features, they should be considered to be within the scope of this specification.

[0252] The above-mentioned embodiments only express several implementation methods of the present invention, and the description thereof is relatively specific and detailed, but it cannot be understood as limiting the scope of the patent of the present invention. It should be pointed out that, for ordinary technicians in this field, several variations and improvements can be made without departing from the concept of the present invention, which all belong to the protection scope of the present invention. Therefore, the protection scope of the patent of the present invention shall be subject to the attached claims.

Claims

1. A phase-shift frequency modulation control method for a bidirectional AC-DC converter with a wide voltage range and low stress, characterized in that: The method comprises: The converter working state is divided into two states: boost and buck. The AC side full-bridge duty cycle D1 is combined with single frequency conversion phase shift modulation as a variable to solve the instantaneous value, average value and maximum value of the inductor current at time t0 in different control modes under different states. According to the switch reset conditions and AC side power factor correction conditions required for each working mode, the phase shift modulation in the boost or buck state, the external phase shift angle δ of the AC side matrix bridge and the DC side full bridge and the duty cycle D2 of the DC side full bridge under each working mode control are calculated respectively, and two working modes that can simultaneously meet the required soft switching characteristics and power factor correction in the boost or buck state are selected; Calculate the instantaneous value, maximum value, effective value and AC side current of the inductor current in two working modes of boost or buck; Through the instantaneous value, maximum value, effective value of the inductor current and the AC side current of the two working modes of the two working modes of the two states of boost or buck, as well as the external phase shift angle δ and the DC side switch duty cycle D2, the working range, mode switching conditions and AC side matrix bridge duty cycle D1 range that meet the two working modes of the two working modes of the two states of boost or buck are obtained respectively; According to the value range of the AC side matrix bridge duty cycle D1 in the two working modes of the boost or buck states, mathematical models of the maximum value of the inductor current, the AC side full bridge duty cycle D1 and the switching frequency f are established respectively, and the appropriate AC side full bridge duty cycle D1 and the switching frequency f are selected through the obtained mathematical model to optimize the maximum value of the inductor current; The converter state is selected according to the voltage relationship on both sides of the transformer, and the converter phase-shift frequency modulation control is realized by optimizing the working range of each working mode in each state, the AC side full-bridge duty cycle D1 and the switching frequency f for the maximum value of the inductor current.

2. The phase-shift frequency modulation control method for a bidirectional AC-DC converter with wide voltage range and low stress according to claim 1, characterized in that: When the converter is in the boost state, the AC side matrix bridge duty cycle D1 and the DC side switch duty cycle D2 satisfy the following relationship: 0≤D1≤1&0≤D2≤1&D1≥D2, When the converter is in the step-down state, the AC side matrix bridge duty cycle D1 and the DC side switch duty cycle D2 satisfy the following relationship: 0≤D1≤1&0≤D2≤1&D1≤D2.

3. The phase-shift frequency modulation control method for a bidirectional AC-DC converter with wide voltage range and low stress according to claim 2, characterized in that: Screen out two working modes that can simultaneously meet the required soft switching characteristics and power factor correction in the boost or buck state, among which, The external phase shift angle δ and the DC side switch duty cycle D2 in working mode 1 are: The external phase shift angle δ and the DC side switch duty cycle D2 in working mode 2 are: Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

4. The wide voltage range and low stress bidirectional AC-DC converter phase shift frequency modulation control method according to claim 3 is characterized in that: The operating ranges, mode switching conditions and required soft switching ranges of the two operating modes are: Working mode range of working mode 1: Soft switching range in operating mode 1: Working mode range of working mode 2: Wherein, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, and T is the converter switching period; Soft switching range of working mode 2: Among them, i L (t0) is the instantaneous value of the inductor current at time t0, i L (t1) is the instantaneous value of the inductor current at time t0, i L (t2) is the instantaneous value of the inductor current at time t0; Switching conditions between working mode 1 and working mode 2: When δ-D2≤1, the converter operates in mode 1; when δ-D2>1, the converter operates in mode 2.

5. The wide voltage range and low stress bidirectional AC-DC converter phase shift frequency modulation control method according to claim 4, characterized in that: The boundary value range of the AC side matrix bridge duty cycle D1 in working mode 1 is as follows: Boundary 1: Boundary 2: Boundary 3: Boundary 4: Boundary 5: D1＝1 Working mode 1 realizes full AC range soft switching. The AC side matrix bridge duty cycle D1 is as follows: Boundary 6: Boundary 7: Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance value of the transformer leakage inductance; The intersection of the boundary range of the AC side matrix bridge duty cycle D1 of working mode 1 and the AC side matrix bridge duty cycle D1 of working mode 1 realizing full AC range soft switching is taken as the final value range of the AC side matrix bridge duty cycle D1 of working mode 1; The boundary value range of the AC side matrix bridge duty cycle D1 in working mode 2 is as follows: Boundary 1: Boundary 2: Boundary 3: Boundary 4: D1＝1 Working mode 2 realizes full AC range soft switching. The AC side matrix bridge duty cycle D1 is as follows: Boundary 5: Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance value of the transformer leakage inductance; The intersection of the boundary range of the AC side matrix bridge duty cycle D1 of working mode 2 and the AC side matrix bridge duty cycle D1 of working mode 2 to achieve full AC range soft switching is taken as the final value range of the AC side matrix bridge duty cycle D1 of working mode 2.

6. The wide voltage range and low stress bidirectional AC-DC converter phase shift frequency modulation control method according to claim 5, characterized in that: According to the value range of the AC side matrix bridge duty cycle D1 of each working mode in each state, mathematical models of the maximum value of the inductor current, the AC side full bridge duty cycle D1 and the switching frequency f are established respectively, and the appropriate AC side full bridge duty cycle D1 and the switching frequency f are selected through the obtained mathematical model to optimize the maximum value of the inductor current, including: According to the obtained AC side matrix bridge duty cycle D1 value range, the inductor current maximum value in each mode can be simplified and a mathematical model of the inductor current maximum value and other control variables can be established; among which, The maximum value expression of the inductor current in working mode 1 is: Substitute the external phase shift angle δ of working mode 1 and the DC side switch duty cycle D2 into the above formula and simplify it: The maximum value expression of the inductor current in working mode 2 is: Substitute the external phase shift angle δ of working mode 2 and the DC side switch duty cycle D2 into the above formula and simplify it: i=I B max The maximum value of working mode 2 is obtained by mathematical model as a constant value I B , working mode 1 is a ternary function that is only related to the values ​​of D1, f and ωt; According to the obtained mathematical model calculation, the appropriate AC side matrix bridge duty cycle D1 and switching frequency f are selected to optimize the maximum value of the inductor current.

7. The phase-shift frequency modulation control method for a bidirectional AC-DC converter with wide voltage range and low stress according to claim 1, characterized in that: The converter state selection is performed according to the voltage magnitude relationship between the two sides of the transformer, specifically: When v i ≤nV o , that is, 0≤D2≤D1≤1, the converter works in the boost state; When v i ≥nV o , that is, 0≤D1≤D2≤1, the converter works in the step-down state; Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, and n is the ratio of the primary side to the secondary side of the high-frequency transformer.

8. The wide voltage range and low stress bidirectional AC-DC converter phase shift frequency modulation control method according to claim 6, characterized in that: The AC side duty cycle D1 and switching frequency f of the optimized boost state working mode 1 and working mode 2 should be: D1＝1 The AC side duty cycle D1 and switching frequency f of the optimized step-down working mode 1 and working mode 2 are: Buck state working mode 1: Buck state working mode 2: D1＝1 Among them, v i is the AC bridge input voltage value, V o is the output voltage of the DC bridge, δ is the phase shift angle between the AC bridge and the DC bridge, D1 is the duty cycle of the AC bridge switch, D2 is the duty cycle of the DC bridge switch, n is the ratio of the primary side to the secondary side of the high-frequency transformer, f is the converter switching frequency, I ref is the reference current value of the AC side, I ref =Isinωt, I represents the peak value of the AC side current, I B It represents the reset initial value of the inductor current in each switching cycle, and L is the equivalent inductance of the transformer leakage inductance.

9. A bidirectional AC-DC converter phase-shift frequency modulation control device with wide voltage range and low stress, characterized in that: The device comprises: Calculation module 1 is used to divide the converter working state into two states: boost and buck, and combine the AC side full-bridge duty cycle D1 as a variable with single-frequency phase-shift modulation to solve the instantaneous value, average value and maximum value of the inductor current at time t0 in different control modes under different states; The mode screening module is used to calculate the external phase shift angle δ of the AC side matrix bridge and the DC side full bridge and the duty cycle D2 of the DC side full bridge under the control of each working mode of the phase shift modulation in the boost or buck state according to the switch reset condition and AC side power factor correction condition required for each working mode, and screen out two working modes in the boost or buck state that can simultaneously meet the required soft switching characteristics and power factor correction; The calculation module 2 is used to calculate the instantaneous value, maximum value, effective value and AC side current of the inductor current in two working modes of the boost or buck state; The calculation module three is used to obtain the working range, mode switching condition and AC side matrix bridge duty cycle D1 range of the two working modes ... A selection and optimization module is used to establish mathematical models of the maximum value of the inductor current, the AC side full-bridge duty cycle D1, and the switching frequency f according to the value range of the AC side matrix bridge duty cycle D1 of the two working modes of the boost or buck states, and select appropriate AC side full-bridge duty cycle D1 and switching frequency f through the obtained mathematical model to optimize the maximum value of the inductor current; The PWM modulation module is used to select the converter state according to the voltage relationship on both sides of the transformer, and realize the phase-shift frequency modulation control of the converter by optimizing the working range of each working mode in each state, the AC side full-bridge duty cycle D1 and the switching frequency f for the maximum value of the inductor current.

10. A storage medium storing computer-readable instructions, characterized in that: When the computer-readable instructions are executed by one or more processors, the one or more processors execute the steps of the phase-shift frequency modulation control method for a wide voltage range and low stress bidirectional AC-DC converter as claimed in any one of claims 1 to 8.

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