Double-loop control method, system and equipment of power electronic converter and medium

Abstract

The invention provides a double-loop control method, system and device of a power electronic converter and a medium, and relates to the field of power electronic converters, and the method comprises the steps: obtaining a voltage frequency domain model and a current loop controller frequency domain model of the converter, and obtaining a current loop transfer function; obtaining a current frequency domain model, and performing linearization processing at the steady-state working point to obtain a linearization model; substituting the linearization model into the current loop transfer function to obtain a relation between the reference current and the output quantity so as to obtain a voltage loop control coupling term and further obtain a decoupling controller; and performing voltage loop and current loop control on the power electronic converter through the decoupling controller. Compared with the prior art, the current loop transfer function is deduced through the voltage frequency domain model and the current loop controller frequency domain model of the power electronic converter, coupling terms introduced by links such as the current loop and a filter for the voltage loop can be clarified, and the decoupling control effect of the d-axis component and the q-axis component is optimized.

Description

Technical Field

[0001] This invention application relates to the field of power electronic converters, and more particularly to a dual-loop control method, system, device, and medium for power electronic converters. Background Technology

[0002] In recent years, power electronic converters have been widely used in new energy power generation. In order to enable the converter voltage and current to track the given parameters quickly and accurately, dual-loop control of voltage and current is usually required to independently control its d-axis and q-axis components.

[0003] However, current technology assumes that the current loop has an extremely fast control response, and the converter output current can track the voltage loop output reference current value in real time without overshoot. This means that the voltage loop output reference current and the converter output current transfer function are equivalent to 1, ignoring the influence of the current loop, filters, and other components on the voltage loop. Therefore, the decoupling control effect of the d-axis component and the q-axis component is not good. Summary of the Invention

[0004] This invention application provides a dual-loop control method, system, device, and medium for power electronic converters to solve the technical problem of how to optimize the decoupling control effect of the d-axis and q-axis components.

[0005] To address the aforementioned technical problems, this invention provides a dual-loop control method for a power electronic converter, comprising:

[0006] Obtain the voltage frequency domain model of the power electronic converter, and obtain the current loop controller frequency domain model of the power electronic converter;

[0007] Based on the voltage frequency domain model and the current loop controller frequency domain model, the current loop transfer function is obtained;

[0008] Obtain the current frequency domain model of the power electronic converter, and linearize the current frequency domain model at the steady-state operating point to obtain a linearized model; substitute the linearized model into the current loop transfer function to obtain the relationship between the reference current and the output quantity;

[0009] Based on the relationship between the reference current and the output, a voltage loop control coupling term is obtained, and then a decoupling controller is obtained based on the voltage loop control coupling term; the power electronic converter is controlled by the decoupling controller for voltage loop and current loop.

[0010] As a preferred embodiment, obtaining the voltage frequency domain model of the power electronic converter includes:

[0011] Obtain the voltage time-domain model of the power electronic converter, and perform a Laplace transform on the voltage time-domain model to obtain the voltage frequency-domain model.

[0012] As a preferred embodiment, the voltage time-domain model is expressed as:

[0013]

[0014] Among them, e d e is the d-axis component of the voltage at the neutral point of the power electronic converter. q L represents the q-axis component of the voltage at the neutral point of the power electronic converter. f For filter inductance, R f The parasitic resistance of the filter inductor is given by ω0, where ω0 is the rated angular frequency of the power grid, and u is the value of ω. sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld i represents the d-axis component of the filter inductor current. lq Let t be the q-axis component of the filter inductor current, and t be time.

[0015] As a preferred embodiment, the current loop transfer function is specifically:

[0016]

[0017] Where, △i ld * For the small perturbation component of the d-axis reference current, Δi lq * Let be the small perturbation component of the q-axis reference current, s be the Laplace operator, and G be the small perturbation component of the q-axis reference current. cl Let Δi be the transfer function of the proportional-integral controller in the current loop. ld For the small perturbation component of the d-axis component of the filter inductor current, Δi lq This is the small perturbation component of the q-axis component of the filter inductor current.

[0018] As a preferred embodiment, obtaining the current frequency domain model of the power electronic converter includes:

[0019] Obtain the current time-domain model of the power electronic converter, and perform a Laplace transform on the current time-domain model to obtain the current frequency-domain model.

[0020] As a preferred embodiment, the current time-domain model is expressed as:

[0021]

[0022] Among them, i ld i represents the d-axis component of the filter inductor current. lq C is the q-axis component of the filter inductor current. f Here, t is the filter capacitor, ω0 is the rated angular frequency of the power grid, and i is the frequency of the filter capacitor.od i represents the d-axis component of the grid-connected current. oq For the q-axis component of the grid-connected current, u sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld This represents the d-axis component of the filter inductor current.

[0023] As a preferred embodiment, the relationship between the reference current and the output quantity is specifically as follows:

[0024]

[0025] Where, △i ld * For the small perturbation component of the d-axis reference current, i lq * Classification of small perturbations to the q-axis reference current, Δu sd For the small disturbance component of the d-axis component of the power electronic converter, Δu sq G represents the small perturbation component of the q-axis component of the power electronic converter, s is the Laplace operator, and G... cl Let L be the transfer function of the proportional-integral controller in the current loop. f For filter inductance, R f C is the parasitic resistance of the filter inductor. f For the filter capacitor, Δi od For the small disturbance component of the d-axis component of the grid-connected current, Δi oq This represents the small disturbance component of the q-axis component of the grid-connected current.

[0026] Accordingly, this invention application also provides a dual-loop control system for a power electronic converter, including a model acquisition module, a current transfer function acquisition module, a relationship acquisition module, and a dual-loop control module; wherein,

[0027] The model acquisition module is used to acquire the voltage frequency domain model of the power electronic converter and the current loop controller frequency domain model of the power electronic converter.

[0028] The current transfer function acquisition module is used to obtain the current loop transfer function based on the voltage frequency domain model and the current loop controller frequency domain model.

[0029] The relationship acquisition module is used to acquire the current frequency domain model of the power electronic converter, and to linearize the current frequency domain model at the steady-state operating point to obtain a linearized model; the linearized model is then substituted into the current loop transfer function to obtain the relationship between the reference current and the output quantity.

[0030] The dual-loop control module is used to obtain a voltage loop control coupling term based on the relationship between the reference current and the output quantity, and then obtain a decoupling controller based on the voltage loop control coupling term; the decoupling controller is used to perform voltage loop and current loop control on the power electronic converter.

[0031] As a preferred embodiment, the model acquisition module acquires the voltage frequency domain model of the power electronic converter, including:

[0032] The model acquisition module acquires the voltage time-domain model of the power electronic converter, performs a Laplace transform on the voltage time-domain model, and obtains the voltage frequency-domain model.

[0033] As a preferred embodiment, the voltage time-domain model is expressed as:

[0034]

[0035] Among them, e d e is the d-axis component of the voltage at the neutral point of the power electronic converter. q L represents the q-axis component of the voltage at the neutral point of the power electronic converter. f For filter inductance, R f The parasitic resistance of the filter inductor is given by ω0, where ω0 is the rated angular frequency of the power grid, and u is the value of ω. sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld i represents the d-axis component of the filter inductor current. lq Let t be the q-axis component of the filter inductor current, and t be time.

[0036] As a preferred embodiment, the current loop transfer function is specifically:

[0037]

[0038] Where, △i ld * For the small perturbation component of the d-axis reference current, Δi lq * Let be the small perturbation component of the q-axis reference current, s be the Laplace operator, and G be the small perturbation component of the q-axis reference current. cl Let Δi be the transfer function of the proportional-integral controller in the current loop. ld For the small perturbation component of the d-axis component of the filter inductor current, Δi lq This is the small perturbation component of the q-axis component of the filter inductor current.

[0039] As a preferred embodiment, the relationship acquisition module acquires the current frequency domain model of the power electronic converter, including:

[0040] The relationship acquisition module obtains the current time-domain model of the power electronic converter, performs a Laplace transform on the current time-domain model, and obtains the current frequency-domain model.

[0041] As a preferred embodiment, the current time-domain model is expressed as:

[0042]

[0043] Among them, i ld i represents the d-axis component of the filter inductor current. lq C is the q-axis component of the filter inductor current. f Here, t is the filter capacitor, ω0 is the rated angular frequency of the power grid, and i is the frequency of the filter capacitor. od i represents the d-axis component of the grid-connected current. oq For the q-axis component of the grid-connected current, u sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld This represents the d-axis component of the filter inductor current.

[0044] As a preferred embodiment, the relationship between the reference current and the output quantity is specifically as follows:

[0045]

[0046] Where, △i ld * For the small perturbation component of the d-axis reference current, i lq * Classification of small perturbations to the q-axis reference current, Δu sd For the small disturbance component of the d-axis component of the power electronic converter, Δu sq G represents the small perturbation component of the q-axis component of the power electronic converter, s is the Laplace operator, and G... cl Let L be the transfer function of the proportional-integral controller in the current loop. f For filter inductance, R f C is the parasitic resistance of the filter inductor. f For the filter capacitor, Δi od For the small disturbance component of the d-axis component of the grid-connected current, Δi oq This represents the small disturbance component of the q-axis component of the grid-connected current.

[0047] Accordingly, this application also provides a terminal device, including a processor, a memory, and a computer program stored in the memory and configured to be executed by the processor, wherein the processor executes the computer program to implement the dual-loop control method of the power electronic converter.

[0048] Accordingly, this application also provides a computer-readable storage medium, the computer-readable storage medium including a stored computer program, wherein, when the computer program is executed, it controls the device where the computer-readable storage medium is located to execute the dual-loop control method of the power electronic converter.

[0049] Compared with the prior art, this invention application has the following beneficial effects:

[0050] This invention provides a dual-loop control method, system, device, and medium for a power electronic converter, comprising: obtaining a voltage frequency domain model of the power electronic converter and a current loop controller frequency domain model of the power electronic converter; obtaining a current loop transfer function based on the voltage frequency domain model and the current loop controller frequency domain model; obtaining a current frequency domain model of the power electronic converter and linearizing the current frequency domain model at the steady-state operating point to obtain a linearized model; substituting the linearized model into the current loop transfer function to obtain the relationship between a reference current and an output quantity; obtaining a voltage loop control coupling term based on the relationship between the reference current and the output quantity, and then obtaining a decoupling controller based on the voltage loop control coupling term; and performing voltage loop and current loop control on the power electronic converter through the decoupling controller. Compared to existing solutions that treat the current loop transfer function as 1, this invention derives the current loop transfer function using the voltage frequency domain model of the power electronic converter and the frequency domain model of the current loop controller. This clarifies the coupling terms introduced by the current loop, filters, and other components into the voltage loop. By substituting the linearized model obtained by linearizing the current frequency domain model at the steady-state operating point into the current loop transfer function, the relationship between the reference current and the output is obtained, thus revealing the voltage loop control coupling terms. Based on these voltage loop control coupling terms, a decoupled controller is derived, which optimizes the decoupling control effect of the d-axis and q-axis components, improves the accuracy and stability of the voltage and current dual-loop control, and ensures the transient and steady-state performance and control accuracy of the controller. Attached Figure Description

[0051] Figure 1 This is a flowchart illustrating an embodiment of the dual-loop control method for a power electronic converter provided in this invention application.

[0052] Figure 2 This is a schematic diagram of a prior art converter topology.

[0053] Figure 3 This is a schematic diagram of a prior art technology for voltage and current dual-loop control.

[0054] Figure 4 This is a schematic diagram illustrating the principle of one embodiment of the voltage outer loop control provided in this application.

[0055] Figure 5 This is a simulation waveform diagram of voltage and current dual-loop control using existing technology.

[0056] Figure 6 This is a simulation waveform diagram of an embodiment of the dual-loop control method for a power electronic converter provided in this application.

[0057] Figure 7 This is a schematic diagram of an embodiment of the dual-loop control system of the power electronic converter provided in this application. Detailed Implementation

[0058] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0059] Example 1

[0060] According to relevant technical records, some related technologies obtain the grid voltage and input it into a pre-constructed inverter main circuit control model that includes improved feedforward decoupling and proportional-integral stages, so as to output a stable voltage.

[0061] Other related technologies achieve decoupling control of voltage source inverters by adjusting the parameters of the voltage loop and current loop proportional-integral controllers. This method adjusts the parameters of the voltage loop and current loop proportional-integral controllers with the goal of decoupling the control loop. In the current loop control, the influence of current cross-coupling terms on active and reactive currents is eliminated, and in the voltage loop control, the influence of voltage cross-coupling terms on active and reactive voltages is eliminated.

[0062] The topology and control structure of the inverter are as follows Figure 2 and 3 As shown, where L f For filter inductance; i ld ,i lq U represents the dq-axis component of the filter inductor current, and u represents the output current of the converter. sd u sq e represents the dq-axis component of the output voltage. d e q i represents the dq-axis component of the converter neutral point voltage; ld * i lq * These are the d-axis and q-axis reference currents, respectively, and the voltage loop output; G CL For the transfer function of the current loop PI controller; G VLThis is the transfer function for the voltage loop PI controller.

[0063] In the aforementioned related technologies, the voltage loop output reference current and the converter output current transfer function are equivalent to 1, ignoring the influence of current loop, filter, and other components on the voltage loop. Therefore, the decoupling control effect of the d-axis and q-axis components is poor, resulting in the controller's transient and steady-state performance failing to meet design goals, and poor control accuracy. Simultaneously, the dq-axis variables are not completely decoupled, making independent control of the dq-axis variables impossible, leading to mutual interference between dq-axis controls and affecting the stability of the control system.

[0064] In addition, the related technologies achieve control decoupling by adjusting the parameters of the proportional-integral controller in the voltage loop and current loop, but there is still a risk that the response speed and accuracy of the proportional-integral controller cannot be guaranteed, resulting in a reduction in control performance.

[0065] For one or more of the above technical issues, please refer to Figure 1 , Figure 1 This invention application provides a dual-loop control method for a power electronic converter, comprising steps S101 to S104. The details of each step are as follows:

[0066] Step S101: Obtain the voltage frequency domain model of the power electronic converter and the current loop controller frequency domain model of the power electronic converter.

[0067] In this embodiment, Kirchhoff's voltage equation for the grid connection point of the power electronic converter in the dq coordinate system can be established first, and the d-axis component u of the filter capacitor voltage (i.e., the output voltage of the power electronic converter) in the dq rotating coordinate system can be obtained. sd and q-axis component u sq and the d-axis component i of the filter inductor current ld and q-axis component i lq Assuming the parameters of the three-phase system are consistent, according to Kirchhoff's laws, the voltage equation in the dq coordinate system is obtained as the voltage time-domain model:

[0068]

[0069] Among them, e d e is the d-axis component of the voltage at the neutral point of the power electronic converter. q L represents the q-axis component of the voltage at the neutral point of the power electronic converter. f For filter inductance, R f The parasitic resistance of the filter inductor is given by ω0, where ω0 is the rated angular frequency of the power grid, and u is the value of ω. sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld i represents the d-axis component of the filter inductor current.lq Let t be the q-axis component of the filter inductor current, and t be time.

[0070] Furthermore, a Laplace transform is performed on the voltage time-domain model to obtain the voltage frequency-domain model, as shown below:

[0071]

[0072] In the formula, s is the Laplace operator.

[0073] In some embodiments, the frequency domain model of the current loop controller of the power electronic converter can be as follows, which also represents the relationship between the output voltage and the reference current:

[0074]

[0075] In the formula, i ld * i is the d-axis reference current. lq * q-axis reference current (i ld * and i lq * (This is also the voltage loop output), k ip k is the proportional coefficient of the proportional-integral controller in the current loop. ii G represents the integral coefficient of the proportional-integral controller in the current loop. cl This is the transfer function of the proportional-integral controller for the current loop.

[0076] Step S102: Based on the voltage frequency domain model and the current loop controller frequency domain model, obtain the current loop transfer function.

[0077] Specifically, based on the voltage frequency domain model and the current loop controller frequency domain model mentioned above, small-signal linearization is performed to obtain the current loop transfer function.

[0078] The process of small-signal linearization can be understood as follows: all voltage and current variables are expressed in the form x = x0 + Δx, where x0 is the steady-state value of the corresponding variable and Δx represents the small disturbance component of the corresponding variable. Substituting x = x0 + Δx into the expression that needs to be linearized for small-signal, since x0 satisfies the original expression relationship, x0 can be eliminated. After eliminating x0, the expression after small-signal linearization is obtained.

[0079] This allows us to obtain the current loop transfer function:

[0080]

[0081] Where, △i ld *For the small perturbation component of the d-axis reference current, Δi lq * Let be the small perturbation component of the q-axis reference current, s be the Laplace operator, and G be the small perturbation component of the q-axis reference current. cl Let Δi be the transfer function of the proportional-integral controller in the current loop. ld For the small perturbation component of the d-axis component of the filter inductor current, Δi lq This is the small perturbation component of the q-axis component of the filter inductor current.

[0082] It is understandable that if the current loop transfer function is designed to be 1 according to existing technology, and the voltage loop is designed accordingly, the effects of the filter inductor, parasitic resistance and current loop controller will be ignored, thus making it impossible to achieve complete decoupling of the dq axis.

[0083] Step S103: Obtain the current frequency domain model of the power electronic converter, and linearize the current frequency domain model at the steady-state operating point to obtain a linearized model; substitute the linearized model into the current loop transfer function to obtain the relationship between the reference current and the output quantity.

[0084] In a preferred embodiment, Kirchhoff's current equations for the grid connection point of the power electronic converter in the dq coordinate system can be established first, and the d-axis component u of the filter capacitor voltage (i.e., the output voltage of the power electronic converter) in the dq rotating coordinate system can be obtained. sd and q-axis component u sq and the d-axis component i of the filter inductor current ld and q-axis component i lq Assuming the parameters of the three-phase system are consistent, according to Kirchhoff's current law, the current equation in the dq coordinate system is obtained as the current time-domain model:

[0085]

[0086] Among them, i ld i represents the d-axis component of the filter inductor current. lq C is the q-axis component of the filter inductor current. f Here, t is the filter capacitor, ω0 is the rated angular frequency of the power grid, and i is the frequency of the filter capacitor. od i represents the d-axis component of the grid-connected current. oq For the q-axis component of the grid-connected current, u sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld This represents the d-axis component of the filter inductor current.

[0087] Furthermore, based on the current time-domain model, a Laplace transform is performed to obtain the current frequency-domain model, as shown below:

[0088]

[0089] Where s is the Laplace operator.

[0090] Furthermore, the current frequency domain model can be linearized at the steady-state operating point to obtain a linearized model:

[0091]

[0092] In the formula, △i od For the small disturbance component of the d-axis component of the grid-connected current, Δi oq For the small disturbance component of the q-axis component of the grid-connected current, Δi ld For the small perturbation component of the d-axis component of the filter inductor current, Δi lq For the small perturbation component of the q-axis component of the filter inductor current, Δu sd For the small disturbance component of the d-axis component of the power electronic converter, Δu sq This refers to the small disturbance component of the q-axis component of the power electronic converter.

[0093] Furthermore, by substituting the linearized model into the current loop transfer function, the relationship between the reference current and the output can be obtained:

[0094]

[0095] In the formula, △i ld * For the small perturbation component of the d-axis reference current, Δi lq * This represents the small perturbation component of the q-axis reference current.

[0096] Step S104: Based on the relationship between the reference current and the output quantity, a voltage loop control coupling term is obtained, and then based on the voltage loop control coupling term, a decoupling controller is obtained; the power electronic converter is controlled by the decoupling controller for voltage loop and current loop.

[0097] Based on the relationship between the reference current and the output quantity, it can be determined that the voltage outer loop coupling terms after the current loop (the aforementioned voltage loop control coupling terms) include:

[0098] The outer loop control coupling term for the d-axis voltage is:

[0099]

[0100] The outer loop control coupling term for the q-axis voltage is:

[0101]

[0102] In practice, R f and Lf Since it is not equal to 0, it can be seen that if the current loop transfer function is regarded as 1 to derive the voltage loop coupling term according to the existing technology, the influence of filter inductance, parasitic resistance and current loop will be ignored, thus making it impossible to achieve complete decoupling.

[0103] This implementation differs from existing technical solutions. The voltage loop control coupling term mentioned above includes a specific current loop transfer function, reflecting the coupling terms introduced by the current loop and filter for the voltage loop. The voltage loop decoupling controller designed accordingly can achieve complete decoupling in the actual control system.

[0104] For example, by adding the voltage loop control coupling term to the voltage loop controller through linear operation to cancel out its influence, the control law of the decoupled controller can be designed as follows:

[0105]

[0106] This decoupling controller takes into account the current loop transfer function, including the current loop control and the effect of the filter on the voltage loop, thus achieving complete decoupling of the voltage loop.

[0107] Furthermore, the designed decoupling controller control law can be applied to the voltage outer loop controller for verification. Specifically, by substituting the control law of the decoupling controller into the relationship between the reference current and the output, the transfer function from the reference voltage to the actual voltage can be obtained as follows:

[0108]

[0109] In the formula, G vl The transfer function for the voltage loop proportional-integral controller. This refers to the small disturbance component of the d-axis reference voltage component of the power electronic converter. This refers to the small disturbance component of the q-axis reference voltage component of the power electronic converter.

[0110] As can be seen from the above formula, without changing the proportional-integral control parameters, the coupling between control loops is completely canceled out by linear operation. At this time, the proportional-integral parameters can be tuned directly with speed and accuracy as the objectives to ensure control performance.

[0111] The transfer function described above demonstrates complete decoupling of the dq-axis voltage and current variables, achieving independent control of the voltage variables. The voltage outer loop control block diagram of this invention is shown below. Figure 4 As shown (for illustrative purposes only, the figure shows l) f That is, the filter inductor L in this embodiment f The current inner loop control can employ existing methods. The simulation waveforms for the existing methods and the dual-loop control method of this embodiment are shown below. Figure 5 and 6As shown in the figure (for illustrative purposes only; under the same operating conditions, the d-axis voltage reference value is 0.93 and the q-axis voltage reference value is 0), it can be seen that the method described in this embodiment can completely eliminate the coupling between the voltage and current dual closed-loop control loops, realize independent control of the dq-axis voltage variables, and thus greatly improve the control accuracy and stability.

[0112] Accordingly, such as Figure 7 As shown, this invention application also provides a dual-loop control system 700 for a power electronic converter, including a model acquisition module 701, a current transfer function acquisition module 702, a relationship acquisition module 703, and a dual-loop control module 704; wherein,

[0113] The model acquisition module 701 is used to acquire the voltage frequency domain model of the power electronic converter and the current loop controller frequency domain model of the power electronic converter.

[0114] The current transfer function acquisition module 702 is used to obtain the current loop transfer function based on the voltage frequency domain model and the current loop controller frequency domain model.

[0115] The relationship acquisition module 703 is used to acquire the current frequency domain model of the power electronic converter, and to linearize the current frequency domain model at the steady-state operating point to obtain a linearized model; the linearized model is then substituted into the current loop transfer function to obtain the relationship between the reference current and the output quantity.

[0116] The dual-loop control module 704 is used to obtain a voltage loop control coupling term based on the relationship between the reference current and the output quantity, and then obtain a decoupling controller based on the voltage loop control coupling term; and to perform voltage loop and current loop control on the power electronic converter through the decoupling controller.

[0117] As a preferred embodiment, the model acquisition module 701 acquires the voltage frequency domain model of the power electronic converter, including:

[0118] The model acquisition module 701 acquires the voltage time-domain model of the power electronic converter, performs a Laplace transform on the voltage time-domain model, and obtains the voltage frequency-domain model.

[0119] As a preferred embodiment, the voltage time-domain model is expressed as:

[0120]

[0121] Among them, e d e is the d-axis component of the voltage at the neutral point of the power electronic converter. q L represents the q-axis component of the voltage at the neutral point of the power electronic converter. f For filter inductance, R fThe parasitic resistance of the filter inductor is given by ω0, where ω0 is the rated angular frequency of the power grid, and u is the value of ω. sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld i represents the d-axis component of the filter inductor current. lq Let t be the q-axis component of the filter inductor current, and t be time.

[0122] As a preferred embodiment, the current loop transfer function is specifically:

[0123]

[0124] Where, △i ld * For the small perturbation component of the d-axis reference current, Δi lq * Let be the small perturbation component of the q-axis reference current, s be the Laplace operator, and G be the small perturbation component of the q-axis reference current. cl Let Δi be the transfer function of the proportional-integral controller in the current loop. ld For the small perturbation component of the d-axis component of the filter inductor current, Δi lq This is the small perturbation component of the q-axis component of the filter inductor current.

[0125] As a preferred embodiment, the relationship acquisition module 703 acquires the current frequency domain model of the power electronic converter, including:

[0126] The relationship acquisition module 703 acquires the current time-domain model of the power electronic converter, performs a Laplace transform on the current time-domain model, and obtains the current frequency-domain model.

[0127] As a preferred embodiment, the current time-domain model is expressed as:

[0128]

[0129] Among them, i ld i represents the d-axis component of the filter inductor current. lq C is the q-axis component of the filter inductor current. f Here, t is the filter capacitor, ω0 is the rated angular frequency of the power grid, and i is the frequency of the filter capacitor. od i represents the d-axis component of the grid-connected current. oq For the q-axis component of the grid-connected current, u sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld This represents the d-axis component of the filter inductor current.

[0130] As a preferred embodiment, the relationship between the reference current and the output quantity is specifically as follows:

[0131]

[0132] Where, △i ld * For the small perturbation component of the d-axis reference current, i lq * Classification of small perturbations to the q-axis reference current, Δu sd For the small disturbance component of the d-axis component of the power electronic converter, Δu sq G represents the small perturbation component of the q-axis component of the power electronic converter, s is the Laplace operator, and G... cl Let L be the transfer function of the proportional-integral controller in the current loop. f For filter inductance, R f C is the parasitic resistance of the filter inductor. f For the filter capacitor, Δi od For the small disturbance component of the d-axis component of the grid-connected current, Δi oq This represents the small disturbance component of the q-axis component of the grid-connected current.

[0133] Accordingly, this application also provides a terminal device, including a processor, a memory, and a computer program stored in the memory and configured to be executed by the processor, wherein the processor executes the computer program to implement the dual-loop control method of the power electronic converter.

[0134] The processor can be a Central Processing Unit (CPU), or other general-purpose processors, digital signal processors (DSPs), application-specific integrated circuits (ASICs), field-programmable gate arrays (FPGAs), or other programmable logic devices, discrete gate or transistor logic devices, discrete hardware components, etc. A general-purpose processor can be a microprocessor or any conventional processor. The processor is the control center of the terminal, connecting various parts of the terminal via various interfaces and lines.

[0135] The memory can be used to store the computer program. The processor implements various functions of the terminal by running or executing the computer program stored in the memory and calling data stored in the memory. The memory may mainly include a program storage area and a data storage area. The program storage area may store the operating system, at least one application program required for a function (such as sound playback function, image playback function, etc.), etc.; the data storage area may store data created according to the use of the mobile phone (such as audio data, phonebook, etc.). In addition, the memory may include high-speed random access memory, and may also include non-volatile memory, such as hard disk, memory, plug-in hard disk, smart media card (SMC), secure digital (SD) card, flash card, at least one disk storage device, flash memory device, or other volatile solid-state storage device.

[0136] Accordingly, this application also provides a computer-readable storage medium, the computer-readable storage medium including a stored computer program, wherein, when the computer program is executed, it controls the device where the computer-readable storage medium is located to execute the dual-loop control method of the power electronic converter.

[0137] If the module integrated into the dual-loop control system of the power electronic converter is implemented as a software functional unit and sold or used as an independent product, it can be stored in a computer-readable storage medium. Based on this understanding, all or part of the processes in the above embodiments of the present invention can also be implemented by a computer program instructing related hardware. The computer program can be stored in a computer-readable storage medium, and when executed by a processor, it can implement the steps of the various method embodiments described above. The computer program includes computer program code, which can be in the form of source code, object code, executable files, or certain intermediate forms. The computer-readable medium can include: any entity or device capable of carrying the computer program code, a recording medium, a USB flash drive, a portable hard drive, a magnetic disk, an optical disk, a computer memory, a read-only memory (ROM), a random access memory (RAM), an electrical carrier signal, a telecommunication signal, and a software distribution medium, etc.

[0138] Compared with the prior art, this invention application has the following beneficial effects:

[0139] This invention provides a dual-loop control method, system, device, and medium for a power electronic converter, comprising: obtaining a voltage frequency domain model of the power electronic converter and a current loop controller frequency domain model of the power electronic converter; obtaining a current loop transfer function based on the voltage frequency domain model and the current loop controller frequency domain model; obtaining a current frequency domain model of the power electronic converter and linearizing the current frequency domain model at the steady-state operating point to obtain a linearized model; substituting the linearized model into the current loop transfer function to obtain the relationship between a reference current and an output quantity; obtaining a voltage loop control coupling term based on the relationship between the reference current and the output quantity, and then obtaining a decoupling controller based on the voltage loop control coupling term; and performing voltage loop and current loop control on the power electronic converter through the decoupling controller. Compared to existing solutions that treat the current loop transfer function as 1, this invention derives the current loop transfer function using the voltage frequency domain model of the power electronic converter and the frequency domain model of the current loop controller. This clarifies the coupling terms introduced by the current loop, filters, and other components into the voltage loop. By substituting the linearized model obtained by linearizing the current frequency domain model at the steady-state operating point into the current loop transfer function, the relationship between the reference current and the output is obtained, thus revealing the voltage loop control coupling terms. Based on these voltage loop control coupling terms, a decoupled controller is derived, which optimizes the decoupling control effect of the d-axis and q-axis components, improves the accuracy and stability of the voltage and current dual-loop control, and ensures the transient and steady-state performance and control accuracy of the controller.

[0140] The specific embodiments described above further illustrate the purpose, technical solution, and beneficial effects of the present invention. It should be understood that the above descriptions are merely specific embodiments of the present invention and are not intended to limit the scope of protection of the present invention. In particular, it should be noted that any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the scope of protection of the present invention for those skilled in the art.

Claims

1. A dual-loop control method for a power electronic converter, characterized in that, include: Obtain the voltage frequency domain model of the power electronic converter, and obtain the current loop controller frequency domain model of the power electronic converter; Based on the voltage frequency domain model and the current loop controller frequency domain model, the current loop transfer function is obtained; Obtain the current frequency domain model of the power electronic converter, and linearize the current frequency domain model at the steady-state operating point to obtain a linearized model; Substituting the linearized model into the current loop transfer function, the relationship between the reference current and the output is obtained. Based on the relationship between the reference current and the output, a voltage loop control coupling term is obtained, and then a decoupling controller is obtained based on the voltage loop control coupling term; the power electronic converter is controlled by the decoupling controller for voltage loop and current loop.

2. The dual-loop control method for a power electronic converter as described in claim 1, characterized in that, The process of obtaining the voltage frequency domain model of the power electronic converter includes: Obtain the voltage time-domain model of the power electronic converter, and perform a Laplace transform on the voltage time-domain model to obtain the voltage frequency-domain model.

3. The dual-loop control method for a power electronic converter as described in claim 2, characterized in that, The voltage time-domain model is expressed as follows: Among them, e d e is the d-axis component of the voltage at the neutral point of the power electronic converter. q L represents the q-axis component of the voltage at the neutral point of the power electronic converter. f For the filter inductor, R f The parasitic resistance of the filter inductor is given by ω0, where ω0 is the rated angular frequency of the power grid, and u is the value of ω. sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld i represents the d-axis component of the filter inductor current. lq Let t be the q-axis component of the filter inductor current, and t be time.

4. The dual-loop control method for a power electronic converter as described in claim 3, characterized in that, The current loop transfer function is specifically as follows: Where, △i ld * For the small perturbation component of the d-axis reference current, Δi lq * Let be the small perturbation component of the q-axis reference current, s be the Laplace operator, and G be the small perturbation component of the q-axis reference current. cl Let Δi be the transfer function of the proportional-integral controller in the current loop. ld For the small perturbation component of the d-axis component of the filter inductor current, Δi lq This is the small perturbation component of the q-axis component of the filter inductor current.

5. The dual-loop control method for a power electronic converter as described in claim 1, characterized in that, The process of obtaining the current frequency domain model of the power electronic converter includes: Obtain the current time-domain model of the power electronic converter, and perform a Laplace transform on the current time-domain model to obtain the current frequency-domain model.

6. The dual-loop control method for a power electronic converter as described in claim 5, characterized in that, The current time-domain model is specifically as follows: Among them, i ld i represents the d-axis component of the filter inductor current. lq C is the q-axis component of the filter inductor current. f Here, t is the filter capacitor, ω0 is the rated angular frequency of the power grid, and i is the frequency of the filter capacitor. od i represents the d-axis component of the grid-connected current. oq For the q-axis component of the grid-connected current, u sd For the d-axis voltage component of the power electronic converter, u sq For the q-axis voltage component of the power electronic converter, i ld This represents the d-axis component of the filter inductor current.

7. The dual-loop control method for a power electronic converter as described in claim 6, characterized in that, The relationship between the reference current and the output quantity is as follows: Where, △i ld * For the small perturbation component of the d-axis reference current, Δi lq * For the small perturbation component of the q-axis reference current, Δu sd For the small disturbance component of the d-axis component of the power electronic converter, Δu sq G represents the small perturbation component of the q-axis component of the power electronic converter, s is the Laplace operator, and G... cl Let L be the transfer function of the proportional-integral controller in the current loop. f For the filter inductor, R f C is the parasitic resistance of the filter inductor. f For the filter capacitor, Δi od For the small disturbance component of the d-axis component of the grid-connected current, Δi oq This represents the small disturbance component of the q-axis component of the grid-connected current.

8. A dual-loop control system for a power electronic converter, characterized in that, It includes a model acquisition module, a current transfer function acquisition module, a relationship acquisition module, and a dual-loop control module; among which, The model acquisition module is used to acquire the voltage frequency domain model of the power electronic converter and the current loop controller frequency domain model of the power electronic converter. The current transfer function acquisition module is used to obtain the current loop transfer function based on the voltage frequency domain model and the current loop controller frequency domain model. The relationship acquisition module is used to acquire the current frequency domain model of the power electronic converter, and to linearize the current frequency domain model at the steady-state operating point to obtain a linearized model; the linearized model is then substituted into the current loop transfer function to obtain the relationship between the reference current and the output quantity. The dual-loop control module is used to obtain a voltage loop control coupling term based on the relationship between the reference current and the output quantity, and then obtain a decoupling controller based on the voltage loop control coupling term; the decoupling controller is used to perform voltage loop and current loop control on the power electronic converter.

9. A terminal device, characterized in that, The device includes a processor, a memory, and a computer program stored in the memory and configured to be executed by the processor, wherein the processor, when executing the computer program, implements the dual-loop control method for a power electronic converter as described in any one of claims 1 to 7.

10. A computer-readable storage medium, characterized in that, The computer-readable storage medium includes a stored computer program, wherein, when the computer program is executed, it controls the device containing the computer-readable storage medium to perform the dual-loop control method for a power electronic converter as described in any one of claims 1 to 7.

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