Frequency tracking device and method for transmitting end of wireless charging system

Abstract

The frequency tracking device and method for the transmitting end of the wireless charging system are simple in structure, low in cost, small in calculation amount, stable and reliable in operation and high in efficiency. The frequency tracking device for the transmitting end of the wireless charging system comprises a sampling module (1), a high-speed comparator (2), a DSP control module (3) and a PI adjusting module (4), the method comprises the following steps: a, signal acquisition; b, capturing at a zero-crossing moment; c, calculating a phase difference; d, closed-loop frequency adjustment; and e, repeating the steps a-d to realize real-time frequency tracking. The method is applied to the technical field of wireless charging.

Description

Technical Field

[0001] This invention relates to the field of wireless charging technology, and in particular to a frequency tracking device and method for the transmitter of a wireless charging system. Background Technology

[0002] In charging systems, magnetic field coupling technology has become the mainstream method for wireless energy transmission due to its high efficiency and convenience. This process typically uses a coil as a medium to convert electromagnetic waves generated at the ground (transmitter) into alternating current (AC), which is then rectified into stable direct current (DC) to supply various vehicle-mounted loads. However, in specific fault conditions, such as when an emergency stop to energy output is required, simply relying on communication methods to notify the ground to halt power transmission can lead to safety hazards or even serious accidents due to the time delay in signal transmission and response. Therefore, developing a rapid emergency circuit-breaking mechanism embedded within the vehicle-mounted terminal is particularly urgent.

[0003] Wireless charging systems are a type of contactless power transmission device based on the principle of magnetic coupling resonance. Due to their high efficiency and ease of use, they have been widely used in electric vehicles, consumer electronics, and other fields. The system mainly consists of two sets of coupled coils: a ground-based (transmitter) coil and a vehicle-mounted (receiver) coil. The transmitter coil generates an alternating magnetic field driven by a high-frequency inverter, while the receiver coil receives energy through electromagnetic induction and provides stable DC power to the vehicle load or battery after rectification and filtering.

[0004] In actual operation, factors such as vehicle parking position deviation, coil spacing changes, or environmental interference can cause dynamic drift in the system coupling coefficient and resonant frequency, leading to system detuning, reduced transmission efficiency, and even safety issues. Therefore, to ensure stable and efficient system operation under a wide range of conditions, a frequency tracking and impedance matching mechanism capable of real-time sensing of the resonant state and adaptive adjustment of control parameters is urgently needed.

[0005] Furthermore, the component parameters in the resonant network (such as coil self-inductance and compensation capacitor values) may drift due to temperature changes, aging, or manufacturing tolerances. Taking a bilateral LCC compensation topology as an example, in a symmetrical design, each side contains multiple components such as resonant inductors and series / parallel capacitors. The parameters of each component can shift towards inductive or capacitive directions, resulting in a high degree of diversity in the overall detuning direction of the system (theoretically, dozens of combinations). These complex drift modes will affect the total impedance phase angle of the system to varying degrees, making it difficult for traditional fixed-frequency control to maintain optimal operating conditions.

[0006] To address these challenges, existing technologies typically employ a phase-locked loop (PLL) control strategy. This strategy involves detecting the phase difference between a specific electrical quantity at the transmitter (such as the resonant current and the midpoint voltage) and adjusting the inverter's operating frequency in real time to maintain this phase difference constant, thereby achieving dynamic compensation for system detuning. Essentially, this method utilizes the frequency selectivity of the resonant network to control the phase of the equivalent impedance through frequency adjustment, ultimately achieving automatic impedance matching and efficiency optimization of the system.

[0007] Currently, common frequency tracking methods mainly include the following categories: 1. Measure the magnitude of active and reactive power and calculate the power angle to perform the frequency tracking process.

[0008] 2. Measure the orthogonal and perpendicular quantities of the resonant current and analyze its Euler angle changes to perform frequency tracking.

[0009] 3. Determine whether the resonant network system is capacitive or inductive by the phase relationship between the resonant current and the midpoint voltage, and use this to change the frequency.

[0010] 5. Perform FFT algorithm on the resonant current to analyze its harmonic composition, and use this to change the frequency.

[0011] 6. The frequency tracking process is performed by injecting frequency jumps and observing the output using the perturbation and observation method.

[0012] The aforementioned prior art has the following disadvantages: 1. Active and reactive power measurement relies on complex hardware circuits, resulting in high costs. In existing technologies, to obtain active and reactive power and calculate the power angle accordingly, complex external sampling and processing circuits (such as multipliers, filters, and phase detection modules) are typically required. Such hardware structures not only increase the system size but also significantly increase the overall manufacturing cost.

[0013] 2. Detection of the orthogonal components of the resonant current is complex and difficult to implement in high-frequency systems. Phase analysis of resonant currents through orthogonal decomposition (i.e., extraction of in-phase and quadrature components) has traditionally relied on additional analog hardware circuitry. While digital methods such as second-order generalized integrators (SOGI) can simplify hardware design, in wireless charging systems operating at frequencies up to 200 kHz, such algorithms are difficult to implement stably and accurately due to limitations in controller latency and bandwidth.

[0014] 3. Frequency regulation methods based on phase discrimination have limited control capabilities and are prone to causing system oscillations. Some solutions adjust the operating frequency by judging the capacitive or inductive nature of the system impedance. While simple to implement, this only allows for coarse mode switching and cannot achieve active, continuous adjustment of the phase difference. Furthermore, this method is prone to misjudgment near the zero-phase crossover point, causing repeated frequency jumps and resulting in system oscillations, thus affecting stability.

[0015] 4. The FFT-based resonant current analysis algorithm is complex and requires high computational resources. While using Fast Fourier Transform (FFT) to perform spectral analysis on resonant currents can extract phase information, its algorithmic complexity is high, requiring numerous floating-point operations and placing high demands on processor performance. Therefore, this method is difficult to apply in real-time to resource-constrained embedded systems or low-cost controllers.

[0016] 5. The perturbation-observation method is sensitive to system timing and bandwidth, limiting its application scenarios. The perturbation-observation method is commonly used in automotive wireless charging systems, especially in architectures where the main controller is located at the receiver, as it avoids latency issues associated with long-distance communication. However, this method has strict requirements on control timing and system bandwidth, and is only applicable to specific types of resonant topologies, making it difficult to implement independently at the transmitter, thus limiting its versatility and deployment flexibility. Summary of the Invention

[0017] The technical problem to be solved by the present invention is to overcome the shortcomings of the prior art and provide a frequency tracking device and method for the transmitter of a wireless charging system that is simple in structure, low in cost, low in computational load, stable and reliable in operation, and highly efficient.

[0018] The technical solution adopted in this invention is a frequency tracking device for the transmitter end of a wireless charging system. This device is disposed at the transmitter coil end of the wireless charging system and includes... The sampling module is used to collect the resonant current signal of the resonant cavity at the transmitter of the wireless charging system; A high-speed comparator is used for zero-crossing detection of the resonant current signal, converting the resonant current signal into a square wave signal with a steep rising edge and zero-crossing information, and sending it to the eCAP pin of the DSP control module. The DSP control module has an enhanced capture unit and an enhanced pulse width modulation unit built in. The enhanced capture unit is used to capture the zero-crossing time of the square wave signal and measure the phase difference between the voltage at the zero-crossing time and the midpoint voltage of the inverter at the transmitting end. The enhanced pulse width modulation unit is used to output an adjustable frequency PWM drive signal to the power drive at the transmitting end of the wireless charging system. The power drive drives the full-bridge high-frequency inverter at the transmitting end to provide high-frequency power to the resonant network at the transmitting end. The PI adjustment module, integrated within the DSP control module, is used to dynamically adjust the PWM frequency based on the phase difference measured by the enhanced capture unit as feedback, and sends the adjusted PWM frequency to the power drive at the transmitter to achieve automatic frequency tracking.

[0019] Furthermore, the sampling module uses a current transformer or a voltage divider resistor, and the resonant current signal is obtained after being amplified by an amplifier. The gain of the amplifier is adjustable from 1 to 10 times.

[0020] Furthermore, the DSP control module uses a TMS320F28035 DSP chip, and the enhanced capture unit supports rising edge triggered capture with a capture accuracy of ≤1ns.

[0021] Furthermore, the high-speed comparator uses a comparator of model LMV7219.

[0022] Furthermore, the transmitter resonant network is compatible with LCC, SS, and LCL resonant topologies, and can be adapted to different topologies through the configuration of charging system parameters.

[0023] A method for tracking the frequency of a wireless charging system transmitter using a frequency tracking device as described above, the method comprising the following steps: a. Signal acquisition: The sampling module acquires the resonant current signal of the resonant cavity at the transmitter of the wireless charging system and inputs it into the high-speed comparator; b. Zero-crossing moment capture: The high-speed comparator converts the resonant current signal into a square wave signal. The enhanced capture unit of the DSP control module is triggered by the rising edge to capture the zero-crossing moment of the square wave signal and record it to the capture register. c. Phase Difference Calculation: The DSP control module reads the time base counter register value of the enhanced pulse width modulation unit to obtain the midpoint voltage timing value of the inverter at the transmitting end. Simultaneously, the DSP control module reads the capture register value of the enhanced capture unit to obtain the resonant current timing value. The phase difference between the midpoint voltage and the resonant current is calculated using an interval judgment method. i E and will i E Normalize to the range [-π, π]. d. Closed-loop frequency adjustment: Adjust the phase difference i E Phase difference θ with the target ref The difference is input to the PI adjustment module, which outputs a frequency adjustment amount fe. Combined with the reference frequency f0, the target control frequency fc = f0 + fe is obtained. After limiting the target control frequency fc, the carrier frequency of the enhanced pulse width modulation unit is updated. e. Repeat steps a to d to achieve real-time frequency tracking.

[0024] Furthermore, in step c, the phase difference between the midpoint voltage and the resonant current is calculated using the interval judgment method. i E Specifically: A square wave containing the zero-crossing information of the resonant current ILrp is obtained through the hardware section. Triggered by the rising edge of the eCAP module, the internal clock of the eCAP module starts timing. This clock is reset to zero after the next rising edge trigger and starts timing again. Therefore, the phase angle of the resonant current and the midpoint voltage can be calculated using the corresponding register values ​​of the eCAP and ePWM modules. Specifically: c1. Calculate the phase values ​​of the midpoint voltage and the resonant current: The value at which the enhanced pulse width modulation unit starts timing for the midpoint voltage vpi is denoted as: , The value at which the rising edge of the resonant current ILrp is captured by the enhanced capture unit to start timing is denoted as: , The period value of the full-bridge high-frequency inverter is denoted as: , Then the phase value θ of the midpoint voltage vpi v for: , Phase value θ of resonant current ILrp i for: ; c2. Phase angle difference calculation: Assume the range of the phase angle difference is (0, 2). π ), then we have , Define the phase difference between the midpoint voltage vpi and the resonant current Ilrp as follows: i E The calculation process is as follows: ; c3, Normalized phase difference θ EN calculate: Based on the relationship between the midpoint voltage vpi and the resonant current ILrp, we have: , , cos(θ v -θ i ) and sin(θ v -θ i This constitutes a unit vector: , Then vector The direction angle is a normalized angle, and the normalized phase difference θ is extracted using the atan2 function. EN : , The normalized phase difference θ EN Phase difference i E The arcsine of the sine value itself is then: i E =θ EN .

[0025] Furthermore, in step c, the phase difference is... i E Normalization to the range [-π, π] specifically means: when i E When ∈(π,2π), phase difference i E Normalization i E -2π; when i E When ∈(-2π,-π), phase difference i E Normalization i E +2π; when i E When ∈[-π,π], maintain phase difference i E constant.

[0026] Furthermore, in step d, the target control frequency fc is limited, and the limiting range is ±20% of the inherent resonant frequency of the resonant network.

[0027] Compared with the prior art, the beneficial effects of the present invention are: 1. Simplified hardware and reduced cost: Only one high-speed comparator is needed as external hardware, and all other functions are implemented internally by the DSP. There is no need for complex circuits such as multipliers and filters, which greatly reduces hardware costs. 2. Strong high-frequency adaptability: It adopts a lightweight algorithm of "zero-crossing capture + interval judgment" to avoid complex operations such as SOGI and FFT. It has low response latency and better stability than existing solutions in high-frequency scenarios of 200-500kHz. 3. High control precision and no oscillation: The phase difference is continuously adjusted through PI closed-loop control, rather than discrete mode switching. There is no misjudgment near zero phase and the frequency fluctuation range is small. 4. Independent operation of the transmitter: It only relies on the local voltage and current signals of the transmitter, without the need for communication feedback from the receiver, and can independently complete frequency tracking, improving the flexibility of system deployment; 5. Good topology compatibility: It can be adapted to mainstream resonant topologies such as LCC, SS, and LCL through software configuration without modifying the hardware structure, making it highly versatile; 6. Convenient engineering implementation: Based on mature DSP peripherals (eCAP, ePWM), the algorithm flow is clear, the amount of code is small, and it is easy to debug and mass produce. Attached Figure Description

[0028] Figure 1 This is a simplified structural diagram of the device of the present invention; Figure 2 It is a waveform diagram for calculating the phase values ​​of the midpoint voltage and resonant current; Figure 3 This is a flowchart for calculating phase angle difference; Figure 4 This is a diagram of the frequency loop construction. Detailed Implementation

[0029] like Figure 1~Figure 4 As shown, the frequency tracking device for the transmitter of the wireless charging system of the present invention includes... Sampling module 1 is used to collect the resonant current signal of the resonant cavity at the transmitter of the wireless charging system; High-speed comparator 2 is used for zero-crossing detection of the resonant current signal, converting the resonant current signal into a square wave signal with a steep rising edge and zero-crossing information, and sending it to the eCAP pin of DSP control module 3. The DSP control module 3 has an enhanced capture unit eCAP and an enhanced pulse width modulation unit ePWM. The enhanced capture unit eCAP is used to capture the zero-crossing moment of the square wave signal and measure the phase difference between the voltage at the zero-crossing moment and the midpoint voltage of the inverter at the transmitting end. The enhanced pulse width modulation unit ePWM is used to output an adjustable frequency PWM drive signal to the power drive at the transmitting end of the wireless charging system. The power drive drives the full-bridge high-frequency inverter at the transmitting end to provide high-frequency power to the resonant network at the transmitting end. The PI adjustment module 4, integrated within the DSP control module 3, is used to dynamically adjust the PWM frequency based on the phase difference measured by the enhanced capture unit eCAP as feedback, and sends the adjusted PWM frequency to the power drive at the transmitter to achieve automatic frequency tracking.

[0030] Specifically, the sampling module 1 uses a current transformer or voltage divider resistors, and the resonant current signal is obtained after amplification by an amplifier. The gain of the amplifier is adjustable from 1 to 10 times. The DSP control module 3 uses a TMS320F28035 or a similar DSP chip with eCAP and ePWM peripherals. The enhanced capture unit eCAP supports rising edge triggered capture with a capture accuracy ≤1ns. The high-speed comparator 2 uses an LMV7219 comparator. The transmitter resonant network is compatible with LCC, SS, and LCL resonant topologies, and adaptation to different topologies is achieved through charging system parameter configuration.

[0031] A method for tracking the frequency of a wireless charging system transmitter using the frequency tracking device described above, the method comprising the following steps: a. Signal acquisition: The sampling module 1 acquires the resonant current signal of the resonant cavity of the wireless charging system transmitter and inputs it into the high-speed comparator 2; b. Zero-crossing moment capture: The high-speed comparator 2 converts the resonant current signal into a square wave signal. The enhanced capture unit eCAP of the DSP control module 3 is triggered by the rising edge to capture the zero-crossing moment of the square wave signal and record it to the capture register. c. Phase Difference Calculation: The DSP control module 3 reads the time base counter register value of the enhanced pulse width modulation unit (ePWM) to obtain the midpoint voltage timing value of the inverter at the transmitting end. Simultaneously, the DSP control module 3 reads the capture register value of the enhanced capture unit (eCAP) to obtain the resonant current timing value. The phase difference between the midpoint voltage and the resonant current is calculated using an interval judgment method. i E and will i E Normalize to the range [-π, π]. d. Closed-loop frequency adjustment: Adjust the phase difference i E Phase difference θ with the target ref The difference is input to the PI adjustment module 4, which outputs the frequency adjustment amount fe. Combined with the reference frequency f0, the target control frequency fc = f0 + fe is obtained. After limiting the target control frequency fc, the carrier frequency of the enhanced pulse width modulation unit ePWM is updated. e. Repeat steps a to d to achieve real-time frequency tracking.

[0032] Specifically, in step c, the phase difference between the midpoint voltage and the resonant current is calculated using the interval judgment method. i E Specifically: c1. Calculate the phase values ​​of the midpoint voltage and the resonant current: The value at which the enhanced pulse width modulation unit (ePWM) starts timing for the midpoint voltage vpi is denoted as: , The value at which the rising edge of the resonant current ILrp is captured by the enhanced capture unit eCAP to start timing is denoted as: , The period value of the full-bridge high-frequency inverter is denoted as: , Then the phase value θ of the midpoint voltage vpi v for: , Phase value θ of resonant current ILrp i for: ; The timing values ​​of the above EPWM and eCAP modules can be directly obtained from the corresponding registers inside the DSP (for the DSC28035, the register that records the starting timing value of the EPWM module is stored in the Time Base Count Register TBCTR, the value of the eCAP module that starts timing when the rising edge is captured is stored in the nth timestamp capture register CAPn, and the inverter's cycle value is stored in the Time Base Period Register TBPRD).

[0033] c2. Phase angle difference calculation: Assume the range of the phase angle difference is (0, 2). π ), then we have , Define the phase difference between the midpoint voltage vpi and the resonant current ILrp as follows: i E The calculation process is as follows: ; c3, Normalized phase difference θ EN calculate: Based on the relationship between the midpoint voltage vpi and the resonant current ILrp, we have: , , cos(θ v -θ i ) and sin(θ v -θ i This constitutes a unit vector: , Then vector The direction angle is a normalized angle, and the normalized phase difference is extracted using the atan2 function. i EN : , The normalized phase difference θ EN Phase difference i E The arcsine of the sine value itself is then: i E = i EN .

[0034] After obtaining the phase difference, the frequency value can be calculated in the frequency loop, converted into the corresponding period value, and then the PFM process can be performed. The frequency loop is constructed as follows: Figure 4 As shown.

[0035] First, the measured phase difference i E Phase angle θ with the target ref Subtracting them, we get the angle difference δe = θ E- θ ref The error signal is amplified by the proportional gain G to obtain the amplified error δE=G·δe, where the gain G needs to be tuned according to the system sampling ratio and control response requirements.

[0036] Subsequently, δE is input to the PI controller, and after proportional-integral calculation, it outputs a frequency adjustment value, fe. This adjustment value is added to the reference frequency f0 (usually the resonant frequency of the resonant network or the rated operating frequency) to obtain the target control frequency: fc = f0 + fe. Finally, upper and lower frequency limits are applied to fc, and the frequency limiting circuit restricts it to the allowable range, generating the final output frequency command, which is used to update the carrier frequency of the PWM module, realizing closed-loop frequency regulation.

[0037] In step c, the phase difference i E Normalization to the range [-π, π] specifically means: when i E When ∈(π,2π), phase difference i E Normalization i E -2π; when i E When ∈(-2π,-π), phase difference i E Normalization i E +2π; when i E When ∈[-π,π], maintain phase difference i E constant.

[0038] In step d, the target control frequency fc is limited, and the limiting range is ±20% of the inherent resonant frequency of the resonant network.

[0039] The present invention will now be described in more detail with reference to specific embodiments.

[0040] I. Device Parameter Settings

[0041] This embodiment uses a wireless charging system with an LCC-LCC resonant topology. The transmitter parameters are as follows: Resonant network: LCC topology (resonant inductor L1=100μH, resonant capacitor C1=100nF, matching capacitor C2~200nF); Full-bridge inverter: uses Si MOSFETs (model IRF840), with a switching frequency range of 80–120kHz; Zero-crossing detection section: current transformer (ratio 1:1000), high-speed comparator (model LMV7219, response time 5ns). DSP control module: TMS320F28035 (clock frequency 60MHz), enhanced capture unit eCAP configured as rising edge triggered, enhanced pulse width modulation unit ePWM carrier mode is symmetrical counting; PI control module: proportional gain Kp=2, integral gain Ki=50.

[0042] II. Method Execution Flow

[0043] Initialization: The DSP control module loads initial parameters, sets the ePWM initial frequency f0 = 100kHz (corresponding to TBPRD = 300), and the target phase difference θ. ref =0 rad; Zero-crossing capture: The resonant current ILrp is sampled by the current transformer and converted into a square wave signal by the LMV7219 comparator. The eCAP unit captures the rising edge of the square wave and stores the timing value in the CAP1 register; at the same time, it reads the TBCTR register value (Tvpi) of the ePWM. Phase difference calculation: Inverter period T = 1 / f0 = 10μs, calculate θ v =2π×Tvpi / 10μs, θ i =2π×CAP1 / 10μs, i E =θ v -θ i ;like i E =3π / 2 (∈(π,2π)), then the normalized angle is θ EN =3π / 2-2π=-π / 2; Frequency adjustment: Angle difference δe = i E- θ ref =-π / 2-0=-π / 2, PI output fe=2×(-π / 2)+50×∫(-π / 2)dt=-π+(-25π×t), assuming t=0.001s, fe≈-3.14-0.0785≈-3.218kHz, target frequency fc=100-3.218=96.782kHz, update TBPRD=Fclk / (2×fc)=60e6 / (2×96782)≈310; Cyclic Update: Repeat the above steps every PWM cycle (≈10.33μs) to dynamically adjust the frequency to θ. EN It approaches 0 rad.

[0044] III. Experimental Verification Results

[0045] In the charging system of this embodiment, the method of the present invention was compared with existing phase discrimination methods and FFT analysis methods. The results are shown in the table below:

[0046] Experiments show that the method of the present invention is superior to existing solutions in terms of hardware cost, high-frequency response, and control accuracy, and can meet the practical application requirements of wireless charging systems.

[0047] This invention detects the zero-crossing point of the resonant current, records the time using the DSP's capture-capture function (eCAP), and combines this with the value of the PWM timer to directly calculate the phase difference between the current and voltage in an interrupt. This phase information is then used to adjust the operating frequency, achieving automatic frequency tracking and soft-switching control.

[0048] This method features simple hardware, low computational cost, and fast response, and can operate efficiently in high-frequency wireless charging systems, making it particularly suitable for applications with high cost and reliability requirements.

[0049] Finally, it should be emphasized that the above description is only a preferred embodiment of the present invention and is not intended to limit the present invention. For those skilled in the art, the present invention can have various changes and modifications. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the protection scope of the present invention.

Claims

1. A frequency tracking device for the transmitter of a wireless charging system, characterized in that: The device includes The sampling module (1) is used to collect the resonant current signal of the resonant cavity of the transmitter of the wireless charging system; A high-speed comparator (2) is used for zero-crossing detection of the resonant current signal, converting the resonant current signal into a square wave signal with a steep rising edge and zero-crossing information, and sending it to the eCAP pin of the DSP control module (3). The DSP control module (3) has an enhanced capture unit (eCAP) and an enhanced pulse width modulation unit (ePWM) built in. The enhanced capture unit (eCAP) is used to capture the zero-crossing time of the square wave signal and measure the phase difference between the voltage at the zero-crossing time and the midpoint voltage of the inverter at the transmitting end. The enhanced pulse width modulation unit (ePWM) is used to output an adjustable frequency PWM drive signal to the power drive at the transmitting end of the wireless charging system. The power drive drives the full-bridge high-frequency inverter at the transmitting end to provide high-frequency power to the resonant network at the transmitting end. The PI adjustment module (4) is integrated into the DSP control module (3) and is used to dynamically adjust the PWM frequency based on the phase difference measured by the enhanced capture unit (eCAP) as feedback, and send the adjusted PWM frequency to the power drive of the transmitter to achieve automatic frequency tracking.

2. The frequency tracking device for the transmitter of a wireless charging system according to claim 1, characterized in that: The sampling module (1) uses a current transformer or a voltage divider resistor, and the resonant current signal is obtained after being amplified by an amplifier. The gain of the amplifier is adjustable from 1 to 10 times.

3. The frequency tracking device for the transmitter of a wireless charging system according to claim 1, characterized in that: The DSP control module (3) uses the TMS320F28035 DSP chip, and the enhanced capture unit (eCAP) supports rising edge triggered capture with a capture accuracy of ≤1ns.

4. The frequency tracking device for the transmitter of a wireless charging system according to claim 1, characterized in that: The high-speed comparator (2) is an LMV7219 comparator.

5. The frequency tracking device for the transmitter of a wireless charging system according to claim 1, characterized in that: The transmitter resonant network is compatible with LCC, SS, and LCL resonant topologies, and can be adapted to different topologies through the configuration of charging system parameters.

6. A method for frequency tracking of a wireless charging system transmitter using a frequency tracking device as described in any one of claims 1 to 5, characterized in that, The method includes the following steps: a. Signal acquisition: The sampling module (1) acquires the resonant current signal of the resonant cavity of the wireless charging system transmitter and inputs it into the high-speed comparator (2). b. Zero-crossing moment capture: The high-speed comparator (2) converts the resonant current signal into a square wave signal. The enhanced capture unit (eCAP) of the DSP control module (3) is triggered by the rising edge to capture the zero-crossing moment of the square wave signal and record it to the capture register. c. Phase difference calculation: The DSP control module (3) reads the time base counter register value of the enhanced pulse width modulation unit (ePWM) to obtain the midpoint voltage timing value of the inverter at the transmitter. At the same time, the DSP control module (3) reads the capture register value of the enhanced capture unit (eCAP) to obtain the resonant current timing value. The phase difference θ between the midpoint voltage and the resonant current is calculated by the interval judgment method. E and θ E Normalize to the range [-π, π]. d. Closed-loop frequency adjustment: Adjust the phase difference θ E Phase difference θ with the target ref The difference is input to the PI adjustment module (4), and the frequency adjustment amount fe is output. Combined with the reference frequency f0, the target control frequency fc = f0 + fe is obtained. After limiting the target control frequency fc, the carrier frequency of the enhanced pulse width modulation unit (ePWM) is updated. e. Repeat steps a to d to achieve real-time frequency tracking.

7. The frequency tracking method for the transmitter of the wireless charging system according to claim 6, characterized in that, In step c, the phase difference θ between the midpoint voltage and the resonant current is calculated using the interval judgment method. E Specifically: c1. Calculate the phase values ​​of the midpoint voltage and the resonant current: The value at which the enhanced pulse width modulation unit (ePWM) starts timing for the midpoint voltage vpi is denoted as: , The value at which the rising edge of the resonant current ILrp is captured by the enhanced capture unit (eCAP) to start timing is denoted as: , The period value of the full-bridge high-frequency inverter is denoted as: , Then the phase value θ of the midpoint voltage vpi v for: , Phase value θ of resonant current ILrp i for: ; c2. Phase angle difference calculation: Assuming the range of the phase angle difference is (0, 2π), then we have , Define the phase difference between the midpoint voltage vpi and the resonant current ILrp as θ. E The calculation process is as follows: ； c3, Normalized phase difference θ EN calculate: Based on the relationship between the midpoint voltage vpi and the resonant current ILrp, we have: ， ， cos(θ v -θ i ) and sin(θ v -θ i This constitutes a unit vector: , Then vector The direction angle is a normalized angle, and the normalized phase difference θ is extracted using the atan2 function. EN : ， The normalized phase difference θ EN Phase difference θ E The arcsine of the sine value itself is then: θ E =θ EN .

8. The frequency tracking method for the transmitter of the wireless charging system according to claim 7, characterized in that, In step c, the phase difference θ E Normalization to the range [-π, π] specifically means: when θ E When ∈(π,2π), the phase difference θ E Normalized to θ E -2π; when θ E When ∈(-2π,-π), the phase difference θ E Normalized to θ E +2π; when θ E When ∈[-π,π], maintain the phase difference θ E constant.

9. The frequency tracking method for the transmitter of the wireless charging system according to claim 6, characterized in that, In step d, the target control frequency fc is limited, and the limiting range is ±20% of the inherent resonant frequency of the resonant network.

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