Two-degree-of-freedom active disturbance rejection control method for dual three-phase motors driven by nine-switch current sources

Abstract

This invention relates to a two-degree-of-freedom active disturbance rejection (ADNR) control method for a dual three-phase motor driven by a nine-switch current source, belonging to the field of permanent magnet motor control technology. The invention addresses the coupling problem between tracking and disturbance rejection performance in traditional ADNR control for dual three-phase permanent magnet synchronous motors based on current source inverters. The invention includes: establishing a mathematical model of the dq-axis current loop of a nine-switch current source inverter in a synchronous rotating coordinate system, incorporating both periodic and aperiodic disturbances; designing an improved extended state observer, removing the error correction term for the current tracking signal and its differential signal update law in the traditional extended state observer, and designing the disturbance update law as a proportional-integral-repetitive control structure; constructing a two-degree-of-freedom ADNR controller based on the control characteristics of the current source inverter, and independently adjusting the reference tracking performance and disturbance rejection performance based on the output of the improved extended state observer, achieving complete decoupling between the two.

Description

Technical Field

[0001] This invention relates to the field of permanent magnet motor control technology, and in particular to a two-degree-of-freedom active disturbance rejection control method for a dual three-phase motor driven by a nine-switch current source. Background Technology

[0002] Current-source inverters, due to their inductive energy storage on the input side, possess high temperature resistance, long lifespan, and accurate current regulation. On the output side, a parallel filter capacitor and the motor inductor form a second-order LC low-pass filter to improve current waveform quality. Their inherent current control characteristics and short-circuit protection capabilities offer unique advantages in applications such as motor drives. Dual three-phase permanent magnet synchronous motors, due to their high power density, high efficiency, and excellent control performance, are widely used in high-end fields such as electric vehicles, industrial servo systems, and aerospace. However, in actual operation, the current control of dual three-phase permanent magnet synchronous motors based on current-source inverters is affected by various disturbances, mainly including periodic disturbances caused by permanent magnet flux harmonics and switching current overlap, as well as non-periodic disturbances caused by parameter changes and cross-coupling terms. Periodic disturbances introduce... Although the AC output side of a current-source inverter has a second-order LC filter structure, it cannot eliminate odd harmonics. These disturbances can lead to increased current tracking error, exacerbated torque ripple, and in severe cases, even system oscillation, thus limiting further improvements in system performance.

[0003] To suppress the aforementioned disturbances, Active Disturbance Rejection Control (ADRC) has been introduced into the control of permanent magnet synchronous motors. ADRC uses an Extended State Observer (ESO) to estimate and compensate for internal and external disturbances in real time, effectively improving the robustness of the system. However, traditional ADRC methods have the following limitations: First, traditional ESOs have limited observation bandwidth for disturbances, making it difficult to effectively suppress periodic disturbances; second, there is a coupling between the reference tracking performance and disturbance suppression performance of traditional ADRC, resulting in a complex parameter tuning process and making it difficult to simultaneously optimize dynamic response and steady-state accuracy, thus limiting further improvements in controller performance.

[0004] To address the problem of suppressing periodic disturbances, existing technologies typically combine resonant controllers, repetitive controllers, and ADRC (Advanced Dynamic Control) mechanisms. For example, a resonant element is embedded in the control law to enhance the suppression capability for disturbances at specific frequencies. However, these methods often lead to increased coupling between the control law and the disturbance transfer function, more complex parameter tuning, and a tendency to amplify disturbances near the resonant frequency. On the other hand, regarding the performance decoupling problem, although some studies have proposed separating tracking and disturbance rejection by modifying the ESO (Electronic System Oscillator) structure or introducing cascaded observers, this often comes at the cost of increasing system order or computational complexity, making it impractical for engineering implementation.

[0005] Therefore, how to effectively suppress the periodic disturbances of current-source inverters while maintaining the advantages of ADRC disturbance estimation, and achieve complete decoupling of reference tracking and disturbance suppression, has become a key problem that urgently needs to be solved in the field of current control of permanent magnet synchronous motors based on current-source inverters. Summary of the Invention

[0006] To address the problems of insufficient periodic disturbance suppression, coupling of tracking and disturbance rejection performance, and complex parameter tuning in traditional active disturbance rejection control (ADRC) for dual three-phase permanent magnet synchronous motors based on current source inverters, this invention provides a two-degree-of-freedom ADRC method for dual three-phase motors driven by a nine-switch current source. This method achieves complete decoupling of reference tracking performance and disturbance suppression performance solely through improvements to the ESO, thereby enhancing the current loop's ability to suppress periodic and non-periodic disturbances.

[0007] The present invention provides a two-degree-of-freedom active disturbance rejection control method for a dual three-phase motor driven by a nine-switch current source, comprising the following steps:

[0008] S1. Establish a mathematical model of the dq-axis current loop of a nine-switch current source inverter in a synchronous rotating coordinate system, including periodic and non-periodic disturbances.

[0009] S2. Design an improved extended state observer, remove the error correction term of the current tracking signal and its differential signal update law in the traditional extended state observer, and design the disturbance update law as a proportional-integral-repetitive control structure.

[0010] S3. Construct a two-degree-of-freedom active disturbance rejection controller based on the control characteristics of a current source inverter. Based on the output of the improved extended state observer, independently adjust the reference tracking performance and disturbance rejection performance to achieve complete decoupling between the two.

[0011] Preferably, in step S1, the mathematical model of the dq-axis current loop in the synchronous rotating coordinate system is:

[0012] .

[0013] In the formula, These represent the d-axis, q-axis, x-axis, and y-axis currents on the output side of a current-source inverter. These represent the motor's d-axis, q-axis, x-axis, and y-axis currents, respectively. For filtering capacitors, For motor resistance, These are the d-axis and q-axis inductances of the motor, respectively. For the x-axis and y-axis inductance, This represents the number of pole pairs of the motor. The mechanical angular velocity of the motor. These are the voltages of the motor's d-axis and q-axis, respectively.

[0014] Rewriting the above current equations, we obtain the mathematical model of the current loop considering lumped disturbances as follows:

[0015] .

[0016] In the formula, These represent the controller gains for the d-axis, q-axis, x-axis, and y-axis, respectively. , These are lumped disturbances along the d-axis, q-axis, x-axis, and y-axis, including periodic disturbances caused by switching superposition regions, flux harmonics, etc., and non-periodic disturbances caused by parameter changes, cross-coupling terms, etc.

[0017] Preferably, in step S1, the periodic disturbances unique to the current source inverter introduce... Second harmonic It is a positive integer. , The main sources of the subharmonics are: current harmonics caused by the switching superposition region, back electromotive force distortion caused by magnetic flux harmonics, and harmonic currents generated by the parallel resonance of the filter capacitor and the motor inductance.

[0018] The current in phase A of the motor is:

[0019] .

[0020] In the formula, These are the amplitudes of the fundamental wave, the 5th harmonic, and the 7th harmonic, respectively. The electric angular velocity of the motor. These are the fundamental, 5th, and 7th harmonic phases, respectively.

[0021] Preferably, in step S2, the discrete form of the improved extended state observer is:

[0022] . .

[0023] In the formula, z is a complex variable. The errors are the d-axis and q-axis, respectively. They are respectively Actual values ​​of d-axis and q-axis currents at time t. They are respectively Current observations along the d-axis and q-axis at time t. They are respectively Current observations along the d-axis and q-axis at time t. They are respectively The d-axis and q-axis perturbation observations at time t, They are respectively The integral components of the d-axis and q-axis perturbation observations at time 1. They are respectively Error compensation for the integral components of the d-axis and q-axis perturbation observations at time t. They are respectively The d-axis and q-axis perturbation observations at time t, They are respectively The integral components of the d-axis and q-axis perturbation observations at time 1. They are respectively Error compensation for the integral components of the d-axis and q-axis perturbation observations at time t. They are respectively Reference currents on the output side of the d-axis and q-axis current source inverter at time t. Sampling time, For observer gain, For repetitive controllers, , For the proportional gain of the repetitive controller, For the feedback gain of the repetitive controller, The sampling period delay of the repetitive controller. This is for delay compensation of the number of sampling periods for the repetitive controller.

[0024] Preferably, in step S2, the update law of the current tracking signal and its differential signal of the improved extended state observer is:

[0025] . .

[0026] In the formula, For the formula that includes the parameters of the improved extended state observer, .

[0027] Preferably, in step S3, the state error feedback control law of the two-degree-of-freedom active disturbance rejection control is:

[0028] .

[0029] In the formula, They are respectively The d-axis and q-axis currents on the output side of the current source inverter at any given time. They are respectively Reference values ​​of d-axis and q-axis currents at time t. These are the proportional gain and derivative gain of the control law, respectively.

[0030] Preferably, in step S3, the transfer function of the reference tracking performance in the two-degree-of-freedom active disturbance rejection controller... Only with control law parameters The transfer function related to disturbance suppression performance Only with the parameters of the improved extended state observer This is related to achieving complete decoupling between reference tracking performance and disturbance suppression performance;

[0031] The transfer function for reference tracking performance is:

[0032] .

[0033] The transfer function for disturbance suppression performance is:

[0034] .

[0035] In the formula, They are respectively Actual values ​​of d-axis and q-axis currents at time t. They are respectively Actual values ​​of d-axis and q-axis perturbations at time points.

[0036] Preferably, the method is applicable to the current control of a dual three-phase permanent magnet synchronous motor drive system based on a current source inverter.

[0037] The beneficial effects of this invention are:

[0038] (1) The improved extended state observer and two-degree-of-freedom control structure are applied to the current source inverter drive system. By improving the ESO structure, the repetitive control is integrated with the ESO, which significantly improves the estimation and suppression capabilities of periodic disturbances and overcomes the performance limitations of traditional ESO under periodic disturbances.

[0039] (2) Through the two-degree-of-freedom design, the reference tracking performance and disturbance suppression performance are completely decoupled, simplifying the parameter tuning process and enabling the system dynamic response and steady-state accuracy to be optimized independently.

[0040] (3) The proposed method effectively improves the control performance of the current loop without significantly increasing the system order and computational complexity, making it easy to implement and apply in engineering.

[0041] (4) Through simulation and experimental verification, the proposed method shows superior performance in terms of periodic and non-periodic disturbance suppression and reference tracking. Attached Figure Description

[0042] Figure 1 This is the topology diagram of the dual three-phase permanent magnet synchronous motor drive system based on a nine-switch current source inverter of the present invention;

[0043] Figure 2 This is a block diagram of the two-degree-of-freedom active disturbance rejection current control structure based on a nine-switch current source inverter of the present invention, with the d-axis as the axis.

[0044] Figure 3 This is a block diagram of the two-degree-of-freedom active disturbance rejection current control structure based on a nine-switch current source inverter of the present invention, with the q-axis as the axis.

[0045] Figure 4 This is a simplified structural block diagram of the two-degree-of-freedom active disturbance rejection current control based on a nine-switch current source inverter of the present invention, with the d-axis as shown;

[0046] Figure 5 This is a simplified block diagram of the two-degree-of-freedom active disturbance rejection current control structure based on a nine-switch current source inverter of the present invention, with the q-axis as an example.

[0047] Figure 6 This is the q-axis current waveform of a traditional ADRC circuit.

[0048] Figure 7 It is a traditional ADRC FFT analysis plot;

[0049] Figure 8 This is a q-axis current waveform diagram of the method proposed in this invention;

[0050] Figure 9 This is an FFT analysis graph of the method proposed in this invention;

[0051] Figure 10 This is the q-axis current waveform diagram of a traditional ADRC inductor mismatch;

[0052] Figure 11 This is a waveform diagram of the q-axis current due to inductor mismatch in the method proposed in this invention;

[0053] Figure 12 This is a waveform of the q-axis current after reducing the control rate parameter using the method proposed in this invention;

[0054] Figure 13 This is a waveform of the q-axis current after the control rate parameter is improved by the method proposed in this invention;

[0055] Figure 14 This is a waveform of the q-axis current after reducing and improving the ESO parameters using the method proposed in this invention;

[0056] Figure 15 This is a waveform of the q-axis current after the method proposed in this invention improves the ESO parameters. Detailed Implementation

[0057] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0058] It should be noted that, unless otherwise specified, the embodiments and features described in the present invention can be combined with each other.

[0059] The present invention will be further described below with reference to the accompanying drawings and specific embodiments, but this is not intended to limit the scope of the invention.

[0060] Specific Implementation Method 1: The following is combined with... Figures 1 to 15 This embodiment describes a two-degree-of-freedom active disturbance rejection control method for a dual three-phase motor driven by a nine-switch current source, comprising the following steps:

[0061] S1. Establish a mathematical model of the dq-axis current loop of a nine-switch current source inverter in a synchronous rotating coordinate system, including periodic and non-periodic disturbances.

[0062] S2. Design an improved extended state observer, remove the error correction term of the current tracking signal and its differential signal update law in the traditional extended state observer, and design the disturbance update law as a proportional-integral-repetitive control structure.

[0063] S3. Construct a two-degree-of-freedom active disturbance rejection controller based on the control characteristics of a current source inverter. Based on the output of the improved extended state observer, independently adjust the reference tracking performance and disturbance rejection performance to achieve complete decoupling between the two.

[0064] like Figure 1 As shown, the nine-switch current source inverter-dual three-phase permanent magnet synchronous motor drive system involved in this invention comprises the following main components:

[0065] DC power supply: Provides a stable DC bus current.

[0066] DC side inductor As an input energy storage element in a current-source inverter, it is used to smooth the DC current. .

[0067] Nine-switch current source inverter: topology as follows Figure 1As shown, it contains nine power switching transistors. (Such as IGBT or MOSFET), it is divided into three bridge arms, and each bridge arm controls two motor phases.

[0068] LC output filter: Parallel filter capacitor for each phase output Together with the stator inductance of the motor, they form a second-order low-pass filter network.

[0069] Dual three-phase permanent magnet synchronous motor (DTP-PMSM): It has two sets of three-phase windings ABC and UVW, with a spatial phase difference of 30° between the two sets of windings, and each set of windings is connected in a star configuration.

[0070] Controller: The control algorithm is implemented using a digital signal processor (such as TI TMS320F28379D) or FPGA.

[0071] The current control of a dual three-phase permanent magnet synchronous motor based on a current-source inverter is affected by various disturbances, including the periodic disturbances unique to the current-source inverter. Second harmonic It is a positive integer. , The main sources of the subharmonics are: current harmonics caused by the switching superposition region, back electromotive force distortion caused by magnetic flux harmonics, and harmonic currents generated by the parallel resonance of the filter capacitor and the motor inductance.

[0072] The current in phase A of the motor is:

[0073] .

[0074] In the formula, These are the amplitudes of the fundamental wave, the 5th harmonic, and the 7th harmonic, respectively. The electric angular velocity of the motor. These are the fundamental, 5th, and 7th harmonic phases, respectively.

[0075] Figure 1 The topology of a current-source inverter is shown. A filter capacitor is connected in parallel on the output side of the current-source inverter; therefore, the output current of the current-source inverter includes the motor stator current and the filter capacitor current. The current equation of the current-source inverter output side in the synchronous rotating coordinate system is as follows:

[0076] .

[0077] In the formula, These represent the d-axis, q-axis, x-axis, and y-axis currents on the output side of a current-source inverter. These represent the d-axis, q-axis, x-axis, and y-axis currents flowing through the filter capacitor, respectively. These represent the motor's d-axis, q-axis, x-axis, and y-axis currents, respectively. For filtering capacitors, This represents the number of pole pairs of the motor. The mechanical angular velocity of the motor. These represent the voltages along the d-axis, q-axis, x-axis, and y-axis of the motor, respectively.

[0078] The voltage equations of a dual three-phase permanent magnet synchronous motor in the synchronous rotating coordinate system are as follows:

[0079] .

[0080] In the formula, For motor resistance, These are the d-axis and q-axis inductances of the motor, respectively. For the x-axis and y-axis inductance, This represents the fundamental flux linkage amplitude of the permanent magnet.

[0081] Substituting the motor voltage equation into the current equation of the current source inverter output side, we obtain the mathematical model of the current loop of the current source inverter output side current in the synchronous rotating coordinate system with respect to the motor current as follows:

[0082] .

[0083] After simplification, the second-order system can be obtained as follows:

[0084] .

[0085] Unify periodic and non-periodic disturbances into lumped disturbances. The second-order state equation is obtained as follows:

[0086] .

[0087] In the formula, These represent the controller gains for the d-axis, q-axis, x-axis, and y-axis, respectively. , These are lumped disturbances along the d-axis, q-axis, x-axis, and y-axis, including periodic disturbances caused by switching superposition regions, flux harmonics, etc., and non-periodic disturbances caused by parameter changes, cross-coupling terms, etc.

[0088] Taking the dq axis as an example, the improved discretization form of ESO is as follows:

[0089] . .

[0090] In the formula, z is a complex variable. The errors are the d-axis and q-axis, respectively. They are respectively Actual values ​​of d-axis and q-axis currents at time t. They are respectively Current observations along the d-axis and q-axis at time t. They are respectively Current observations along the d-axis and q-axis at time t. They are respectively The d-axis and q-axis perturbation observations at time t, They are respectively The integral components of the d-axis and q-axis perturbation observations at time 1. They are respectively Error compensation for the integral components of the d-axis and q-axis perturbation observations at time t. They are respectively The d-axis and q-axis perturbation observations at time t, They are respectively The integral components of the d-axis and q-axis perturbation observations at time 1. They are respectively Error compensation for the integral components of the d-axis and q-axis perturbation observations at time t. They are respectively Reference currents on the output side of the d-axis and q-axis current source inverter at time t. Sampling time, For observer gain, For repetitive controllers, , For the proportional gain of the repetitive controller, For the feedback gain of the repetitive controller, The sampling period delay of the repetitive controller. This is for delay compensation of the number of sampling periods for the repetitive controller.

[0091] make The improved ESO expression can be rewritten as: . .

[0092] Transforming the above equation, we can obtain the formulas for motor dq-axis current tracking, its differential signal, and disturbance tracking signal as follows:

[0093] . .

[0094] The state error feedback control rate is:

[0095] .

[0096] Substituting the formulas for current tracking, its differential signal, and disturbance tracking signal into the above equation, we can finally obtain:

[0097] .

[0098] In the formula, , They are respectively The d-axis and q-axis currents on the output side of the current source inverter at any given time. They are respectively Reference values ​​of d-axis and q-axis currents at time t. Here are the control law parameters, namely the proportional gain and the derivative gain.

[0099] The simplified block diagram of the current loop active disturbance rejection control structure can be drawn from the control law formula, as follows: Figure 4 and Figure 5 As shown. Based on the simplified current loop structure diagram, the transfer function of the reference tracking performance can be derived as follows:

[0100] .

[0101] The transfer function for disturbance suppression performance is:

[0102] .

[0103] In the formula, They are respectively Actual values ​​of d-axis and q-axis currents at time t. They are respectively Actual values ​​of d-axis and q-axis perturbations at time points.

[0104] As can be seen from the two transfer functions, Includes only control law parameters , Includes only parameters for improved ESO This indicates that the reference tracking performance depends only on the control law parameters and not on the improved ESO parameters; similarly, the disturbance suppression performance depends only on the improved ESO parameters and not on the control law parameters. Therefore, the proposed ADRC controller is a two-degree-of-freedom controller, which fully decouples the reference tracking performance and the disturbance suppression performance.

[0105] Figure 6 , 7Tables 8 and 9 show a comparison of the q-axis current waveforms and FFT analysis between the traditional ADRC and the method proposed in this invention. It can be seen that the traditional ADRC has a steady-state ripple of 3.61A, a THD of 8.83%, a settling time of 0.124s, and an overshoot of 10.27%. The method proposed in this invention has a steady-state ripple of 0.96A, a THD of 1.33%, a settling time of 0.106s, and an overshoot of 2.28%. Because the traditional ADRC has limited ESO bandwidth, it cannot effectively suppress periodic disturbances, resulting in larger current steady-state ripples and THD. In contrast, the method proposed in this invention effectively suppresses periodic disturbances by improving the repetitive control loop in the ESO, significantly reducing current ripple and improving steady-state accuracy. Furthermore, the method proposed in this invention has smaller overshoot and shorter settling time, demonstrating superior reference tracking capability.

[0106] Figure 10 , 11 The paper presents a comparison of the q-axis current waveforms of traditional ADRC and the method proposed in this invention under inductor parameter mismatch. Regarding aperiodic disturbance suppression, when the system is subjected to aperiodic disturbances such as inductor mismatch, the q-axis current response of traditional ADRC exhibits a large drop and a long recovery time; while the method proposed in this invention achieves rapid recovery and minimal fluctuations by improving the disturbance estimation and feedforward compensation of ESO.

[0107] Figure 12 , 13 The graphs show the reference tracking performance verification of the method proposed in this invention under different control rate parameters. Figure 14 , 15 The diagram illustrates the disturbance suppression performance verification of the proposed method under varying ESO parameters. When the control law parameters change, the settling time increases from 0.106 s to 0.113 s, and the overshoot increases from 2.28% to 2.39%, while the steady-state fluctuation remains essentially unchanged. This only affects the reference tracking performance and has no significant impact on the disturbance suppression performance. When the improved ESO parameters change, the steady-state current fluctuation increases from 0.96 A to 1.01 A, while the overshoot and settling time remain essentially unchanged. This only affects the disturbance suppression performance and has no significant impact on the reference tracking performance, fully verifying the two-degree-of-freedom characteristics of the proposed method.

[0108] In summary, this invention effectively solves the problems of insufficient periodic disturbance suppression and coupling of tracking and anti-disturbance performance in the current loop of a dual three-phase permanent magnet synchronous motor drive system based on a nine-switch current source inverter by improving the extended state observer structure and the two-degree-of-freedom controller design. It has important theoretical value and engineering application prospects.

[0109] While the invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the invention. Therefore, it should be understood that many modifications can be made to the exemplary embodiments, and other arrangements can be designed without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood that different dependent claims and features described herein can be combined in ways different from those described in the original claims. It is also understood that features described in conjunction with individual embodiments can be used in other described embodiments.

Claims

1. A dual three-phase motor nine-switch current source drive two-degree-of-freedom active disturbance rejection control method, characterized in that, The method comprises the following steps: S1, establishing a dq-axis current loop mathematical model of a nine-switch current source type inverter containing periodic disturbance and non-periodic disturbance in a synchronous rotating coordinate system; S2, designing an improved extended state observer, deleting an error correction term of a current tracking signal and a differential signal update law thereof in a traditional extended state observer, and designing a disturbance update law as a proportional-integral-repetitive control structure; S3, constructing a two-degree-of-freedom active disturbance rejection controller based on a control characteristic of the current source type inverter, independently adjusting a reference tracking performance and a disturbance suppression performance based on an output of the improved extended state observer, and realizing complete decoupling of the two performances; In step S1, the dq-axis current loop mathematical model in the synchronous rotating coordinate system is: wherein, are motor d-axis, q-axis, x-axis and y-axis currents, respectively, are motor d-axis, q-axis, x-axis and y-axis currents, respectively, is a filter capacitor, is a motor resistance, are motor d-axis and q-axis inductances, respectively, are x-axis and y-axis inductances, is a motor pole pair number, is a motor mechanical angular velocity, are motor d-axis and q-axis voltages, respectively. The dq-axis current loop mathematical model is rewritten to obtain a current loop mathematical model considering lumped disturbance: wherein respectively d-axis, q-axis, x-axis and y-axis controller gains, , respectively d-axis, q-axis, x-axis and y-axis lumped disturbances, including periodic disturbances caused by the switching overmodulation region, flux linkage harmonics and parameter variations, and non-periodic disturbances caused by cross-coupling terms; The periodic disturbance introduced by the current source inverter in step S1 sub-harmonics, is a positive integer, , The sub-harmonics mainly come from: current harmonics caused by switching overcurrent area, back electromotive force distortion caused by flux linkage harmonics, and harmonic currents generated by parallel resonance of filter capacitor and motor inductance. Wherein, the current of the motor A phase is: wherein are the fundamental, 5th, 7th harmonic amplitudes, respectively, is the motor electrical angular velocity, are the fundamental, 5th, 7th harmonic phases, respectively. In step S2, a discrete form of the improved extended state observer is: In the formula, For complex variables, The errors are the d-axis and q-axis, respectively. They are respectively Actual values ​​of d-axis and q-axis currents at time t. They are respectively Current observations along the d-axis and q-axis at time t. They are respectively Current observations along the d-axis and q-axis at time t. They are respectively The d-axis and q-axis perturbation observations at time t. They are respectively The integral components of the d-axis and q-axis perturbation observations at time 1. They are respectively Error compensation for the integral components of the d-axis and q-axis perturbation observations at time t. They are respectively The d-axis and q-axis perturbation observations at time t, They are respectively The integral components of the d-axis and q-axis perturbation observations at time 1. They are respectively Error compensation for the integral components of the d-axis and q-axis perturbation observations at time t. They are respectively Reference currents on the output side of the d-axis and q-axis current source inverter at time t. Sampling time, For observer gain, For repetitive controllers, , For the proportional gain of the repetitive controller, For the feedback gain of the repetitive controller, The sampling period delay of the repetitive controller. This is for delay compensation of the number of sampling periods for the repetitive controller.

2. The dual three-phase motor nine-switch current source drive two-degree-of-freedom active disturbance rejection control method according to claim 1, characterized in that, In step S2, the current tracking signal and the differential signal update law thereof of the improved extended state observer are: In the formula, is an intermediate variable for characterizing the improved extended state observer parameters; 。 3. The dual three-phase motor nine-switch current source drive two-degree-of-freedom active disturbance rejection control method according to claim 2, characterized in that, In step S3, a state error feedback control law of the two-degree-of-freedom active disturbance rejection control is: In the formula, respectively d-axis and q-axis currents at the output side of the current source type inverter, respectively d-axis and q-axis current reference values at the time, respectively 4. The dual three-phase motor nine-switch current source drive two-degree-of-freedom active disturbance rejection control method according to claim 3, characterized in that, In step S3, the transfer function of the reference tracking performance in the two-degree-of-freedom active disturbance rejection controller is only related to the control law parameters The transfer function of the disturbance rejection performance is only related to the improved extended state observer parameters The complete decoupling of the reference tracking performance and the disturbance rejection performance is achieved; Wherein, a transfer function of the reference tracking performance is: A transfer function of the disturbance suppression performance is: In the formula, are respectively d-axis and q-axis current actual values at the time point, are respectively d-axis and q-axis disturbance actual values at the time point.

5. The dual three-phase motor nine-switch current source drive two-degree-of-freedom active disturbance rejection control method of claim 1, wherein, The method is suitable for current control of a dual three-phase permanent magnet synchronous motor driving system based on a current source type inverter.

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