Portable load adaptive bidirectional charging and discharging circuit module

Abstract

This invention relates to the field of power electronics technology, specifically to a portable load-adaptive bidirectional charge-discharge circuit module. The module includes a battery and a load port; the first half-bridge is composed of gallium nitride (GaN) switches, and the second half-bridge is composed of silicon-based field-effect transistors (MOSFETs); the main winding of a coupling inductor connects the switching nodes of the two half-bridges, and the induced voltage output from the secondary winding is rectified and filtered to provide bias power; an auxiliary resonant branch, including an auxiliary switch and a resonant capacitor, is connected in parallel across the switches, with the auxiliary switch's on-time preceding that of the first GaN switch; the control unit determines the DC current and high-frequency ripple component based on the shunt resistor and the secondary winding signal, and generates a PWM signal based on the port voltage to control the on and off of all switches. This invention achieves self-biased power supply and high-precision composite current detection, while reducing switching losses and improving energy conversion efficiency.

Description

Technical Field

[0001] This invention relates to the field of power electronics technology. More specifically, this invention relates to a portable load-adaptive bidirectional charge-discharge circuit module. Background Technology

[0002] Bidirectional charge-discharge circuits are responsible for bidirectional energy dispatch between the battery and the load or power source. They typically employ a four-switch non-isolated bidirectional Buck-Boost converter topology, enabling voltage buck-boost conversion without polarity reversal. To reduce the size of passive components like inductors and capacitors and increase power density, current technologies usually require increasing the circuit's switching frequency. However, in silicon-based power device circuits, increasing the switching frequency significantly increases switching losses, reducing overall conversion efficiency and introducing heat dissipation issues, which contradicts the requirements for thinner and lighter portable devices.

[0003] To address the aforementioned issues, existing technologies employ wide-bandgap semiconductor devices such as gallium nitride (GaN) to replace silicon-based MOSFETs, leveraging their low on-resistance and switching losses to support high operating frequencies. Furthermore, soft-switching techniques, such as zero-voltage or zero-current switching, are utilized by adding an auxiliary resonant network to create soft-switching conditions for the main switch, thereby reducing switching losses.

[0004] However, the aforementioned improvements to the existing technology have the following drawbacks: First, gallium nitride devices are expensive and have stringent requirements for the drive circuit; second, soft-switching solutions require more auxiliary components and control, and the implementation range of soft switching is limited by changes in load and input voltage, making it difficult to maintain high efficiency under a wide range of operating conditions; third, the circuit uses a series sampling resistor for rapid detection of inductor current, which introduces additional power losses; finally, the control system and drive circuit require a stable and reliable auxiliary power supply, and setting up an additional auxiliary power converter will occupy valuable space and cost in portable devices, reducing integration. Summary of the Invention

[0005] The purpose of this invention is to propose a portable load adaptive bidirectional charging and discharging circuit module to solve the problem that the selection of switching devices, topology, control strategy and auxiliary functions cannot be optimized in a coordinated manner at the system level in the current field; to this end, the present invention provides a solution in one aspect.

[0006] This invention provides a portable load-adaptive bidirectional charge-discharge circuit module, comprising: a battery port and a load port; a first half-bridge circuit connected to the battery port; the first half-bridge circuit being composed of a first gallium nitride switch and a second gallium nitride switch; a second half-bridge circuit connected to the load port; the second half-bridge circuit being composed of a first silicon-based metal-oxide-semiconductor field-effect transistor and a second silicon-based metal-oxide-semiconductor field-effect transistor; a coupled inductor including a main winding and a secondary winding, the main winding being connected between the switching nodes of the first half-bridge circuit and the switching nodes of the second half-bridge circuit; one end of the secondary winding being grounded, and the other end being used to output an induced voltage proportional to the rate of change of the main winding current, which, after rectification and filtering, provides bias power to the control unit; and an auxiliary winding being connected in series. The auxiliary resonant branch, consisting of an auxiliary switch and a resonant capacitor, is connected in parallel across the second gallium nitride switch in the first half-bridge circuit. The turn-on timing of the auxiliary switch is set to precede that of the first gallium nitride switch in the first half-bridge circuit. The control unit's signal acquisition terminal is connected to the battery port, the load port, the shunt resistor connected in series in the main winding circuit, and the secondary winding of the coupling inductor. Based on the voltage drop signal of the shunt resistor and the induced voltage signal of the secondary winding, the control unit jointly determines the DC component and high-frequency ripple component of the main circuit current, and, in conjunction with the voltages of the battery port and the load port, generates a set of pulse width modulation signals to control the turn-on and turn-off timing of all switching devices in the first half-bridge circuit, the second half-bridge circuit, and the auxiliary resonant branch.

[0007] Preferably, the switching node of the first half-bridge circuit is the connection point between the source of the first gallium nitride switch and the drain of the second gallium nitride switch; the switching node of the second half-bridge circuit is the connection point between the source of the first silicon-based metal-oxide-semiconductor field-effect transistor and the drain of the second silicon-based metal-oxide-semiconductor field-effect transistor.

[0008] Preferably, the induced voltage output from the secondary winding of the coupled inductor passes sequentially through a half-wave rectifier circuit composed of Schottky diodes, a π-type filter circuit composed of two capacitors and one inductor, and a low-dropout linear regulator to generate a stable DC voltage to power the control unit.

[0009] Preferably, the turns ratio of the main winding to the auxiliary winding of the coupled inductor is set to 20:1 or 10:1.

[0010] Preferably, the auxiliary switch in the auxiliary resonant branch is an enhancement-mode p-channel metal-oxide-semiconductor field-effect transistor; a fast recovery diode is connected in reverse parallel across the resonant capacitor to provide a discharge path for the energy stored on the resonant capacitor after the auxiliary switch is turned off, and to clamp the reverse voltage on the resonant capacitor.

[0011] Preferably, the control unit has a built-in digital signal processor. The voltage drop signal of the shunt resistor is sampled after passing through a second-order Butterworth low-pass filter to calculate the DC component of the main circuit current. The induced voltage of the secondary winding of the coupled inductor is sampled after passing through a capacitively coupled high-pass filter and combined with the mutual inductance coefficient of the coupled inductor to calculate the high-frequency ripple component of the main circuit current.

[0012] Preferably, the control unit adopts a phase-shift control strategy, which controls the direction and power of energy transmission by adjusting the phase difference of the pulse width modulation signal between the first half-bridge and the second half-bridge; the drive signal of the auxiliary switch maintains a fixed lead time with the drive signal of the first gallium nitride switch.

[0013] Preferably, the portable load adaptive bidirectional charge and discharge circuit module further includes an over-temperature protection circuit, which includes a negative temperature coefficient thermistor. The negative temperature coefficient thermistor is in close contact with the surface of the main winding of the coupled inductor. The negative temperature coefficient thermistor is connected in series with a fixed resistor to divide the voltage and then connected to the analog comparator input terminal of the control unit. When the detected temperature exceeds a preset threshold, the control unit puts all switching devices in the off state.

[0014] Preferably, a 5-ohm ferrite bead is connected in series in the gate drive circuit of the first gallium nitride switch; and a 10-ohm ferrite bead is connected in series in the gate drive circuit of the second gallium nitride switch, to suppress electromagnetic interference generated by different switches during the switching process.

[0015] Preferably, the DC component and high-frequency ripple component of the main circuit current are jointly determined based on the voltage drop signal of the shunt resistor and the induced voltage signal of the secondary winding, and a set of pulse width modulation signals is generated in combination with the voltages of the battery port and the load port, specifically as follows:

[0016] The control unit sets a target voltage reference value for the battery port or load port according to the operating mode; the operating mode is either charging mode or discharging mode; the control unit acquires the actual voltage of the battery port or load port and compares it with the target voltage reference value to generate a voltage error signal; the voltage error signal is compensated by a first proportional-integral controller to generate a target reference value for the main circuit current; the control unit acquires the voltage drop signal of the shunt resistor to determine the DC component of the main circuit current, and acquires the induced voltage signal of the secondary winding of the coupled inductor to determine the high-frequency ripple component of the main circuit current, and superimposes the two to obtain the actual voltage of the main circuit. The current feedback value is compared with the actual current feedback value to generate a current error signal. The current error signal is compensated by a second proportional-integral controller to generate a phase shift angle for adjusting power transmission. Based on the phase shift angle, the control unit generates two sets of complementary pulse width modulation signals with a duty cycle of 50%, which are used to drive the first half-bridge and the second half-bridge, respectively. There is a phase difference between the two sets of signals determined by the phase shift angle. The control unit generates a drive signal for the auxiliary switch, whose turn-on timing has a preset lead time relative to the turn-on timing of the first gallium nitride switch of the first half-bridge to achieve soft switching operation.

[0017] The beneficial effects of this invention are as follows: This invention utilizes the secondary winding of the coupled inductor to provide self-biased power supply to the control unit, reducing circuit size and material costs. Simultaneously, this secondary winding is also used to extract the high-frequency ripple component of the main circuit current, which, combined with the DC component detected by the shunt resistor, constitutes a composite current sampling scheme, improving the completeness of current detection and control accuracy. The added auxiliary resonant branch establishes a zero-voltage turn-on condition for the gallium nitride switch in the first half-bridge, reducing switching losses, improving overall energy conversion efficiency, and reducing device thermal stress. Differential gate drive suppression for different switching transistors and temperature monitoring of key inductor components enhance the electromagnetic compatibility of circuit operation and the reliability of long-term operation. Attached Figure Description

[0018] Figure 1 The schematic diagram illustrates the circuit diagram of the portable load adaptive bidirectional charge and discharge circuit module in this embodiment. Detailed Implementation

[0019] The technical solutions of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention.

[0020] like Figure 1As shown, a portable load-adaptive bidirectional charge-discharge circuit module in this embodiment includes: a battery port and a load port; a first half-bridge circuit 1 connected to the battery port; and a second half-bridge circuit 2 connected to the load port. The first half-bridge circuit 1 is composed of a first gallium nitride switch T11 and a second gallium nitride switch T12; the second half-bridge circuit 2 is composed of a first silicon-based metal-oxide-semiconductor field-effect transistor T21 and a second silicon-based metal-oxide-semiconductor field-effect transistor T22.

[0021] For example, the battery port uses an XT60 connector for connecting lithium-ion battery packs with a nominal voltage range of 12V to 24V. The load port uses a USB Type-C connector that supports the Universal Serial Bus Power Delivery protocol and provides an output voltage of 5V to 20V.

[0022] In the first half-bridge circuit 1 connected to the battery port, a gallium nitride switch with a higher switching frequency and lower on-resistance is used to handle high voltage and reduce switching losses. In the second half-bridge circuit 2 connected to the load port, a silicon-based metal-oxide-semiconductor field-effect transistor with a lower cost is used to balance performance and reduce cost.

[0023] Specifically, such as Figure 1 As shown, the drain of the first gallium nitride switch T11 is connected to the positive terminal of the battery port, and the source is connected to the switch node N1; the drain of the second gallium nitride switch T12 is connected to the switch node N1, and the source is grounded.

[0024] The drain of the first silicon-based metal-oxide-semiconductor field-effect transistor T21 is connected to the positive terminal of the load port, and the source is connected to the switching node N2; the drain of the second silicon-based metal-oxide-semiconductor field-effect transistor T22 is connected to the switching node N2, and the source is grounded.

[0025] In an optional embodiment, the switching node N1 of the first half-bridge circuit 1 is the connection point between the source of the first gallium nitride switch T11 and the drain of the second gallium nitride switch T12; the switching node N2 of the second half-bridge circuit 2 is the connection point between the source of the first silicon metal oxide semiconductor field-effect transistor T21 and the drain of the second silicon metal oxide semiconductor field-effect transistor T22.

[0026] Specifically, in the first half-bridge circuit 1, the drain of the upper gallium nitride switch T11 is connected to the positive terminal of the battery port, and the source of the lower gallium nitride switch T12 is grounded. This connection method constitutes a standard totem-pole half-bridge structure. The switching node N1 of the first half-bridge circuit 1 is also the core node for energy exchange with the main winding of the coupled inductor. In discharge mode, the voltage of the switching node N1 of the first half-bridge circuit 1 switches rapidly between the battery voltage and the ground level at a frequency of several hundred kilohertz.

[0027] In the second half-bridge circuit 2, the drain of the upper transistor, the first silicon-oxide-semiconductor field-effect transistor T21, is connected to the positive terminal of the load port, while the source of the lower transistor, the second silicon-oxide-semiconductor field-effect transistor T22, is grounded. The switching node N2 of the second half-bridge circuit 2 is connected to the other end of the coupling inductor. In charging mode, the voltage at the switching node N2 of the second half-bridge circuit 2 switches between the load voltage and ground level. By coordinating the voltage waveforms of the two switching nodes, bidirectional energy flow between the battery and the load is achieved.

[0028] The portable load adaptive bidirectional charging and discharging circuit module in this embodiment also includes: a coupling inductor 3, an auxiliary resonant branch 4, and a control unit 5.

[0029] like Figure 1 As shown, the coupling inductor 3 includes a main winding and a secondary winding. The main winding is connected between the switching node N1 of the first half-bridge circuit 1 and the switching node N2 of the second half-bridge circuit 2. One end of the secondary winding is grounded, and the other end is used to output an induced voltage proportional to the rate of change of the current of the main winding, which is then rectified and filtered to provide bias power to the control unit.

[0030] In an optional embodiment, the turns ratio of the primary winding to the secondary winding of the coupled inductor can be set to 20:1 or 10:1.

[0031] In an optional embodiment, the induced voltage output from the secondary winding of the coupled inductor passes sequentially through a half-wave rectifier circuit composed of Schottky diodes, a π-type filter circuit, and a low-dropout linear regulator to generate a stable DC voltage to power the control unit; wherein, the π-type filter circuit consists of two capacitors and one inductor.

[0032] For example, the turns ratio of the main winding to the secondary winding of the coupled inductor is set to 10:1. When the switching frequency of the main circuit is 200 kHz, the secondary winding induces a pulsed AC voltage with a peak value of approximately 8V at the same frequency. This induced voltage is fed to a Schottky diode (model SS14) for half-wave rectification to obtain a pulsating DC voltage. In this process, the low forward voltage drop characteristic of the Schottky diode is utilized to maximize energy retention.

[0033] The pulsating DC voltage then passes through a π-type filter network consisting of a 10 microfarad input capacitor, a 22 microhenry inductor, and a 47 microfarad output capacitor. This π-type filter network can filter out the switching frequency and harmonic components, suppressing the voltage ripple to within 200 millivolts, forming a relatively smooth but still unregulated DC voltage.

[0034] Then, the DC power is input to a low dropout linear regulator of model RT9193 and outputs a stable 3.3V DC voltage to power the digital signal processor and peripheral sampling circuits in the control unit, ensuring the stability of the control core in an electromagnetic environment.

[0035] In other words, in this circuit, by adding a secondary winding with a turns ratio of 20:1 or 10:1 during the winding of the main inductor, an alternating magnetic field generated on the inductor when the main circuit switches is used to induce an AC voltage in the secondary winding. This voltage is half-wave rectified by a Schottky diode, then filtered by a π-type filter consisting of a 10 μF capacitor, a 100 μH inductor, and another 10 μF capacitor, and input to a low-dropout linear regulator with an output voltage of 3.3V, thereby providing a clean and stable operating power supply for the control unit.

[0036] like Figure 1 As shown, the auxiliary resonant branch 4 includes an auxiliary switch and a resonant capacitor connected in series; the auxiliary resonant branch 4 is connected in parallel across the second gallium nitride switch T12 of the first half-bridge circuit 1; wherein, the turn-on timing of the auxiliary switch precedes the turn-on timing of the first gallium nitride switch T11 of the first half-bridge circuit 1.

[0037] In an optional embodiment, the auxiliary switch in the auxiliary resonant branch 4 is an enhancement-mode p-channel metal-oxide-semiconductor field-effect transistor; a fast recovery diode is connected in reverse parallel across the resonant capacitor to provide a discharge path for the energy stored on the resonant capacitor after the auxiliary switch is turned off, clamping the reverse voltage on the resonant capacitor. The fast recovery diode can be a MUR120.

[0038] Specifically, the source of the p-channel MOSFET is connected to the drain of the second gallium nitride switch T12, i.e., the switching node N1 of the first half-bridge circuit 1. The drain is connected to the negative terminal of the battery port through a resonant capacitor, making the drive circuit design relatively simple. During the dead time of the first half-bridge circuit 1, and approximately 50 nanoseconds before the first gallium nitride switch T11 is about to turn on, the control unit drives the enhancement-mode p-channel MOSFET to turn on. At this time, the resonant capacitor resonates through the enhancement-mode p-channel MOSFET and the main winding of the coupling inductor 3, pulling the voltage of the switching node N1 of the first half-bridge circuit 1 to zero, creating conditions for the zero-voltage turn-on of the first gallium nitride switch T11.

[0039] After the brief resonant period ends, the enhancement-mode p-channel MOSFET turns off. Due to the continuous effect of the inductor current, the voltage across the resonant capacitor may experience reverse overshoot. At this time, the fast recovery diode connected in anti-parallel to the resonant capacitor turns on, clamping the reverse voltage across the capacitor to a diode voltage drop of approximately 0.7V. This also provides a fast release path for the remaining resonant energy, preventing excessive reverse voltage from damaging the enhancement-mode p-channel MOSFET and ensuring consistent initial resonant conditions for each switching cycle.

[0040] Therefore, in this circuit, a series branch consisting of an enhancement-mode p-channel metal-oxide-semiconductor field-effect transistor as an auxiliary switch and a resonant capacitor is connected in parallel across the two ends of the second gallium nitride switch T12 in the first half-bridge circuit 1. During the dead time of each switching cycle, when the second gallium nitride switch T12 is turned off and before the first gallium nitride switch T11 is turned on, the control unit briefly turns on the auxiliary switch. At this time, the resonant capacitor resonates with the main inductor, quickly pulling the voltage of the switching node N1 of the first half-bridge circuit 1 to ground potential, thereby creating a zero-voltage turn-on condition for the first gallium nitride switch that is about to be turned on, reducing turn-on losses.

[0041] The signal acquisition terminal of the control unit 5 is connected to the battery port, the load port, the shunt resistor connected in series in the main winding circuit, and the secondary winding of the coupling inductor. The control unit determines the DC component and high-frequency ripple component of the main circuit current based on the voltage drop signal of the shunt resistor and the induced voltage signal of the secondary winding, and generates a set of pulse width modulation signals in combination with the voltages of the battery port and the load port to control the turn-on and turn-off timing of all switching devices in the first half-bridge circuit 1, the second half-bridge circuit 2, and the auxiliary resonant branch 4.

[0042] In an optional embodiment, the control unit has a built-in digital signal processor. The voltage drop signal of the shunt resistor is sampled after passing through a second-order Butterworth low-pass filter to calculate the DC component of the main circuit current. The induced voltage of the secondary winding of the coupled inductor is sampled after passing through a capacitively coupled high-pass filter and combined with the mutual inductance coefficient of the coupled inductor to calculate the high-frequency ripple component of the main circuit current.

[0043] For example, a 5 milliohm shunt resistor is connected in series with the negative terminal of the battery port. The current signal flowing through this shunt resistor contains a DC component and high-frequency ripple. This signal passes through a second-order Butterworth low-pass filter with a Sallen-Key topology built from an operational amplifier, with the filter's cutoff frequency set to 1 kHz. This filter attenuates the ripple at the 200 kHz switching frequency by more than 60 dB, thus outputting a smooth DC voltage signal. A 12-bit analog-to-digital converter inside the digital signal processor samples this signal at a frequency of 10 kHz to calculate the average charge and discharge current of the battery.

[0044] The induced voltage in the secondary winding of the coupled inductor is proportional to the rate of change of the primary winding current, reflecting current ripple information. This induced voltage is filtered out by a passive capacitive high-pass filter with a cutoff frequency of 10 kHz to remove DC bias and low-frequency noise, and then sampled at high speed by another analog-to-digital converter of the digital signal processor. Based on the sampled value and the known mutual inductance coefficient M, the digital signal processor reconstructs the high-frequency ripple waveform of the primary circuit current in real time through digital integration, which is used to achieve cycle-by-cycle peak current control.

[0045] This circuit employs a composite current detection scheme. One path uses a 5 milliohm shunt resistor. The current signal flowing through this shunt resistor is amplified 100 times by an instrumentation amplifier and then passed through a second-order active low-pass filter with a cutoff frequency of 5 kHz to obtain the average current of the main circuit, i.e., the DC component. The other path acquires the AC voltage induced by the secondary winding of the coupled inductor. This voltage is proportional to the rate of change of the main winding current. After being shaped by a high-speed comparator, it is used to monitor the peak and valley information of the current ripple in real time, i.e., the high-frequency ripple component. The digital signal processor inside the control unit integrates the two current information to implement an internal and external dual closed-loop control algorithm based on predictive current control mode. It generates four pulse width modulation signals and one auxiliary pulse width modulation signal to drive the four switches and the auxiliary switch in the first half-bridge circuit 1 and the second half-bridge circuit 2, respectively.

[0046] In an optional embodiment, the control unit 5 employs a phase-shift control strategy, adjusting the phase difference of the pulse width modulation signal between the first half-bridge circuit 1 and the second half-bridge circuit 2 to control the direction and power of energy transmission; the drive signal of the auxiliary switch maintains a fixed lead time with the drive signal of the first gallium nitride switch T11.

[0047] Specifically, the duty cycles of the upper and lower transistors in the first half-bridge circuit 1 and the second half-bridge circuit 2 are close to 50%. When the battery discharges to the load, the control unit 5 causes the pulse width modulation signal of the first half-bridge circuit 1 to lead the pulse width modulation signal of the second half-bridge circuit 2, increasing the phase difference from 0 degrees to 90 degrees, and the transmission power also increases from zero to its maximum value. Conversely, when charging the battery from the load, the pulse width modulation signal of the second half-bridge circuit 2 leads the first half-bridge circuit 1, and the charging power is controlled by adjusting the phase difference.

[0048] The auxiliary switch is used to implement soft switching for the first gallium nitride switch T11; therefore, the auxiliary switch is turned on before the first gallium nitride switch T11 is turned on. The control unit 5 sets the rising edge of the pulse width modulation drive signal of the auxiliary switch to always lead the rising edge of the drive signal of the first gallium nitride switch T11 by a certain time, for example, 45 nanoseconds. This lead time is calculated based on the parameters of the auxiliary resonant branch 4 and preset in the firmware of the digital signal processor to ensure reliable triggering of the zero-voltage turn-on condition under all operating conditions.

[0049] In an optional embodiment, the portable load adaptive bidirectional charge and discharge circuit module further includes: an over-temperature protection circuit, the over-temperature protection circuit including a negative temperature coefficient thermistor, the negative temperature coefficient thermistor being in close contact with the surface of the main winding of the coupled inductor; the negative temperature coefficient thermistor and a fixed resistor are connected in series to divide the voltage and then connected to the analog comparator input terminal of the control unit; when the detected temperature exceeds a preset threshold, the control unit controls all switching devices to be in the off state.

[0050] For example, a negative temperature coefficient (NTC) thermistor with a Beta of 3950 and a resistance of 10 kΩ at 25°C is selected. This NTC thermistor is tightly bonded to the hottest point of the core or winding of the coupled inductor using thermal grease. This NTC thermistor is connected in series with a 10 kΩ resistor between a 3.3V reference power supply and ground, forming a temperature-dependent voltage divider circuit.

[0051] The voltage at the voltage divider point is fed into an analog comparator integrated within the control unit. The other input of this comparator is connected to a reference voltage set by a digital potentiometer, corresponding to a temperature threshold of 85 degrees Celsius. As the temperature of the coupled inductor rises, the thermistor resistance decreases, and the voltage at the voltage divider point rises. If this voltage exceeds the reference voltage, the analog comparator output flips, triggering a non-maskable interrupt. In the interrupt service routine, the control unit immediately disables all pulse-width modulation output channels, turning off all switching devices and achieving hardware-level rapid over-temperature protection.

[0052] In an optional embodiment, a 5-ohm ferrite bead is connected in series in the gate drive circuit of the first gallium nitride switch; and a 10-ohm ferrite bead is connected in series in the gate drive circuit of the second gallium nitride switch, to suppress electromagnetic interference generated by different switches during the switching process.

[0053] The first gallium nitride switch, acting as the upper transistor, is susceptible to the Miller plateau effect during switching, which can affect the drive circuit. Furthermore, its high switching speed makes it prone to high-frequency oscillations. Therefore, in the circuit of this invention, a 5-ohm ferrite bead is connected in series in the gate drive circuit of the first gallium nitride switch and placed close to the gate pin. This ferrite bead provides moderate damping without significantly increasing the turn-on / turn-off delay, suppressing oscillations caused by parasitic inductance and capacitance in the gate circuit, and ensuring a clean gate voltage waveform without overshoot.

[0054] The second gallium nitride switch, acting as the lower transistor, has its source grounded, resulting in a relatively simple drive circuit. However, it may still experience ringing due to other parasitic parameters in the circuit during turn-off. Therefore, in the circuit of this invention, a ferrite bead with an equivalent resistance of 10 ohms at high frequencies is connected in series in the gate drive circuit. This slightly larger resistance bead slows down the switching edge rate, creating a slight asymmetry with the switching rate of the upper transistor. This asymmetry helps to avoid the spectral peaks of high-frequency harmonics, thereby optimizing electromagnetic compatibility performance.

[0055] In an optional embodiment, the DC component and high-frequency ripple component of the main circuit current are jointly determined based on the voltage drop signal of the shunt resistor and the induced voltage signal of the secondary winding, and a set of pulse width modulation signals is generated in combination with the voltages of the battery port and the load port, specifically as follows:

[0056] The control unit sets the target voltage reference value of the battery port or load port according to the operating mode; the operating mode is either charging mode or discharging mode.

[0057] The control unit acquires the actual voltage of the battery port or load port and compares it with the target voltage reference value to generate a voltage error signal; the voltage error signal is compensated by the first proportional-integral controller to generate a target reference value for the main circuit current.

[0058] The control unit collects the voltage drop signal of the shunt resistor to determine the DC component of the main circuit current, and collects the induced voltage signal of the coupled inductor secondary winding to determine the high-frequency ripple component of the main circuit current. The two are superimposed to obtain the actual current feedback value of the main circuit.

[0059] The target reference value of the generated main circuit current is compared with the actual current feedback value to generate a current error signal; the current error signal is compensated by a second proportional-integral controller to generate a phase shift angle for adjusting power transmission.

[0060] Based on the phase shift angle, the control unit generates two sets of complementary pulse width modulation signals with a duty cycle close to 50%, which are used to drive the first half-bridge circuit and the second half-bridge circuit, respectively. The phase difference between the two sets of signals is determined by the phase shift angle. The control unit generates a drive signal to drive the auxiliary switch, and its turn-on timing is ahead of the turn-on timing of the first gallium nitride switch of the first half-bridge circuit by a preset time to achieve soft switching operation.

[0061] Specifically, if the current mode is charging, and the goal is to charge a 3.7V battery to 4.2V, then the target voltage reference value is 4.2V. The controller monitors the actual battery voltage in real time, for example, if it is currently 3.8V. It compares the target voltage reference value with the actual voltage, obtaining a voltage error signal of 0.4V. This voltage error signal is input to the first proportional-integral controller. Based on the magnitude and duration of the voltage error signal, the controller calculates the most suitable charging current. For example, to reach the target as quickly as possible, a 2-amp main loop current is needed. The generated target reference value of 2A for the main loop current is then passed to the next level of control.

[0062] Upon receiving a command requiring 2A of current, the controller needs to control the actual current of the circuit to achieve the target. The actual current of the main circuit is obtained in two ways: the average current is obtained using the shunt resistor, for example, 1.9A; simultaneously, the high-frequency ripple of the current is obtained using the induced voltage of the coupled inductor's secondary winding, for example, fluctuations of ±0.1A. The determination of the high-frequency ripple component of the main circuit current utilizes the principle of electromagnetic induction. When a rapidly changing ripple current flows through the main winding, it generates a changing magnetic field. This changing magnetic field induces a voltage in the secondary winding, and the magnitude of this induced voltage is proportional to the rate of change of the main winding current. Since the DC component is constant and its rate of change is zero, it does not induce a voltage in the secondary winding. The control unit only needs to collect the voltage signal induced in the secondary winding to obtain information proportional to the high-frequency ripple current. Combined with the known inductor parameters, the specific value and shape of the ripple component can be calculated.

[0063] Superimposing the two values ​​yields an actual current feedback value of 1.9A ± 0.1A. The controller compares this actual current feedback value with the target reference value, generating a current error signal of 0.1A. This current error signal is then fed into the second proportional-integral controller to calculate a specific execution parameter, namely the phase shift angle. More specifically, the controller calculates the phase shift angle based on the magnitude and duration of the current error signal.

[0064] Suppose that the phase shift angle needs to be adjusted from 15 degrees to 16 degrees to increase the current from 1.9A to 2A. Based on the 16-degree phase shift angle, the control unit generates two sets of drive signals to control the switches in the first and second half-bridge circuits. There is a slight time difference, or phase difference, between the switching actions of the two sets of signals, which determines the amount of power transferred. The PWM generator within the control unit generates two sets of square wave signals: one to drive the first half-bridge circuit at the battery end, and the other to drive the second half-bridge circuit at the load end. When the phase shift angle is zero, the rising and falling edges of the two sets of signals are perfectly synchronized. When the controller calculates a non-zero phase shift angle, such as 20 degrees, the PWM generator delays the start-up time of one set of signals relative to the other by a certain amount of time. This time delay is equal to the proportion of the 20-degree angle in the entire switching cycle (360 degrees). Furthermore, a drive signal that is always slightly ahead is generated for the auxiliary switch to ensure less loss during the main switch's switching, achieving soft switching.

[0065] In the description of this specification, "multiple" means at least two, such as two, three or more, etc., unless otherwise expressly and specifically defined.

[0066] While various embodiments of the invention have been shown and described in this specification, it will be apparent to those skilled in the art that such embodiments are provided by way of example only. Many modifications, alterations, and alternatives will occur to those skilled in the art without departing from the spirit and essence of the invention.

Claims

1. A portable load-adaptive bidirectional charging and discharging circuit module, characterized in that, include: Battery port and load port; The first half-bridge circuit connected to the battery port; The first half-bridge circuit consists of a first gallium nitride switch and a second gallium nitride switch; The second half-bridge circuit connected to the load port; The second half-bridge circuit is composed of a first silicon-based metal-oxide-semiconductor field-effect transistor and a second silicon-based metal-oxide-semiconductor field-effect transistor. The system includes a coupled inductor consisting of a main winding and a secondary winding. The main winding is connected between the switching nodes of the first half-bridge circuit and the second half-bridge circuit. One end of the secondary winding is grounded, and the other end is used to output an induced voltage that is proportional to the rate of change of the main winding current. After rectification and filtering, the induced voltage provides bias power to the control unit. An auxiliary resonant branch, including an auxiliary switch and a resonant capacitor connected in series, is connected in parallel across the second gallium nitride switch of the first half-bridge circuit. The turn-on timing of the auxiliary switch is set to precede the turn-on timing of the first gallium nitride switch of the first half-bridge circuit. The control unit's signal acquisition terminal is connected to the battery port, load port, shunt resistor connected in series in the main winding circuit, and the secondary winding of the coupling inductor. Based on the voltage drop signal from the shunt resistor and the induced voltage signal from the secondary winding, the control unit jointly determines the DC component and high-frequency ripple component of the main circuit current. Combined with the voltages at the battery port and load port, it generates a set of pulse width modulation signals to control the on / off timing of all switching devices in the first half-bridge circuit, the second half-bridge circuit, and the auxiliary resonant branch. Specifically, the generated set of pulse width modulation signals is as follows: The control unit sets the target voltage reference value for the battery port or load port according to the operating mode; the operating mode is either charging mode or discharging mode. The control unit acquires the actual voltage at the battery port or load port and compares it with the target voltage reference value to generate a voltage error signal. The voltage error signal is compensated by the first proportional-integral controller to generate the target reference value of the main circuit current. The control unit collects the voltage drop signal of the shunt resistor to determine the DC component of the main circuit current, and collects the induced voltage signal of the coupled inductor secondary winding to determine the high-frequency ripple component of the main circuit current. The two are superimposed to obtain the actual current feedback value of the main circuit. The target reference value of the generated main circuit current is compared with the actual current feedback value to generate a current error signal. The current error signal is compensated by the second proportional-integral controller to generate a phase shift angle for adjusting power transmission; The control unit generates two sets of complementary pulse width modulation signals with a duty cycle of 50% based on the phase shift angle, which are used to drive the first half-bridge circuit and the second half-bridge circuit respectively. The phase difference between the two sets of signals is determined by the phase shift angle. The control unit generates a drive signal to drive the auxiliary switch, and its turn-on timing is ahead of the turn-on timing of the first gallium nitride switch in the first half-bridge circuit by a preset time, so as to realize soft switching operation.

2. The portable load adaptive bidirectional charging and discharging circuit module according to claim 1, characterized in that, The switching node of the first half-bridge circuit is the connection point between the source of the first gallium nitride switch and the drain of the second gallium nitride switch; the switching node of the second half-bridge circuit is the connection point between the source of the first silicon-based metal-oxide-semiconductor field-effect transistor and the drain of the second silicon-based metal-oxide-semiconductor field-effect transistor.

3. The portable load adaptive bidirectional charging and discharging circuit module according to claim 1, characterized in that, The induced voltage output from the secondary winding of the coupled inductor passes sequentially through a half-wave rectifier circuit composed of Schottky diodes, a π-type filter circuit composed of two capacitors and one inductor, and a low-dropout linear regulator to generate a stable DC voltage to power the control unit.

4. The portable load adaptive bidirectional charge and discharge circuit module according to claim 1, characterized in that, The turns ratio of the main winding to the auxiliary winding of the coupled inductor is set to 20:1 or 10:

1.

5. The portable load adaptive bidirectional charge and discharge circuit module according to claim 1, characterized in that, The auxiliary switch in the auxiliary resonant branch is an enhancement-mode p-channel metal-oxide-semiconductor field-effect transistor; a fast recovery diode is connected in reverse parallel across the resonant capacitor to provide a discharge path for the energy stored on the resonant capacitor after the auxiliary switch is turned off, and to clamp the reverse voltage on the resonant capacitor.

6. The portable load adaptive bidirectional charge and discharge circuit module according to claim 1, characterized in that, The control unit has a built-in digital signal processor. The voltage drop signal of the shunt resistor is sampled after passing through a second-order Butterworth low-pass filter to calculate the DC component of the main circuit current. The induced voltage of the secondary winding of the coupled inductor is sampled after passing through a capacitively coupled high-pass filter and combined with the mutual inductance coefficient of the coupled inductor to calculate the high-frequency ripple component of the main circuit current.

7. The portable load adaptive bidirectional charge and discharge circuit module according to claim 1, characterized in that, The control unit employs a phase-shift control strategy, which controls the direction and power of energy transmission by adjusting the phase difference of the pulse width modulation signal between the first half-bridge and the second half-bridge; the drive signal of the auxiliary switch maintains a fixed lead time with the drive signal of the first gallium nitride switch.

8. The portable load adaptive bidirectional charge and discharge circuit module according to claim 1, characterized in that, It also includes an over-temperature protection circuit, which includes a negative temperature coefficient thermistor. The negative temperature coefficient thermistor is in close contact with the surface of the main winding of the coupled inductor. The negative temperature coefficient thermistor is connected in series with a fixed resistor to divide the voltage and then connected to the analog comparator input terminal of the control unit. When the detected temperature exceeds a preset threshold, the control unit puts all switching devices in the off state.

9. The portable load adaptive bidirectional charge and discharge circuit module according to claim 1, characterized in that, A 5-ohm magnetic bead is connected in series in the gate drive circuit of the first gallium nitride switch; A 10-ohm magnetic bead is connected in series in the gate drive circuit of the second gallium nitride switch to suppress electromagnetic interference generated by different switches during the switching process.

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