Overmodulation method for isolating direct-current bus double-inverter power supply open-winding permanent magnet synchronous motor

Abstract

The invention discloses an overmodulation method for isolating a direct-current bus double-inverter power supply open-winding permanent magnet synchronous motor. Comprising the steps of calculating a modulation coefficient according to reference voltage output by a control loop; dividing an overmodulation area according to the modulation coefficients, and determining superposition components and weight coefficients corresponding to different modulation coefficients; deducing zero sequence voltage needing to be injected according to the amplitude and the phase of the superposed component; according to the superposition component and the weight coefficient after the zero-sequence voltage is injected, weighted calculation is carried out to obtain the corrected three-phase voltage of the double inverters; distributing the corrected three-phase voltage of the double inverters to the two inverters; and finally, the obtained three-phase reference voltages of the two inverters are subjected to carrier wave comparison to obtain a driving signal, so that the required output voltage is modulated. The method is realized based on a carrier wave comparison mode, switching loss and calculation burden can be reduced while the DC bus voltage utilization rate is improved, and the method is easy to deploy and apply.

Description

TECHNICAL FIELD

[0001] The present application belongs to the field of AC motor and drive control, more particularly, to a method for overmodulation of open-winding permanent magnet synchronous motor fed by dual-inverter with isolated DC bus. BACKGROUND

[0002] Dual-inverter with isolated DC bus fed open-winding permanent magnet synchronous motor system has higher voltage utilization, better harmonic characteristics and potential fault tolerance, and has been widely used in electric vehicles, aerospace, high-power transmission and other fields. Especially for electric vehicles, it is very suitable to adopt this topology because two isolated DC power supplies can be easily obtained by using battery power supply. In addition, in order to meet the wider speed range of electric vehicles under the capacity limitation of power battery, the voltage utilization of dual-inverter with isolated DC bus can be further enhanced by modulation technology.

[0003] Overmodulation technology makes the inverter operate in the nonlinear region and eventually reach the six-step region, at which time the output voltage of the inverter will be 10% higher than in the linear region, thereby increasing the speed and output torque of the motor and enhancing the motor operating performance. This is of great significance to expand the speed range of electric vehicles and reduce the voltage requirements of power batteries.

[0004] However, there is little research on overmodulation strategies for dual-inverter with isolated DC bus topology. The main reason is that the dual-inverter with isolated DC bus topology has similar voltage vector distribution characteristics as a single inverter and does not need to consider zero-sequence current suppression, so the overmodulation strategy of this topology is not much different from that of the traditional single inverter. The dual-inverter with isolated DC bus topology can be regarded as two independent two-level inverters, and the traditional space vector pulse width modulation is used to generate the required voltage. From the perspective of the two inverters, the traditional overmodulation strategy can be easily extended to the dual-inverter with isolated DC bus topology. However, directly copying the overmodulation strategy of the traditional single inverter to the dual-inverter with isolated DC bus topology will have two problems that cannot be solved.

[0005] The first problem is how to reduce switching loss in the overmodulation region. The dual-inverter with isolated DC bus is composed of six bridge arms, and the number of switching actions using the traditional modulation algorithm is twice that of a single inverter, which will increase the switching loss. The second problem is how to reduce the computational complexity of the overmodulation algorithm in the dual-inverter with isolated DC bus. Existing overmodulation algorithms are basically space vector-based modulation algorithms, which cannot avoid the cumbersome steps of segmenting the reference voltage vector, determining the sector and calculating the vector action time, and there are also a large number of unavoidable trigonometric function calculations. Especially in the dual-inverter with isolated DC bus topology, the computational complexity will further increase. SUMMARY

[0006] In view of the defects of the prior art, the purpose of the present application is to provide an overmodulation method for an open-winding permanent magnet synchronous motor powered by an isolated DC bus dual-inverter, which realizes the overmodulation strategy through carrier comparison, and can improve the utilization rate of DC bus voltage while reducing switching loss and computational burden.

[0007] To achieve the above-mentioned purpose, the present application realizes an overmodulation method for an open-winding permanent magnet synchronous motor powered by an isolated DC bus dual-inverter, comprising the following steps: (1) calculating a modulation coefficient according to the output of the current loop dq and the shaft reference voltage; (2) dividing the overmodulation region according to the modulation coefficient, and determining two groups of superimposed components and weight coefficients corresponding to different modulation coefficients; (3) deriving the corresponding zero sequence voltage according to the amplitude and phase of the superimposed components, and injecting the zero sequence voltage into the three-phase voltage of the dual-inverter corresponding to the superimposed components; (4) calculating the three-phase voltage of the dual-inverter after correction by weighting according to the three-phase voltage of the dual-inverter after injecting the zero sequence voltage and the weight coefficient; (5) redistributing the three-phase voltage of the dual-inverter after correction to obtain the three-phase reference voltage of the two inverters; (6) obtaining the drive signal through carrier comparison for the three-phase reference voltage allocated to the two inverters respectively, so as to modulate the required output voltage.

[0008] Further, the isolated DC bus dual-inverter topology is composed of two three-phase two-level inverters, each inverter is powered by an independent DC power supply, and the voltages of the two DC power supplies are the same. The AC connection of the isolated DC bus dual-inverter connects the stator winding, and supplies power to the three-phase open-winding permanent magnet synchronous motor. This configuration does not have a zero sequence loop, and does not need to consider the suppression of zero sequence current, so the voltage vector can be used more flexibly to better transition to the overmodulation region and the square wave region.

[0009] Further, the modulation coefficient M is calculated according to the formula . is the reference voltage vector amplitude of the dual-inverter, is the output phase voltage fundamental amplitude of the dual-inverter under six-beat working condition.

[0010] Further, the overmodulation region is divided into two kinds, the first kind is when the modulation coefficient M satisfies 0.9069 M ≤0.9514, which is called overmodulation region I; the second kind is when the modulation coefficient M satisfies M> 0.9514, which is called overmodulation region II.

[0011] Further, the overmodulation region II is divided into two cases, the first case is that when the dual-inverter runs in overmodulation region II and finally transits to six-step modulation, M satisfying 0.9514 M ≤ 1; the second case is that when the dual-inverter runs in overmodulation region II and finally transits to twelve-step modulation, M satisfying 0.9514 M ≤ 0.966. Six-step modulation is that the action time of 6 voltage vectors located at the vertices of the hexagon is 1 / 6 of the fundamental period, at this time the voltage utilization reaches the maximum, but the current distortion is also the largest; twelve-step modulation is that the action time of 12 voltage vectors located on the boundary of the hexagon is 1 / 12 of the fundamental period, at this time the voltage utilization is reduced by 3.4% compared with six-step modulation, but the current distortion is smaller.

[0012] Further, the superimposed component is determined according to the modulation coefficient. The superimposed component corresponding to the dual-inverter running in overmodulation region I is V sin and V hex ; the superimposed component corresponding to the dual-inverter running in overmodulation region II and finally transiting to six-step modulation is V hex and V six ; the superimposed component corresponding to the dual-inverter running in overmodulation region II and finally transiting to twelve-step modulation is V hex and V twe . V sin The corresponding voltage vector locus is a circle with a radius of , which is calculated according to the formula . V hex The corresponding voltage vector locus is the voltage vector on the boundary of the hexagon, which is calculated according to the formula . V six The corresponding voltage vector locus is that six vectors with an amplitude of 4 V dc / 3 are located at the vertices of the hexagon, which is calculated according to the formula , where θ is the phase angle of the reference voltage vector, and m is the sector in which the reference voltage vector is located. V twe The corresponding voltage vector locus is twelve voltage vectors on the boundary of the hexagon, which contains six vectors with an amplitude of 4 Vdc The vector of / 3 lies at the vertices of the hexagon and at the six points with a magnitude of 2. V dc / The vector lies between the vertices of the hexagon.

[0013] Furthermore, the superimposed components V sin Corresponding zero-sequence voltage According to the formula Calculated; V hex Corresponding zero-sequence voltage According to the formula Calculated; V six and V twe The corresponding zero-sequence voltages and According to the formula Calculated. and These are the three-phase reference voltages of the dual inverters. ( x = a , b , c The maximum and minimum values ​​of ).

[0014] Furthermore, the corrected three-phase voltage of the dual inverter is calculated by weighting the three-phase voltage of the dual inverter after injecting zero-sequence voltage and weighting coefficients. Three corresponding expressions for the three-phase voltage of the corrected dual inverter can be derived based on the different modulation regions. The three-phase voltage of the corrected dual inverter corresponding to the dual inverter operating in over-modulation zone one... ( x = a , b , c According to the formula Calculations show that and They are dq Shaft reference voltage, k 1 represents the weighting coefficient corresponding to the dual inverter operating in overmodulation zone 1; the three-phase voltage of the dual inverter after correction, when operating in overmodulation zone 2 and finally transitioning to six-step modulation, is calculated according to the formula. Calculations show that k 2 represents the weighting coefficient corresponding to the dual inverters operating in over-modulation zone 2 and eventually transitioning to six-step modulation. It is related to The sign function; the corrected three-phase voltage of the dual inverter operating in over-modulation zone two and eventually transitioning to twelve-step modulation is based on the formula. The calculation is as follows, k 3 is the weight coefficient corresponding to the operation of the double-inverter in the over-modulation two-zone and the final transition to twelve-step modulation, x a b c The symbol function is related to the sign function. θ

[0015] Further, the three-phase voltage of the modified double-inverter is redistributed to obtain the three-phase reference voltages of the two inverters by judging the positive and negative of the three-phase voltage of the modified double-inverter. The obtained three-phase reference voltages of the two inverters are obtained by the carrier comparison method to obtain the driving signal, so as to modulate the required output voltage to complete the over-modulation.

[0016] The application further provides an electronic device, comprising a computer readable storage medium and a processor. The computer readable storage medium is used for storing executable instructions. The processor is used for reading the executable instructions stored in the computer readable storage medium and executing the above method.

[0017] The application further provides a computer readable storage medium, which stores computer instructions, and the computer instructions are used for making the processor execute the above method.

[0018] The application further provides a computer program product, which comprises computer programs or instructions, and the computer programs or instructions are executed by the processor to realize the above method.

[0019] The application provides an over-modulation method for an open-winding permanent magnet synchronous motor powered by a double-inverter isolated DC bus, which has the following beneficial effects: 1. The over-modulation method is based on the carrier comparison method, and the three-phase reference voltage is directly modified by using the strictly derived three-phase voltage expression of the double-inverter, which avoids the sector judgment, vector action time calculation and a large number of trigonometric function calculations, unlike the traditional over-modulation strategy that needs to be segmented and modified according to the position of the reference voltage in the space vector.

[0020] 2. The over-modulation method extends the idea of clamping bridge arms to the over-modulation region. The zero-sequence voltage needed to be injected in the over-modulation region is derived, and the three-phase reference voltages of the two inverters are obtained by the voltage distribution method to reduce the switching loss and improve the efficiency of the system.

[0021] ​​​​​3、The overmodulation method proposed in the application can finally transit to six-step modulation with higher voltage utilization or twelve-step modulation with smaller output voltage harmonics, both of which improve the output capacity of the isolated DC bus dual-inverter and provide more choices for practical applications. BRIEF DESCRIPTION OF DRAWINGS

[0022] Figure 1 is a schematic block diagram of the isolated DC bus dual-inverter power supply open-winding permanent magnet synchronous motor system and overmodulation method of the application. Figure 2 is a voltage vector trajectory diagram corresponding to different superimposed components of the application.

[0023] Figure 3 is the A-phase voltage waveform of the modified dual-inverter, the A-phase reference voltage waveform of the two inverters and the voltage vector trajectory diagram when the modulation coefficient is 0.93 and 0.95.

[0024] Figure 4 is the A-phase voltage waveform of the modified dual-inverter, the A-phase reference voltage waveform of the two inverters and the voltage vector trajectory diagram when the modulation coefficient is 0.98 and 1.

[0025] Figure 5 is the A-phase voltage waveform of the modified dual-inverter, the A-phase reference voltage waveform of the two inverters and the voltage vector trajectory diagram when the modulation coefficient is 0.96 and 0.966.

[0026] Figure 6 is a three-phase pole voltage waveform diagram of the two inverters when the modulation coefficient is 0.93 and 0.98.

[0027] Figure 7 is the dynamic experimental process of the isolated DC bus dual-inverter transiting from the linear region to the six-step modulation or the twelve-step modulation. DETAILED DESCRIPTION

[0028] In order to make the objectives, technical solutions and advantages of the application clearer, the application will be further described in detail below with reference to the drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the application and do not limit the application. In addition, the technical features involved in each embodiment of the application described below can be combined with each other as long as they do not conflict with each other.

[0029] Figure 1 is a schematic block diagram of the isolated DC bus dual-inverter power supply open-winding permanent magnet synchronous motor system and overmodulation method of the application. The isolated DC bus dual-inverter topology consists of two three-phase two-level inverters, each of which is powered by an independent DC power supply, and the voltages of the two DC power supplies are the same, both of which are The AC side of the isolated DC bus dual inverter is connected to the stator winding to power a three-phase open-winding permanent magnet synchronous motor. This configuration eliminates the zero-sequence loop and eliminates the need to consider zero-sequence current suppression, thus allowing for more flexible use of voltage vectors to better transition to the overmodulation and square wave regions. The overmodulation method includes the following steps: (1) Based on the output of the current loop dq Calculate the modulation coefficient using the shaft reference voltage; (2) Divide the overmodulation region according to the modulation coefficient, and determine the two sets of superimposed components and weighting coefficients corresponding to different modulation coefficients; (3) Derive the corresponding zero-sequence voltage based on the amplitude and phase of the superimposed component, and inject the zero-sequence voltage into the three-phase voltage of the dual inverter corresponding to the superimposed component; (4) The corrected three-phase voltage of the dual inverter is calculated by weighting the three-phase voltage of the dual inverter after the zero-sequence voltage is injected and the weighting coefficient. (5) The three-phase voltages of the corrected dual inverters are redistributed to obtain the three-phase reference voltages of the two inverters; (6) The three-phase reference voltages allocated to the two inverters are used to obtain the drive signal by carrier comparison, thereby modulating the required output voltage.

[0030] Specifically, the modulation coefficient M for:

[0031] in, It is the reference voltage vector magnitude of the dual inverter. It is the fundamental amplitude of the output phase voltage of the dual inverter under six-step operation.

[0032] Specifically, the overmodulation region is divided into two types. The first type is when the modulation coefficient... M Satisfies 0.9069 < M ≤0.9514, this is called overmodulation zone one; the second is when the modulation coefficient M satisfy M >0.9514, this is called the overmodulation zone two.

[0033] Specifically, the overmodulation two-zone configuration is divided into two cases. The first case is when the dual inverters operate in the overmodulation two-zone and eventually transition to six-step modulation. M Satisfies 0.9514 < M ≤1; The second case is when the dual inverters are operating in over-modulation zone two and eventually transition to twelve-step modulation. M Satisfies 0.9514 < M≤0.966. Six-step modulation involves the six voltage vectors located at the vertices of the hexagon acting for 1 / 6 of the fundamental period, at which point the voltage utilization rate reaches its maximum, but the current distortion is also the maximum. Twelve-step modulation involves the twelve voltage vectors located on the boundaries of the hexagon acting for 1 / 12 of the fundamental period, at which point the voltage utilization rate is reduced by 3.4% compared to six-step modulation, but the current distortion is smaller.

[0034] Specifically, the superimposed components are determined based on the modulation coefficients, and the voltage vector trajectories corresponding to different superimposed components are as follows: Figure 2 As shown. When M When the linear modulation ratio is 0.9069, the reference voltage vector trajectory is the inscribed circle of a hexagon, which corresponds to the voltage vector at the maximum linear modulation ratio. V sin :

[0035] in, θ It is the phase angle of the reference voltage vector.

[0036] when M When = 0.9514, the reference voltage vector trajectory is the voltage vector on the hexagon. V hex :

[0037] in, m It is the sector where the reference voltage vector is located.

[0038] When the dual inverters operate in over-modulation zone 1, a portion of the reference voltage vector will exceed the modulation range of the hexagon. Therefore, according to the superposition principle, a phase angle with the same as the reference voltage vector is selected. V sin and V hex The reference voltage vector is modified as a superposition component. Weighting coefficients. k 1 and the modified reference voltage vector V mod They are respectively:

[0039]

[0040] when M When the value is 1, the reference voltage vector trajectory is the six vertices of a hexagon, which are the six voltage vectors with the largest amplitudes. V six :

[0041] When the dual inverters operate in overmodulation zone 2 and eventually transition to six-step modulation, the superimposed component becomes one with the same phase angle as the reference voltage vector. V hex and closest to the reference voltage vector V six Weighting coefficients k 2 and the modified reference voltage vector V mod They are respectively:

[0042]

[0043] when M When = 0.966, the reference voltage vector trajectory is twelve voltage vectors on the hexagonal boundary. V twe It contains six amplitudes of 4 V dc The vector of / 3 lies at the vertices of the hexagon and at the six points with a magnitude of 2. V dc / The vector lies between the vertices of the hexagon. When the dual inverter operates in over-modulation zone 2 and eventually transitions to 12-step modulation, the superimposed components become... V hex and closest to the reference voltage vector V twe Weighting coefficients k 3 and the modified reference voltage vector V mod They are respectively:

[0044]

[0045] Therefore, by simply replacing the superposition component in the above formula with a relationship expressed using the three-phase reference voltage of the dual inverter, the modified relationship between the three-phase voltage and the three-phase reference voltage output by the control loop can be obtained.

[0046] Specifically, to replace the superimposed component, it is necessary to calculate the zero-sequence voltage corresponding to the superimposed component and the three-phase voltage of the dual inverter after injecting the zero-sequence voltage corresponding to the superimposed component. The superimposed component... V sin , V hex , V six and V twe Corresponding zero-sequence voltage , , , They are respectively:

[0047]

[0048]

[0049] in, and These are the three-phase reference voltages of the dual inverters. ( x = a , b , c The maximum and minimum values ​​of ).

[0050] Specifically, the corrected three-phase voltage of the dual inverter is calculated by weighting the three-phase voltage of the dual inverter after injecting zero-sequence voltage and weighting coefficients. Three corresponding expressions for the corrected three-phase voltage of the dual inverter can be derived based on the different modulation regions. The corrected three-phase voltage of the dual inverter operating in over-modulation zone one corresponds to... ( x = a , b , c )for:

[0051]

[0052] in, and They are dq Shaft reference voltage.

[0053] The three-phase voltage of the dual inverter after the over-modulation zone 2 and the final transition to the six-step modulation zone is corrected. ( x = a , b , c )for:

[0054] in, It is related to The sign function satisfies the following relationship:

[0055] The three-phase voltage of the dual inverter after the over-modulation zone 2 and the final transition to the 12-step modulation zone is corrected. ( x = a , b ,c )for:

[0056] in, It is related to θ The sign function satisfies the following relationship:

[0057] The above formula can be used to simply express the three-phase voltage of the dual inverter in the overmodulation region, avoiding the tedious steps of calculating the modulation angle and holding angle, segmenting the reference voltage vector, determining the sector, and calculating the vector action time.

[0058] Specifically, in order to reduce the number of switching actions, by allocating The method used to obtain the three-phase reference voltages of the two inverters is as follows:

[0059] in, V ref1_x ( x = a , b , c () is the three-phase reference voltage of inverter 1. V ref2_x ( x = a , b , c ) is the three-phase reference voltage of inverter 2.

[0060] Specifically, the obtained V ref1_x and V ref2_x The driving signal is obtained by carrier comparison, thereby modulating the required output voltage to complete the overmodulation strategy and reduce switching losses and computation.

[0061] The overmodulation method for an open-winding permanent magnet synchronous motor powered by a dual inverter on an isolated DC bus, provided by this invention, was verified through simulation and experiments. The simulation and experimental operating conditions were consistent: the two DC-side power supply voltages were set to 75V, the switching frequency was 10kHz, and the load torque of the open-winding permanent magnet synchronous motor was 3N·m. By giving different reference speeds, the operating range of the dual inverter was adjusted from the linear region to the overmodulation region.

[0062] Modulation coefficient M The corrected A-phase voltage of the dual inverter at values ​​of 0.93 and 0.95. Simulated waveforms, A-phase reference voltages of the two inverters V ref1_a and Vref2_a The simulated waveforms and voltage vector trajectories under the two modulation coefficients are as follows: Figure 3 As shown, the dual inverters are currently operating in overmodulation zone one. As the modulation coefficient increases, It is closer to a trapezoidal wave, and the output voltage vector trajectory is more approximately hexagonal. V ref1_a and V ref2_a The value will be clamped to zero for half a fundamental cycle, thereby reducing the number of switching operations of the dual inverter.

[0063] Modulation coefficient M When it is 0.98 and 1 , V ref1_a and V ref2_a The simulated waveforms and voltage vector trajectories under the two modulation coefficients are as follows: Figure 4 As shown, the dual inverters are currently operating in over-modulation zone two and eventually transition to six-step modulation. As the modulation coefficient increases, It is closer to a square wave, and the output voltage vector trajectory is closer to the six vertices of a hexagon.

[0064] Modulation coefficient M When it is 0.96 and 0.966 , V ref1_a and V ref2_a The simulated waveforms and voltage vector trajectories under the two modulation coefficients are as follows: Figure 5 As shown, the dual inverters are currently operating in over-modulation zone two and eventually transition to twelve-step modulation. As the modulation coefficient increases, It is closer to a three-level square wave, and the output voltage vector trajectory is closer to the 12 points on the hexagonal boundary. V ref1_a and V ref2_a Both need to be zero to generate a phase A voltage with a voltage magnitude of zero, thereby generating a multi-level voltage waveform.

[0065] Modulation coefficient M The three-phase voltages of inverter 1 at values ​​of 0.93 and 0.98 ( V XO , X = A , B , C ) and the three-phase voltage of inverter 2 ( , = , , The experimental waveform is as follows:Figure 6 As shown, every 60 electrical degrees, only half of the switching transistors are switched, while the others are clamped. Furthermore, the overmodulation method corrects the voltage vector to the hexagonal boundary, at which point five of the six bridge arms are clamped, and only one will switch. Therefore, switching losses are significantly reduced.

[0066] The dynamic experimental process of transitioning the dual inverter from the linear region to six-step or twelve-step modulation by increasing the motor speed is as follows: Figure 7 As shown, the rotational speeds are arranged from top to bottom. n Phase A current i a and line voltage V AB Initially, the motor speed is 1140 rpm, and the dual inverters operate in the linear region. The overmodulation method of this invention allows the dual inverters to smoothly transition to six-step or twelve-step modulation without current surges and with stable speed. Furthermore, the maximum speed under six-step modulation is 1378 rpm, and the maximum speed under twelve-step modulation is 1318 rpm, but the phase current waveform quality is better under twelve-step modulation. Therefore, the overmodulation method of this invention provides more options for practical applications, allowing selection of either higher-speed six-step modulation or better-current-quality twelve-step modulation, improving the output capability of the isolated DC bus dual inverters compared to the linear region.

[0067] The overmodulation method provided by this invention for an open-winding permanent magnet synchronous motor powered by a dual inverter on an isolated DC bus is based on carrier comparison, which improves the utilization rate of DC bus voltage while reducing switching losses and computational burden.

[0068] The present invention also relates to an electronic device, including a memory and a processor, wherein the memory stores a computer program, and the processor executes the computer program to implement the steps of the method described above.

[0069] The electronic device can be a desktop computer, laptop, handheld computer, or cloud server, etc. The processor can be a Central Processing Unit (CPU), or other general-purpose processors, digital signal processors (DSPs), application-specific integrated circuits (ASICs), field-programmable gate arrays (FPGAs), or other programmable logic devices, discrete gate or transistor logic devices, discrete hardware components, etc. The memory can be used to store computer programs and / or modules. The processor performs various functions of the electronic device by running or executing the computer programs and / or modules stored in the memory, and by accessing data stored in the memory.

[0070] The present invention also relates to a computer-readable storage medium having a computer program stored thereon, which, when executed by a processor, implements the steps of the method described above.

[0071] Specifically, the memory may include high-speed random access memory, as well as non-volatile memory, such as hard disks, RAM, plug-in hard disks, smart media cards (SMC), secure digital (SD) cards, flash cards, at least one disk storage device, flash memory device, or other volatile solid-state storage devices.

[0072] This invention provides a computer program product or computer program that includes computer instructions stored in a computer-readable storage medium. A processor of a computer device reads the computer instructions from the computer-readable storage medium and executes the computer instructions, causing the computer device to perform the steps of the method described in the above embodiments of this invention.

[0073] The technical features of the embodiments described above can be combined arbitrarily. For the sake of brevity, not all possible combinations of the technical features in the above embodiments are described. However, as long as the combination of these technical features does not contradict each other, it should be considered within the scope of this specification. It should be noted that the terms "in one embodiment," "for example," and "again" in this invention are intended to illustrate the invention and are not intended to limit the invention.

[0074] Those skilled in the art will readily understand that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A method for overmodulation of a doubly-fed open-winding permanent magnet synchronous machine powered by an isolated DC bus dual-inverter topology, the isolated DC bus dual-inverter topology comprising two three-phase two-level inverters, each inverter being powered by an independent DC power source, the two DC power sources having the same voltage, the AC output of the isolated DC bus dual-inverter being connected to the stator windings of a three-phase open-winding permanent magnet synchronous machine, characterized in that, The method comprises the following steps: (1) According to the output of the current loop dq The modulation coefficient is calculated based on the shaft reference voltage. (2) dividing the overmodulation region according to the modulation coefficient, and determining two groups of superimposed components and weight coefficients corresponding to different modulation coefficients; (3) deriving the corresponding zero sequence voltage according to the amplitude and phase of the superimposed components, and injecting the zero sequence voltage into the three-phase voltage of the double inverter corresponding to the superimposed components; (4) calculating the three-phase voltage of the corrected double inverter by weighting according to the three-phase voltage of the double inverter after injecting the zero sequence voltage and the weight coefficient; (5) redistributing the three-phase voltage of the corrected double inverter to obtain the three-phase reference voltage of the two inverters; (6) obtaining the driving signal through carrier comparison of the three-phase reference voltage allocated to the two inverters respectively, so as to modulate the required output voltage.

2. The overmodulation method of claim 1, wherein, The modulation coefficient M According to the formula The calculation is, The reference voltage vector amplitude of the double inverter, The output phase voltage fundamental amplitude of the double inverter in six-pulse working condition.

3. The overmodulation method of claim 2, wherein, The overmodulation region is divided into two, the first is when the modulation coefficient M satisfies 0.9069 M ≤0.9514, which is called overmodulation region one; the second is when the modulation coefficient M satisfies M >0.9514, which is called overmodulation region two; the overmodulation region two is divided into two cases, the first case is when the dual-inverter runs in overmodulation region two and finally transits to six-step modulation, M satisfies 0.9514 M ≤1; the second case is when the dual-inverter runs in overmodulation region two and finally transits to twelve-step modulation, M satisfies 0.9514 M ≤0.

966.

4. The overmodulation method of claim 3, wherein, The superimposed component is determined according to the modulation coefficient: the superimposed component corresponding to the double-inverter operating in the over-modulation region I is V sin and V hex ; the superimposed component corresponding to the double-inverter operating in the over-modulation region II and finally transitioning to six-step modulation is V hex and V six ; the superimposed component corresponding to the double-inverter operating in the over-modulation region II and finally transitioning to twelve-step modulation is V hex and V twe ; V sin The corresponding voltage vector locus is a circle with a radius of , and the phase angle of the voltage vector is the same as that of the reference voltage vector; V hex The corresponding voltage vector locus is a voltage vector on the boundary of a hexagon, and the phase angle of the voltage vector is the same as that of the reference voltage vector; V six The corresponding voltage vector locus is six vectors with a magnitude of 4 V dc / 3 located at the vertices of the hexagon; V twe The corresponding voltage vector locus is twelve voltage vectors on the boundary of a hexagon, including six vectors with a magnitude of 4 V dc / 3 located at the vertices of the hexagon and six vectors with a magnitude of 2 V dc / located between the vertices of the hexagon.

5. The overmodulation method of claim 4, wherein, The superimposed component V sin The corresponding zero sequence voltage According to the formula Calculated; V hex The corresponding zero sequence voltage According to the formula Calculated; V six And V twe The corresponding zero sequence voltage And According to the formula Calculated, wherein And The maximum and minimum values of the three-phase reference voltage of the double inverter, respectively.

6. The overmodulation method of claim 1, wherein, The three-phase voltage of the corrected double inverter is calculated by weighting according to the three-phase voltage of the double inverter after injecting the zero sequence voltage and the weight coefficient, and three corresponding three-phase voltage expressions of the corrected double inverter can be derived according to the different modulation regions.

7. The overmodulation method of claim 6, wherein, The three-phase reference voltage of the two inverters is obtained by redistributing the three-phase voltage of the corrected double inverter, which comprises judging the positive and negative of the three-phase voltage of the corrected double inverter to determine the three-phase reference voltage of the two inverters. 8.An electronic device comprising a memory and a processor, the memory storing a computer program, wherein, The processor executes the computer program to realize the steps of the method of any one of claims 1 to 7.

9. A computer readable storage medium having stored thereon a computer program, characterized in that, The computer program is executed by the processor to realize the steps of the method of any one of claims 1 to 7.

10. A computer program product comprising computer programs or instructions, characterized in that, The computer program or instructions are executed by the processor to realize the steps of the method of any one of claims 1 to 7.

Images