A bandpass filter of an asymmetrically tapered via array

Abstract

The application relates to a band-pass filter of an asymmetrically graded via array, belonging to the technical field of microwave radio frequency passive devices, comprising a SIW rectangular resonant cavity which comprises four resonant cavities, has coupling windows between adjacent resonant cavities, and has different arrangement modes of metalized vias in longitudinal metalized via arrays in different resonant cavities; different arrangement modes of metalized vias in longitudinal metalized via arrays in different coupling windows, the spacing between metalized vias in longitudinal metalized via arrays in the coupling windows is smaller than the minimum spacing between metalized vias in the resonant cavities, the longitudinal metalized via array of at least one coupling window is arranged in a trapezoidal gradient, and the setting positions or arrangement modes of the transverse metalized via arrays in different coupling windows are different; the band-pass filter can more simply and effectively solve the problems of strong stray mode, large insertion loss and narrow adjustable range of passband bandwidth of a traditional SIW band-pass filter.

Description

Technical Field

[0001] This invention relates to the field of microwave radio frequency passive device technology, and in particular to a bandpass filter with an asymmetric gradient via array. Background Technology

[0002] As 5G communication, X-band radar, satellite communication, and other systems continue to evolve towards higher frequencies, smaller sizes, and greater integration, RF front-ends are placing increasingly stringent demands on filters in terms of insertion loss, frequency selectivity, out-of-band rejection, and impedance matching. Substrate integrated waveguides (SIWs) combine the high quality factor (Q value) of traditional metallic waveguides with the ease of integration of planar circuits, making them one of the mainstream implementation solutions for microwave and millimeter-wave devices.

[0003] Traditional SIW bandpass filters typically include a dielectric substrate, a top metal layer disposed on the upper surface of the dielectric substrate, and a bottom metal layer disposed on the lower surface of the dielectric substrate. The dielectric substrate is usually constructed with a metallized via array. Each metallized via in the metallized via array is connected to the top metal layer and the bottom metal layer, and the metallized via array adopts a periodic metallized via array. The metallized vias are arranged in a continuous, equally spaced (uniform) and symmetrical manner along the length direction of the dielectric substrate. The metal via array, the top metal layer, and the bottom metal layer together constitute the SIW rectangular resonant cavity. However, in traditional SIW bandpass filters, the waveguide sidewalls, which are equivalent to two rows of metallized vias, have discrete periodic characteristics, making them highly susceptible to exciting high-order spurious modes. Spurious resonances couple with the dominant mode, leading to a significant increase in passband ripple, insufficient out-of-band suppression depth, and severe degradation of far-end stopband performance. Furthermore, traditional SIW bandpass filters have relatively high overall insertion loss. On one hand, the discrete periodic characteristics of the waveguide sidewalls, equivalent to metallized vias, make electromagnetic energy leakage through gaps between vias easy, introducing additional radiation loss. On the other hand, the inter-cavity coupling window has limited ability to control the inter-cavity coupling strength, making it difficult to achieve a good balance between narrowband filtering characteristics and insertion loss, and hindering the flexible realization of the target passband bandwidth. Moreover, under narrowband filtering design conditions, the inter-cavity energy transmission efficiency decreases, and transmission loss increases significantly. Simultaneously, conventional dielectric substrates have high dielectric losses at high frequencies. The superposition of multiple losses makes it difficult to reduce the overall insertion loss of traditional SIW bandpass filters, limiting their application in high-performance RF systems.

[0004] To overcome the technical shortcomings of traditional SIW bandpass filters, such as strong spurious modes, high insertion loss, and narrow adjustable passband bandwidth, existing technologies typically employ improved approaches such as introducing defective ground structures or using half-mode waveguide structures. For example, Chinese patent CN118841727A discloses a millimeter-wave SIW broadband bandpass filter with a defective ground structure, CN103682534B discloses a dielectric waveguide filter with a defective ground-loaded magnetic coupling structure, CN114156617B discloses a dual-mode SIW rectangular cavity dual-band filter with a complementary split-ring resonator, and CN104466316A discloses a 2X-band defective junction structure-half-mode substrate integrated waveguide filter. All of these technologies improve frequency response characteristics by constructing defective ground structures or half-mode waveguide structures, aiming to achieve the technical effects of spurious suppression, low loss, and widened passband bandwidth. However, in practice, both defective ground structures and half-mode waveguide structures inherently suffer from high structural complexity. Introducing these structures into traditional SIW bandpass filters not only requires complex structural and parameter design of the entire filter but also increases the complexity of subsequent manufacturing processes, thereby driving up manufacturing costs. Therefore, how to more simply and effectively solve the problems of strong spurious modes, high insertion loss, and narrow adjustable passband bandwidth in traditional SIW bandpass filters remains a pressing technical challenge in this field. Summary of the Invention

[0005] To address the technical problems of strong spurious modes and high insertion loss in traditional SIW bandpass filters, this invention provides a bandpass filter that employs an asymmetric gradient composite metallized via array. Without increasing the number of cavity stages, expanding the volume, or introducing any additional complex structures, it achieves improved field constraint. This not only effectively suppresses spurious modes and enhances out-of-band suppression, but also effectively reduces insertion loss, improves passband flatness, and expands the passband bandwidth.

[0006] A bandpass filter with an asymmetric gradient via array includes a dielectric substrate, a top metal layer disposed on the upper surface of the dielectric substrate, and a bottom metal layer disposed on the lower surface of the dielectric substrate. The dielectric substrate is constructed with a metallized via array, each metallized via in the array being connected to the top metal layer and the bottom metal layer, respectively. The metal via array, the top metal layer, and the bottom metal layer together constitute a SIW rectangular resonant cavity. The SIW rectangular resonant cavity includes four resonant cavities, with adjacent resonant cavities coupled through coupling windows. Each resonant cavity includes longitudinal metallized via arrays constructed on the upper and lower sides of the dielectric substrate and linearly arranged along the length of the dielectric substrate. The longitudinal metallized via arrays within different resonant cavities... The arrangement of the metallized vias in the arrays is different; the coupling windows include vertical arrays of metallized vias constructed on the upper and lower sides of the dielectric substrate and arranged linearly along the length of the dielectric substrate, and horizontal arrays of metallized vias constructed between the vertical arrays of metallized vias and arranged linearly along the width of the dielectric substrate; the arrangement of the metallized vias in the vertical arrays of metallized vias in different coupling windows is different, the spacing between the metallized vias in the vertical arrays of metallized vias in the coupling windows is smaller than the minimum spacing between the metallized vias in the resonant cavity, and the vertical arrays of metallized vias in at least one coupling window are arranged in a trapezoidal gradient; the setting position or arrangement of the horizontal arrays of metallized vias in different coupling windows is different. In this scheme, by employing a dense arrangement of metallized vias with a spacing smaller than the minimum spacing of the resonant cavities in the coupling window region, the field confinement effect in the window region is enhanced, the interstage coupling strength is improved, and the insertion loss is effectively reduced. A trapezoidal gradient arrangement of longitudinal metallized vias within at least one coupling window achieves a smooth transition of the equivalent waveguide wall characteristic impedance along the propagation direction, eliminating abrupt reflection interfaces and improving the voltage standing wave ratio while expanding the passband bandwidth. By varying the arrangement of longitudinal and transverse metallized via arrays within different coupling windows, the periodic boundary conditions of the structure are actively disrupted, making it impossible to satisfy the phase matching conditions of higher-order spurious modes, thereby significantly improving the out-of-band suppression level. Furthermore, by varying the arrangement of longitudinal metallized via arrays within different resonant cavities, the overall geometric symmetry of the filter is broken, eliminating the symmetrical boundary conditions upon which even-order higher-order modes rely for excitation, further purifying the passband spectrum and improving passband flatness. This solution utilizes four aperture placement mechanisms—encryption, gradient, non-uniformity, and asymmetry—to work together synergistically. Without increasing the number of cavity stages, expanding the filter volume, or introducing any additional complex structures, it achieves multiple technical effects simultaneously and more simply and effectively, including reduced insertion loss, extended passband bandwidth, enhanced out-of-band suppression, and improved passband ripple. This breakthrough overcomes the technical bottleneck of traditional SIW filters where various performance indicators are mutually restrictive and mutually exclusive, resulting in a comprehensive performance improvement that is unpredictable by those skilled in the art based on existing technologies.

[0007] Furthermore, the SIW rectangular resonant cavity includes a first coupling window, a second coupling window, and a third coupling window arranged sequentially at intervals along the transmission direction. In the first coupling window, the spacing between the metallized vias in the longitudinal metallized via array is less than the minimum spacing between metallized vias in the resonant cavity. The second coupling window contains k metallized vias, where k is an odd number and k≠1. Centered on the middle metallized via, the metallized vias on its left and right sides are arranged in a trapezoidal gradient. The third coupling window contains at least three metallized vias, centered on a metallized via that offset from the center. The metallized vias on its left and right sides are arranged asymmetrically. In this scheme, by controlling the spacing between the metallized vias in the longitudinal metallized via array within the first coupling window to be smaller than the minimum spacing (SVP) between metallized vias in the resonant cavity, this spacing is made smaller than the spacing between the metallized vias in the longitudinal metallized via array within the upstream resonant cavity, thus achieving the purpose of denser arrangement. By configuring the longitudinal metallized via array within the second coupling window in a trapezoidal gradient arrangement, the characteristic impedance of the SIW equivalent waveguide wall changes smoothly along the propagation direction, which can eliminate obvious reflection interfaces. At the same time, the gradient arrangement can also broaden the equivalent waveguide wall's support for transmission modes (dominant mode TE). 10 The wide frequency response range allows for better impedance matching across a wider frequency band, which helps reduce the voltage standing wave ratio and increase the passband bandwidth. By configuring the longitudinal metallized via array within the third coupling window in an asymmetrical arrangement, the geometric symmetry of the entire filter can be completely broken, causing even-order modes to lose the "symmetric boundary conditions" for excitation, thus making it impossible to establish stable resonance. This not only further purifies the spectral environment and eliminates obvious spurious resonance peaks, but also effectively improves the out-of-band suppression effect at the far end.

[0008] Preferably, within the third coupling window, the spacing between adjacent metallized vias in the longitudinal metallized via array gradually increases along the direction away from the center. This allows the metallized via spacing within the third coupling window to gradually transition to the standard metallized via spacing, resulting in smoother and less abrupt wave transmission.

[0009] Furthermore, within the coupling window, the transverse metallized via array is aligned with one of the metallized vias in the longitudinal metallized via array.

[0010] Preferably, the longitudinal metallized via arrays on both sides of the same resonant cavity are arranged symmetrically; the longitudinal metallized via arrays on both sides of the same coupling window are also arranged symmetrically. This simplifies the structure.

[0011] Furthermore, the metallized vias in the transverse metallized via array within the coupling window are non-uniformly arranged along the width direction of the dielectric substrate, and the spacing between two adjacent metallized vias is less than or equal to the minimum spacing between metallized vias within the resonant cavity. In this scheme, by non-uniformly arranging the metallized vias in the transverse metallized via array, the periodic boundary conditions in the width direction can be disrupted. When the periodicity is disrupted, the phase matching conditions of the higher-order modes that originally required strict phase matching to establish a standing wave are no longer satisfied. The energy cannot form a stable resonance within the cavity, but is scattered to other directions or converted into radiation loss. The resonant frequencies of these higher-order modes are effectively pushed away from the working passband, causing them to disappear or significantly attenuate on the S-parameter curve, exhibiting a significantly enhanced effect of far-end out-of-band suppression. Furthermore, by adjusting the spacing between two adjacent metallized vias... The spacing is configured to be less than or equal to the minimum spacing (SVP) between metallized vias in the resonant cavity, which enables the denser arrangement of metallized vias in the transverse metallized via array. The denser arrangement can increase the equivalent transverse electric wall density in the coupling window region, making the metal boundaries on both sides of the window denser. The electric field energy can be more effectively confined in the window region, thereby enhancing the electromagnetic coupling coefficient between adjacent resonant cavities. A stronger coupling coefficient means that, under the same transmission distance, the field transition distance required for energy to be transferred from the previous cavity to the next cavity is shorter, and radiation loss and dielectric loss are significantly reduced.

[0012] Preferably, the spacing between adjacent metallized vias in the lateral metallized via array gradually decreases from the lower or upper side of the dielectric substrate to the middle of the dielectric substrate. This results in a larger coupling window width, which not only enhances interstage coupling strength and extends passband bandwidth, but also reduces radiation loss and dielectric loss during energy transfer between stages, thus further reducing insertion loss.

[0013] Further, the SIW rectangular resonant cavity includes a first resonant cavity, a second resonant cavity, a third resonant cavity, and a fourth resonant cavity. There is a first coupling window between the first resonant cavity and the second resonant cavity, a second coupling window between the second resonant cavity and the third resonant cavity, and a third coupling window between the third resonant cavity and the fourth resonant cavity. Among them, the longitudinal metallized via array in the first resonant cavity includes three metallized vias, and the distance between adjacent two metallized vias is svp, where svp = 2.384 mm; the longitudinal metallized via array in the first coupling window includes three metallized vias, and the distance between adjacent two metallized vias in the longitudinal metallized via array is a, where a < svp; the longitudinal metallized via array in the second resonant cavity includes three metallized vias, and along the transmission direction, the distances between adjacent two metallized vias are svp and f in sequence, where f > svp; the longitudinal metallized via array in the second coupling window includes five metallized vias. Taking the metallized via in the middle as the center, along the direction away from the center, the distances between adjacent two metallized vias are a and b in sequence, where b > a; the longitudinal metallized via array in the third resonant cavity includes three metallized vias, and along the transmission direction, the distances between adjacent two metallized vias are f and svp in sequence; the longitudinal metallized via array in the third coupling window includes five metallized vias. Taking the second metallized via along the transmission direction as the center, the distance between the metallized vias on the left side of the center is a, and the distances between the metallized vias on the right side of the center are c, d, and e in sequence, where c > b, d > c, and e > d; the longitudinal metallized via array in the fourth resonant cavity includes one metallized via. In this solution, by controlling the svp size to 2.384 mm, it is possible to break through the technical bottleneck that it is difficult to balance the passband bandwidth, insertion loss, and standing wave ratio in the prior art. On the premise of ensuring good out-of-band rejection performance, comprehensive collaborative optimization of the passband characteristics is achieved, and its comprehensive filtering performance is better.

[0014] Further, the transverse metallized via array in the first coupling window includes two metallized vias, and the distance between the two metallized vias is b; the longitudinal metallized via array in the first resonant cavity is aligned with the longitudinal metallized via array in the first coupling window, and the distance between them is svp; the transverse metallized via array in the second coupling window includes three metallized vias, and along the direction from the upper side to the lower side, the distances between the three metallized vias are svp and b in sequence; the distance between the transverse metallized via array and the upper longitudinal metallized via array is svp; the transverse metallized via array in the third coupling window includes two metallized vias, the distance between the two metallized vias is b, and the distance between the transverse metallized via array and the longitudinal metallized via array located on the lower side is svp.

[0015] Furthermore, a = 1.8mm. In this scheme, by controlling the spacing a to 1.8mm, the passband bandwidth, insertion loss, and voltage standing wave ratio can be synergistically optimized without changing the center frequency. This overcomes the technical contradiction of the trade-off between various performance indicators in the prior art, and achieves a more significant comprehensive filtering effect.

[0016] Furthermore, b = 2.0mm. When adjusting the spacing b, various indicators are mutually restrictive. Out-of-band rejection improves with increasing b, but loss and port matching initially improve but then deteriorate. The passband will exhibit three states: missing, flat, and distorted. In this solution, by controlling the spacing b to 2mm, the overall RF performance of the filter is better and the effect is more significant.

[0017] Furthermore, c = 2.1mm. In this scheme, the spacing c = 2.1mm is a critical threshold value. Exceeding the threshold will cause a significant reduction in passband bandwidth. By selecting the spacing c as 2.1mm, the risk of bandwidth degradation can be effectively avoided, and multiple RF performance parameters can be balanced and matched.

[0018] Preferably, d=2.2mm; e=2.3mm; f=3.352mm.

[0019] Furthermore, the transverse metallized via arrays within two adjacent coupling windows are respectively disposed on the upper and lower sides of the dielectric substrate. In this scheme, the transverse metallized via arrays within two adjacent coupling windows are respectively disposed on different sides of the dielectric substrate.

[0020] Furthermore, the top metal layer includes an input region, a resonant region, and an output region connected in sequence. The resonant region includes a metal layer, and the metallized via array is constructed on the dielectric substrate in the region corresponding to the resonant region. The input region includes an input microstrip feed line and an input transition structure, with both ends of the input transition structure connected to one end of the input microstrip feed line and one end of the metal layer, respectively. The input transition structure is constructed as a trapezoidal structure. The output region includes an output microstrip feed line and an output transition structure, with both ends of the output transition structure connected to one end of the output microstrip feed line and one end of the metal layer, respectively. The output transition structure is constructed as a trapezoidal structure. Preferably, the other end of the input microstrip feed line extends to the end of the dielectric substrate to serve as an input port; the other end of the output microstrip feed line extends to the end of the dielectric substrate to serve as an output port.

[0021] Beneficial effects: Compared with the prior art, the bandpass filter of the asymmetric gradient via array provided by the present invention adopts an asymmetric gradient composite metallized via array. Through the synergistic cooperation of four via arrangement mechanisms, namely encryption, gradient, non-uniformity and asymmetry, the electromagnetic field constraint is optimized without increasing the number of cavity stages, increasing the volume, or introducing any additional complex structures. It can effectively suppress spurious modes, improve out-of-band suppression capability, reduce insertion loss, improve passband flatness, and effectively expand the passband bandwidth. Attached Figure Description

[0022] To more clearly illustrate the technical solutions of the embodiments of the present invention, the accompanying drawings used in the embodiments will be briefly introduced below. It should be understood that the following drawings only show some embodiments of the present invention and should not be regarded as a limitation on the scope. For those skilled in the art, other related drawings can be obtained based on these drawings without creative effort.

[0023] Figure 1 This is a three-dimensional structural diagram of a bandpass filter provided in Embodiment 1 of the present invention.

[0024] Figure 2 for Figure 1 Front view.

[0025] Figure 3 for Figure 1 Top view.

[0026] Figure 4 This is a schematic diagram showing the division of the resonant cavity and coupling window in a bandpass filter provided in Embodiment 1 of the present invention.

[0027] Figure 5 This is a schematic diagram of the structure of the first resonant cavity in a bandpass filter provided in Embodiment 1 of the present invention.

[0028] Figure 6 This is a schematic diagram of the structure of the first coupling window in a bandpass filter provided in Embodiment 1 of the present invention.

[0029] Figure 7 This is a schematic diagram of the structure of the second resonant cavity in a bandpass filter provided in Embodiment 1 of the present invention.

[0030] Figure 8 This is a schematic diagram of the structure of the second coupling window in a bandpass filter provided in Embodiment 1 of the present invention.

[0031] Figure 9 This is a schematic diagram of the structure of the third resonant cavity in a bandpass filter provided in Embodiment 1 of the present invention.

[0032] Figure 10This is a schematic diagram of the structure of the third coupling window in a bandpass filter provided in Embodiment 1 of the present invention.

[0033] Figure 11 This is a schematic diagram of the bandpass filter used as a control group in Embodiment 1 of the present invention. In the diagram, all metallized vias are uniformly arranged.

[0034] Figure 12 The image shows the S-parameter simulation curves of the bandpass filter provided in Embodiment 1 of the present invention.

[0035] Figure 13 This is a voltage standing wave ratio (VSWR) curve of the bandpass filter provided in Embodiment 1 of the present invention.

[0036] Figure 14 The return loss curves of each bandpass filter provided in Embodiment 2 of the present invention are shown.

[0037] Figure 15 The insertion loss curves of each bandpass filter provided in Embodiment 2 of the present invention are shown.

[0038] Figure 16 The image shows the voltage standing wave ratio (VSWR) curves of each bandpass filter provided in Embodiment 4 of the present invention.

[0039] Figure 17 The return loss curves of each bandpass filter provided in Embodiment 4 of the present invention are shown.

[0040] Figure 18 The insertion loss curves of each bandpass filter provided in Embodiment 4 of the present invention are shown.

[0041] The markings in the figure are as follows: Metal top layer 11, dielectric substrate 12, metal bottom layer 13; Input region 2, input microstrip feed line 21, input transition structure 22; Resonant region 3, metal layer 31, vertical metallized via array 32, horizontal metallized via array 33, metallized via 34; Output region 4, output microstrip feed line 41, output transition structure 42; First resonant cavity 51, second resonant cavity 52, third resonant cavity 53, fourth resonant cavity 54; First coupling window 61, second coupling window 62, third coupling window 63. Detailed Implementation

[0042] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. The components of the embodiments of the present invention described and shown in the accompanying drawings can generally be arranged and designed in various different configurations. Therefore, the following detailed description of the embodiments of the present invention provided in the accompanying drawings is not intended to limit the scope of the claimed invention, but merely to illustrate selected embodiments of the invention. All other embodiments obtained by those skilled in the art based on the embodiments of the present invention without inventive effort are within the scope of protection of the present invention.

[0043] Example 1

[0044] like Figure 1 As shown, the present invention provides a bandpass filter with an asymmetric gradient via array, comprising a dielectric substrate 12, a top metal layer 11 disposed on the upper surface of the dielectric substrate 12, and a bottom metal layer 13 disposed on the lower surface of the dielectric substrate 12. The dielectric substrate 12, the top metal layer 11, and the bottom metal layer 13 form a three-layer stacked structure.

[0045] like Figure 1 and Figure 2 As shown, the dielectric substrate 12 can be made of Rogers 5880 high-frequency laminate, with a relative permittivity εr = 2.2 and a loss tangent tanδ ≤ 0.0009. The dielectric substrate 12 has a cuboid structure. For ease of description, in this embodiment, the length direction of the dielectric substrate 12 is the longitudinal direction (or Y direction), the width direction of the dielectric substrate 12 is the transverse direction (or X direction), and the two sides along the length direction of the dielectric substrate 12 are respectively referred to as the upper side and the lower side of the dielectric substrate 12, as shown. Figure 1 and Figure 2 As shown.

[0046] like Figure 1 and Figure 2 As shown, in this embodiment, the metal substrate 13 completely covers the lower surface of the dielectric substrate 12, and the metal substrate 13 can serve as a complete radio frequency ground plane.

[0047] In this embodiment, the metal top layer 11 includes an input region 2, a resonant region 3, and an output region 4 connected in sequence, wherein, as shown... Figure 1 and Figure 3 As shown, the resonant region 3 includes a rectangular metal layer 31. The width of the metal layer 31 is the same as the width of the dielectric substrate 12, and the length of the metal layer 31 is less than the length of the dielectric substrate 12. The metal layer 31 is preferably constructed in the middle position of the dielectric substrate 12 in order to reserve space for the input region 2 and the output region 4 connected to both sides.

[0048] like Figure 1 and Figure 3 As shown, in this embodiment, the input region 2 includes an etched input microstrip feed line 21 and an input transition structure 22. The two ends of the input transition structure 22 are connected to one end of the input microstrip feed line 21 and one end of the metal layer 31, respectively. The input transition structure 22 serves as a gradual transition. In implementation, the input transition structure 22 can preferably be constructed as a trapezoidal structure, such as... Figure 1 and Figure 3 As shown. The other end of the input microstrip feed line 21 can extend to the end of the dielectric substrate 12 to serve as an input port.

[0049] Similarly, such as Figure 1 and Figure 3 As shown, in this embodiment, the output region 4 includes an etched output microstrip feed line 41 and an output transition structure 42. The two ends of the output transition structure 42 are connected to one end of the output microstrip feed line 41 and one end of the metal layer 31, respectively. The output transition structure 42 serves as a gradual transition. In implementation, the output transition structure 42 can preferably be constructed as a trapezoidal structure, such as... Figure 3 As shown. The other end of the output microstrip feed line 41 can extend to the end of the dielectric substrate 12 to serve as an output port.

[0050] like Figure 3 and Figure 4 As shown, in this embodiment, a metallized via array is constructed on the dielectric substrate 12 corresponding to the resonant region 3. The metallized via array includes multiple metallized vias 34 arranged along the length direction of the dielectric substrate 12 and multiple metallized vias 34 arranged along the width direction of the dielectric substrate 12. Each metallized via 34 penetrates the dielectric substrate 12 perpendicularly along the thickness direction of the dielectric substrate 12, and the upper and lower ends of each metallized via 34 are reliably connected to the metal layer 31 of the top metal layer 11 and the metal bottom layer 13, respectively. In this embodiment, the metal via array, the top metal layer 11, and the bottom metal layer 13 together constitute a SIW rectangular resonant cavity. The SIW rectangular resonant cavity includes a multi-stage resonant cavity defined by the metallized vias 34. Adjacent resonant cavities are coupled through coupling windows. The coupling windows are mainly used to control the inter-stage energy coupling strength, wherein, as shown... Figure 3 and Figure 4 As shown, each resonant cavity includes longitudinal metallized via arrays 32 constructed on the upper and lower sides of the dielectric substrate 12 and linearly arranged along the length of the dielectric substrate 12. The longitudinal metallized via arrays 32 on both sides of the same resonant cavity are symmetrically arranged. The arrangement of the metallized vias 34 in the longitudinal metallized via arrays 32 in different resonant cavities is different, which is beneficial to enhance coupling, reduce insertion loss, and effectively eliminate stray resonance peaks in the passband. At the same time, as Figure 3 and Figure 4As shown, the coupling window includes not only the vertical metallized via array 32 constructed on the upper and lower sides of the dielectric substrate 12 and arranged linearly along the length direction of the dielectric substrate 12, but also the horizontal metallized via array 33 constructed between the vertical metallized via array 32 and arranged linearly along the width direction of the dielectric substrate 12. The horizontal metallized via array 33 is aligned with one of the metallized vias 34 in the vertical metallized via array 32. In this embodiment, the longitudinal metallized via arrays 32 on both sides of the same coupling window are symmetrically arranged. The arrangement of the metallized vias 34 in the longitudinal metallized via arrays 32 within different coupling windows is different. The spacing between the metallized vias 34 in the longitudinal metallized via arrays 32 within the coupling window is smaller than the minimum spacing between the metallized vias 34 in the resonant cavity, so as to achieve a dense arrangement. The metallized vias 34 in the longitudinal metallized via arrays 32 within at least one coupling window are arranged in a trapezoidal gradient, so that the characteristic impedance of the SIW equivalent waveguide wall changes smoothly along the propagation direction, eliminating obvious reflection interfaces, and at the same time, it can broaden the equivalent waveguide wall to support transmission modes (dominant mode TE). 10 The frequency response range allows the impedance to maintain a good matching state over a wider frequency band; the different placement or arrangement of the transverse metallized via array 33 in different coupling windows helps to reduce the voltage standing wave ratio and increase the passband bandwidth.

[0051] It is understood that, in this embodiment, the arrangement includes, but is not limited to, the number of metallized vias 34 and the spacing between adjacent metallized vias 34. In this embodiment, the inner diameter of each metallized via 34 can preferably be constructed to be the same, such as... Figure 3 As shown, for example, the inner diameter of the metallized through-hole 34 is 1.192 mm.

[0052] For ease of understanding, this embodiment provides a preferred implementation method for detailed description, such as... Figure 3 and Figure 4As shown, the filter defines a four-stage SIW rectangular resonant cavity through a metallized via 34. The four-stage SIW rectangular resonant cavities are arranged in a linear cascade. For ease of description, along the direction from the input region 2 to the output region 4 (i.e., along the transmission direction (or propagation direction), which will not be elaborated further), the four-stage SIW rectangular resonant cavity includes a first resonant cavity 51, a second resonant cavity 52, a third resonant cavity 53, and a fourth resonant cavity 54. There is a first coupling window 61 between the first resonant cavity 51 and the second resonant cavity 52, a second coupling window 62 between the second resonant cavity 52 and the third resonant cavity 53, and a third coupling window 63 between the third resonant cavity 53 and the fourth resonant cavity 54. The first coupling window 61, the second coupling window 62, and the third coupling window 63 are used to adjust the interstage energy coupling strength, respectively. Specifically, in one embodiment, the thickness of the dielectric substrate 12 can be 0.762 mm, and the width of the dielectric substrate 12 can be 19.65 mm; the thickness of the metal bottom layer 13 can be 0.035 mm; the width of the metal bottom layer 13 can be 19.65 mm; the width of the metal layer 31 in the metal top layer 11 can be 19.65 mm; the output microstrip feed line 41 and the output transition structure 42 are constructed at the middle position along the width direction of the dielectric substrate 12; similarly, the input microstrip feed line 21 and the input transition structure 22 are also constructed at the middle position along the width direction of the dielectric substrate 12.

[0053] like Figure 3 As shown, Figure 3 This is a top view of a bandpass filter provided in this embodiment. In the figure, W50 = 1.8 mm, representing the linewidth of the input microstrip feed line 21; l50 = 8 mm, representing the line length of the input microstrip feed line 21; lt = 6.85 mm, representing the length of the input transition structure 22; and wt = 5.5 mm, representing the width of the end of the input transition structure 22 away from the input microstrip feed line 21. In this embodiment, the input region 2 and the output region 4 are symmetrically arranged at both ends of the resonant region 3. The structure and size parameters of the output region 4 can be the same as those of the input region 2, which is beneficial for simplifying the structure. It is understood that in other embodiments, the structure and size parameters of the output region 4 can also be different from those of the input region 2, which will not be described in detail here.

[0054] For ease of description, in this embodiment, the spacing of the uniformly arranged metallized vias 34 within the resonant cavity is represented by SVP. This spacing is also the minimum spacing between the metallized vias 34 in the resonant cavity. In this embodiment, SVP is 2.384 mm; Figure 4 and Figure 5As shown, the longitudinal metallized via array 32 in the first resonant cavity 51 includes three metallized vias 34, and the spacing between two adjacent metallized vias 34 is svp; the spacing between the two longitudinal metallized via arrays 32 in the first resonant cavity 51 is w_siw=15.65mm; each metallized via 34 in the two longitudinal metallized via arrays 32 corresponds to one along the width direction of the dielectric substrate 12.

[0055] like Figure 4 and Figure 6 As shown, the vertical metallized via array 32 in the first coupling window 61 includes three metallized vias 34. The spacing 'a' between any two adjacent metallized vias 34 is less than svp (i.e., less than 2.384 mm) to achieve a denser arrangement, which is beneficial for enhancing coupling and reducing insertion loss. For example, in one embodiment, the spacing 'a' between any two adjacent metallized vias 34 in the vertical metallized via array 32 is 1.8 mm. Figure 6 As shown. The transverse metallized via array 33 in the first coupling window 61 includes two metallized vias 34, and the spacing b between the two metallized vias 34 is also less than svp (i.e., less than 2.384 mm), thus achieving a denser arrangement. This denser arrangement increases the equivalent transverse electric wall density in the coupling window region, making the metal boundaries on both sides of the window more compact. The electric field energy can be more effectively confined within the window region, thereby enhancing the electromagnetic coupling coefficient between adjacent resonant cavities. A stronger coupling coefficient means that, for the same transmission distance, the required field transition distance is shorter when energy is transferred from one cavity to the next, significantly reducing radiation loss and dielectric loss. For example, in one embodiment, the spacing b between the two metallized vias 34 in the transverse metallized via array 33 is 2 mm. Figure 6 As shown; simultaneously, the transverse metallized via array 33 is aligned with the metallized via 34 located at the middle position in the longitudinal metallized via array 32, for example, as Figure 6 As shown, the horizontal metallized via array 33 is aligned with the second metallized via 34 from the left in the vertical metallized via array 32, and the spacing between the horizontal metallized via array 33 and the vertical metallized via array 32 located below is svp (it can be understood that when the number of metallized vias 34 in the vertical metallized via array 32 is even, the horizontal metallized via array 33 preferentially aligns with the metallized via 34 near the middle position); correspondingly, as Figure 6 As shown, the distance w_w1 between the horizontal metallized via array 33 and the vertical metallized via array 32 located on the upper side is 11.266mm, that is, the window width of the first coupling window 61 is 11.266mm; the window length w_l1 is 3.6mm.

[0056] In this embodiment, the longitudinal metallized via array 32 in the first resonant cavity 51 is aligned with the longitudinal metallized via array 32 in the first coupling window 61, and the distance between them is svp.

[0057] like Figure 4 and Figure 7 As shown, in this embodiment, the longitudinal metallized via array 32 in the second resonant cavity 52 includes three metallized vias 34. Along the transmission direction (i.e., the direction from input region 2 to output region 4), the spacing between adjacent metallized vias 34 is svp and f, where svp = 2.384 mm and f = 3.352 mm. That is, the spacing between adjacent metallized vias 34 in the longitudinal metallized via array 32 in the second resonant cavity 52 gradually increases, achieving a non-uniform and asymmetric arrangement, which is beneficial for suppressing spurious modes. This results in bandwidth expansion while out-of-band suppression increases instead of decreasing. The spacing between two longitudinal metallized via arrays 32 in the second resonant cavity 52 is w_siw = 15.65 mm.

[0058] In this embodiment, the longitudinal metallized via array 32 in the second resonant cavity 52 is aligned with the longitudinal metallized via array 32 in the first coupling window 61, and the distance between them is greater than svp. For example, the distance between the longitudinal metallized via array 32 in the second resonant cavity 52 and the longitudinal metallized via array 32 in the first coupling window 61 can be 2.968 mm.

[0059] In implementation, the longitudinal metallized via array 32 within the second coupling window 62 includes k metallized vias 34, where k is preferably an odd number and k≠1. Centered on the middle metallized via 34, the metallized vias 34 on its left and right sides are arranged in a trapezoidal gradient. For example, along the direction away from the center, the spacing between adjacent metallized vias 34 gradually increases, achieving a longitudinal gradient arrangement. In conventional schemes, the via spacing in the propagation direction is a constant value, resulting in periodic abrupt changes in the characteristic impedance of the equivalent waveguide wall along the propagation direction. Each abrupt change produces a small reflection, and when these reflections are superimposed in phase, they form significant echo peaks (i.e., poor voltage standing wave ratios) at certain frequency points. Furthermore, the equivalent waveguide wall corresponding to a constant spacing has a fixed cutoff frequency and fixed dispersion characteristics, limiting the passband bandwidth. This embodiment employs a longitudinally gradient arrangement, which allows the characteristic impedance of the SIW equivalent waveguide wall to change smoothly along the propagation direction, thus eliminating obvious reflection interfaces. Simultaneously, the gradient arrangement also broadens the equivalent waveguide wall's range for transmission modes (dominant mode TE). 10The wider frequency response range allows for better impedance matching across a broader frequency band, which helps reduce the voltage standing wave ratio (VSWR) (e.g., from 1.31 in conventional solutions to 1.21) and increase the passband bandwidth (from 180MHz to 200MHz). For example, ... Figure 4 and Figure 8 As shown, the longitudinal metallized via array 32 in the second coupling window 62 includes five metallized vias 34. With the middle metallized via 34 as the center, the spacing between the metallized vias 34 in directions away from the center is a = 1.8 mm and b = 2 mm, respectively. In this embodiment, the longitudinal metallized via array 32 in the second resonant cavity 52 is aligned with the longitudinal metallized via array 32 in the second coupling window 62, and the spacing between them is svp.

[0060] like Figure 8 As shown, the horizontal metallized via array 33 is aligned with the metallized via 34 located in the middle position of the vertical metallized via array 32, and the spacing between the horizontal metallized via array 33 and the vertical metallized via array 32 located on the upper side is svp. Along the direction from top to bottom, the spacing between each metallized via 34 in the transverse metallized via array 33 within the second coupling window 62 can be gradually reduced, achieving a transversely non-uniform arrangement. This can disrupt the periodic boundary conditions in the width direction. When the periodicity is disrupted, the phase matching conditions of the higher-order modes that originally required strict phase matching to establish the standing wave are no longer met. The energy cannot form a stable resonance in the cavity, but is scattered to other directions or converted into radiation loss. The resonant frequencies of these higher-order modes are effectively pushed away from the working passband, causing them to disappear or attenuate significantly on the S-parameter curve, exhibiting a significantly enhanced effect of far-end out-of-band suppression. For example, the suppression reaches 31.28 dB at 7.79 GHz, while the conventional uniform scheme is only 24.35 dB at 7.79 GHz, with the suppression capability improved by about 6.93 dB (a relative enhancement of about 28.5%). For example, in one embodiment, the transverse metallized via array 33 in the second coupling window 62 may include three metallized vias 34, with the spacing between the three metallized vias 34 along the direction from top to bottom being: svp = 2.384 mm, b = 2 mm, etc. Figure 8As shown, the spacing w_w2 between the transverse metallized via array 33 and the longitudinal metallized via array 32 located on the lower side is 8.882 mm, that is, the window width of the first coupling window 61 is 8.882 mm, which makes the window width of the second coupling window 62 larger. A larger window width means that the opening between the two resonant cavities is larger, and the electromagnetic field can more easily enter from one cavity to the other, increasing the interstage coupling coefficient. This can achieve the effect of enhancing the interstage coupling strength, which not only reduces radiation loss and dielectric loss when energy is transferred between stages, but also further promotes the realization of a low insertion loss of 0.7655 dB; it also helps to broaden the passband bandwidth (from 180 MHz to 200 MHz). It can be understood that in this embodiment, the window length is w_l2 = 7.6 mm.

[0061] like Figure 4 and Figure 9 As shown, in this embodiment, the longitudinal metallized via array 32 in the third resonant cavity 53 may also include three metallized vias 34. Along the transmission direction, the spacing between two adjacent metallized vias 34 is f and svp, respectively, where f = 3.352 mm and svp = 2.384 mm. That is, the spacing between two adjacent metallized vias 34 in the longitudinal metallized via array 32 within the third resonant cavity 53 gradually decreases. In this embodiment, the second resonant cavity 52 and the third resonant cavity 53 are symmetrically arranged on both sides of the second coupling window 62.

[0062] In this embodiment, the longitudinal metallized via array 32 in the third resonant cavity 53 is aligned with the longitudinal metallized via array 32 in the second coupling window 62, and the distance between them is svp.

[0063] In implementation, the number of metallized vias 34 in the vertical metallized via array 32 within the third coupling window 63 can preferably be equal to the number of metallized vias 34 in the vertical metallized via array 32 within the second coupling window 62. The alignment positions of the horizontal metallized via array 33 and the vertical metallized via array 32 within the third coupling window 63 differ from those in the second coupling window 62. Furthermore, within the third coupling window 63, with the metallized vias 34 aligned with the horizontal metallized via array 33 in the vertical metallized via array 32 as the center, the metallized vias 34 on both sides are arranged with asymmetrical spacing. The overall structure of a conventional SIW filter is symmetrical, and the via array is also symmetrically arranged (i.e., the spacing sequence on the left and right sides is exactly the same). The symmetrical structure naturally supports the separation of odd and even modes, where even modes (such as TE) are separated. 20 TE 30The field distribution of the filter (e.g., symmetric, symmetrical) is easily excited by symmetrical structures and generates spurious responses in or near the passband. In practice, it has been found that even with a gradually varying and non-uniform arrangement, if the left and right sides are perfectly symmetrical, some even-order higher modes (whose electric field distribution is symmetrical about the central plane) may still be excited because they satisfy the boundary conditions of the symmetrical structure. In this embodiment, by constructing an asymmetrical arrangement within the third coupling window 63, the geometric symmetry of the entire filter can be completely broken, causing these even-order modes to lose the "symmetric boundary conditions" for excitation, thus preventing them from establishing stable resonances. This not only further purifies the spectral environment and helps control the passband ripple at an excellent level of 0.43dB (with no obvious spurious resonance peaks), but also provides additional improvements in near-end and far-end out-of-band suppression (e.g., suppression reaches 16.22dB at 9.15GHz, basically on par with the conventional scheme's 17.66dB, while far-end suppression is significantly enhanced).

[0064] For example, such as Figure 4 and Figure 10 As shown, the vertical metallized via array 32 in the third coupling window 63 includes five metallized vias 34. The horizontal metallized via array 33 is aligned with the second metallized via 34 of the vertical metallized via array 32. Centered on the second metallized via 34 along the transmission direction, the two sides are arranged with asymmetrical spacing. The spacing between the metallized vias 34 on the left side is a=1.8mm, and the spacing between the metallized vias 34 on the right side is c=2.1mm, d=2.2mm, and e=2.3mm respectively. That is, in the case of dense arrangement, the spacing between two adjacent metallized vias 34 gradually increases along the direction away from the center, which is beneficial to achieving better results.

[0065] like Figure 10 As shown, the horizontal metallized via array 33 is closer to the lower vertical metallized via array 32, and the distance between them is 2.384 mm. Along the direction from the lower to the upper side, the distance between each metallized via 34 in the horizontal metallized via array 33 within the third coupling window 63 is preferably less than 2.384 mm to achieve a denser (i.e., more compact) effect; however, it can also be equal to 2.384 mm. For example, in one embodiment, the horizontal metallized via array 33 in the third coupling window 63 includes two metallized vias 34, the distance between the two metallized vias 34 is b = 2 mm, and the distance between the horizontal metallized via array 33 and the lower vertical metallized via array 32 is svp; correspondingly, as... Figure 10As shown, the spacing w_w3 between the horizontal metallized via array 33 and the vertical metallized via array 32 located on the upper side is 11.266mm, that is, the window width of the third coupling window 63 is 11.266mm; the window length w_l3 is 8.4mm.

[0066] In this embodiment, the longitudinal metallized via array 32 in the third resonant cavity 53 is aligned with the longitudinal metallized via array 32 in the third coupling window 63, and the distance between them is 2.968 mm, which is greater than svp.

[0067] In this embodiment, the longitudinal metallized via array 32 in the fourth resonant cavity 54 may include one or more metallized vias 34. For example, Figure 4 As shown, the longitudinal metallized via array 32 in the fourth resonant cavity 54 may include a metallized via 34, which is aligned with the longitudinal metallized via array 32 in the third coupling window 63, and the spacing between them is svp, to achieve a smooth transition.

[0068] In implementation, as an example, the length of the entire resonant region 3 is l_siw = 54.57 mm. In implementation, the transverse metallized via arrays 33 within adjacent coupling windows are respectively disposed on the upper and lower sides of the dielectric substrate 12; that is, the transverse metallized via arrays 33 within adjacent coupling windows are respectively disposed on different sides of the dielectric substrate 12. In implementation, the spacing between adjacent transverse metallized via arrays 33 can be determined according to requirements; as an example, such as... Figure 3 As shown, the gap between two adjacent transverse metallized via arrays 33 can be 16.688 mm.

[0069] To verify the performance of the bandpass filter provided in this embodiment, this embodiment further employs ANSYS HFSS high-frequency electromagnetic simulation software to simulate the bandpass filter using the finite element method. The specific simulation parameters include: a solution frequency of 8.44 GHz; a sweep frequency range of 7~11 GHz with a step size of 0.001 GHz; a convergence condition of a maximum of 25 iterations and ΔS ≤ 0.02, where ΔS represents the maximum change in S-parameters between two consecutive adaptive iterations; S-parameters are complex parameters describing the energy relationship between incident and reflected waves between RF network ports, used to characterize the network's transmission and reflection characteristics, including return loss Si. 11 and insertion loss S 21During the calculation, a smaller ΔS indicates that the S-parameters of the re-refined mesh hardly change, resulting in stable and reliable results. Furthermore, in the simulation experiment, the radiation boundary includes an air box with a size greater than or equal to λ / 4. The air box refers to the virtual computational region surrounding the entire three-dimensional filter model, used to simulate the far-field radiation boundary of electromagnetic waves, where λ represents the wavelength of the electromagnetic wave.

[0070] The results of the simulation experiment are as follows Figure 12 and Figure 13 As shown, where, Figure 12 This is a simulation curve of the S-parameters of the bandpass filter provided in this embodiment, where the horizontal axis represents frequency (unit: GHz), the vertical axis represents the amplitude of the S-parameters (unit: dB), and the red curve represents the return loss S. 11 The purple curve represents the insertion loss S. 21 The center frequency of the bandpass filter is 8.44 GHz, the passband bandwidth is 200 MHz, the minimum insertion loss S21 in the passband is 0.77 dB, which is very low; the passband ripple is 0.43 dB, which is excellent, and there are no obvious spurious resonance peaks.

[0071] Figure 13 The graph shows the voltage standing wave ratio (VSWR) of the bandpass filter provided in this embodiment, where the horizontal axis represents frequency (unit: GHz) and the vertical axis represents voltage standing wave ratio (VSWR). The graph shows that at the center frequency of 8.44 GHz, the VSWR is 1.21, very close to the ideal value of 1. The passband starts at 8.34 GHz and cuts off at 8.54 GHz. Within the entire passband (i.e., within the range of 8.34 GHz to 8.54 GHz), the maximum VSWR does not exceed 1.78. The stopband attenuation includes 31.28 dB suppression at 7.79 GHz and 16.22 dB suppression at 9.15 GHz. Therefore, the bandpass filter provided by this invention exhibits excellent impedance matching characteristics and low signal reflection throughout the entire passband.

[0072] Furthermore, this embodiment also provides a bandpass filter as a control group. This bandpass filter also uses the same Rogers 5880 substrate as the dielectric substrate 12 and is configured with the same number of metallized vias 34 to form a metallized via array. A four-stage SIW rectangular resonant cavity is formed through the metallized via array. The relevant dimensional parameters are the same as those of the above-mentioned bandpass filter. The difference from the above-mentioned bandpass filter is that in the bandpass filter used as a control group, the metallized vias 34 are uniformly arranged, such as... Figure 11 As shown in Table 1, the same ANSYS HFSS high-frequency electromagnetic simulation software was used to simulate the bandpass filter, with the same simulation parameters.

[0073] Table 1. Performance Comparison of the Invention and Traditional Uniform Hole Distribution Scheme

[0074] As shown in Table 1, under the same design framework, the control group's insertion loss was 0.86 dB, the center frequency VSWR was 1.31, and the passband bandwidth was only 180 MHz. However, this invention, through an asymmetric gradient aperture strategy, effectively reduced the insertion loss from 0.86 dB to 0.77 dB, a reduction of approximately 10.5%; simultaneously optimized the center frequency VSWR from 1.31 to 1.21; expanded the passband bandwidth from 180 MHz to 200 MHz, an increase of approximately 11%; and significantly enhanced the far-end out-of-band rejection (24.35 dB at 7.79 GHz vs. 31.28 dB at 7.79 GHz), achieving unexpected technical effects.

[0075] In the bandpass filter provided in this embodiment, the metallized vias 34 in the metallized via array are arranged using an asymmetric, gradient via placement strategy, which has the following advantages:

[0076] First, on both sides of the common coupling window between two adjacent resonant cavities, the spacing of the metallized vias 34 of the first coupling window 61 is reduced from SVP (2.384 mm) to a smaller, denser spacing (1.8 mm). The denser region covers at least one via period on each side of the first coupling window 61, making the via density near the first coupling window 61 significantly higher than that in areas far from the window. The via period is typically the spacing between two adjacent vias (i.e., via spacing) along the propagation direction (i.e., longitudinal direction), and the via density refers to the number of metallized vias 34 per unit area. The equivalent transverse electric wall density of the first coupling window 61 is 0.56 holes / mm, which is about 33.3% higher than that of the uniform scheme (the via period is svp, which is 2.384mm and the via density is about 0.42 holes / mm). This enhances the electric field confinement, and the interstage coupling coefficient increases from 0.0117 in the uniform scheme to 0.0119. This reduces the radiation loss and dielectric loss during energy transfer, and the insertion loss in the passband decreases from 0.86dB in the uniform scheme to 0.77dB, a relative reduction of about 10.5%.

[0077] Secondly, along the transmission direction, the metallized vias 34 in this invention also adopt a longitudinally gradient arrangement: with the metallized via 34 located in the middle of the second coupling window 62 as the center, the spacing between the vias on both sides increases monotonically in the direction away from the center. This longitudinally gradient arrangement enables the characteristic impedance of the SIW equivalent waveguide wall to transition smoothly along the propagation direction, which can eliminate the reflection superposition caused by the periodic abrupt change in impedance in the conventional constant spacing scheme, and also broadens the equivalent waveguide wall for the main mode TE. 10The frequency response range of the filter is improved, and the passband bandwidth can be extended from 180MHz to 200MHz, which is about 11% of the original range. The voltage standing wave ratio is optimized from 1.31 to 1.21. Overall, it achieves a comprehensive performance improvement with low insertion loss, wide passband and excellent port matching.

[0078] Third, in the direction perpendicular to the transmission direction (i.e., laterally), the present invention employs a laterally non-uniform arrangement of 34 metallized vias. This non-uniform design can break the periodic boundary conditions of the metallized via array in the width direction, allowing higher-order spurious modes (such as TE) to be more easily dispersed. 20 TE 30 The phase matching conditions required for resonance cannot be met, making it difficult to establish a stable standing wave distribution. As a result, the resonant frequencies of these higher-order spurious modes are shifted out of the operating passband. Specifically, this manifests as a significant enhancement in far-end out-of-band suppression. For example, the suppression reaches 31.28 dB at 7.79 GHz, while the conventional uniform scheme only achieves 24.35 dB at 7.79 GHz, representing an improvement of approximately 6.93 dB (a relative enhancement of approximately 28.5%), which is very significant.

[0079] Fourth, this invention employs an asymmetrical spacing arrangement: Centered on the metallized vias 34 aligned with the horizontal metallized via array 33 within the vertical metallized via array 32 of the third coupling window 63, the metallized vias 34 on both sides of this center are asymmetrically arranged. This asymmetrical arrangement includes differences in the number of metallized vias 34, the spacing between them, and the variation pattern of the via spacing, thus exhibiting asymmetry. For example, in this embodiment, only one metallized via 34 is configured on the left side of the center, with a spacing of 1.8 mm between adjacent metallized vias 34; three metallized vias 34 are configured on the right side of the center, with spacings of 2.1 mm, 2.2 mm, and 2.3 mm respectively, starting from the center and moving away from it. This breaks the mirror symmetry at the entire bandpass filter level. Compared to conventional symmetrical structures, this asymmetrical spacing arrangement can further suppress even-order high-order modes (TE). 20 The excitation enables excellent bandpass ripple control at 0.43dB, and further improvements are achieved in near- and far-end out-of-band suppression.

[0080] The four structural features described above constitute a unified, systematic composite aperture strategy. In traditional SIW filter design, reducing insertion loss, improving voltage standing wave ratio (VSWR), expanding bandwidth, and enhancing out-of-band rejection (OBS) often constrain each other. For example, to reduce insertion loss, enhanced coupling is usually required, but this often leads to narrower bandwidth or worsened OBS rejection. To improve return loss (reduce VSWR), it may be necessary to sacrifice some bandwidth or increase the number of cavity stages (increase volume). However, this invention breaks these constraints at the physical root through the synergistic effect of the four structural features described above: the encryption mechanism enhances coupling and reduces insertion loss, but without a gradient mechanism to smooth impedance, enhanced coupling may introduce greater reflections. The gradient mechanism precisely solves this problem, allowing low insertion loss and low VSWR to coexist. While the gradient mechanism broadens the bandwidth, it may make out-of-band spurious modes more easily excited. The non-uniform arrangement and left-right asymmetric layout suppress spurious modes at the source, so that while the bandwidth is expanded, the out-of-band suppression does not decrease but increases. The non-uniform arrangement and asymmetric layout work together to eliminate spurious resonance peaks in the passband. While maintaining the single-layer PCB full-planar structure and without introducing defect ground structure (DGS), half-mode waveguide (HMSIW) or any additional etching units, this invention achieves a comprehensive performance leap with an insertion loss reduction of about 10.5%, passband ripple controlled at 0.43dB, VSWR optimized from 1.31 to 1.21, passband bandwidth expanded from 180MHz to 200MHz, and significantly enhanced far-end out-of-band suppression.

[0081] Example 2

[0082] Based on the bandpass filter structure provided in Example 1, this example further explores the impact of different values ​​of the base spacing SVP on the performance of the bandpass filter. Specifically, this example sets up four control groups: control group 1, control group 2, control group 3, and control group 4.

[0083] In control group 1, svp=2.184mm, the bandpass filter structure and other parameters are the same as in example 1;

[0084] In control group 2, svp=2.284mm, the bandpass filter structure and other parameters are the same as in example 1;

[0085] In control group 3, svp=2.484mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0086] In control group 4, svp=2.584mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0087] These bandpass filters were simulated using the ANSYS HFSS high-frequency electromagnetic simulation software described above, with the same simulation parameters. The simulation results are as follows. Figure 14 and Figure 15 As shown, where Figure 14 The graph shows the return loss curves for each bandpass filter. Figure 15 Table 2 shows the insertion loss curves for each bandpass filter and a comparison of their performance metrics.

[0088] Table 2 Performance Comparison

[0089] As shown in Table 2, when the SVP size is 2.384 mm, the filter passband bandwidth reaches 200 MHz, significantly better than the control group schemes. Simultaneously, the insertion loss is as low as 0.77 dB and the voltage standing wave ratio (VSWR) is only 1.21, both optimal values ​​for each group. It is noteworthy that when the SVP size deviates from 2.384 mm, whether decreasing to below 2.284 mm or increasing to above 2.484 mm, the passband bandwidth decreases significantly. Control group 1 even fails to form a flat passband, while the passband bandwidth of controls 3 and 4 drops sharply to 120 MHz. Simultaneously, the insertion loss and VSWR also show a deteriorating trend. Therefore, the bandpass filter architecture provided in Example 1, by controlling the SVP size to 2.384 mm, overcomes the technical bottleneck of the prior art where passband bandwidth, insertion loss, and VSWR are difficult to balance. While ensuring good out-of-band suppression performance, it achieves comprehensive synergistic optimization of passband characteristics, and its overall filtering performance is significantly better than the control schemes.

[0090] Example 3

[0091] Based on the bandpass filter structure provided in Example 1, this example further explores the influence of the spacing 'a' between the metallized vias in the longitudinal metallized via array within the first coupling window on the performance of this bandpass filter; such as Figure 6 As shown, the vertical metallized via array within the first coupling window includes three metallized vias. For ease of description, the distance between the metallized via at the middle position and the metallized via on the left is a (left), and the distance between the metallized via at the middle position and the metallized via on the right is a (right).

[0092] First, in this embodiment, the spacing a(right) of the right encryption zone is maintained at 1.8mm, and a = 1.8mm is also maintained at the remaining positions of the bandpass filter. Four control groups are set: control group 5, control group 6, control group 7, and control group 8.

[0093] In control group 5, a(left) = 1.6 mm, and the bandpass filter structure and other parameters were the same as in Example 1;

[0094] In control group 6, a(left) = 1.7 mm, and the bandpass filter structure and other parameters were the same as in Example 1;

[0095] In control group 7, a(left) = 1.9 mm, and the bandpass filter structure and other parameters were the same as in Example 1;

[0096] In control group 8, a(left) = 2.0 mm, and the bandpass filter structure and other parameters were the same as in Example 1;

[0097] It can be understood that a (left) corresponds to a in Example 1; these bandpass filters were simulated using the ANSYS HFSS high-frequency electromagnetic simulation software described above, with the same simulation parameters as above, and the results of the simulation experiment are shown in Table 3.

[0098] Table 3. Parameter scanning results of the left-side encryption interval a (left) of the first coupling window.

[0099] As shown in Table 3, when a(left) = 1.8mm, the passband bandwidth reaches its maximum value of 200MHz, the voltage standing wave ratio (VSWR) is as low as 1.21, and the far-end out-of-band rejection reaches a peak of 31.28dB@7.79GHz, exhibiting the best overall performance. Although the insertion loss is slightly lower (0.75dB) when a(left) = 1.6mm, its VSWR deteriorates to 1.37, and the bandwidth narrows to 170MHz; when a(left) = 1.9mm, the bandwidth drops sharply to 140MHz. Specifically, with the center frequency of each group remaining at 8.44GHz, setting the spacing a(left) to 1.8mm achieves a passband bandwidth of 200MHz, the maximum value for each group, significantly outperforming the control group schemes. The effect of varying spacing 'a' (left) on passband bandwidth exhibits a non-monotonic characteristic: as 'a' (left) gradually increases from 1.6 mm to 1.8 mm, the passband bandwidth continuously improves from 170 MHz to 200 MHz, while the voltage standing wave ratio (VSWR) significantly improves from 1.37 to 1.21; however, when 'a' (left) continues to increase to 1.9 mm, the passband bandwidth drops sharply to 140 MHz, and the VSWR also deteriorates; when 'a' (left) further increases to 2.0 mm, although the passband bandwidth recovers somewhat, the insertion loss increases significantly to 1.17 dB, and the passband transmission performance deteriorates markedly. Therefore, by controlling the spacing 'a' (left) at 1.8 mm, without changing the center frequency, a synergistic optimization of passband bandwidth, insertion loss, and VSWR is achieved, overcoming the technical contradiction of the inverse relationship between performance indicators in existing technologies, and resulting in a better overall filtering effect.

[0100] Secondly, in this embodiment, the spacing a(left) of the left encryption zone is maintained at 1.8mm, and a = 1.8mm is also maintained at the remaining positions of the bandpass filter. Four control groups are set up: control group 9, control group 10, control group 11, and control group 12.

[0101] In control group 9, a(right) = 1.6 mm, and the bandpass filter structure and other parameters are the same as in Example 1;

[0102] In control group 10, a(right) = 1.7 mm, and the bandpass filter structure and other parameters are the same as in Example 1;

[0103] In control group 11, a(right) = 1.9 mm, and the bandpass filter structure and other parameters are the same as in Example 1;

[0104] In control group 12, a(right) = 2.0 mm, and the bandpass filter structure and other parameters are the same as in Example 1;

[0105] It can be understood that a (right) corresponds to a in Example 1; these bandpass filters were simulated using the ANSYS HFSS high-frequency electromagnetic simulation software described above, with the same simulation parameters as above, and the results of the simulation experiment are shown in Table 4.

[0106] Table 4. Scanning results of the right-side encryption interval a (right) of the first coupling window.

[0107] As shown in Table 4, the optimal overall performance is achieved when a(right) = 1.8mm: bandwidth 200MHz, VSWR = 1.21, and far-end out-of-band rejection 31.28dB@7.79GHz. Although the far-end out-of-band rejection is slightly higher (31.46dB) when a(right) = 2.0mm, the bandwidth narrows to 170MHz, and the near-end out-of-band rejection deteriorates to 16.07dB. Specifically, the center frequency of each filter group is stable at 8.44GHz, the insertion loss is consistent at 0.77dB, and the voltage standing wave ratio, far-end out-of-band rejection, and near-end out-of-band rejection levels are similar. By setting the spacing a(right) to 1.8mm, the filter passband bandwidth reaches 200MHz, significantly better than the control group schemes. When the spacing a(right) deviates from this specific value, whether decreasing to 1.7mm or 1.6mm, or increasing to 1.9mm or 2.0mm, the passband bandwidth decreases significantly. It can be seen that by controlling the spacing a (right) to 1.8 mm, the passband bandwidth can be maximized without compromising insertion loss, voltage standing wave ratio and out-of-band suppression performance.

[0108] Based on the comparison results in Tables 3 and 4, when the spacing between the encryption zones on both sides of the first coupling window is symmetrical and both are 1.8 mm, that is, when a = 1.8 mm, the filter achieves the optimal overall performance: center frequency 8.44 GHz, passband bandwidth 200 MHz, insertion loss 0.77 dB, voltage standing wave ratio 1.21, far-end out-of-band rejection 31.28 dB @ 7.79 GHz, and near-end out-of-band rejection 16.22 dB @ 9.15 GHz.

[0109] Example 4

[0110] Based on the bandpass filter structure provided in Example 1, this example further explores the influence of different values ​​of spacing b on the performance of the bandpass filter. Specifically, this example sets up four control groups: control group 13, control group 14, control group 15, and control group 16.

[0111] In control group 13, b=1.5mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0112] In control group 14, b=1.7mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0113] In control group 15, b=2.3mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0114] In control group 16, b=2.5mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0115] These bandpass filters were simulated using the ANSYS HFSS high-frequency electromagnetic simulation software described above, with the same simulation parameters. The simulation results are as follows. Figure 16 , Figure 17 and Figure 18 As shown, where Figure 17 The graph shows the return loss curves for each bandpass filter. Figure 18 Table 5 shows the insertion loss curves for each bandpass filter and a comparison of their performance metrics.

[0116] Table 5 Performance Comparison

[0117] Based on the comprehensive parameter scanning data, it can be seen that, according to the above bandpass filter structure, b=2.0mm is the optimal value for electrical performance. At this value, the insertion loss is the lowest (0.77dB), the voltage standing wave ratio (VSWR) is optimal (1.21), and the passband is flat with a significant bandwidth (200MHz). As the b value increases from 1.5mm to 2.5mm, the center frequency continuously shifts (8.33→8.53GHz), the far-end out-of-band rejection continuously improves (25.06→36.28dB), and the near-end rejection also improves (14.61→17.96dB). However, the insertion loss and VSWR show a trend of first decreasing and then increasing. Except for the point where b=2mm, the return loss at the other points is extremely poor, or even non-existent. This indicates that at the other four points, the port matching is poor, and the passband shape changes from "unformed" to "flat" and then to "skewed distortion," as shown in the image. Figure 16 As shown, it can be seen that the various indicators of the filter have a contradictory relationship of mutual constraint as b changes, and there is no single monotonic optimization law. Based on the filter structure provided in Example 1, it is found that b=2.0mm is the optimal balance point. At b=2.0mm, multiple key RF performances of the filter can be optimized synchronously, and the effect is optimal. This optimal effect cannot be achieved at other points.

[0118] Example 5

[0119] Based on the bandpass filter structure provided in Example 1, this example further explores the influence of different values ​​of the spacing c within the third coupling window on the performance of the bandpass filter. Specifically, this example sets up four control groups: control group 17, control group 18, control group 19, and control group 20.

[0120] In control group 17, c=1.8mm, the bandpass filter structure and other parameters were the same as in Example 1;

[0121] In control group 18, c=1.9mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0122] In control group 19, c=2.0mm, the bandpass filter structure and other parameters were the same as in Example 1;

[0123] In control group 20, c=2.2mm, the bandpass filter structure and other parameters are the same as in Example 1;

[0124] These bandpass filters were simulated using the ANSYS HFSS high-frequency electromagnetic simulation software described above, with the same simulation parameters as above. The results of the simulation experiments are shown in Table 5.

[0125] Table 6 Performance Comparison

[0126] As shown in Table 6, the center frequency of each filter group remained stable at 8.44 GHz. When the spacing c was in the range of 1.8 mm to 2.1 mm, the filters could achieve a passband bandwidth of 200 MHz. Once the spacing c increased to 2.2 mm, the passband bandwidth immediately dropped to 170 MHz. In Example 1, the spacing c was limited to 2.1 mm. While maintaining a full bandwidth of 200 MHz and low insertion loss, a good voltage standing wave ratio was obtained, and the out-of-band suppression performance at both the far and near ends remained excellent. The above results indicate that there is a critical threshold value for the spacing c, which is 2.1 mm. Exceeding this threshold will cause a significant reduction in passband bandwidth. In implementation, by selecting the spacing c at the critical threshold (e.g., 2.1 mm), the risk of bandwidth degradation can be effectively avoided, and multiple RF performance parameters can be balanced and matched.

[0127] The above description is merely a specific embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any changes or substitutions that can be easily conceived by those skilled in the art within the scope of the technology disclosed in the present invention should be included within the scope of protection of the present invention.

Claims

1. A bandpass filter with an asymmetric gradient via array, comprising a dielectric substrate, a top metal layer disposed on the upper surface of the dielectric substrate, and a bottom metal layer disposed on the lower surface of the dielectric substrate; The dielectric substrate is constructed with an array of metallized vias, each of which is connected to a top metal layer and a bottom metal layer. The array of metallized vias, together with the top and bottom metal layers, constitutes a SIW rectangular resonant cavity. Its characteristic is... The SIW rectangular resonant cavity includes four resonant cavities. Adjacent resonant cavities are coupled through coupling windows. Each resonant cavity includes an array of longitudinally metallized vias constructed on the upper and lower sides of the dielectric substrate and arranged linearly along the length of the dielectric substrate. The arrangement of the metallized vias in the longitudinally metallized via arrays in different resonant cavities is different. The coupling window includes a vertical array of metallized vias constructed on the upper and lower sides of the dielectric substrate and arranged linearly along the length of the dielectric substrate, and a horizontal array of metallized vias constructed between the vertical arrays of metallized vias and arranged linearly along the width of the dielectric substrate. The arrangement of metallized vias in the longitudinal metallized via arrays within different coupling windows is different. The spacing between the metallized vias in the longitudinal metallized via arrays within the coupling window is smaller than the minimum spacing between the metallized vias in the resonant cavity, and the longitudinal metallized via arrays in at least one coupling window are arranged in a trapezoidal gradient. The setting position or arrangement of the transverse metallized via arrays within different coupling windows is different.

2. The bandpass filter with an asymmetric gradient aperture array according to claim 1, characterized in that, The SIW rectangular resonant cavity includes a first coupling window, a second coupling window, and a third coupling window arranged sequentially at intervals along the transmission direction, wherein... The spacing between the metallized vias in the longitudinal metallized via array within the first coupling window is all smaller than the minimum spacing between the metallized vias in the resonant cavity. The vertical metallized via array in the second coupling window includes k metallized vias, where k is an odd number and k≠1. With the metallized via located in the middle as the center, the metallized vias on the left and right sides are arranged in a trapezoidal gradient. The vertical metallized via array within the third coupling window includes at least three metallized vias, with the metallized vias on the left and right sides arranged asymmetrically, centered on the off-center metallized via.

3. A bandpass filter with an asymmetric gradient aperture array according to claim 2, characterized in that, Within the third coupling window, the spacing between adjacent metallized vias in the longitudinal metallized via array gradually increases along the direction away from the center.

4. A bandpass filter with an asymmetric gradient aperture array according to claim 1, characterized in that, Within the coupling window, the transverse metallized via array is aligned with one of the metallized vias in the longitudinal metallized via array.

5. A bandpass filter with an asymmetric gradient aperture array according to claim 1, characterized in that, The longitudinal metallized via arrays on both sides of the same resonant cavity are arranged symmetrically; the longitudinal metallized via arrays on both sides of the same coupling window are arranged symmetrically.

6. A bandpass filter with an asymmetric gradient aperture array according to claim 1, characterized in that, The metallized vias in the transverse metallized via array within the coupling window are non-uniformly arranged along the width direction of the dielectric substrate, and the spacing between two adjacent metallized vias is less than or equal to the minimum spacing between metallized vias within the resonant cavity.

7. A bandpass filter with an asymmetric gradient aperture array according to claim 6, characterized in that, From the bottom or top side of the dielectric substrate to the middle of the dielectric substrate, the spacing between two adjacent metallized vias in the transverse metallized via array gradually decreases.

8. A bandpass filter with an asymmetric gradient aperture array according to claim 2, characterized in that, The SIW rectangular resonant cavity includes a first resonant cavity, a second resonant cavity, a third resonant cavity, and a fourth resonant cavity. A first coupling window exists between the first and second resonant cavities, a second coupling window exists between the second and third resonant cavities, and a third coupling window exists between the third and fourth resonant cavities. The longitudinal metallized via array in the first resonant cavity includes three metallized vias, and the spacing between any two adjacent metallized vias is svp, where svp = 2.384 mm. The vertical metallized via array in the first coupling window includes three metallized vias, and the spacing between any two adjacent metallized vias in the vertical metallized via array is 'a', where 'a' is the distance between the metallized vias. <svp; The longitudinal metallized via array in the second resonant cavity includes three metallized vias. Along the transmission direction, the spacing between two adjacent metallized vias is svp and f, respectively, where f>svp. The vertical metallized via array in the second coupling window includes five metallized vias. With the metallized via located in the middle as the center, the spacing between two adjacent metallized vias along the direction away from the center is a and b respectively, where b>a. The longitudinal metallized via array in the third resonant cavity includes three metallized vias, and the spacing between two adjacent metallized vias along the transmission direction is f and svp, respectively. The vertical metallized via array in the third coupling window includes five metallized vias. Centered on the second metallized via along the transmission direction, the spacing between the metallized vias to the left of the center is a, and the spacing between the metallized vias to the right of the center is c, d, e, respectively, where c>b, d>c, and e>d. The longitudinal metallized via array in the fourth resonant cavity includes a metallized via.

9. A bandpass filter with an asymmetric gradient aperture array according to claim 8, characterized in that, The transverse metallized via array in the first coupling window includes two metallized vias with a spacing of b between them; the longitudinal metallized via array in the first resonant cavity is aligned with the longitudinal metallized via array in the first coupling window, and the spacing between them is svp. The transverse metallized via array in the second coupling window includes three metallized vias. Along the direction from the top to the bottom, the spacing between the three metallized vias is svp and b, respectively. The spacing between the transverse metallized via array and the upper longitudinal metallized via array is svp. The transverse metallized via array in the third coupling window includes two metallized vias with a spacing b between them, and a spacing svp between the transverse metallized via array and the longitudinal metallized via array located on the lower side.

10. A bandpass filter with an asymmetric gradient aperture array according to any one of claims 8-9, characterized in that, a=1.8mm; b=2.0mm; c=2.1mm; d=2.2mm; e=2.3mm; f=3.352mm.

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