Magnetic resonance wireless power supply device

Abstract

To provide a magnetic resonance type wireless power supply device capable of performing a selection fixing of at least one of two resonance loops with a method for not applying a limitation to a layout of a power supply coil and a power reception coil in regardless of an environment of a circumference.SOLUTION: A magnetic resonance type wireless power supply device comprises: a power supply side resonance circuit that contains a power supply coil; and a power reception resonance circuit that contains a power reception coil, and performs both of the power supply coil and the power reception coil to be in a magnetic resonance by using a PT (Parity-Time) symmetry. In a power supply side resonance circuit 1, an inverter 3 and a current sensor 4 are connected, and a timing that is a time when switching elements 33 and 34 in an internal part of the inverter 3 are turned on / turned off is determined on the basis of a pulse generated by a signal after a phase relation is adjusted by a phase adjustment circuit 30, and thus any one of two resonance loops is selected and fixed.SELECTED DRAWING: Figure 9

Description

Technical Field

[0001] The present invention relates to a magnetic resonance wireless power supply device that uses parity-time symmetry (hereinafter referred to as "PT symmetry") to perform non-contact power supply by magnetically resonating a power supply coil and a power receiving coil.

Background Art

[0002] Conventionally, several known technologies exist for wireless power supply technology, such as electromagnetic induction methods and magnetic field resonance methods. Among these, the wireless power supply technology of the electromagnetic induction method is used, for example, for charging mobile phones. The coils are arranged vertically, that is, based on the same principle as a transformer, power can be transmitted only when the distance (transmission distance) between the power supply coil and the power receiving coil is very close.

[0003] However, the wireless power supply technology of the electromagnetic induction method has a problem that since the transmission distance is as short as about several millimeters, the distance between the power supply coil and the power receiving coil cannot be increased. Also, if the positions of the power supply coil and the power receiving coil deviate or separate even slightly, charging and power supply cannot be performed, that is, it is vulnerable to positional deviation, so it is difficult to apply to artificial devices attached inside the human body such as artificial hearts or devices with multi-directional rotation or shaft misalignment such as robotic arms.

[0004] In addition, the wireless power supply technology of the magnetic field resonance method has a transmission distance of about several centimeters to several meters, so the distance between the power supply coil and the power receiving coil can be increased compared to the electromagnetic induction method, and it is approaching a practical level. However, power cannot be transmitted unless the distance between the power supply coil and the power receiving coil is fixed. Whether it approaches or moves away from that distance, or even if the angle is formed, the transmission efficiency decreases and the required power cannot be transmitted, that is, this method is also sensitive to positional deviation, so it is difficult to apply to devices with multi-directional rotation or shaft misalignment such as robotic arms (rotating power supply target).

[0005] Here, as one of the wireless power supply methods, there is a technology of wireless power supply using Parity-Time symmetry (hereinafter referred to as "PT symmetry"). This wireless power supply using PT symmetry is a physical system with a non-Hermitian Hamiltonian and is a new concept of wireless power supply first announced in 2017 (see Non-Patent Document 1).

[0006] When PT symmetry is conserved, the eigenenergy of the Hamiltonian becomes a real number. Therefore, although it is a non-Hermitian system, it behaves as if the energy of the system is conserved. In this case, the energy transmitted between the power supply side resonance circuit and the power receiving side resonance circuit per unit time no longer depends on the coupling coefficient between the two resonance circuits. As a result, even if the transmission distance changes or there is a misalignment in the positions of the power supply coil and the power receiving coil, as long as PT symmetry is conserved, the transmission power and the power transmission efficiency will always be kept constant.

[0007] Also, for the purpose of suppressing fluctuations in the transmission power and the power transmission efficiency with respect to changes in the transmission distance, many methods of automatically adjusting the frequency of the inverter used as an AC power supply have been proposed (see Non-Patent Document 2). However, in those control methods, it is not possible to make the transmission power and the power transmission efficiency completely independent of the transmission distance. This is because all of those systems are not designed as non-Hermitian physical systems and do not conserve PT symmetry.

[0008] And the wireless power supply using PT symmetry is obtained by replacing the AC power supply in the conventional magnetic field resonance type wireless power supply technology with an inverter that behaves electrically in the same way as a negative resistor, that is, an inverter that behaves as a negative resistor.

[0009] This is a well-known technique as disclosed, for example, in Patent Document 1. However, to explain it in more detail, an inverter that behaves as a negative resistance is an inverter in which the switching frequency and voltage amplitude are not fixed in advance. It has a circuit configuration in which the switching frequency is determined by the apparent resonance frequency of the wireless power supply circuit as seen from the output terminal of the inverter, and the switching frequency follows the change in the apparent resonance frequency of the wireless power supply circuit that can change due to changes in the transmission distance or positional misalignment of the coil at a fast response speed. Here, the wireless power supply circuit refers to a circuit including both the power supply side resonance circuit and the power receiving side resonance circuit, and all the subsequent circuits connected thereto. Also, the apparent resonance frequency means the substantial resonance frequency taking into account the interaction between the power supply side resonance circuit and the power receiving side resonance circuit since they interact with each other.

[0010] In addition, in wireless power supply using PT symmetry, the aforementioned inverter is used by self-exciting oscillation. There are two modes for oscillation, and PT symmetry can be preserved in either mode. Here, these two modes will be explained in more detail. FIG. 1 is a diagram showing an equivalent circuit of the S-P topology, which is the same as FIG. 14 in Patent Document 1.

[0011] As shown in FIG. 1, the power supply side resonance circuit 1 and the power receiving side resonance circuit 2 of the S-P topology can be represented as a complex resonance circuit coupled by a mutual inductance k m L. Also, L in FIG. 1 represents the self-inductance of each of the power supply coil 11 and the power receiving coil 21. L(1 - k m ) represents the leakage inductance of each of the power supply coil 11 and the power receiving coil 21. Also, r 1 ’ and r 2 ’ represent the winding resistances of each of the power supply coil 11 and the power receiving coil 21. r crepresents the iron loss equivalent resistance. C represents the capacitance (capacitance) of the capacitor. In the circuit shown in FIG. 1, there are two resonance loops (the resonance loop of loop I and the resonance loop of loop II) through which the resonance current flowing through the double resonance circuit can circulate. Also, regarding the frequency, the resonance frequency of loop I is ω l , and the resonance frequency of loop II is ω h .

[0012] However, as shown in the experimental results of Non-Patent Document 1, during operation, the two oscillation modes (resonance loops) may switch unintentionally. When the switching of the oscillation mode (resonance loop) occurs, the operation becomes unstable and power transmission itself becomes difficult. Therefore, a method for surely selecting one of the oscillation modes (resonance loops) was required.

[0013] Therefore, as one method for solving such a problem, in Patent Document 1, in a magnetic resonance type wireless power supply device that uses PT symmetry and performs non-contact power supply by magnetically resonating a power supply coil and a power receiving coil, a method of selecting one of the resonance loops by adjusting the magnitude of the Q value (selectivity) of the two resonance loops by coil design has been proposed.

Prior Art Documents

Patent Documents

[0014]

Patent Document 1

Non-Patent Documents

[0015]

Non-Patent Document 1

Non-Patent Document 2

Non-Patent Document 3

Non-Patent Document 4

Summary of the Invention

Problems to be Solved by the Invention

[0016] However, in a magnetic resonance wireless power supply device as shown in Patent Document 1 or the like, for example, a method of selecting one of the resonance loops by adjusting the Q values (selectivity) of the two resonance loops by coil design has been adopted. Therefore, there has been an application problem that restrictions occur in the design of the power supply coil and the power receiving coil, and the coils cannot have free shapes and dimensions.

[0017] The present invention has been made to solve the above problems. In a magnetic resonance wireless power supply device that uses PT symmetry and performs non-contact power supply by magnetically resonating a power supply coil and a power receiving coil, it is not affected by the surrounding environment, and in a method that does not impose restrictions on the design of the power supply coil and the power receiving coil, an object is to provide a magnetic resonance wireless power supply device capable of selectively fixing one of the two resonance loops.

Means for Solving the Problems

[0018] In order to achieve the above object, the present invention provides a magnetic resonance type wireless power supply device that includes a power supply side resonance circuit including a power supply coil and a power receiving side resonance circuit including a power receiving coil, utilizes Parity-Time symmetry, and performs non-contact power supply by magnetically resonating the power supply coil and the power receiving coil. When the power supply side resonance circuit and the power receiving side resonance circuit are regarded as a complex resonance circuit coupled to each other by mutual inductance, there are two resonance loops (the resonance loop of Loop I and the resonance loop of Loop II) through which the resonance current flowing through the complex resonance circuit can circulate. The power supply side resonance circuit is connected to an inverter and a sensor that detects the current or magnetic field of the power supply coil. The inverter includes a phase adjustment circuit that can adjust the phase relationship between the alternating current voltage applied to the power supply side resonance circuit and the alternating current flowing through the power supply coil based on the current phase based on the current of the power supply coil detected by the sensor or the magnetic field phase based on the magnetic field of the power supply coil. The switching timing, which is the time to turn on or off the switching element inside the inverter, is determined based on a pulse generated by a signal after the phase relationship is adjusted by the phase adjustment circuit, so that one of the two resonance loops is selectively fixed.

Effect of the Invention

[0019] According to the present invention, in a magnetic resonance type wireless power supply device that utilizes PT symmetry and performs non-contact power supply by magnetically resonating a power supply coil and a power receiving coil, by providing a phase adjustment circuit for adjusting the phase relationship between the alternating current voltage applied to the power supply side resonance circuit and the alternating current flowing through the power supply coil, it is possible to selectively fix one of the two resonance loops in a manner that is not affected by the surrounding environment and does not impose restrictions on the design of the power supply coil and the power receiving coil.

Brief Description of the Drawings

[0020]

Figure 1

Figure 2

Figure 3

Figure 4

Figure 5

Figure 6

Figure 7

Figure 8

Figure 9

Figure 10

Figure 11

Figure 12

Figure 13

Figure 14

Figure 15

Figure 16

Figure 17

Figure 18

Embodiments for Carrying Out the Invention

[0021] The present invention relates to a magnetic resonance type wireless power supply device that uses parity-time symmetry (hereinafter referred to as "PT symmetry") to perform non-contact power supply by magnetically resonating a power supply coil and a power receiving coil. Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[0022] Embodiment 1. FIG. 2 is a conceptual diagram showing two typical circuit configurations of a magnetic field resonance type wireless power supply using PT symmetry. In any of the circuit examples shown in FIGS. 2(a) and 2(b), the power supply-side resonance circuit 1 is connected to an inverter 3 as an AC power supply. On the other hand, a load resistor R L is connected to the load of the power receiving-side resonance circuit 2. In these circuits, L 1 represents the self-inductance of the power supply coil 11, C 1 represents the capacitance of the power supply-side capacitor, L 2 represents the self-inductance of the power receiving coil 21, C 2 represents the capacitance of the power receiving-side capacitor. Also, r 1 and r2 represent the resistance components included in the power supply side resonance circuit 1 and the power receiving side resonance circuit 2, respectively.

[0023] The circuit example shown in Fig. 2(a) is called the S-S topology because the coil and capacitor of the power supply side resonance circuit 1 are connected in series, and the coil and capacitor of the power receiving side resonance circuit 2 are also connected in series. In the circuit example shown in Fig. 2(b), since the coil and capacitor of the power supply side resonance circuit 1 are connected in series and the coil and capacitor of the power receiving side resonance circuit 2 are connected in parallel, it is called the S-P topology. An example of wireless power supply that preserves PT symmetry in the S-S topology is disclosed in Non-Patent Document 3, and an example of wireless power supply that preserves PT symmetry in the S-P topology is disclosed in Non-Patent Document 4.

[0024] Here, although it is repetitive, taking the equivalent circuit of the S-P topology shown in Fig. 1 as an example, the power supply side resonance circuit 1 and the power receiving side resonance circuit 2 of the S-P topology can be represented as a complex resonance circuit coupled by mutual inductance k m L. Also, L in Fig. 1 represents the self-inductance of each of the power supply coil 11 and the power receiving coil 21. L(1 - k m ) represents the leakage inductance of each of the power supply coil 11 and the power receiving coil 21. Also, r 1 ’ and r 2 ’ represent the winding resistances of each of the power supply coil 11 and the power receiving coil 21. r c represents the iron loss equivalent resistance. C represents the capacitance (electrostatic capacitance) of the capacitor. In the circuit shown in Fig. 1, there are two resonance loops (the resonance loop of Loop I and the resonance loop of Loop II) through which the resonance current flowing through the complex resonance circuit can circulate. Also, regarding the frequency, let the resonance frequency of Loop I be ω l , and the resonance frequency of Loop II be ω h .

[0025] Also, in the wireless power supply device using PT symmetry in Embodiment 1 of the present invention, as described above, the inverter is self-oscillated and used, and there are two modes in the oscillation, and in both modes, PT symmetry can be preserved, which is the same as the conventional case. As shown in FIGS. 1 and 2, it goes without saying that the power supply side resonance circuit 1 includes the power supply coil 11, and the power reception side resonance circuit 2 includes the power reception coil 21.

[0026] And in any of the topologies shown in FIGS. 2(a) and 2(b), the wireless power supply circuit seen from the output terminal of the inverter 3 forms a double resonance circuit, and there are two resonance loops. However, the exact mathematical formulas of the two resonance loops cannot be derived from the circuit diagrams shown in FIG. 2. Regarding this, strictly speaking, it can be obtained from the coupled mode theory, and the derivation process is as shown in Non-Patent Document 4 (as shown by one of the applicants of the present application in Non-Patent Document 4). Here, the detailed explanation and derivation process are omitted, and only the results are described. They are as shown in equations (1) and (2).

[0027] [Number]

[0028] [Number]

[0029] That is, there is a resonance loop having the resonance frequency of ω shown in equation (1), and a resonance loop having the resonance frequency of ω shown in equation (2). And these two resonance loops are in the relationship of ω > ω. ω is the natural angular resonance frequency when the power supply side resonance circuit and the power reception side resonance circuit are far apart and not completely coupled, and can be expressed by equation (3). Note that Γ and Γ in equations (1) and (2) h of the resonance loop with the resonance frequency, and the resonance loop with the resonance frequency of ω shown in equation (2) exists. And these two resonance loops are in the relationship of ω l > ω h > ω l of. ω 0 is the natural angular resonance frequency when the power supply side resonance circuit and the power reception side resonance circuit are far apart and not completely coupled, and can be expressed by equation (3). Note that Γ in equations (1) and (2) 20 and ΓL All of these are parameters (CMT parameters) in the coupling mode theory. However, as detailed in Patent Document 1 (as shown by one of the applicants of the present application in Patent Document 1 and Non-Patent Document 4), since they have little relevance in the description of the present invention and the definition explanation would be lengthy, detailed explanations are omitted here.

[0030]

Number

[0031] Also, k is a parameter called the coupling rate, which is a parameter related to the number of times energy is exchanged per unit time between the power supply side resonance circuit and the power receiving side resonance circuit. Furthermore, the coupling rate k can be expressed as in Equation (4) using the magnetic coupling coefficient k m between the power supply coil and the power receiving coil.

[0032]

Number

[0033] Here, for example, in the case of the S-P topology in Fig. 2(b), when Equations (1) and (2) are applied, they can be expressed as the following Equations (5) and (6).

[0034]

Number

[0035]

Number

[0036] Note that the above Equations (5) and (6) can also be approximately expressed as Equations (7) and (8). In Patent Document 1, these approximate equations, Equations (7) and (8), are used in the explanation.

[0037] [Number]

[0038] [Number]

[0039] From equations (5) and (6), the resonance frequency ω of loop II h and the resonance frequency ω of loop I l are found to be functions of only the magnetic coupling coefficient k m . In the case of a complex coil shape, it is difficult to formulate the relationship between the magnetic coupling coefficient k m and the distance d (transmission distance) between the power supply coil and the power receiving coil. Generally, it is obtained by numerical calculation using a computer. However, the relationship in which the magnetic coupling coefficient k m is inversely proportional to the transmission distance d between the coils is universal. Fig. 3 is a graph of the numerical calculation results showing a typical example of the relationship between the magnetic coupling coefficient k m and the transmission distance d between two coils (power supply coil and power receiving coil).

[0040] As shown in Fig. 3, since the relationship between the magnetic coupling coefficient k m and the transmission distance d between two coils (power supply coil and power receiving coil) can be numerically calculated, from equations (5) and (6), the resonance frequency ω of loop II h and the resonance frequency ω of loop I l and the relationship between the transmission distance d between two coils (power supply coil and power receiving coil) can also be calculated. Fig. 4 is a graph showing the relationship between the resonance frequency ω of loop II h and the resonance frequency ω of loop I l and the transmission distance d between two coils (power supply coil and power receiving coil).

[0041] As shown in Fig. 4, the resonance frequency ω of loop II h tends to decrease as the transmission distance d between the two coils increases, and the resonance frequency ω of loop I lIt increases in the direction of increasing transmission distance d. In the example shown in FIG. 4, the transmission distance d between the two coils is around 70 mm, and the resonance frequencies of the two resonance loops match. For transmission distances after the match, since PT symmetry is not preserved, it is excluded from the discussion of the present invention.

[0042] As described above, in wireless power feeding with PT symmetry, since the inverter is used with self-excitation oscillation, the resonance frequency ω of loop II h and the resonance frequency ω of loop I l Whichever resonance loop is selected, self-excitation oscillation occurs. However, when there is a variation in the transmission distance d during operation, if the resonance frequency ω of loop II h and the resonance frequency ω of loop I l suddenly and unintentionally switch, the frequency of self-excitation oscillation will suddenly (discontinuously) change greatly, and thus power transmission will be interrupted at this time. Also, since there are differences in the magnitude of the transmitted power and the power transmission efficiency depending on the mode, there is also a problem that a step occurs in the magnitude of the transmitted power due to the switching between the resonance frequency ω of loop II h and the resonance frequency ω of loop I l

[0043] Therefore, in this invention, based on the idea that even if the transmission distance d varies, if one of the resonance loops is fixed, the frequency of self-excitation oscillation changes smoothly (continuously), enabling stable wireless power feeding, a circuit configuration is provided with a phase adjustment circuit for advancing / delaying the phase of the voltage with respect to the current of the power feeding coil 11 for the purpose of selecting and fixing either one of the resonance loops having the resonance frequency ω of loop II h or the resonance loop having the resonance frequency ω of loop I l

[0044] First, the circuit configuration of an inverter without a phase adjustment circuit such as the invention of the present application will be described. For an inverter used in PT-symmetric wireless power supply, a class-D inverter or a class-E inverter is used. FIG. 5 is a circuit diagram when a class-D inverter is connected to a power supply-side resonance circuit as an inverter used in PT-symmetric wireless power supply. Actually, there is a power-receiving-side resonance circuit that is wirelessly powered from the power supply-side resonance circuit 1 shown in FIG. 5, and a load resistor connected to the power-receiving-side resonance circuit, but the illustration is omitted in FIG. 5. FIG. 6 is a schematic diagram showing the phase relationship of each waveform of the power supply-side resonance circuit 1 shown in FIG. 5.

[0045] As described in the prior art, wireless power supply using PT symmetry is obtained by replacing an AC power supply in the conventional magnetic resonance type wireless power supply technology with an inverter that behaves electrically like a negative resistor, that is, an inverter that behaves as a negative resistor.

[0046] An inverter that behaves as a negative resistor is an inverter whose switching frequency and voltage amplitude are not fixed in advance, and has a circuit configuration in which the switching frequency is determined by the apparent resonance frequency of the wireless power supply circuit as seen from the output terminal of the inverter, and the switching frequency follows the change in the apparent resonance frequency of the wireless power supply circuit that can change due to changes in the transmission distance or positional deviation between two coils (power supply coil and power receiving coil) with a fast response speed. And a wireless power supply circuit includes all circuits including the power supply-side resonance circuit, the power-receiving-side resonance circuit, and circuits connected thereto. Also, the apparent resonance frequency means the substantial resonance frequency taking into account the interaction between the power supply-side resonance circuit and the power-receiving-side resonance circuit because they interact with each other.

[0047] As shown in FIG. 5, an inverter 3 and a current sensor 4 are connected to the power supply side resonance circuit 1, and the inverter 3 and the current sensor 4 realize a negative resistance circuit. Further, as elements constituting the inverter 3, there are a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34. The high-side FET 33 and the low-side FET 34 are switching elements inside the inverter 3.

[0048] In the comparator 31 in FIG. 5, a detection signal of the alternating current i flowing through the power supply coil 11 detected by the current sensor 4 is input. When an input pulse, which is the output of the comparator 31, is input to the gate driver 32, two-phase output pulses (a first-phase output pulse and a second-phase output pulse) with a phase difference of 180° for alternately energizing / non-energizing (turning on / turning off) the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, are output from the gate driver 32. The input pulse input to the gate driver 32 is created based on the alternating current i flowing through the power supply coil 11 1 or the alternating magnetic field generated in the power supply coil 11. 1

[0049] At this time, the switching timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, is determined based on the input pulse to the gate driver 32. Here, the "switching timing" refers to the time when the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, are turned on or off. Then, the detection signal of the alternating current i 1 detected by the current sensor 4 installed between the inverter 3 and the power supply side resonance circuit 1 is input to the comparator 31 to generate the aforementioned input pulse.

[0050] ​That is, the drive current of the power supply coil 11 in the power supply side resonance circuit 1, which is the output of the inverter 3, is fed back to the control side of the inverter 3 to control the output current of the inverter 3, and this is called "feedback control". Here, the detection signal of the alternating current i 1 is an alternating voltage signal in phase with the alternating current i 1 . In principle, since the alternating current i 1 and the input voltage v 1 of the power supply side resonance circuit are in phase (no phase difference), self-excited oscillation occurs due to positive feedback. This self-excited oscillation excites the power supply coil to realize wireless power supply.

[0051] However, in an actual circuit, the alternating current i 1 and the input voltage v 1 will not be in phase. Actually, since a time delay occurs in the process of the voltage signal propagating through the circuit, the input voltage v 1 lags behind the alternating current i 1 . This phase difference will be referred to as the initial phase difference. The phase compensation circuit described in Patent Document 1 compensates for this initial phase difference and is a circuit for making the alternating current i 1 and the input voltage v 1 in phase, and has a significantly different purpose from the invention of the present application.

[0052] And, as described above, making the alternating current i 1 and the input voltage v 1 in phase is theoretically correct, but actually, when the alternating current i 1 and the input voltage v 1 are made in phase, the problem that the switching between the two resonance loops as described above frequently occurs will occur. Conversely, when an intentional phase difference is provided between the alternating current i 1 and the input voltage v 1 , it has been found by experiments that it stabilizes in one of the resonance loops.

[0053] Fig. 7 shows the experimental results of the phase relationship between the alternating current i 1 and the input voltage v 1 . Fig. 7(a) shows the alternating current i 1For the input voltage v 1 When it is a lagging phase, Fig. 7(b) shows the alternating current i 1 and the input voltage v 1 When they are in phase, Fig. 7(c) shows the alternating current i 1 For the input voltage v 1 When it is a leading phase is shown.

[0054] As shown in Fig. 7(a), when the input voltage v 1 lags behind the alternating current i 1 , the resonance loop of the resonance frequency ω l of loop I is selected. As shown in Fig. 7(b), when the alternating current i 1 and the input voltage v 1 are almost in phase, the resonance loop becomes unstable. Also, as shown in Fig. 7(c), when the input voltage v 1 leads the alternating current i 1 , the resonance loop of the resonance frequency ω h of loop II is selected.

[0055] Next, the principle of selecting the resonance loop by adjusting the phase relationship between the alternating current i 1 and the input voltage v 1 will be explained. As described above, the inverter of the wireless power supply device in Embodiment 1 of this invention self-oscillates by positive feedback of the current waveform of the alternating current i 1 . Therefore, this inverter can be regarded as an AC power supply with a fixed oscillation frequency. As shown in Fig. 5, the inverter 3 is connected to the power supply-side resonance circuit 1. Then, the power-receiving-side resonance circuit 2 (see Figs. 1 and 2) and the load resistor R L (see Figs. 1 and 2) are also indirectly connected to the inverter 3 by magnetic coupling.

[0056] The circuit seen from the output terminal of the inverter 3 forms a resonance circuit, and Fig. 8 shows a simple equivalent circuit representing this resonance circuit. Fig. 8 is a circuit diagram showing the equivalent circuit of wireless power supply connected to the inverter. Here, let the resistance component of the circuit be R, the inductance component be L, and the capacitance component be C. When the inverter self-oscillates at the resonance frequency ω c the magnitudes of the inductive reactance and the capacitive reactance must satisfy the following equation (9).

[0057]

Equation

[0058] Therefore, the reactance component of the resonance circuit as seen from the output terminal of the inverter becomes zero, and it is equivalent to a circuit in which only the resistor R is connected to the inverter. This is the original equivalent circuit when PT symmetry is preserved. And as described above, in the original equivalent circuit, only the resistance component R is connected to the inverter, so the alternating current i 1 and the input voltage v 1 are in phase. However, in this invention, the phase difference between the alternating current i 1 and the input voltage v 1 which are originally in phase is forcibly created.

[0059] For example, as shown in Fig. 7(a), when the input voltage v 1 is delayed with respect to the alternating current i 1 the equivalent circuit connected to the inverter becomes capacitive. In other words, when the input voltage v 1 is delayed with respect to the alternating current i 1 the resonance frequency ω deviates slightly from the original resonance frequency ω c and the equivalent circuit connected to the inverter becomes capacitive.

[0060] On the other hand, as shown in Fig. 7(c), when the input voltage v 1 is advanced with respect to the alternating current i 1 the resonance frequency ω deviates slightly from the original resonance frequency ω c and the equivalent circuit connected to the inverter becomes inductive. By the way, the impedance Z of the equivalent circuit connected to the inverter can be expressed by the following equation (10). Also, the inductive condition is equation (11).

[0061] [Number]

[0062] [Number]

[0063] Also, from equation (9), since the inductance component L can be expressed as in equation (12), substituting equation (12) into equation (11), the inductive condition becomes as in equation (13).

[0064] [Number]

[0065] [Number]

[0066] Thus, for the alternating current i 1 advancing the input voltage v 1 at that time, the resonance frequency ω is slightly higher than the original resonance frequency ω c .

[0067] As described above, the resonance frequency ω c is the resonance frequency ω of loop II h or the resonance frequency ω of loop I l . Also, since ω h > ω l is in the relationship, a sufficient condition for satisfying the relationship ω > ω shown in equation (13) c is ω > ω h . However, since ω is a frequency slightly higher than ω h , ω becomes a frequency that hardly changes from ω h . That is, for the alternating current i 1 advancing the input voltage v 1 at that time, the resonance frequency ω of loop IIh will select the resonance loop of

[0068] On the other hand, the capacitive condition is given by equation (14). Also, from equations (9) and (14), the capacitive condition becomes as in equation (15).

[0069]

Equation

[0070]

Equation

[0071] Thus, when the input voltage v 1 lags behind the alternating current i 1 , the resonance frequency ω at that time is slightly lower than the original resonance frequency ω c .

[0072] As described above, the resonance frequency ω c is the resonance frequency ω h of loop II or the resonance frequency ω l of loop I. Also, since ω h > ω l , a sufficient condition to satisfy the relationship ω < ω c shown in equation (15) is ω < ω l . However, since ω is a frequency slightly lower than ω l , ω becomes a frequency that hardly changes from ω l . That is, when the input voltage v 1 lags behind the alternating current i 1 , the resonance loop of the resonance frequency ω l of loop I will be selected.

[0073] Therefore, an actual circuit example will be described. As described above, the alternating current i 1 and the input voltage v 1When intentionally providing a phase difference, it has been confirmed that it can be stabilized in any of the resonance loops (for experimental results, refer to FIGS. 16 to 18 described later). Therefore, for the purpose of selecting and fixing the resonance loop of the resonance frequency ω h of loop II, when advancing the input voltage v 1 with respect to the alternating current i 1 , a phase advance circuit may be inserted before inputting the alternating current detected by the current sensor into the comparator. However, since the phase advance circuit also functions as a high-pass filter and preferentially passes high-frequency noise components, the S / N ratio deteriorates and it is not suitable for stable operation.

[0074] On the other hand, the phase delay circuit also functions as a low-pass filter and blocks high-frequency noise components, so it is suitable for stable operation. Of course, since it is a phase delay circuit, it cannot create a phase advance circuit. Therefore, in an actual circuit, as shown in FIG. 9, a phase adjustment circuit 30 is configured by combining a preamplifier 36 and a phase delay circuit 35 to realize a phase advance. Thereby, it is possible to select and fix either one of the two resonance loops and realize a stable operation.

[0075] FIG. 9 is a circuit diagram showing an example of an inverter with a phase adjustment circuit and a power supply side resonance circuit in Embodiment 1 of the present invention. As shown in FIG. 9, an inverter 3 is connected to the power supply side resonance circuit 1. Regarding the elements constituting the inverter 3, namely, a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34, it is the same as the circuit diagram shown in FIG. 5.

[0076] In addition, in FIG. 9, only the power supply side resonance circuit 1, the inverter 3 connected thereto, and the current sensor 4 are illustrated. However, on the premise that, similar to FIGS. 1 and 2, it includes a power supply side resonance circuit 1 including a power supply coil 11 and a power receiving side resonance circuit 2 including a power receiving coil 21, and utilizes PT symmetry (Parity-Time symmetry) to perform non-contact power supply by magnetically resonating the power supply coil 11 and the power receiving coil 21. When the power supply side resonance circuit 1 and the power receiving side resonance circuit 2 are regarded as a complex resonance circuit coupled to each other by mutual inductance, there exist two resonance loops (the resonance loop of Loop I and the resonance loop of Loop II) through which the resonance current flowing through the complex resonance circuit can circulate.

[0077] However, in the case of the circuit in FIG. 9, that is, an example of the circuit in Embodiment 1 of the present invention, a phase adjustment circuit 30 including a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 is installed before the input to the comparator 31 provided inside the inverter 3. Based on detecting the current or magnetic field of the power supply coil 11, the alternating current i 1 is delayed by more than 180° by the phase adjustment circuit 30 (inverting amplifier circuit 36 and phase delay circuit 35) before being input to the comparator 31, and phase control is performed to control it as an advancing phase. Thereby, the "resonance loop of Loop II" among the two resonance loops can be selectively fixed. 1 That is, if the phase adjustment circuit 30 (preamplifier (inverting amplifier circuit) 36 and phase delay circuit 35) in FIG. 9 is not installed, the alternating current i flowing through the power supply coil 11 detected by the current sensor 4 is input to the comparator 31 of the inverter 3.

[0078] 1 ​When the detection signal of [[ID=]] is input and the input pulse, which is the output of the comparator 31, is input to the gate driver 32, two-phase output pulses (the first-phase output pulse and the second-phase output pulse) with a phase difference of 180° for alternately energizing / non-energizing (turning on / turning off) the high-side FET 33 and the low-side FET 34, which are the switching elements inside the inverter 3, are output from the gate driver 32. The input pulse input to the gate driver 32 is the alternating current i flowing through the power supply coil 11 1 or is created based on the alternating magnetic field generated in the power supply coil 11.

[0079] However, in Fig. 9, before the input to the comparator 31, a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 are installed, and based on detecting the current or magnetic field of the power supply coil 11, the alternating current i 1 Before being input to the comparator 31, the alternating current i 1 is delayed by more than 180° by the inverting amplifier circuit 36 and the phase delay circuit 35, and a signal is input to the comparator 31. When the input pulse, which is the output of the comparator 31, is input to the gate driver 32, two-phase output pulses (the first-phase output pulse and the second-phase output pulse) with a phase difference of 180° for alternately energizing / non-energizing (turning on / turning off) the high-side FET 33 and the low-side FET 34, which are the switching elements inside the inverter 3, are output from the gate driver 32, and the input pulse input to the gate driver 32 is the alternating current i flowing through the power supply coil 11 1 or is created based on the alternating magnetic field generated in the power supply coil 11, which is the same as the inverter 3 shown in Fig. 5.

[0080] At this time, the switching timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, is determined based on the input pulse to the gate driver 32. Here, the "switching timing" is, as described above, the time to turn on or off the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3. And the alternating current i 1 detected by the current sensor 4 installed between the inverter 3 and the power supply-side resonance circuit 1 is input to the comparator 31 as a signal whose phase has been adjusted by the preamplifier (inverting amplifier circuit) 36 and the phase delay circuit 35, and the above-mentioned input pulse is generated.

[0081] That is, in the case of FIG. 9, the switching timing of the output pulses alternately output from the gate driver 32 to the high-side FET 33 and the low-side FET 34, which are switching elements inside the inverter 3, is based on the current phase or magnetic field phase of the power supply coil 11 detected by the current sensor 4, and is a signal after phase control by feedforward control of an advancing phase or a delaying phase, that is, a signal after the phase relationship has been adjusted by the phase adjustment circuit 30. That is, it is determined based on the input pulse generated by inputting the signal to the comparator 31.

[0082] Here, since the preamplifier 36 is an inverting amplifier circuit, the alternating current i 1When the detection signal is input to the preamplifier 36, as shown in FIG. 10, the phase of the output signal is inverted by 180° with respect to the input signal. FIG. 10 is a schematic diagram showing the voltage waveforms of the respective parts of the phase adjustment circuit 30 (preamplifier 36 and phase delay circuit 35) shown in FIG. 9. Thereafter, at the output of the phase delay circuit (low-pass filter) 35 using an operational amplifier, the phase further lags. For example, when a phase delay of 130° is generated in the phase delay circuit 35, the phase delay including the preamplifier 36 is 180 + 130 = 310°. That is, since it is delayed by one cycle (360°) and the phase is advanced by 360 - 310 = 50°, phase advance can be equivalently realized. That is, the alternating current i 1 is delayed by more than 180° and input to the comparator 31.

[0083] Also, in FIG. 9, as an example of a negative resistance circuit connected to the power supply side resonance circuit 1, the inverter 3 and the current sensor 4 are used for explanation. However, instead of the current sensor 4, a magnetic sensor may be used. Since the current phase and the magnetic field phase coincide, the current may be detected by the current sensor 4, the current phase may be detected, and the alternating current may be detected based on the current phase. Alternatively, the magnetic field may be detected by the magnetic sensor, the magnetic field phase may be detected, and the alternating current may be detected based on the magnetic field phase. That is, as the sensor constituting the negative resistance circuit together with the inverter 3, any sensor that can detect the current or magnetic field of the power supply coil 11 in the power supply side resonance circuit 1 may be used. This also applies to FIGS. 5 and 8 and FIGS. 11 and later described below.

[0084] That is, the circuit shown in FIG. 9 is a device that detects the current or magnetic field of the power supply coil 11 for the oscillation of the inverter 3 connected to the power supply side resonance circuit 1, detects the zero cross of the voltage or current of the power supply coil 11 based on the current or magnetic field, and self-excites. And the resonance frequency ω of loop II hFor the purpose of selectively fixing the resonance loop having it (the "resonance loop of Loop II"), in order to advance the phase of the voltage with respect to the current of the power supply coil 11, the alternating current i detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 1 Before inputting it into the comparator 31 in the inverter 3, a preamplifier (inverting amplifier circuit) 36 and a phase delay circuit 35 are inserted and delayed by more than 180°, so as to control it as an advancing phase. That is, it rotates the delay phase by 180° or more to realize the advancing phase. As a result, it is possible to solve the problem that the advancing phase control is vulnerable to noise and the operation becomes unstable.

[0085] In this way, the power supply side resonance circuit 1 is connected with the inverter 3 and a sensor (current sensor 4 in FIG. 9) for detecting the current or magnetic field of the power supply coil 11. Based on the current phase (or the magnetic field phase based on the magnetic field of the power supply coil 11) detected by the sensor, the inverter 3 performs phase control by feedback control of the advancing phase or the delay phase. That is, the power supply side resonance circuit 1 is provided with a phase adjustment circuit 30 that can adjust the phase relationship between the alternating voltage applied to the power supply side resonance circuit 1 and the alternating current flowing through the power supply coil 11. The switching timing, which is the time to turn on or off the switching element inside the inverter 3, is determined based on the pulse generated by the signal after the phase relationship is adjusted by the phase adjustment circuit 30, whereby one of the two resonance loops is selectively fixed.

[0086] Also, in the case of the embodiment (circuit) shown in FIG. 9, a comparator 31 is provided inside the inverter 3, and a phase delay circuit 35 as a phase adjustment circuit 30 is installed before the input to the comparator 31. Then, the inverter 3 adjusts the phase relationship as an advancing phase that advances the phase of the AC voltage applied to the power supply-side resonance circuit 1 with respect to the AC current flowing through the power supply coil 11 by delaying the AC current by more than 180° and inputting it to the comparator 31 before the AC current detected by the sensor (current sensor 4 in FIG. 9) based on the current or magnetic field of the power supply coil 11 is input to the comparator 31 for the purpose that the phase delay circuit 35 selects the "resonance loop of loop II" among the two resonance loops.

[0087] In the embodiment (circuit) shown in FIG. 9, as a specific example of the phase adjustment circuit 30, a low-pass filter using an operational amplifier is used as the phase delay circuit 35. However, if a large phase delay of about 130° is generated by the low-pass filter, the gain (output amplitude / input amplitude) of the low-pass filter becomes too small, and it is also assumed that the amplitude of the output signal becomes too small. In that case, there is a method of using a time delay circuit instead of the low-pass filter.

[0088] FIG. 11 is a circuit diagram showing an example of an inverter having a time delay circuit and a power supply-side resonance circuit in Embodiment 1 of the present invention. As shown in FIG. 11, an inverter 3 is connected to the power supply-side resonance circuit 1. Regarding the elements constituting the inside of the inverter 3, namely, a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34, they are the same as those in the circuit diagrams shown in FIGS. 5 and 9.

[0089] However, in the case of the circuit in FIG. 11, that is, another example of the circuit in Embodiment 1 of the present invention, a preamplifier (inverting amplifier circuit) 36 is installed before the input to the comparator 31 provided inside the inverter 3, and a time delay circuit 37 is installed after the output of the comparator 31, and phase control is performed by controlling the delay time. In the case of FIG. 11, the phase adjustment circuit 30 includes at least the time delay circuit 37.

[0090] That is, the circuit shown in FIG. 11 advances the phase of the voltage with respect to the current of the power supply coil 11 for the purpose of selectively fixing a resonance loop (the "resonance loop of Loop II") having a resonance frequency ω h of Loop II. The alternating current i 1 detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is input to the comparator 31 in the inverter 3, and a time delay circuit 37 is inserted after the output of the comparator 31, and phase control is performed by controlling the delay time.

[0091] By connecting in series multiple time delay circuits 37 each formed by combining a NOT-type logic element with a Schmitt trigger and an RC integrating circuit at the subsequent stage of the comparator 31, a large phase delay of about 130° is generated. That is, real-time control by a NOT circuit and an RC integrating circuit is realized for the control of the switch signal input to the comparator 31. As a result, the problem that the phase delay of the analog circuit has frequency dependence can be solved. Note that the logic element does not necessarily have to be provided with a Schmitt trigger. Also, as the logic element, a buffer circuit can be used instead of the NOT type.

[0092] As described above, in the embodiment (circuit) shown in FIG. 11, a comparator 31 is provided inside the inverter 3, and an alternating current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 11) is configured to be input to the comparator 31. Further, a time delay circuit 37 as a phase adjustment circuit 30 is installed after the output of the comparator 31. The inverter 3 performs control to adjust the phase relationship by controlling the delay time by the time delay circuit 37, whereby one of the two resonance loops is selectively fixed.

[0093] In the actual circuit examples of FIGS. 9 and 11, by attaching the current sensor 4 to the connection line of the power supply coil 11 to detect the current of the power supply coil 11 (the current of the power supply side resonance circuit 1), the phase of the alternating current i 1 is detected. The same effect can also be achieved by attaching a magnetic sensor (not shown) near the power supply coil 11 to detect the magnetic field of the power supply coil 11 (the magnetic field of the power supply side resonance circuit 1). As the magnetic sensor, in addition to a Hall element, it may also be a magnetic sensor called a loop coil in which an electric wire is formed in a loop shape. In this case, the inverter 3 and the magnetic sensor constitute the aforementioned negative resistance circuit.

[0094] As described above, in FIG. 9, the phase is controlled by inserting a phase delay circuit (low-pass filter) 35 before the comparator 31, and in FIG. 11, the phase is controlled by inserting a time delay circuit 37 after the comparator 31. As another assumed embodiment, instead of the time delay circuit 37 shown in FIG. 11, a circuit that can control only the phase called an all-pass filter may be used.

[0095] FIG. 12 is a circuit diagram showing an example of an inverter having an all-pass filter and a power supply side resonance circuit in Embodiment 1 of the present invention. As shown in FIG. 12, an inverter 3 is connected to the power supply side resonance circuit 1. Regarding the elements constituting the inside of the inverter 3, namely, a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34, they are the same as those in the circuit diagrams shown in FIGS. 5, 9, and 11.

[0096] However, in the case of the circuit of FIG. 12, that is, yet another example of the circuit in Embodiment 1 of the present invention, a preamplifier (inverting amplifier circuit) 36 is installed before the input to the comparator 31 provided inside the inverter 3, and an all-pass filter 38 is installed after the output of the comparator 31, and phase control is performed by this all-pass filter 38.

[0097] That is, the circuit shown in FIG. 12 is for the purpose of selecting and fixing a resonance loop (the "resonance loop of Loop II") having a resonance frequency ω h of Loop II. In order to advance the phase of the voltage with respect to the current of the power supply coil 11, the alternating current i 1 detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is input to the comparator 31 inside the inverter 3, and an all-pass filter 38 is inserted after the output of the comparator 31, and phase control is performed by this all-pass filter 38.

[0098] Thus, in the embodiment (circuit) shown in FIG. 12, a comparator 31 is provided inside the inverter 3, and the alternating current based on the current of the power supply coil 11 or the magnetic field detected by the sensor (current sensor 4 in FIG. 12) is configured to be input to the comparator 31, and after the output of the comparator 31, an all-pass filter 38 as a phase adjustment circuit 30 is installed. And the inverter 3 performs control to adjust the phase relationship by the all-pass filter 38, whereby one of the two resonance loops is selected and fixed.

[0099] Furthermore, as shown in FIG. 13, a method of controlling the phase using a PLL (phase-locked loop) 39 can also be considered instead of the time delay circuit 37 shown in FIG. 11 and the all-pass filter 38 shown in FIG. 12. FIG. 13 is a circuit diagram showing an example of an inverter having a PLL (phase-locked loop) and a power supply-side resonance circuit in Embodiment 1 of the present invention. As shown in FIG. 13, an inverter 3 is connected to the power supply-side resonance circuit 1. Regarding the elements constituting the inside of the inverter 3, namely, a comparator 31, a gate driver 32, a high-side FET (field effect transistor) 33, and a low-side FET (field effect transistor) 34, they are the same as those shown in the circuit diagrams of FIGS. 5, 9, 11, and 12.

[0100] However, in the case of the circuit of FIG. 13, that is, yet another example of the circuit in Embodiment 1 of the present invention, a preamplifier (inverting amplifier circuit) 36 is installed before the input to the comparator 31 provided inside the inverter 3, and a PLL (phase-locked loop) 39 is installed after the output of the comparator 31, and phase control is performed by this PLL (phase-locked loop) 39.

[0101] That is, the circuit shown in this FIG. 13 is for the purpose of selectively fixing a resonance loop having a resonance frequency ω h of “resonance loop of Loop II”. In order to advance the phase of the voltage with respect to the current of the power supply coil 11, the alternating current i 1 detected by using the current sensor 4 or a magnetic sensor (not shown) of the power supply coil 11 is input to the comparator 31 inside the inverter 3. After the output of the comparator 31, a PLL (phase-locked loop) 39 is inserted, and phase control is performed by controlling the advance and delay of the phase by this PLL (phase-locked loop) 39.

[0102] Thus, in the embodiment (circuit) shown in FIG. 13, a comparator 31 is provided inside the inverter 3, and an alternating current based on the current or magnetic field of the power supply coil 11 detected by a sensor (current sensor 4 in FIG. 13) is configured to be input to the comparator 31. And after the output of the comparator 31, a PLL (phase-locked loop) 39 as a phase adjustment circuit 30 is installed. Then, the inverter 3 performs control to adjust the phase relationship by advancing or delaying the phase of the alternating voltage applied to the power supply side resonance circuit 1 with respect to the alternating current flowing through the power supply coil 11 by the PLL (phase-locked loop) 39. As a result, one of the two resonance loops is selected and fixed.

[0103] FIG. 14 is a block diagram showing an example of the specific internal function of the PLL (phase-locked loop) 39. As shown in FIG. 14, the PLL (phase-locked loop) 39 includes at least a PFD (phase detector) 91 and a VCO (voltage controlled oscillator) 92. And the PFD (phase detector) 91 calculates the phase difference between the input signal serving as the reference frequency and the feedback signal of the output from the VCO (voltage controlled oscillator) 92 whose frequency changes according to the voltage, and inputs the phase difference to the VCO (voltage controlled oscillator) 92, thereby synchronizing the phases of the input signal and the output signal. Therefore, by applying some offset voltage to the input voltage of the VCO (voltage controlled oscillator) 92, the range of the frequency of the output signal of the PLL (phase-locked loop) 39 can be restricted.

[0104] For example, if the resonance frequency ω h of loop II is in the frequency range of 70 kHz to 90 kHz, and the resonance frequency ω l of loop I is in the frequency range of 50 kHz to 70 kHz, then if the frequency range of the output signal of the PLL (phase-locked loop) 39 is set to 70 kHz to 90 kHz, the resonance frequency ω l of loop I will not operate. Therefore, the resonance frequency ω hThe stability of the operation is enhanced. That is, by controlling the input voltage of the VCO (Voltage Controlled Oscillator) 92, restricting the oscillation frequency of the VCO (Voltage Controlled Oscillator) 92, and restricting the frequency range of the output signal of the PLL (Phase Locked Loop) 39, it is possible to enhance the operation stability when either of the two resonance loops is selected.

[0105] Here, a signal obtained by inverting, by a preamplifier (inverting amplifier circuit) 36, an AC signal based on the current detected by the current sensor 4 shown in FIG. 13 is input to the comparator 31, and the input pulse generated in the comparator 31 is the input signal of the PLL (Phase Locked Loop) 39. Therefore, when the output signal of the PLL (Phase Locked Loop) 39 is input to the gate driver 32, compared with the case of positive feedback without phase control, the frequency or phase can be controlled by controlling the input voltage of the VCO (Voltage Controlled Oscillator) 92 (controlling the control value of the VCO 92). Utilizing this, as the resonance frequency of the power supply side resonance circuit 1, the resonance frequency ω of loop II h or the resonance frequency ω of loop I l can be selectively fixed to any one of them. As another method, a limit (restriction) is imposed on the oscillation frequency variation range of the VCO (Voltage Controlled Oscillator) 92 in the PLL (Phase Locked Loop) 39 so that it does not oscillate at all at the resonance frequency ω of loop I h or so that it only oscillates at the resonance frequency ω of loop I l It is also possible to control in such a way.

[0106] In this way, by limiting (restricting) the variation range of the oscillation frequency of the VCO (Voltage Controlled Oscillator) 92 in the PLL (Phase Locked Loop) 39 so that it cannot oscillate at all at the resonance frequency ω of loop I l it is inevitable that only the resonance frequency ω of loop II h is selected, so it is possible to forcibly select and fix it to the resonance frequency ω of loop II h That is, by using the PLL (phase-locked loop) 39 that imposes a frequency limit on the VCO (voltage-controlled oscillator) 92, the output frequency of the PLL (phase-locked loop) 39 can be restricted. As a result, the resonance loop can be fixed in either direction, preventing unstable switching from occurring.

[0107] This can prevent the problem that the current loop randomly switches due to, for example, fluctuations in the positions of the power supply coil and the power receiving coil during wireless power supply. As described above, by adding the PLL (phase-locked loop) 39, not only can phase control be performed by the PLL, but also phase fluctuations and the like can be reduced, enabling stable operation to be expected. Thus, there is also the advantage that the stability of the operation can be improved.

[0108] Furthermore, as shown in FIG. 15, instead of the time delay circuit 37 shown in FIG. 11, the all-pass filter 38 shown in FIG. 12, or the PLL (phase-locked loop) 39 shown in FIG. 13, a method of controlling the phase using both the all-pass filter 38 and the PLL (phase-locked loop) 39 is also conceivable. FIG. 15 is a circuit diagram showing an example of an inverter having an all-pass filter and a PLL (phase-locked loop) and a power supply-side resonance circuit in Embodiment 1 of this invention. As shown in FIG. 15, an inverter 3 is connected to the power supply-side resonance circuit 1. Regarding the elements constituting the inverter 3, namely, the comparator 31, the gate driver 32, the high-side FET (field-effect transistor) 33, and the low-side FET (field-effect transistor) 34, they are the same as those shown in the circuit diagrams of FIGS. 5, 9, 11, 12, and 13.

[0109] FIG. 16 is a graph showing the measured waveforms of the alternating current i 1 and the input voltage v 1 when adjusted so that the input voltage v 1 is in a leading phase with respect to the alternating current i 1 in an actual device in Embodiment 1 of this invention. The dashed line represents the alternating current i 1 and the solid line represents the input voltage v 1is shown. That is, FIG. 16 shows the phase difference between the alternating voltage applied to the power supply side resonance circuit 1 and the alternating current flowing through the power supply coil 11. Further, FIG. 17 is a graph showing the result of measuring the change in the resonance frequency when the distance between the power supply coil 11 provided in the power supply side resonance circuit 1 and the power receiving coil 21 provided in the power receiving side resonance circuit 2, that is, the transmission distance d between these two coils, is changed in the phase relationship of FIG. 16. The ● mark indicates the experimental result, and the broken line indicates the calculation result.

[0110] In the experiment, when the phase difference shown in FIG. 16 is 30 to 35 degrees, that is, when the alternating voltage v 1 leads the alternating current i 1 by 30 to 35 degrees in phase, as shown in FIG. 17, it was confirmed that the resonance frequency ω h of loop II is stably selected and fixed. Looking at FIG. 17, it can be seen that at any transmission distance, that is, even when the transmission distance d changes, the resonance loop of the resonance frequency ω h of loop II is always selected and fixed.

[0111] FIG. 18 is a graph showing the result of measuring the transmission power when the transmission distance d between the two coils is changed. The ● mark indicates the experimental result, and the broken line indicates the calculation result. Looking at FIG. 18, it can be confirmed that power supply is always possible at any transmission distance, that is, even when the transmission distance d changes.

[0112] As described above, according to this invention, in a magnetic resonance type wireless power feeding device that performs non-contact power feeding by using PT symmetry to magnetically resonate a power feeding coil and a power receiving coil, by providing a phase adjustment circuit for adjusting the phase, it becomes possible to select and fix a resonance loop without being affected by the surrounding environment and without imposing restrictions on the design of the power feeding coil and the power receiving coil. That is, without being affected by the surrounding environment and without being affected by the selection of a frequency suitable for wireless power feeding, the core shape and material of the coil, etc. (no adjustment by the coil design is required), without imposing restrictions on the design of the power feeding coil and the power receiving coil, it is possible to select and fix any one of the two resonance loops, and the coil can have a free shape and dimensions.

[0113] Note that within the scope of the invention of this application, it is possible to modify any component of the embodiment or omit any component of the embodiment.

Industrial Applicability

[0114] The magnetic resonance type wireless power feeding device of this invention can be widely applied to wireless power feeding in a wide variety of environments, such as not only short-distance wireless power feeding like charging a mobile phone, but also environments with water around like an underwater drone, and environments with a lot of metal around like power transmission to equipment in a factory.

Explanation of Reference Numerals

[0115] 1 Power feeding side resonance circuit 2 Power receiving side resonance circuit 3 Inverter 4 Current sensor 11 Power feeding coil 21 Power receiving coil 30 Phase adjustment circuit 31 Comparator 32 Gate driver 33 High-side FET 34 Low-side FET 35 Phase delay circuit 36 Preamplifier (inverting amplifier circuit) 37-time delay circuit 38 all-pass filter 39 PLL (phase-locked loop) 91 PFD (phase detector) 92 VCO (voltage-controlled oscillator)

Claims

1. A magnetic resonance wireless power supply device that includes a power supply side resonance circuit including a power supply coil and a power reception side resonance circuit including a power reception coil, utilizes parity-time symmetry, and performs non-contact power supply by magnetically resonating the power supply coil and the power reception coil, when the power supply side resonance circuit and the power reception side resonance circuit are regarded as a complex resonance circuit coupled to each other by mutual inductance, there are two resonance loops (the resonance loop of loop I and the resonance loop of loop II) through which the resonance current flowing through the complex resonance circuit can circulate, a sensor for detecting the current or magnetic field of the power supply coil is connected to the power supply side resonance circuit, the inverter includes a phase adjustment circuit that can adjust the phase relationship between the alternating current voltage applied to the power supply side resonance circuit and the alternating current flowing through the power supply coil based on the current phase based on the current of the power supply coil detected by the sensor or the magnetic field phase based on the magnetic field of the power supply coil, the switching timing, which is the time to turn on or off the switching element inside the inverter, is determined based on a pulse generated by a signal after the phase relationship is adjusted by the phase adjustment circuit, so that any one of the two resonance loops is selected and fixed A magnetic resonance wireless power supply device characterized by this.

2. A comparator is provided inside the inverter, and a phase delay circuit as the phase adjustment circuit is installed before the input to the comparator, the inverter performs control to adjust the phase relationship as an advanced phase that advances the phase of the alternating current voltage applied to the power supply side resonance circuit with respect to the alternating current flowing through the power supply coil by delaying the alternating current by more than 180° and inputting it to the comparator before the alternating current based on the current or magnetic field of the power supply coil detected by the sensor is input to the comparator for the purpose of selecting the "resonance loop of loop II" among the two resonance loops by the phase delay circuit The magnetic resonance wireless power supply device according to claim 1, characterized by this.

3. A comparator is provided inside the inverter, and an alternating current based on the current or magnetic field of the power supply coil detected by the sensor is configured to be input to the comparator. And a time delay circuit as the phase adjustment circuit is installed after the output of the comparator. The inverter performs control to adjust the phase relationship by controlling the delay time by the time delay circuit. The magnetic resonance type wireless power supply device according to claim 1, characterized in that.

4. A comparator is provided inside the inverter, and an alternating current based on the current or magnetic field of the power supply coil detected by the sensor is configured to be input to the comparator. And an all-pass filter as the phase adjustment circuit is installed after the output of the comparator. The inverter performs control to adjust the phase relationship by the all-pass filter. The magnetic resonance type wireless power supply device according to claim 1, characterized in that.

5. A comparator is provided inside the inverter, and an alternating current based on the current or magnetic field of the power supply coil detected by the sensor is configured to be input to the comparator. And a PLL (phase-locked loop) as the phase adjustment circuit is installed after the output of the comparator. The inverter performs control to adjust the phase relationship by advancing or delaying the phase of the alternating voltage applied to the power supply side resonance circuit with respect to the alternating current flowing through the power supply coil by the PLL (phase-locked loop). The magnetic resonance type wireless power supply device according to claim 1, characterized in that.

6. The PLL (phase-locked loop) includes at least a VCO (voltage controlled oscillator), and by controlling the input voltage of the VCO (voltage controlled oscillator), the oscillation frequency of the VCO (voltage controlled oscillator) is restricted, and the frequency range of the output signal of the PLL (phase-locked loop) is restricted. The magnetic resonance type wireless power supply device according to claim 5, characterized in that.

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