Composite attenuator

Abstract

The present invention provides a composite attenuator capable of applying a desired level of attenuation to a transmission signal in response to temperature changes, regardless of the operation state of a switching attenuator.　The present invention provides a composite attenuator (A) in which a switching attenuator (5), which switches an attenuation level on or off, and a variable attenuator (4), which allows continuous adjustment of the attenuation level are connected in series, wherein the composite attenuator (A) is provided with a switching control unit (7) that generates a switching signal (Vc) for setting the attenuation level of the switching attenuator (5), and a variable control unit (6) that generates a variable signal (Vm) for setting the attenuation level of the variable attenuator (4), and wherein the variable control unit (6) corrects the variable signal (Vm) in accordance with the switching signal (Vc).

Description

Composite Attenuator

[0001] This application claims priority from Japanese Patent Application No. 2024-004524, filed on Jan. 16, 2024, the contents of which are incorporated herein by reference.

[0002] Patent Document 1 listed below discloses a high-frequency amplifier circuit including a variable attenuator, a bandgap reference current source circuit, and a variable attenuator control voltage generation circuit. In this high-frequency amplifier circuit, the bandgap reference current source circuit outputs a temperature-proportional current and a temperature-fixed current, the variable attenuator control voltage generation circuit outputs a control voltage that changes with a predetermined slope based on the temperature-proportional current and the temperature-fixed current, and the variable attenuator controls the attenuation based on the control voltage, thereby suppressing gain changes due to temperature fluctuations.

[0003] In other words, this high-frequency amplifier circuit is configured so that the attenuation of the variable attenuator increases when the temperature is low by applying a control voltage that exhibits a negative slope with respect to temperature to the gate of the transistor that constitutes the variable attenuator. Since amplifiers generally have a large gain when the temperature is low, by combining the above-mentioned variable attenuator with the amplifier, it is possible to control the gain of the high-frequency amplifier circuit to a constant level and maintain a constant output power level.

[0004] Japanese Patent No. 5765203

[0005] When a relatively large change in attenuation is required, a switching attenuator that switches between providing and not providing attenuation may be used in combination with a variable attenuator that continuously varies the attenuation. In this case, the switching attenuator is cascaded to the variable attenuator in the signal transmission path.

[0006] However, when the transistors constituting the switching attenuator are turned on and the switching attenuator operates as an attenuator, the attenuation of the switching attenuator still has a slight temperature dependency. Therefore, in a composite attenuator in which a switching attenuator is cascaded with a variable attenuator, there is a problem that the way in which the attenuation changes with temperature varies depending on the operating state of the switching attenuator.

[0007] The present invention has been made in view of the above circumstances, and has as its object to provide a composite attenuator that can apply a desired amount of attenuation to a transmission signal in response to temperature changes, regardless of the operating state of a switchable attenuator.

[0008] In order to achieve the above object, a first aspect of the composite attenuator according to the present invention is a composite attenuator in which a switchable attenuator that switches between providing and not providing an attenuation amount and a variable attenuator that can continuously change the attenuation amount are connected in cascade, and the composite attenuator comprises a switch control unit that generates a switch signal that sets the attenuation amount of the switchable attenuator, and a variable control unit that generates a variable signal that sets the attenuation amount of the variable attenuator, and the variable control unit corrects the variable signal in response to the switch signal.

[0009] A second aspect of the composite attenuator according to the present invention is the first aspect, wherein the variable attenuator and the switching attenuator each comprise an input terminal, an output terminal, a transmission path, and a transistor shunt-connected to the transmission path, and the transistor may be controlled by the variable signal.

[0010] A third aspect of the composite attenuator according to the present invention is the above-mentioned second aspect, wherein the variable attenuator and the switchable attenuator may include an impedance matching circuit between the input terminal and the transistor or between the output terminal and the transistor, or both.

[0011] A fourth aspect of the composite attenuator according to the present invention is the composite attenuator of any one of the first to third aspects, wherein the variable control section includes a temperature compensation current mirror circuit, and generates the variable signal so as to suppress fluctuations in the attenuation amount of the variable attenuator due to temperature fluctuations.

[0012] A fifth aspect of the composite attenuator according to the present invention is the composite attenuator of any one of the first to fourth aspects, wherein the variable control section may include a current-voltage conversion circuit, an instrumentation amplifier, and a reference voltage generation circuit.

[0013] A sixth aspect of the composite attenuator according to the present invention is the fourth or fifth aspect, wherein resistors having different temperature characteristic slopes are connected to the input and output sides of the temperature compensation current mirror circuit.

[0014] A seventh aspect of the composite attenuator according to the present invention is any one of the fourth to sixth aspects, wherein the temperature compensating current mirror circuit has a folded current mirror circuit cascade-connected to its input and output sides, each of which is coupled with resistors having different slopes of temperature characteristics.

[0015] An eighth aspect of the composite attenuator according to the present invention is the composite attenuator of any one of the fourth to seventh aspects, wherein the temperature compensating current mirror circuit comprises a first current mirror circuit whose output current has a negative slope with respect to temperature, and a second current mirror circuit whose output current has a positive slope with respect to temperature.

[0016] According to the present invention, it is possible to provide a composite attenuator that can apply a desired amount of attenuation to a transmission signal in response to temperature changes, regardless of the operating state of a switchable attenuator.

[0017] FIG. 1 is a block diagram showing the overall configuration of a composite attenuator according to a first embodiment of the present invention. FIG. 2 is a circuit diagram showing the circuit configuration of a variable attenuator and a switchable attenuator according to the first embodiment of the present invention. FIG. 3 is a circuit diagram showing the circuit configuration of another variable attenuator and a switchable attenuator according to the first embodiment of the present invention. FIG. 4 is a circuit diagram showing the circuit configuration of another variable attenuator and a switchable attenuator according to the first embodiment of the present invention. FIG. 5 is a circuit diagram showing the circuit configuration of another variable attenuator and a switchable attenuator according to the first embodiment of the present invention. FIG. 6 is a circuit diagram showing the circuit configuration of another variable attenuator and a switchable attenuator according to the first embodiment of the present invention. FIG. 7 is a circuit diagram showing the circuit configuration of another variable attenuator and a switchable attenuator according to the first embodiment of the present invention. FIG. 8 is a circuit diagram showing the configuration of a temperature compensating current mirror circuit according to the first embodiment of the present invention. FIG. 9 is a block diagram showing a detailed configuration of a variable control section according to the first embodiment of the present invention. FIG. 10 is a circuit diagram of an instrumentation amplifier showing a detailed configuration of a variable control section according to the first embodiment of the present invention. FIG. 11 is a characteristic diagram showing the temperature characteristic of a control voltage according to the first embodiment of the present invention. Fig. 1 is a characteristic diagram showing the attenuation of a variable attenuator and a switchable attenuator in the first embodiment of the present invention. Fig. 2 is a characteristic diagram showing the attenuation of another variable attenuator and a switchable attenuator in the first embodiment of the present invention. Fig. 3 is a block diagram showing the overall configuration of a composite attenuator in a second embodiment of the present invention. Fig. 4 is a characteristic diagram showing the temperature characteristics of a control voltage in the second embodiment of the present invention. Fig. 5 is a characteristic diagram showing the temperature characteristics of a control voltage in the second embodiment of the present invention. Fig. 6 is a characteristic diagram showing the difference between the attenuation in the second embodiment of the present invention and the attenuation of a comparative example.

[0018] Hereinafter, first and second embodiments of the present invention will be described with reference to the drawings. [First Embodiment] First, the first embodiment will be described with reference to Figs. 1 to 6C. As shown in Fig. 1, a composite attenuator A according to the first embodiment includes an RF input terminal 1, an RF output terminal 2, a transmission line 3 (transmission path), a variable attenuator 4, a switchable attenuator 5, a variable control unit 6, and a switch control unit 7. Of these components, the variable control unit 6 includes a temperature compensation current mirror circuit 6a and a control voltage generation unit 6b.

[0019] The RF input terminal 1 is a connection terminal that receives a high-frequency signal So (RF signal) of a predetermined frequency (predetermined wavelength) from the outside. That is, the RF input terminal 1 is connected to a supply source of the high-frequency signal So outside the composite attenuator A, and is also connected to an input terminal of the variable attenuator 4 inside the composite attenuator A. The high-frequency signal So is a transmission signal in the composite attenuator A.

[0020] The RF output terminal 2 is a connection terminal for externally transferring the high frequency signal So attenuated by the composite attenuator A. That is, the RF output terminal 2 is connected to a supply destination of the high frequency signal So outside the composite attenuator A, and is also connected to an output terminal of the switchable attenuator 5 inside the composite attenuator A.

[0021] The transmission line 3 is provided to connect the RF input terminal 1 and the RF output terminal 2, and the high frequency signal So is transmitted from the RF input terminal 1 to the RF output terminal 2. This transmission line 3 is a high frequency signal line provided to transmit the high frequency signal So from the RF input terminal 1 to the RF output terminal 2 with as little loss as possible. The transmission line 3, through which the high frequency signal So is transmitted as a transmission signal, corresponds to the transmission path in the present invention.

[0022] The variable attenuator 4 is a high-frequency attenuation circuit whose attenuation can be continuously changed, and is equipped with an input terminal, an output terminal, and a control terminal. The input terminal of the variable attenuator 4 is connected to the RF input terminal 1, and the output terminal is connected to the input terminal of the switching attenuator 5. The control terminal of the variable attenuator 4 is connected to the output terminal of the control voltage generator 6b.

[0023] In this variable attenuator 4, the attenuation amount (variable attenuation amount) applied to the high frequency signal So input from the RF input terminal 1 is continuously variably set based on the control voltage Vm input from the control voltage generator 6b. That is, in this variable attenuator 4, the high frequency signal So input from the RF input terminal 1 is attenuated by the variable attenuation amount and output to the switchable attenuator 5.

[0024] The switching attenuator 5 is a high-frequency attenuation circuit that switches between the presence and absence of attenuation, and is equipped with an input terminal, an output terminal, and a control terminal. The input terminal of the switching attenuator 5 is connected to the output terminal of the variable attenuator 4, and the output terminal is connected to the RF output terminal 2. The control terminal of the switching attenuator 5 is also connected to the output terminal of the switching control unit 7 and the correction terminal of the control voltage generation unit 6b.

[0025] In such a switching attenuator 5, the presence or absence of an attenuation amount (switching attenuation amount) to be applied to the high frequency signal So input from the variable attenuator 4 is set based on the switching signal Vc input from the switching control unit 7. That is, this switching attenuator 5 switches between a state in which the high frequency signal So input from the variable attenuator 4 is applied with the switching attenuation amount and output to the RF output terminal 2, and a state in which the high frequency signal So is output to the RF output terminal 2 without being applied with the switching attenuation amount.

[0026] Here, the variable attenuator 4 and the switching attenuator 5 in this embodiment may have either the same circuit configuration or different circuit configurations, and the attenuation amount may be changed continuously or with or without being changed depending on the control signals Vctl1 and Vctl2. Figures 2A to 2G show examples of circuit configurations of the variable attenuator 4 and the switching attenuator 5 that adjust the attenuation amount continuously or with or without being changed based on the control signals Vctl1 and Vctl2, but the same or different circuit configurations from these examples of circuit configurations may be used for the variable attenuator 4 and the switching attenuator 5.

[0027] 2A includes a transmission line g1 provided between an input terminal RFin and an output terminal RFout, and a transistor g2 such as an N-channel MOS field effect transistor, which is provided between the transmission line g1 and ground potential (GND).

[0028] That is, the drain terminal of this transistor g2 is connected to the transmission line g1, the source terminal is connected to the ground potential (GND), and the control signal Vctl1 is input to the gate terminal of this transistor g2. In this variable attenuator 4 and switching attenuator 5, the resistance value (DS resistance value) between the drain terminal and the source terminal of the transistor g2 is adjusted by the control signal Vctl1, thereby setting the variable attenuation or the switching attenuation.

[0029] The variable attenuator 4 and the switchable attenuator 5 shown in Fig. 2B are obtained by adding a resistor g3 having a predetermined resistance value to the variable attenuator 4 and the switchable attenuator 5 shown in Fig. 2A. That is, the first end of the resistor g3 is connected to the source terminal of the transistor g2, and the second end is connected to the ground potential (GND).

[0030] In the variable attenuator 4 and the switching attenuator 5, the resistance obtained by adding the resistance of the resistor g3 to the DS resistance of the transistor g2 acts between the transmission line g1 and the ground potential (GND). The variable attenuation or switching attenuation of the variable attenuator 4 or the switching attenuator 5 is set by adjusting the DS resistance using the control signal Vctl1.

[0031] 2C, the variable attenuator 4 and the switchable attenuator 5 are provided with a second transistor g4 between the transmission line g1 and the ground potential (GND) in addition to the transistor g2 (first transistor). The second transistor g4 is, for example, a P-channel MOS field effect transistor, and is connected in parallel to the first transistor g2.

[0032] That is, the second transistor g4 has a source terminal connected to the transmission line g1, a drain terminal connected to the ground potential (GND), and a gate terminal to which a control signal Vctl2 is input. This control signal Vctl2 is a voltage complementary to the control signal Vctl1 applied to the first transistor g2.

[0033] In such variable attenuator 4 and switchable attenuator 5, the first transistor g2 (N-channel MOS field effect transistor) and the second transistor g4 (P-channel MOS field effect transistor) exhibit complementary behavior with respect to nonlinearity caused by amplitude fluctuations of the transmission signal, and the nonlinearity can be substantially canceled. In other words, the variable attenuator 4 and switchable attenuator 5 of Figure 2C can improve the linearity in the attenuation process of the transmission signal.

[0034] 2D, the variable attenuator 4 and the switchable attenuator 5 have a third transistor g5 (N-channel MOS field effect transistor) provided between a first transistor g2 (N-channel MOS field effect transistor) and ground potential (GND). That is, the source terminal of the first transistor g2 is connected to the drain terminal of the third transistor g5.

[0035] The third transistor g5 has a source terminal connected to ground potential (GND) and a gate terminal to which the control signal Vctl1 is applied, similar to the gate terminal of the first transistor g2. In the variable attenuator 4 and the switchable attenuator 5, the first transistor g2 and the third transistor g5 are connected in series between the transmission line g1 and ground potential (GND).

[0036] Therefore, the voltage-divided transmission signal is applied to the first transistor g2 and the third transistor g5, and the voltage of the transmission signal applied to the first transistor g2 and the third transistor g5 decreases. As a result, the variable attenuator 4 and the switchable attenuator 5 shown in Figure 2D can improve the linearity of the attenuation process of the transmission signal.

[0037] 2E corresponds to a pair of transmission lines g6 and g7. The pair of transmission lines g6 and g7 is provided as a balanced transmission line (differential path) and transmits a transmission signal in the form of a differential signal from input terminals RFinm and RFinp to output terminals RFoutm and RFoutp.

[0038] That is, of the pair of transmission lines g6 and g7, the first transmission line g6 transmits one transmission signal from the input terminal RFinm to the output terminal RFoutm, and the second transmission line g7 transmits the other transmission signal from the input terminal RFinp to the output terminal RFoutp.

[0039] A first transistor g2 (N-channel MOS field effect transistor) is provided between the first transmission line g6 and the ground potential (GND), and a fourth transistor g8 (N-channel MOS field effect transistor) is provided between the second transmission line g7 and the ground potential (GND).

[0040] That is, the fourth transistor g8 has a drain terminal connected to the second transmission line g7, a source terminal connected to ground potential (GND), and a gate terminal to which the control signal Vctl1 is applied. The variable attenuator 4 and switchable attenuator 5 shown in Figure 2E are provided with the first transistor g2 and the fourth transistor g8 corresponding to the pair of transmission lines g6 and g7, so that an appropriate amount of attenuation can be applied to a transmission signal in the form of a differential signal.

[0041] The variable attenuator 4 and the switchable attenuator 5 shown in FIG. 2F are obtained by adding a fifth transistor g9 (N-channel MOS field effect transistor) and a sixth transistor g10 (N-channel MOS field effect transistor) to the variable attenuator 4 and the switchable attenuator 5 shown in FIG. 2A.

[0042] The fifth transistor g9 is provided in the middle of the transmission line g1 between the connection point with the first transistor g2 and the output terminal RFout, and the sixth transistor g10 is provided in the middle of the transmission line g1 between the output side (the output terminal RFout side) of the fifth transistor g9 and ground potential (GND).

[0043] That is, the fifth transistor g9 has a drain terminal connected to the input terminal RFin and the drain terminal of the first transistor g2, a source terminal connected to the output terminal RFout and the drain terminal of the sixth transistor g10, and a gate terminal to which the control signal Vctl2 is applied.

[0044] On the other hand, the sixth transistor g10 has a drain terminal connected to the source terminal of the fifth transistor g9 and the output terminal RFout, a source terminal connected to the ground potential (GND), and a gate terminal to which the control signal Vctl1 is applied, similar to the gate terminal of the first transistor g2.

[0045] The variable attenuator 4 and the switching attenuator 5 constitute a so-called π-type attenuator by the first transistor g2 and the sixth transistor g10 which are shunt-connected to the transmission line g1, and the fifth transistor g9 which is connected in series to the transmission line g1.

[0046] 2F, the control signal Vctl1 applied to the gate terminals of the first transistor g2 and the sixth transistor g10 and the control signal Vctl2 applied to the gate terminal of the fifth transistor g9 are set to complementary voltages, thereby making it possible to match the input and output impedances of the transmission line g1. Note that, for the purpose of matching the input and output impedances, a T-type attenuator may be used instead of the above-mentioned π-type attenuator.

[0047] The variable attenuator 4 and the switching attenuator 5 shown in Fig. 2G have a pair of matching circuits g11 and g12 added to the variable attenuator 4 and the switching attenuator 5 shown in Fig. 2A. That is, the variable attenuator 4 and the switching attenuator 5 have impedance matching circuits both between the input terminal RFin and the first transistor g2 and between the output terminal RFout and the first transistor g2.

[0048] Note that one of the pair of matching circuits g11 and g12 may be omitted (deleted) as necessary. That is, the variable attenuator 4 and the switchable attenuator 5 in the first embodiment are provided with an impedance matching circuit between the input terminal RFin and the first transistor g2 (transistor) or between the output terminal RFout and the first transistor g2 (transistor), or both.

[0049] Of the pair of matching circuits g11 and g12, the first matching circuit g11 is provided midway along the transmission line g1, between the input terminal RFin and the connection point of the first transistor g2, while the second matching circuit g12 is provided midway along the transmission line g1, between the connection point of the first transistor g2 and the output terminal RFout.

[0050] The first matching circuit g11 is an input impedance matching circuit that matches the input impedance of the variable attenuator 4 and the switchable attenuator 5. On the other hand, the second matching circuit g12 is an output impedance matching circuit that matches the output impedance of the variable attenuator 4 and the switchable attenuator 5.

[0051] The variable attenuator 4 and switchable attenuator 5 shown in FIG. 2G are provided with a pair of matching circuits g11 and g12, which suppresses reflection of the transmission signal at the input and output, thereby suppressing the influence on other circuits connected to the input and output sides of the variable attenuator 4 and switchable attenuator 5.

[0052] Here, the variable attenuator 4 and the switching attenuator 5 in the first embodiment may be configured by combining multiple circuits of the types shown in Figures 2A to 2G as necessary. Also, the variable attenuator 4 and the switching attenuator 5 in the first embodiment may be configured by cascading multiple basic attenuators each having the circuit configuration shown in Figures 2A to 2G.

[0053] When a plurality of basic attenuators are connected in cascade to form the variable attenuator 4 and the switching attenuator 5, the transistors in each basic attenuator may have different sizes. For example, if the sizes of the transistors have a relationship of 2n powers, the attenuation can be adjusted efficiently.

[0054] As described above, the variable control unit 6 includes the temperature compensation current mirror circuit 6a and the control voltage generation unit 6b, and is a first control signal generation unit that generates, as a first control signal, the control voltage Vm that sets the variable attenuation amount of the variable attenuator 4. The variable control unit 6 corrects the variable attenuation amount of the variable attenuator 4 based on the switching signal Vc input from the switching control unit 7 to a correction terminal. That is, the variable control unit 6 corrects the control voltage Vm based on the switching signal Vc, thereby applying a desired variable attenuation amount to the high-frequency signal So (transmission signal) regardless of the operating state of the switching attenuator 5.

[0055] The temperature compensation current mirror circuit 6a is a current generating circuit having a pair of output terminals (first and second output terminals). The first output terminal of the temperature compensation current mirror circuit 6a is connected to the first input terminal of the control voltage generating unit 6b, and the second output terminal of the temperature compensation current mirror circuit 6a is connected to the second input terminal of the control voltage generating unit 6b.

[0056] The temperature compensation current mirror circuit 6a generates, as output currents, a first temperature compensation current I1 and a second temperature compensation current I2 that suppress fluctuations in the variable attenuation of the variable attenuator 4 due to fluctuations in the ambient temperature T. That is, the temperature compensation current mirror circuit 6a generates the first temperature compensation current I1 (first output current) whose current value decreases as the ambient temperature T increases, and the second temperature compensation current I2 (second output current) whose current value remains constant even when the ambient temperature T fluctuates, and outputs the first temperature compensation current I1 and the second temperature compensation current I2 to the control voltage generating unit 6b.

[0057] That is, the first temperature-compensated current I1 is a current that has a negative gradient with respect to changes in the ambient temperature T. In contrast, the second temperature-compensated current I2 has a characteristic that its current value does not change even when the ambient temperature T changes. Note that the second temperature-compensated current I2 may have a positive gradient with respect to changes in the ambient temperature T, if necessary.

[0058] Such a temperature-compensating current mirror circuit 6a has the circuit configurations shown in Figures 3A and 3B. Figure 3A shows a circuit configuration of a type in which the first temperature-compensated current I1 or the second temperature-compensated current I2 is output to the control voltage generating unit 6b, while Figure 3B shows a circuit configuration of a type in which the first temperature-compensated current I1 or the second temperature-compensated current I2 flows in from the control voltage generating unit 6b.

[0059] The temperature compensation current mirror circuit 6a includes a pair of current mirror circuits 8 and 9 corresponding to the first temperature compensation current I1 and the second temperature compensation current I2. The pair of current mirror circuits 8 and 9 have the circuit configuration shown in FIG. 3A or 3B. When the circuit configuration shown in FIG. 3A is used, the first current mirror circuit 8 generates the first temperature compensation current I1 and outputs it to the first input terminal of the control voltage generator 6b. The second current mirror circuit 9 generates the second temperature compensation current I2 and outputs it to the second input terminal of the control voltage generator 6b.

[0060] 3A will be described first. The first current mirror circuit 8 includes a first reference terminal 8a, a pair of transistors 8b and 8c (first and second transistors), a pair of resistors 8d and 8e (first and second resistors), and a first output terminal 8f. The first current mirror circuit 8 has a first mirror ratio set by the pair of resistors 8d and 8e.

[0061] 3A, the first reference current source that outputs the first reference current Iref1 is omitted for convenience sake. That is, the first current mirror circuit 8 includes the first reference terminal 8a, a pair of transistors 8b and 8c, a pair of resistors 8d and 8e, and a first output terminal 8f shown in FIG. 3A, as well as the first reference current source.

[0062] The first reference terminal 8 a is connected to the drain and gate terminals of the first transistor 8 b, the gate terminal of the second transistor 8 c, and a first reference current source (not shown). A first reference current Iref1 is input to the first reference terminal 8 a from the first reference current source.

[0063] As shown in the figure, the first transistor 8b is a P-channel MOS field effect transistor. The drain terminal of the first transistor 8b is connected to its own gate terminal, the gate terminal of the second transistor 8c, and the first reference terminal 8a. The source terminal of the first transistor 8b is connected to one end of the first resistor 8d.

[0064] Furthermore, the gate terminal of the first transistor 8b is connected to its own drain terminal, the gate terminal of the second transistor 8c, and the first reference terminal 8a. That is, the first transistor 8b is provided in a diode-connected state with its drain terminal and gate terminal connected to each other. The first transistor 8b operates in the active region by the first reference current Iref1, and its source current is set to a current value substantially equal to the first reference current Iref1.

[0065] The second transistor 8c has a drain terminal connected to the first output terminal 8f and a source terminal connected to the first end of the second resistor 8e. The second transistor 8c also has a gate terminal connected to the gate and drain terminals of the first transistor 8b and to the first reference current source via the first reference terminal 8a. The second transistor 8c operates in the active region with the first reference current Iref1, and the source current is set to a current value substantially equal to the first reference current Iref1.

[0066] The first resistor 8d is a two-terminal element having a first resistance R1(T) that depends on the ambient temperature T. This first resistance R1 has a temperature characteristic in which the resistance decreases as the ambient temperature T increases, i.e., a negative slope. A first end of the first resistor 8d is connected to the source terminal of the first transistor 8b, and a second end of the first resistor 8d is connected to the DC power supply Vcc. When the first transistor 8b operates in the active region based on the first reference current Iref1, a first voltage drop corresponding to the first reference current Iref1 is generated across the first resistor 8d.

[0067] The second resistor 8e is a two-terminal element having a second resistance R2(T) that depends on the ambient temperature T. This second resistance R2 has a temperature characteristic in which the resistance increases as the ambient temperature T increases, i.e., a positive slope. A first end of the second resistor 8e is connected to the source terminal of the second transistor 8c, and a second end of the second resistor 8e is connected to the DC power supply Vcc. That is, in the first current mirror circuit 8, the first resistor 8d and the second resistor 8e, which have different slopes of the temperature characteristics, are connected to the input and output sides.

[0068] The second transistor 8c operates in the active region based on the first reference current, so that the first mirror current, i.e., the first temperature-compensated current I1, corresponding to the first reference current Iref1 and the first mirror ratio, flows through the second resistor 8e, and a first voltage drop corresponding to the first mirror current (first temperature-compensated current I1) occurs across the second resistor 8e.

[0069] The first output terminal 8f is a first output terminal of the first current mirror circuit 8. As shown in the figure, the first output terminal 8f is connected to the drain terminal of the second transistor 8c inside the first current mirror circuit 8, and is connected to the first input terminal of the control voltage generating unit 6b outside the first current mirror circuit 8.

[0070] Here, the drain current of the second transistor 8c is a first mirror current that is set in accordance with the first reference current Iref1 and the first mirror ratio. That is, the first output terminal 8f outputs the first mirror current to the control voltage generator 6b as the first temperature compensation current I1 of the first current mirror circuit 8.

[0071] In such a first current mirror circuit 8, when the gate-source voltage Vgs1 (Iref1) of the first transistor 8b and the gate-source voltage Vgs2 (I1) of the second transistor 8c are used, the following equation (1) is established: Vgs1 (Iref1) + Iref1·R1(T) = Vgs2 (I1) + I1·R2(T) (1)

[0072] Furthermore, if it is assumed that the gate-source voltage Vgs1 (Iref1) of the first transistor 8b and the gate-source voltage Vgs2 (I1) of the second transistor 8c are equal, the following equation (1) holds: I1 / Iref1=R1(T) / R2(T) (2)

[0073] The first temperature compensation current I1 has a smaller current value as the ambient temperature T increases, because the second resistance R2(T) has a positive gradient with respect to the ambient temperature T, the first resistance R1(T) has a negative gradient with respect to the ambient temperature T, and the first reference current Iref1 has no gradient with respect to the ambient temperature T. In other words, the first temperature compensation current I1 has a negative gradient with respect to changes in the ambient temperature T.

[0074] 3A, the second current mirror circuit 9 includes a second reference terminal 9a, a pair of transistors 9b and 9c (third and fourth transistors), a pair of resistors 9d and 9e (third and fourth resistors), and a second output terminal 9f. The second current mirror circuit 9 has a second mirror ratio set by the pair of resistors 9d and 9e.

[0075] 3A, the second reference current source that outputs the second reference current Iref2 is omitted for convenience of illustration. That is, the second current mirror circuit 9 includes the second reference current source in addition to the second reference terminal 9a, the pair of transistors 9b and 9c, the pair of resistors 9d and 9e, and the second output terminal 9f shown in FIG. 3A.

[0076] The second reference terminal 9 a is connected to the drain and gate terminals of the third transistor 9 b, the gate terminal of the fourth transistor 9 c, and a second reference current source (not shown). A second reference current Iref2 is input to the second reference terminal 9 a from the second reference current source.

[0077] As shown in the figure, the third transistor 9b is a P-channel MOS field effect transistor. The drain terminal of the third transistor 9b is connected to its own gate terminal, the gate terminal of the fourth transistor 9c, and the second reference terminal 9a. The source terminal of the third transistor 9b is connected to the first end of the third resistor 9d.

[0078] Furthermore, the third transistor 9b has its gate terminal connected to its own drain terminal, the gate terminal of the fourth transistor 9c, and the second reference terminal 9a. That is, the third transistor 9b is provided in a diode-connected state with its drain terminal and gate terminal connected to each other. The third transistor 9b operates in the active region by the second reference current Iref2, and its source current is set to a current value substantially equal to the second reference current Iref2.

[0079] The fourth transistor 9c has a drain terminal connected to the second output terminal 9f and a source terminal connected to the first end of the fourth resistor 9e. The gate terminal of the fourth transistor 9c is connected to the gate and drain terminals of the third transistor 9b and to the second reference current source via the second reference terminal 9a. The fourth transistor 9c operates in the active region with the second reference current Iref2, and the source current is set to a current value substantially equal to the second reference current Iref2.

[0080] The third resistor 9d is a two-terminal element having a third resistance R3 that is independent of the ambient temperature T. This third resistance R3 has temperature characteristics such that the resistance does not change as the ambient temperature T increases. A first end of the third resistor 9d is connected to the source terminal of the third transistor 9b, and a second end of the third resistor 9d is connected to the DC power supply Vcc. When the third transistor 9b operates in the active region based on the second reference current Iref2, a third voltage drop corresponding to the second reference current Iref2 is generated across the third resistor 9d.

[0081] The fourth resistor 9e is a two-terminal element having a fourth resistance value R4 that is independent of the ambient temperature T. This fourth resistance value R4 has temperature characteristics such that the resistance value does not change with an increase in the ambient temperature T. A first end of the fourth resistor 9e is connected to the source terminal of the fourth transistor 9c, and a second end of the fourth resistor 9e is connected to the DC power supply Vcc.

[0082] The fourth transistor 9c operates in the active region based on the second reference current Iref2, so that the second mirror current, i.e., the second temperature-compensated current I2, which is determined based on the second reference current Iref2 and the second mirror ratio, flows through the fourth resistor 9e. Also, a fourth voltage drop corresponding to the second mirror current (second temperature-compensated current I2) occurs across the fourth resistor 9e.

[0083] The second output terminal 9f is the second output terminal of the second current mirror circuit 9. As shown in the figure, the second output terminal 9f is connected to the drain terminal of the fourth transistor 9c inside the second current mirror circuit 9, and is connected to the second input terminal of the control voltage generating unit 6b outside the second current mirror circuit 9.

[0084] Here, the drain current of the fourth transistor 9c is a second mirror current that is set in accordance with the second reference current Iref2 and the second mirror ratio. That is, the second output terminal 9f outputs the second mirror current to the control voltage generator 6b as the second temperature compensation current I2 of the second current mirror circuit 9.

[0085] In such a second current mirror circuit 9, when the gate-source voltage Vgs3 (Iref2) of the third transistor 9b and the gate-source voltage Vgs4 (I2) of the fourth transistor 9c are used, the following equation (3) is established: Vgs3 (Iref2) + Iref2·R3(T) = Vgs4 (I2) + I2·R4(T) (3)

[0086] Furthermore, if it is assumed that the gate-source voltage Vgs3 (Iref2) of the third transistor 9b and the gate-source voltage Vgs4 (I2) of the fourth transistor 9c are equal, the following equation (4) holds: I2 / Iref2=R3(T) / R4(T) (4)

[0087] The second temperature compensation current I2 has a constant current value even when the ambient temperature T changes, because the third resistance R3(T) and the fourth resistance R4(T) have no gradient with respect to the ambient temperature T, and the second reference current Iref2 has no gradient with respect to the ambient temperature T. In other words, the second temperature compensation current I2 has no gradient with respect to changes in the ambient temperature T.

[0088] 3B, the first current mirror circuit 8 includes a first reference terminal 8a, a pair of transistors 8b and 8c (first and second transistors), a pair of resistors 8d and 8e (first and second resistors), and a first output terminal 8f. The first current mirror circuit 8 has a first mirror ratio set by the pair of resistors 8d and 8e.

[0089] 3B, the first reference current source that outputs the first reference current Iref1 is omitted for convenience. The first current mirror circuit 8 includes the first reference terminal 8a, a pair of transistors 8b and 8c, a pair of resistors 8d and 8e, and a first output terminal 8f, as well as the first reference current source.

[0090] The first reference terminal 8 a is connected to the drain and gate terminals of the first transistor 8 b, the gate terminal of the second transistor 8 c, and a second reference current source (not shown). A first reference current Iref1 is input to the first reference terminal 8 a from the first reference current source.

[0091] As shown in the figure, the first transistor 8b is an N-channel MOS field effect transistor. The drain terminal of the first transistor 8b is connected to its own gate terminal, the gate terminal of the second transistor 8c, and the first reference terminal 8a. The source terminal of the first transistor 8b is connected to a first end of the first resistor 8d.

[0092] Furthermore, the gate terminal of the first transistor 8b is connected to its own drain terminal, the gate terminal of the second transistor 8c, and the first reference terminal 8a. That is, the first transistor 8b is provided in a diode-connected state with its drain terminal and gate terminal connected to each other. The first transistor 8b operates in the active region by the first reference current Iref1, and its source current is set to a current value substantially equal to the first reference current Iref1.

[0093] The second transistor 8c has a drain terminal connected to the first output terminal 8f, a source terminal connected to the first end of the second resistor 8e, and a gate terminal connected to the gate and drain terminals of the first transistor 8b and the first reference current source. The second transistor 8c operates in the active region with the first reference current Iref1, and the source current is set to a current value substantially equal to the first reference current Iref1.

[0094] The first resistor 8d is a two-terminal element having a first resistance R1(T) that depends on the ambient temperature T. This first resistance R1(T) has a temperature characteristic in which the resistance decreases as the ambient temperature T increases, i.e., a negative slope. A first end of the first resistor 8d is connected to the source terminal of the first transistor 8b, and a second end of the first resistor 8d is connected to ground potential (GND). When the first transistor 8b operates in the active region based on the first reference current Iref1, a first voltage drop corresponding to the first reference current Iref1 is generated across the first resistor 8d.

[0095] The second resistor 8e is a two-terminal element having a second resistance value R2(T) that depends on the ambient temperature T. This second resistance value R2(T) has a temperature characteristic in which the resistance value increases as the ambient temperature T increases, i.e., a positive slope. A first end of the second resistor 8e is connected to the source terminal of the second transistor 8c, and a second end of the second resistor 8e is connected to the ground potential (GND). In this first current mirror circuit 8, the first resistor 8d and the second resistor 8e, whose temperature characteristic slopes are different, are connected to the input and output sides.

[0096] When the second transistor 8c operates in the active region based on the first reference current Iref1, a first mirror current corresponding to the first reference current Iref1 and the first mirror ratio flows through the second resistor 8e, and a second voltage drop corresponding to the first mirror current occurs across the second resistor 8e.

[0097] The first output terminal 8f is a first output terminal of the first current mirror circuit 8. As shown in the figure, the first output terminal 8f is connected to the drain terminal of the second transistor 8c inside the first current mirror circuit 8, and is connected to the first input terminal of the control voltage generating unit 6b outside the first current mirror circuit 8.

[0098] The drain current of the second transistor 8c is a first mirror current that is set in accordance with the first reference current Iref1 and the first mirror ratio and has temperature dependency. The first output terminal 8f outputs this first mirror current as a first temperature compensation current I1 to the control voltage generator 6b.

[0099] 3B, the second current mirror circuit 9 includes a second reference terminal 9a, a pair of transistors 9b and 9c (third and fourth transistors), a pair of resistors 9d and 9e (third and fourth resistors), and a second output terminal 9f. The second current mirror circuit 9 has a second mirror ratio set by the pair of resistors 9d and 9e.

[0100] 3B, the second reference current source that outputs the second reference current Iref2 is omitted for convenience. The second current mirror circuit 9 includes the second reference terminal 9a, a pair of transistors 9b and 9c, a pair of resistors 9d and 9e, and a second output terminal 9f, as well as the second reference current source.

[0101] The second reference terminal 9 a is connected to the drain and gate terminals of the third transistor 9 b, the gate terminal of the fourth transistor 9 c, and a second reference current source (not shown), which inputs a second reference current Iref2 to the reference terminal 9 a.

[0102] As shown in the figure, the third transistor 9b is an N-channel MOS field effect transistor. The drain terminal of the third transistor 9b is connected to its own gate terminal, the gate terminal of the fourth transistor 9c, and the second reference terminal 9a. The source terminal of the third transistor 9b is connected to the first end of the third resistor 9d.

[0103] Furthermore, the third transistor 9b has its gate terminal connected to its own drain terminal, the gate terminal of the fourth transistor 9c, and the second reference terminal 9a. That is, the third transistor 9b is provided in a diode-connected state with its drain terminal and gate terminal connected to each other. The third transistor 9b operates in the active region by the second reference current Iref2, and its source current is set to a current value substantially equal to the second reference current Iref2.

[0104] The fourth transistor 9c has a drain terminal connected to the second output terminal 9f and a source terminal connected to the first end of the fourth resistor 9e. The gate terminal of the fourth transistor 9c is connected to the gate and drain terminals of the third transistor 9b and to the second reference current source via the second reference terminal 9a. The fourth transistor 9c operates in the active region with the second reference current Iref2, and the source current is set to a current value substantially equal to the second reference current Iref2.

[0105] The third resistor 9d is a two-terminal element having a third resistance value R3. This third resistance value R3 is a resistance value that does not have temperature dependence. A first end of the third resistor 9d is connected to the source terminal of the third transistor 9b, and a second end of the third resistor 9d is connected to ground potential (GND). When the third transistor 9b operates in the active region based on the second reference current, a third voltage drop corresponding to the second reference current Iref2 is generated across the third resistor 9d.

[0106] The fourth resistor 9e is a two-terminal element having a fourth resistance value R4. This fourth resistance value R4 is a resistance value that does not have temperature dependency. A first terminal of the fourth resistor 9e is connected to the source terminal of the fourth transistor 9c, and a second terminal of the fourth resistor 9e is connected to ground potential (GND).

[0107] The fourth transistor 9c operates in the active region based on the second reference current Iref2, so that a second mirror current corresponding to the second reference current Iref2 and the second mirror ratio flows through the fourth resistor 9e, and a fourth voltage drop corresponding to the second mirror current occurs across the fourth resistor 9e.

[0108] The second output terminal 9f is the second output terminal of the second current mirror circuit 9. As shown in the figure, the second output terminal 9f is connected to the drain terminal of the fourth transistor 9c inside the second current mirror circuit 9, and is connected to the second input terminal of the control voltage generating unit 6b outside the second current mirror circuit 9.

[0109] The drain current of the fourth transistor 9c is a second mirror current that is set in accordance with the second reference current and the second mirror ratio and does not have temperature dependency, and the second output terminal 9f outputs this second mirror current as a second temperature compensation current I2 to the control voltage generator 6b.

[0110] Here, in the first current mirror circuit 8 and the second current mirror circuit 9, a current mirror circuit having the circuit configuration of Fig. 3A and a current mirror circuit having the circuit configuration of Fig. 3B may be cascade-connected. In this case, the current mirror circuit connected in the subsequent stage functions as a folding current mirror circuit that folds back the flow of current.

[0111] The control voltage generating unit 6b generates a control voltage Vm based on the first temperature compensated current I1 and the second temperature compensated current I2 input from the temperature compensation current mirror circuit 6a and the switching signal Vc input from the switching control unit 7.

[0112] As also shown in FIG. 4A, the control voltage generating unit 6b includes a first input terminal 10a, a second input terminal 10b, a correction terminal 10c, and an output terminal 10d, as well as a first current-voltage conversion circuit 10e, a second current-voltage conversion circuit 10f, an instrumentation amplifier 10g, and a reference voltage generating circuit 10h.

[0113] In this control voltage generating unit 6b, the first input terminal 10a is connected to the first output terminal of the temperature compensating current mirror circuit 6a, the second input terminal 10b is connected to the second output terminal of the temperature compensating current mirror circuit 6a, the correction terminal 10c is connected to the output terminal of the switching control unit 7 and the control terminal of the switching attenuator 5, and the output terminal 10d is connected to the control terminal of the variable attenuator 4.

[0114] The first current-voltage conversion circuit 10e has an input terminal connected to the output terminal of the first current mirror circuit 8 in the temperature compensation current mirror circuit 6a, and an output terminal connected to the positive-phase input terminal of the instrumentation amplifier 10g. The first current-voltage conversion circuit 10e converts the first temperature-compensated current I1 input from the first current mirror circuit 8 into a first temperature-compensated voltage V1 and outputs it to the instrumentation amplifier 10g.

[0115] The second current-voltage conversion circuit 10f has an input terminal connected to the output terminal of the second current mirror circuit 9 in the temperature compensation current mirror circuit 6a, and an output terminal connected to the inverting input terminal of the instrumentation amplifier 10g. The second current-voltage conversion circuit 10f converts the second temperature-compensated current I2 input from the second current mirror circuit 9 into a second temperature-compensated voltage V2 and outputs it to the instrumentation amplifier 10g.

[0116] The instrumentation amplifier 10g has a positive-phase input terminal, a negative-phase input terminal, an output terminal, a control terminal, and a reference voltage terminal. The positive-phase input terminal is connected to the output terminal of the first current-voltage conversion circuit 10e, and the negative-phase input terminal is connected to the output terminal of the second current-voltage conversion circuit 10f.

[0117] Furthermore, in the instrumentation amplifier 10g, the output terminal is the output terminal of the control voltage generator 6b and is connected to the control terminal of the variable attenuator 4. Furthermore, in the instrumentation amplifier 10g, the control terminal is the correction terminal of the control voltage generator 6b and is connected to the output terminal of the switching controller 7 and the control terminal of the switching attenuator 5. Furthermore, in the instrumentation amplifier 10g, the reference voltage terminal is connected to the output terminal of the reference voltage generator circuit 10h.

[0118] Such an instrumentation amplifier 10g generates a control voltage Vm based on a first temperature-compensated voltage V1 input from a first current-voltage conversion circuit 10e, a second temperature-compensated voltage V2 input from a second current-voltage conversion circuit 10f, a switching signal Vc input from a switching control unit 7, and a reference voltage Vref input from a reference voltage generation circuit 10h.

[0119] 4B, the instrumentation amplifier 10g includes a first operational amplifier 11a, a second operational amplifier 11b, a first feedback resistor 11c, a second feedback resistor 11d, a variable resistor 11e, a first input resistor 11f, a second input resistor 11g, a third operational amplifier 11h, a bias resistor 11i, and a third feedback resistor 11j.

[0120] The first operational amplifier 11a has a positive-phase input terminal, a negative-phase input terminal, and an output terminal. The positive-phase input terminal of the first operational amplifier 11a is connected to the output terminal of the first current-voltage conversion circuit 10e, and the negative-phase input terminal is connected to a first end of the first feedback resistor 11c and a first end of the variable resistor 11e.

[0121] The output terminal of the first operational amplifier 11a is connected to the second end of the first feedback resistor 11c and the first end of the first input resistor 11f. The first operational amplifier 11a amplifies the first temperature compensation voltage V1 in positive phase based on the resistance values ​​of the first feedback resistor 11c and the variable resistor 11e, and outputs the amplified voltage as the first temperature compensation voltage V1a to the first end of the first input resistor 11f.

[0122] The second operational amplifier 11b has a positive-phase input terminal, a negative-phase input terminal, and an output terminal, similar to the first operational amplifier 11a. The positive-phase input terminal of the second operational amplifier 11b is connected to the output terminal of the second current-voltage conversion circuit 10f, and the negative-phase input terminal is connected to a first end of the second feedback resistor 11d and a second end of the variable resistor 11e.

[0123] The output terminal of the second operational amplifier 11b is connected to the second end of the second feedback resistor 11d and the first end of the second input resistor 11g. The second operational amplifier 11b amplifies the second temperature compensation voltage V2 in positive phase based on the resistance values ​​of the second feedback resistor 11d and the variable resistor 11e, and outputs the amplified voltage as the second temperature compensation voltage V2a to the first end of the second input resistor 11g.

[0124] The first feedback resistor 11c is a two-terminal element having a predetermined resistance value. A first end of the first feedback resistor 11c is connected to a first end of the variable resistor 11e and the inverting input terminal of the first operational amplifier 11a, and a second end of the first feedback resistor 11c is connected to the output terminal of the first operational amplifier 11a and a first end of the first input resistor 11f.

[0125] The second feedback resistor 11d is a two-terminal element having a predetermined resistance value. A first end of the second feedback resistor 11d is connected to the second end of the variable resistor 11e and the inverting input terminal of the second operational amplifier 11b, and a second end of the second feedback resistor 11d is connected to the output terminal of the second operational amplifier 11b and the first end of the second input resistor 11g.

[0126] The variable resistor 11e is a three-terminal element having a predetermined resistance value. A first end of the variable resistor 11e is connected to a first end of the first feedback resistor 11c and the negative-phase input terminal of the first operational amplifier 11a, and a second end of the variable resistor 11e is connected to a first end of the second feedback resistor 11d and the negative-phase input terminal of the second operational amplifier 11b.

[0127] The variable resistor 11e has an intermediate variable terminal connected to the correction terminal 10c. That is, the variable resistor 11e is a resistor whose resistance value (variable resistance value) is variably set by a switching signal Vc input from the switching control unit 7 to the correction terminal 10c.

[0128] The first input resistor 11f is a two-terminal element having a predetermined resistance value. A first end of the first input resistor 11f is connected to the output terminal of the first operational amplifier 11a and the second end of the first feedback resistor 11c, and a second end of the first input resistor 11f is connected to the negative-phase input terminal of the third operational amplifier 11h and the first end of the third feedback resistor 11j.

[0129] The second input resistor 11g is a two-terminal element having a predetermined resistance value. A first end of the second input resistor 11g is connected to the output terminal of the second operational amplifier 11b and the second end of the second feedback resistor 11d, and a second end of the second input resistor 11g is connected to the positive input terminal of the third operational amplifier 11h and the first end of the bias resistor 11i.

[0130] The third operational amplifier 11h has an inverting input terminal, a positive input terminal, and an output terminal, similar to the first operational amplifier 11a and the second operational amplifier 11b described above. In the third operational amplifier 11h, the inverting input terminal is connected to the second end of the first input resistor 11f and the first end of the third feedback resistor 11j, and the positive input terminal is connected to the second end of the second input resistor 11g and the first end of the bias resistor 11i.

[0131] The third operational amplifier 11h has an output terminal connected to the second end of the third feedback resistor 11j and the output terminal 10d. The third operational amplifier 11h differentially amplifies the first temperature compensation voltage V1a input from the first operational amplifier 11a and the second temperature compensation voltage V2a input from the second operational amplifier 11b, and outputs the difference voltage between the first temperature compensation voltage V1a and the second temperature compensation voltage V2a as a control voltage Vm to the output terminal 10d.

[0132] The bias resistor 11i is a two-terminal element having a predetermined resistance value. A first terminal of the bias resistor 11i is connected to the second terminal of the second input resistor 11g and the positive-phase input terminal of the third operational amplifier 11h, and a second terminal of the bias resistor 11i is connected to the output terminal of the reference voltage generating circuit 10h. A predetermined reference voltage Vref is input to the second terminal of the bias resistor 11i from the reference voltage generating circuit 10h.

[0133] The third feedback resistor 11j is a two-terminal element having a predetermined resistance value. A first end of the third feedback resistor 11j is connected to the second end of the first input resistor 11f and the inverting input terminal of the third operational amplifier 11h, and a second end of the third feedback resistor 11j is connected to the output terminal of the third operational amplifier 11h and the output terminal 10d.

[0134] The reference voltage generating circuit 10h has an input terminal connected to the correction terminal 10c and the control terminal of the instrumentation amplifier 10g, and an output terminal connected to the reference voltage terminal of the instrumentation amplifier 10g. The reference voltage generating circuit 10h generates a reference voltage Vref according to the switching signal Vc input to the correction terminal 10c and outputs it to the reference voltage terminal of the instrumentation amplifier 10g.

[0135] The control voltage generator 6b converts the first temperature-compensated current I1 input from the temperature-compensating current mirror circuit 6a into a first temperature-compensated voltage V1 using a first current-voltage converter circuit 10e and inputs the first temperature-compensated voltage V1 to the positive-phase input terminal of the instrumentation amplifier 10g. The control voltage generator 6b also converts the second temperature-compensated current I2 input from the temperature-compensating current mirror circuit 6a into a second temperature-compensated voltage V2 using a second current-voltage converter circuit 10f and inputs the second temperature-compensated voltage V2 to the negative-phase input terminal of the instrumentation amplifier 10g.

[0136] The control voltage generator 6b generates the control voltage Vm by differentially amplifying the first temperature-compensated voltage V1 and the second temperature-compensated voltage V2 using the instrumentation amplifier 10g. When generating the control voltage Vm, the instrumentation amplifier 10g adjusts the ratio between the first temperature-compensated voltage V1 and the second temperature-compensated voltage V2 based on the switching signal Vc, and also adjusts the offset voltage using the reference voltage Vref input from the reference voltage generator circuit 10h.

[0137] The control voltage generating unit 6b generates a control voltage Vm that suppresses fluctuations in the variable attenuation of the variable attenuator 4 due to the operating state of the switching attenuator 5 and that suppresses fluctuations in the variable attenuation of the variable attenuator 4 due to temperature fluctuations, by referring to the first temperature compensation current I1 (first mirror current) and the second temperature compensation current I2 (second mirror current) input from the temperature compensation current mirror circuit 6a in addition to the switching signal Vc input from the switching control unit 7.

[0138] That is, the control voltage generating unit 6b generates the control voltage Vm by referring to the switching signal Vc when generating the control voltage Vm, so that the variable attenuator 4 applies the desired amount of variable attenuation to the high-frequency signal So (transmission signal) regardless of the operating state of the switching attenuator 5.

[0139] Furthermore, the control voltage generator 6b generates the control voltage Vm by referring to the first temperature-compensated current I1 and the second temperature-compensated current I2, thereby suppressing temperature-related fluctuations in the variable attenuation of the variable attenuator 4. Such a control voltage Vm corresponds to the variable signal of the present invention.

[0140] The switching control unit 7 has an output terminal which is connected to the input terminal of the switching attenuator 5 and the correction terminal of the control voltage generating unit 6b. The switching control unit 7 is a second control signal generating unit which generates a switching signal Vc as a second control signal which sets the switching attenuation of the switching attenuator 5. The switching signal Vc is a binary signal which switches between ON / OFF of the switching attenuation of the switching attenuator 5.

[0141] Next, the operation of the composite attenuator A according to the first embodiment will be described in detail with reference to FIGS. 5A, 5B, and 6A to 6C.

[0142] 1, the composite attenuator A has a variable attenuator 4 and a switchable attenuator 5 cascade-connected on a transmission line 3 (transmission path). In such a composite attenuator A, when the attenuation (switchable attenuation) of the switchable attenuator 5 is switched to, for example, zero or a predetermined value based on the switching signal Vc, the total attenuation of the variable attenuator 4 and the switchable attenuator 5 changes depending on whether or not the switchable attenuation is present. In other words, the total attenuation changes depending on the operating state of the switchable attenuator 5.

[0143] In such a situation, the variable control unit 6 receives a switching signal Vc from the switching control unit 7 that controls the switchable attenuator 5. The variable control unit 6 corrects the control voltage Vm according to the switching signal Vc, i.e., the operating state of the switchable attenuator 5, thereby eliminating or suppressing the change in the total attenuation caused by the presence or absence of the switchable attenuation.

[0144] The temperature characteristic of the total attenuation of the composite attenuator A is maintained close to the desired temperature characteristic by such correction of the control voltage Vm in the variable control section 6. That is, according to the composite attenuator A of the first embodiment, it is possible to apply a desired total attenuation to the high-frequency signal (transmission signal) regardless of the operating state of the switchable attenuator 5.

[0145] The variable control section 6 in the composite attenuator A generates a control voltage Vm by differentially processing the first temperature-compensated current I1 and the second temperature-compensated current I2 generated by the temperature-compensating current mirror circuit 6a in the control voltage generating section 6b. Therefore, the control voltage Vm acts on the variable attenuator 4 so as to maintain the variable attenuation amount regardless of fluctuations in the ambient temperature T.

[0146] 5A shows the relationship between the first temperature compensation voltage V1 and the control voltage Vm. As shown in this characteristic diagram, the control voltage Vm has a negative gradient with respect to the ambient temperature T, similar to the first temperature compensation voltage V1, and therefore prevents or suppresses changes in the variable attenuation of the variable attenuator 4 caused by fluctuations in the ambient temperature T.

[0147] 5B shows the relationship between the first temperature-compensated voltage V1, the second temperature-compensated voltage V2, and the control voltage Vm when the second temperature-compensated current I2 has a temperature characteristic (positive gradient) opposite to the temperature characteristic (negative gradient) of the first temperature-compensated voltage V1. In this case, the gradient of the control voltage Vm can be made larger than that shown in FIG. 5A.

[0148] 6A to 6C show the temperature characteristics of the variable attenuation Gm, the temperature characteristics of the switching attenuation Gk, and the temperature characteristics of the total attenuation Gt. As shown in Fig. 6A, the variable attenuation Gm has a temperature characteristic that increases as the ambient temperature T decreases. In contrast, when the transistors constituting the switching attenuator 5 are on, the switching attenuation Gk has a temperature characteristic that fluctuates slightly as the ambient temperature T changes. Furthermore, the total attenuation Gt has a temperature characteristic that is an integration of the variable attenuation Gm and the switching attenuation Gk.

[0149] 6C, the composite attenuator A according to the first embodiment exhibits a variable attenuation Gs that compensates for relatively small temperature variations in the switching attenuation Gk. That is, by correcting the control voltage Vm based on the switching signal Vc, the slope of the variable attenuation Gs can be adjusted, thereby allowing a desired attenuation to be applied to the high-frequency signal (transmission signal) regardless of the operating state of the switching attenuator 5.

[0150] Second Embodiment Next, a second embodiment of the present invention will be described with reference to Figures 7 to 9. In this second embodiment, the same components as those in the first embodiment are denoted by the same reference numerals.

[0151] 7, the composite attenuator B according to the second embodiment includes an RF input terminal 1, an RF output terminal 2, a transmission line 3 (transmission path), two variable attenuators 4A and 4B, four switchable attenuators 5A to 5D, a variable control unit 6A, and a switchable control unit 7A. The variable control unit 6A according to the second embodiment includes a temperature compensation current mirror circuit 6a and a control voltage generating unit 6b'.

[0152] Here, the ratio of the transistor sizes in the third switching attenuator 5C to the fourth switching attenuator 5D is equal to the ratio of the transistor sizes in the first switching attenuator 5A to the second switching attenuator 5B, e.g., 2. Also, the ratio of the transistor sizes in the second switching attenuator 5C to the fourth switching attenuator 5D is e.g., 2.

[0153] Furthermore, the length of the transmission line 3 between the circuit group consisting of the first variable attenuator 4A, the first switching attenuator 5A, and the second switching attenuator 5B and the circuit group consisting of the second variable attenuator 4B, the third switching attenuator 5C, and the fourth switching attenuator 5D (i.e., between the second variable attenuator 5B and the second variable attenuator 4B) is set to a length equivalent to half the wavelength of the high-frequency signal. By setting the length of the transmission line 3 in this manner, mutual impedance matching is achieved between the circuit group consisting of the first variable attenuator 4A, the first switching attenuator 5A, and the second switching attenuator 5B and the circuit group consisting of the second variable attenuator 4B, the third switching attenuator 5C, and the fourth switching attenuator 5D.

[0154] The first variable attenuator 4A is a high-frequency attenuation circuit whose attenuation can be continuously changed, and includes an input terminal, an output terminal, and a control terminal. The input terminal of the first variable attenuator 4A is connected to the RF input terminal 1, and the output terminal is connected to the input terminal of the first switching attenuator 5A. The control terminal of the first variable attenuator 4A is connected to the output terminal of the control voltage generator 6b' and the control terminal of the second variable attenuator 4B.

[0155] The first variable attenuator 4A applies a first variable attenuation amount to the high frequency signal So input from the RF input terminal 1 based on the control voltage Vm input from the control voltage generator 6b′, and outputs the signal to the first switching attenuator 5A. The first variable attenuation amount is an attenuation amount that can take on continuous values, similar to the variable attenuation amount in the first embodiment.

[0156] The first switching attenuator 5A is a high-frequency attenuation circuit that switches between on and off attenuation and includes an input terminal, an output terminal, and a control terminal. The input terminal of the first switching attenuator 5A is connected to the output terminal of the first variable attenuator 4A, and the output terminal is connected to the input terminal of the second switching attenuator 5B. The control terminal of the first switching attenuator 5A is connected to the first output terminal of the switching control unit 7A, the control terminal of the third switching attenuator 5C, and the first correction terminal of the control voltage generation unit 6b'.

[0157] In the first switching attenuator 5A, whether or not to apply an attenuation amount (first switching attenuation amount) to the high-frequency signal So input from the first variable attenuator 4A is set based on the first switching signal Vc1 input from the switching control unit 7A. That is, based on the first switching signal Vc1, the first switching attenuator 5A switches between a state in which the high-frequency signal So input from the variable attenuator 4A is applied with the first switching attenuation amount and output to the second switching attenuator 5B, and a state in which the high-frequency signal So is not applied with the first switching attenuation amount and output to the second switching attenuator 5B.

[0158] The second switching attenuator 5B is a high-frequency attenuation circuit that switches between on and off attenuation and includes an input terminal, an output terminal, and a control terminal. The input terminal of the second switching attenuator 5B is connected to the output terminal of the first switching attenuator 5A, and the output terminal is connected to the input terminal of the second variable attenuator 4B. The control terminal of the second switching attenuator 5B is connected to the second output terminal of the switching control unit 7A, the control terminal of the fourth switching attenuator 5D, and the second correction terminal of the control voltage generation unit 6b'.

[0159] In the second switching attenuator 5B, whether or not to apply an attenuation amount (second switching attenuation amount) to the high-frequency signal So input from the first switching attenuator 5A is set based on the second switching signal Vc2 input from the switching control unit 7A. That is, based on the second switching signal Vc2, the second switching attenuator 5B switches between a state in which the high-frequency signal So is applied with the second switching attenuation amount and output to the second variable attenuator 4B, and a state in which the high-frequency signal So is output to the second variable attenuator 4B without applying the second switching attenuation amount to the high-frequency signal So.

[0160] The second variable attenuator 4B is a high-frequency attenuation circuit whose attenuation can be continuously changed, similar to the first variable attenuator 4A, and includes an input terminal, an output terminal, and a control terminal. The input terminal of the second variable attenuator 4B is connected to the output terminal of the second switching attenuator 5B, and the output terminal is connected to the input terminal of the third switching attenuator 5C. The control terminal of the second variable attenuator 4B is also connected to the output terminal of the control voltage generator 6b' and the control terminal of the first variable attenuator 4A.

[0161] The second variable attenuator 4B applies a second variable attenuation amount to the high-frequency signal So input from the second switching attenuator 5B based on the control voltage Vm input from the control voltage generator 6b′, and outputs the signal to the third switching attenuator 5C. The second variable attenuation amount is an attenuation amount that can take on a continuous value, similar to the first variable attenuation amount.

[0162] The third switching attenuator 5C is a high-frequency attenuation circuit that switches between on and off attenuation and has an input terminal, an output terminal, and a control terminal. The second switching attenuator 5C has an input terminal connected to the output terminal of the second variable attenuator 4B and an output terminal connected to the input terminal of the fourth switching attenuator 5D. The control terminal of the third switching attenuator 5C is connected to the first output terminal of the switching control unit 7A, the control terminal of the first switching attenuator 5A, and the first correction terminal of the control voltage generation unit 6b'.

[0163] In the third switching attenuator 5C, whether or not to apply an attenuation amount (third switching attenuation amount) to the high-frequency signal So input from the second variable attenuator 4B is set based on the first switching signal Vc1 input from the switching control unit 7 A. That is, based on the first switching signal Vc1, the third switching attenuator 5C switches between a state in which the high-frequency signal So is applied with the second switching attenuation amount and output to the fourth switching attenuator 5D, and a state in which the high-frequency signal So is output to the fourth switching attenuator 5D without applying the third switching attenuation amount to the high-frequency signal So.

[0164] The fourth switching attenuator 5D is a high-frequency attenuation circuit that switches between on and off attenuation and includes an input terminal, an output terminal, and a control terminal. The input terminal of the fourth switching attenuator 5D is connected to the output terminal of the third switching attenuator 5C, and the output terminal is connected to the RF output terminal 2. The control terminal of the fourth switching attenuator 5D is also connected to the second output terminal of the switching control unit 7A, the control terminal of the second switching attenuator 5B, and the second correction terminal of the control voltage generation unit 6b'.

[0165] In the fourth switching attenuator 5D, whether or not to apply an attenuation amount (fourth switching attenuation amount) to the high frequency signal So input from the third switching attenuator 5C is set based on the second switching signal Vc2 input from the switching control unit 7 A. That is, based on the second switching signal Vc2, the fourth switching attenuator 5D switches between a state in which the high frequency signal So is applied with the fourth switching attenuation amount and output to the RF output terminal 2, and a state in which the high frequency signal So is output to the RF output terminal 2 without being applied with the fourth switching attenuation amount.

[0166] Among the components of the variable control unit 6A, the temperature compensation current mirror circuit 6a is the same as in the first embodiment. Unlike the control voltage generation unit 6b in the first embodiment, the control voltage generation unit 6b′ has a first correction terminal and a second correction terminal in addition to a first input terminal, a second input terminal, and an output terminal.

[0167] The control voltage generator 6b' has a first input terminal connected to the first output terminal of the temperature compensation current mirror circuit 6a, a second input terminal connected to the second output terminal of the temperature compensation current mirror circuit 6a, and an output terminal connected to the control terminal of the first variable attenuator 4A and the control terminal of the second variable attenuator 4B.

[0168] The first correction terminal of the control voltage generator 6b' is connected to the first output terminal of the switching controller 7A, the control terminal of the first switching attenuator 5A, and the control terminal of the third switching attenuator 5C. The second correction terminal of the control voltage generator 6b' is connected to the second output terminal of the switching controller 7A, the control terminal of the second switching attenuator 5B, and the control terminal of the fourth switching attenuator 5D.

[0169] Such a control voltage generating unit 6b' generates a control voltage Vm based on the first temperature compensation current I1 and the second temperature compensation current I2 input from the temperature compensation current mirror circuit 6a and the first switching signal Vc1 and the second switching signal Vc2 input from the switching control unit 7A.

[0170] The switching control unit 7A has a first output terminal and a second output terminal. In the switching control unit 7A, the first output terminal is connected to the control terminal of the first switching attenuator 5A, the control terminal of the third switching attenuator 5C, and the first correction terminal of the control voltage generation unit 6b'. In addition, in the switching control unit 7A, the second output terminal is connected to the control terminal of the second switching attenuator 5B, the control terminal of the fourth switching attenuator 5D, and the second correction terminal of the control voltage generation unit 6b'.

[0171] The switching control unit 7A generates a first switching signal Vc1 and outputs it to the first switching attenuator 5A, the third switching attenuator 5C, and the control voltage generation unit 6b'. The switching control unit 7A also generates a second switching signal Vc2 and outputs it to the second switching attenuator 5B, the fourth switching attenuator 5D, and the control voltage generation unit 6b'.

[0172] The first switching signal Vc1 is a binary signal that switches between the presence or absence of the first switching attenuation in the first switching attenuator 5A and the presence or absence of the third switching attenuation in the third switching attenuator 5C. The second switching signal Vc2 is a binary signal that switches between the presence or absence of the second switching attenuation in the second switching attenuator 5B and the presence or absence of the fourth switching attenuation in the fourth switching attenuator 5D.

[0173] Next, the operation of the composite attenuator B according to the second embodiment will be described in detail with reference to FIGS. 8A, 8B and 9. FIG.

[0174] In this composite attenuator B, the operation states of the first switchable attenuator 5A and the third switchable attenuator 5C, i.e., the presence or absence (ON / OFF) of the first switchable attenuation and the third switchable attenuation, are switched by a first switch signal Vc1 generated by a switch control unit 7A. Also, in this composite attenuator B, the operation states of the second switchable attenuator 5B and the fourth switchable attenuator 5D, i.e., the presence or absence (ON / OFF) of the second switchable attenuation and the fourth switchable attenuation, are switched by a second switch signal Vc2 generated by a switch control unit 7A.

[0175] The control voltage generating unit 6b' of the variable control unit 6A receives the first switching signal Vc1 and the second switching signal Vc2, and performs correction processing on the control voltage Vm in accordance with the operating states of the first to fourth switching attenuators 5A to 5D. As a result, the control voltage Vm applies a desired amount of attenuation to the high-frequency signal (transmission signal), regardless of the operating states of the first to fourth switching attenuators 5A to 5D.

[0176] 8A and 8B are characteristic diagrams showing the operation of the composite attenuator B according to the second embodiment. Fig. 8A shows the temperature characteristics of the control voltage Vm according to the ON / OFF states of the first switching signal Vc1 and the second switching signal Vc2, and Fig. 8B shows the attenuation of the composite attenuator B according to the ON / OFF states of the first switching signal Vc1 and the second switching signal Vc2. The attenuation of the composite attenuator B increases as the temperature decreases, and it is shown that the offset of the attenuation can be adjusted by controlling the switching of the first to fourth switchable attenuators 5A to 5D.

[0177] 9 is a characteristic diagram showing the difference in attenuation between the composite attenuator B according to the second embodiment and a comparative example. The comparative example is a composite attenuator B in which the first switching signal Vc1 and the second switching signal Vc2 are not input to the variable control section 6A, and therefore the control voltage Vm is not corrected based on the first switching signal Vc1 and the second switching signal Vc2.

[0178] As shown in FIG. 9, in the composite attenuator according to the comparative example, the variation in the amount of attenuation (about 2.5 dB) increases depending on whether the first switching signal Vc1 and the second switching signal Vc2 are set to ON or OFF, but the variation in the amount of attenuation (about 1 dB) of composite attenuator B depending on whether the first switching signal Vc1 and the second switching signal Vc2 are set to ON or OFF is smaller than that of the composite attenuator according to the comparative example.

[0179] According to the second embodiment, it is possible to apply a desired amount of attenuation to a high-frequency signal (transmission signal) regardless of the operating states of the first to fourth switchable attenuators 5A to 5D, and it is also possible to reduce variations in the amount of attenuation due to the ON / OFF settings of multiple switchable signals (first switchable signal Vc1 and second switchable signal Vc2).

[0180] According to the present disclosure, it is possible to provide a composite attenuator that can apply a desired amount of attenuation to a transmission signal in response to temperature changes, regardless of the operating state of a switching attenuator.

[0181] A...composite attenuator, I1...first temperature compensation current, I2...second temperature compensation current, Vc...switching signal, Vm...control voltage (variable signal), So...high frequency signal, 1...RF input terminal, 2...RF output terminal, 3...transmission line (transmission path), 4, 4A, 4B...variable attenuator, 5, 5A, 5B, 5C, 5D...switching attenuator, 6, 6A...variable control section, 6a...temperature compensation current mirror circuit, 6b, 6b'...control voltage generation section, 7, 7A...switching control section, 8...first current mirror circuit, 9...second current mirror circuit, 10a...first input terminal, 10b... Second input terminal, 10c...correction terminal, 1d...output terminal, 10e...first current-voltage conversion circuit, 10f...second current-voltage conversion circuit, 10g...instrumentation amplifier, 10h...reference voltage generation circuit, 11a...first operational amplifier, 11b...second operational amplifier, 11c...first feedback resistor, 11d...second feedback resistor, 11e...variable resistor, 11f...first input resistor, 11g...second input resistor, 11h...third operational amplifier, 11i...bias resistor, 11j...third feedback resistor

Claims

1. A composite attenuator in which a switching attenuator whose attenuation amount switches between presence and absence and a variable attenuator whose attenuation amount can change continuously are connected in cascade, the composite attenuator comprising: a switching control unit that generates a switching signal for setting the attenuation amount of the switching attenuator; and a variable control unit that generates a variable signal for setting the attenuation amount of the variable attenuator, wherein the variable control unit corrects the variable signal according to the switching signal.

2. The composite attenuator according to claim 1, wherein the variable attenuator and the switching attenuator each include an input terminal, an output terminal, a transmission path, and a transistor shunt-connected to the transmission path, and the transistor is controlled by the variable signal.

3. The composite attenuator according to claim 2, wherein the variable attenuator and the switching attenuator each include an impedance matching circuit either on one or both of the sides between the input terminal and the transistor and between the output terminal and the transistor.

4. The composite attenuator according to any one of claims 1 to 3, wherein the variable control unit includes a temperature compensation current mirror circuit and generates the variable signal so as to suppress fluctuations in the attenuation amount of the variable attenuator based on temperature fluctuations.

5. The composite attenuator according to any one of claims 1 to 4, wherein the variable control unit includes a current-voltage conversion circuit, an instrumentation amplifier, and a reference voltage generation circuit.

6. The composite attenuator according to claim 4, wherein resistors having different slopes of temperature characteristics are connected to the input side and the output side of the temperature compensation current mirror circuit.

7. The composite attenuator according to claim 4, wherein a folded-back current mirror circuit in which resistors having different slopes of characteristics with respect to temperature are coupled to the input side and the output side, respectively, is connected in cascade to the temperature compensation current mirror circuit.

8. The composite attenuator according to claim 4, wherein the temperature compensation current mirror circuit includes a first current mirror circuit whose output current has a negative slope with respect to temperature and a second current mirror circuit whose output current has a positive slope with respect to temperature.

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