Counter electromotive force observer, high-power permanent magnet synchronous motor control device and high-power permanent magnet synchronous motor control method

Abstract

The invention discloses a back electromotive force observer and a high-power permanent magnet synchronous motor control device and method. The invention aims to solve the problems of large model discrete error and inaccurate and even divergent back electromotive force observer result when a high-power permanent magnet synchronous motor works in a low-carrier-ratio working condition caused by low switching frequency and the motor operates in a medium-speed region and a high-speed region. A permanent magnet synchronous motor back electromotive force observer coefficient matrix is improved by adopting a Pade approximation method, a high-order discrete model is established, and meanwhile, aiming at the problem of inaccurate position observation caused by a mean value error existing after sampling current calculation, a compensation link is designed to reduce a position observation error, so that the accuracy of position observation is improved. Through the measures, the stable operation of the high-power permanent magnet synchronous motor in the full-speed domain range is ensured.

Description

Technical Field

[0001] The present invention belongs to the technical field of motor control, and in particular relates to a back-electromotive force observer, a high-power permanent magnet synchronous motor control device and a method. Background Art

[0002] Permanent magnet synchronous motors have become the first choice for variable frequency speed control electrical transmission systems due to their small size, high power density, simple structure and good speed control performance, and are widely used in various industries. In recent years, the development and application of high-power permanent magnet synchronous motors and high-speed permanent magnet synchronous motors have been continuously expanded, and permanent magnet synchronous motor speed control systems have gradually become a research hotspot in the field of AC speed control transmission. Permanent magnet synchronous motors can be divided into two types: surface mount type and built-in type according to the different structures of their rotor permanent magnets.

[0003] At present, in the high-performance speed regulation system of permanent magnet synchronous motor, vector control scheme is usually adopted. It is necessary to install detection devices such as encoder, rotary transformer or Hall sensor at one end of the motor shaft to obtain the rotor position angle information. The installation of detection devices increases the system cost, volume and reliability to a certain extent. In order to avoid these problems, the position sensorless control technology of permanent magnet synchronous motor has become a research hotspot in the control system of permanent magnet synchronous motor. At present, the position sensorless control technology of permanent magnet synchronous motor has different control methods in different speed ranges. The position sensorless control technology of permanent magnet synchronous motor in the medium and high speed range mainly observes the rotor position information through back electromotive force or magnetic flux. The position sensorless control technology of permanent magnet synchronous motor in the low speed range mainly obtains the rotor position information through the salient pole characteristics of the motor.

[0004] Permanent magnet synchronous motor position sensorless control technology has been widely used commercially and has achieved commercial value. However, these applications work at higher switching frequencies and smaller control cycles. The ratio of the switching frequency of the inverter to the motor operating base frequency, i.e., the carrier ratio, is within a larger range, showing excellent control performance. However, when the permanent magnet synchronous motor faces high-power or high-speed applications, the carrier ratio will inevitably decrease due to the loss of the switching device and the control performance of the main control chip. Under low carrier ratio conditions, the dual closed-loop vector control system will face the impact of digital control delay, aggravated dq axis coupling, and inaccurate mathematical models of permanent magnet synchronous motors. These problems directly lead to reduced control performance of the dual closed-loop vector control system, and in severe cases, cause instability of the control system.

[0005] In order to solve the problem of reduced current loop control performance caused by severe cross-coupling of the permanent magnet synchronous motor speed control system when the carrier ratio is relatively low, the prior art "Low-carrier ratio current prediction control system and method for high-power permanent magnet synchronous motor" proposes a current PWM prediction control method. Although stable operation under high-power and low-carrier ratio conditions is achieved, the proposed method only considers surface-mounted motors and requires additional hardware to obtain the motor rotor position, which limits the application scenarios.

[0006] In order to solve the problem of poor decoupling performance of the PI regulator and increased control delay in the permanent magnet synchronous motor control system under low carrier ratio operating conditions, which leads to poor current loop performance or even instability, the prior art "A low carrier ratio control method for permanent magnet synchronous motor" designs a current regulator with delay compensation function and adjustable dq-axis dynamic characteristics. Although it effectively improves the stability of the motor control system under low carrier ratio conditions, it requires a complex table lookup process and still requires additional hardware to obtain the motor rotor position.

[0007] Therefore, designing a simple and effective solution to solve the problems faced by the permanent magnet synchronous motor position sensorless control system under low carrier ratio conditions, which is also applicable to both surface mount and embedded permanent magnet synchronous motors, is an urgent problem to be solved in the high-power permanent magnet synchronous motor position sensorless control system. Summary of the invention

[0008] The purpose of the present invention is to provide a back-EMF observer, a high-power permanent magnet synchronous motor control device and method, which are used to solve the problems of digital control delay, aggravated dq axis coupling and inaccurate mathematical model of permanent magnet synchronous motor faced by the position sensorless control system of high-power permanent magnet synchronous motor under low carrier ratio conditions. The present invention improves the coefficient matrix of the back-EMF observer of the permanent magnet synchronous motor by adopting the Padé approximation method, establishes a high-order discrete model, and at the same time, in view of the problem of inaccurate position observation caused by the mean error after the sampled current calculation, the present invention designs a compensation link to reduce the position observation error, and through the above measures, ensures the stable operation of the high-power permanent magnet synchronous motor in the full speed range.

[0009] The present invention is implemented according to the following technical solutions:

[0010] In a first aspect, the present invention provides a back-EMF observer, which improves a coefficient matrix of a back-EMF observer of a permanent magnet synchronous motor by using a Padé approximation method;

[0011] The improved back-EMF observer converts the current signal in the dq coordinate system, the compensated voltage reference signal and the compensated voltage reference signal into the observed back-EMF;

[0012] The improved back-electromotive force observer includes a dq axis current estimation link and a dq axis back-electromotive force estimation link, wherein the dq axis current estimation link is used to obtain an observation current signal matrix in a dq coordinate system at the next moment, and the dq axis back-electromotive force estimation link is used to obtain an observation back-electromotive force signal matrix in a dq coordinate system at the next moment;

[0013] The improved back-EMF observer is expressed as:

[0014]

[0015] in, is the observed dq axis current value at time k+1, is the observed dq axis current value at time k, is the observed value of the dq-axis back EMF at time k+1, is the value of the dq-axis back EMF at the observed time k, is the reference value of the dq axis voltage at time k, i dq (k) is the dq axis current value obtained by the current sensor collecting the three-phase current at time k after Park transformation. The matrices G1 and G2 are feedback gain matrices adaptive to the speed. A and B are coefficient matrices. I is the unit matrix and O is the zero matrix. The coefficient matrix A is expressed as:

[0016] A=A d1 +A d2

[0017] Among them, A d1 is the constant coefficient matrix on the diagonal of the original coefficient matrix After first-order Euler discretization, we get A d2 is the speed coupling coefficient matrix on the anti-diagonal line of the original coefficient matrix After the Padre approximation,

[0018]

[0019] in,

[0020]

[0021]

[0022]

[0023] The coefficient matrix B is expressed as:

[0024]

[0025] Among them, L d , L qis the dq axis inductance of the permanent magnet synchronous motor, R s is the stator resistance of the permanent magnet synchronous motor, ω e is the motor electrical angular velocity, T s is the control cycle of the control program.

[0026] In one implementation, the feedback gain coefficient matrix is ​​set by directly configuring poles in the z domain to ensure the convergence of the observer. The characteristic polynomial det(zI-A1+GL) of the back-EMF observer is improved by using the algebraic properties of the block matrix and is expressed as:

[0027]

[0028] Let det(zI-A1+GL)=det((z-p1)(z-p2)I), and the expression of the feedback gain matrix is ​​obtained according to the coefficient matching principle:

[0029] G1=(1-p1-p2)I+A, G2=(1-p1)(1-p2)I

[0030] Among them, matrix A is the coefficient matrix related to the speed, and the stability of the observer can be guaranteed by configuring the poles within the unit circle.

[0031] In one embodiment, the device includes a Park converter, the above-mentioned back-EMF observer, a phase-locked loop module, a current PI controller, a speed PI controller, and an inverse Park converter with delay compensation;

[0032] The input end of the Park converter is connected to the output signal of the permanent magnet synchronous motor UVW three-phase current sensor respectively, and the angle signal input end of the Park converter is connected to the angle output end of the phase-locked loop; the current signal output of the Park converter is divided into a d-axis current signal i d With the q-axis current signal i q , the dq axis current signal is connected to the input terminal of the improved back-EMF observer;

[0033] The back-EMF observer input terminal is connected to the current PI controller output terminal and the Park converter output terminal respectively, and the improved back-EMF observer output terminal is connected to the phase-locked loop input terminal;

[0034] The output end of the phase-locked loop estimates the rotational speed and the reference rotational speed, and then connects to the input end of the speed PI controller. The output end of the phase-locked loop estimates the rotational speed and connects to the input end of the current PI controller and the inverse Park transformation with delay compensation. The output end of the phase-locked loop estimates the rotor position angle and connects to the input end of the current PI controller, the inverse Park transformation with delay compensation, and the Park transformation.

[0035] The output end of the current PI controller is connected to the input end of the inverse Park transformation with delay compensation;

[0036] The output end of the inverse Park transformation with delay compensation is connected to the input end of the SVPWM inverter.

[0037] In one embodiment, the current PI controller is a complex vector decoupled current PI controller; the complex vector decoupled current PI controller includes a d-axis PI, a d-axis integrator, a q-axis integrator and a q-axis PI; the proportional coefficient of the d-axis PI is k p_id , the integration coefficient is k i_id , the proportional coefficient of the q-axis PI is k p_iq , the integration coefficient is k i_iq , the integration coefficient of the d-axis integrator is k p_id and The coefficient of the q-axis integrator is k p_iq and The product of Equal to the difference between the output of the d-axis PI and the d-axis integrator, the output =Q-axis PI output plus Q-axis integrator output plus The product of ψ and ψ, where ψ is the magnetic flux of the permanent magnet, which is a constant;

[0038] The k p_id =L d *2π*f c , k i_id =R s *2π*f c , where L d is the d-axis inductance of the permanent magnet synchronous motor, R s is the stator resistance of the permanent magnet synchronous motor, f c Sets the cutoff frequency for the current loop bandwidth.

[0039] In one implementation, the compensation function of the inverse Park converter with delay compensation is expressed as:

[0040] It includes angle delay compensation and amplitude compensation. The angle delay compensation is Amplitude compensation is For amplitude compensation, T s is the program control period in the present invention, which is numerically the same as the PWM switching period. is the electrical angular velocity of the motor rotor.

[0041] In one embodiment, the high-power permanent magnet synchronous motor control device based on position sensorless also includes: a current mean error compensation module, whose input end is respectively connected to the output end of the complex vector decoupling current PI controller and the output end of the phase-locked loop to estimate the speed, and whose output end is summed with the dq axis current signal, and the summation result is subtracted from the reference current and the difference is connected to the input end of the complex vector decoupling current PI controller.

[0042] In one embodiment, the current mean error compensation module uses the dq axis voltage reference value at the input end and the estimated rotor speed The current value of dq axis compensation is obtained through calculation; the current mean error compensation calculation formula is:

[0043]

[0044] Among them, i dq_com is the current value of dq axis compensation, is the reference value of dq axis voltage, T s To control the cycle, is the estimated value of the motor rotor speed,

[0045] In one embodiment, the phase-locked loop includes a four-quadrant inverse tangent function, a PI controller and an integrator; the four-quadrant inverse tangent function atan2 is used to calculate the angle error value Angle error value Used to drive the phase-locked loop, angle error value Calculated by PI controller right Integrate to get the motor rotor position angle The proportional coefficient in the PI controller is k p_PLL =2ξω0, the integral coefficient is The integration time constant in the integrator is T s , where ξ is the damping coefficient and ω is the phase-locked loop bandwidth.

[0046] In a second aspect, the present invention provides a control method based on the above-mentioned high-power permanent magnet synchronous motor control device, the method comprising:

[0047] Collect the UVW three-phase current signal of the permanent magnet synchronous motor and obtain the dq axis current value i through the Park converter dq ;

[0048] Improve the dq axis current estimation link in the back EMF observer according to the dq axis current value i dq , voltage reference value Estimated value of back EMF at time k And the predicted dq axis current at time k Calculate the predicted value of the dq axis current at time k+1

[0049] Improve the dq axis back EMF estimation link in the back EMF observer according to the predicted value of the dq axis current at the next moment Predicted dq-axis current at time k And the reference value of voltage Calculate the estimated value of the back EMF at time k+1

[0050] Observations Obtaining the estimated motor rotor position through a phase-locked loop and speed

[0051] The reference value of the dq axis voltage and the observed rotor speed information As the input of the current mean error compensation module, the current value i that needs to be compensated for the current regulation link is calculated dq_com ;

[0052] The estimated speed With reference speed The q-axis reference current is obtained by using the speed PI controller.

[0053] The current value i of the dq axis current signal and the dq axis current signal compensation dq_com The sum is calculated and then the difference is made with the reference current as the input of the complex vector decoupling current PI controller. The reference voltage of the dq axis is obtained by calculation.

[0054] According to the dq axis reference voltage Speed ​​estimate and the rotor position estimate Inverse Park calculation with delay compensation to obtain the reference voltage of αβ axis

[0055] SVPWM is based on the reference voltage of the αβ axis After calculation, the inverter is controlled to output three-phase voltage to the motor, realizing stable operation of the high-power permanent magnet synchronous motor in the full speed range.

[0056] In a third aspect, the present invention provides a high-power permanent magnet synchronous motor control system, the system comprising: the high-power permanent magnet synchronous motor control device mentioned above.

[0057] Beneficial effects of the present invention:

[0058] (1) The present invention uses a high-order discrete coefficient matrix, so that the back-EMF observer still has good performance under low carrier ratio conditions, effectively ensuring the stable operation of the permanent magnet synchronous motor position sensorless control system under low carrier ratio conditions and improving the control performance.

[0059] (2) The present invention uses a gain matrix related to the speed, so that the observer has good performance in a wider speed range, thereby improving the control performance of the entire system.

[0060] (3) The present invention uses a current mean error compensation link to effectively reduce the rotor position observation error caused by the sampled current mean error.

[0061] (4) The present invention uses a complex vector decoupling current PI controller to effectively realize the decoupling of the dq axis current. The implementation method is simple and ensures that the motor has good performance under low carrier ratio conditions.

[0062] (5) The present invention achieves a simple and effective solution to the problems faced by the position sensorless control system of a permanent magnet synchronous motor under low carrier ratio conditions, and is applicable to both surface-mounted and embedded permanent magnet synchronous motors. BRIEF DESCRIPTION OF THE DRAWINGS

[0063] The accompanying drawings are part of the present invention and are used to provide a further understanding of the present invention. The exemplary embodiments of the present invention and their descriptions are used to explain the present invention, but do not constitute an improper limitation of the present invention. Obviously, the accompanying drawings described below are only some embodiments. For ordinary technicians in this field, other accompanying drawings can be obtained based on these accompanying drawings without creative work.

[0064] Figure 1 A principle block diagram of an improved back-EMF observer provided in one embodiment of the present invention;

[0065] Figure 2 In order to realize the two-phase stationary coordinate system αβ and the two-phase synchronous rotating coordinate system d where each signal is located during back EMF observation real q real and the relative relationship diagram between the estimated two-phase synchronous rotating coordinate system dq;

[0066] Figure 3 A structural block diagram of a high-power permanent magnet synchronous motor control device provided by an embodiment of the present invention;

[0067] Figure 4 It is a structural block diagram of a complex vector decoupling current PI controller provided in one embodiment of the present invention;

[0068] Figure 5 A structural block diagram of current mean error compensation provided in one embodiment of the present invention;

[0069] Figure 6 A structural block diagram of a phase-locked loop provided in one embodiment of the present invention;

[0070] Figure 7 is a comparison chart of simulation results, where Figure 7 (a) is the simulation diagram of the improved back-EMF observer with current mean error compensation link. Figure 7 (b) is the simulation diagram of the improved back-EMF observer without the current mean error compensation link.

[0071] It should be noted that these drawings and textual descriptions are not intended to limit the conceptual scope of the present invention in any way, but are intended to illustrate the concept of the present invention for those skilled in the art by referring to specific embodiments. DETAILED DESCRIPTION

[0072] The present invention will be further described below in conjunction with the accompanying drawings.

[0073] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0074] Aiming at the low carrier ratio condition caused by the high-power permanent magnet synchronous motor working at low switching frequency, the model discrete error is large, and the back-EMF observer result is inaccurate or even divergent when the motor runs in the medium and high speed area. The present invention proposes a back-EMF observer, a high-power permanent magnet synchronous motor control device and method.

[0075] Reference Figure 1 As shown, an embodiment of the present disclosure provides a back-EMF observer, which improves the coefficient matrix of the back-EMF observer of the permanent magnet synchronous motor by using the Padé approximation method.

[0076] The general form of the method used to perform the Padé approximation is described as Where P [m / n] (AT s ) is a rational function, is a polynomial of degree m, It is a polynomial of degree n. Usually, B0=I is taken to normalize the denominator polynomial. Substitute the speed coupling coefficient matrix on the anti-diagonal line in the original coefficient matrix into the above formula, take m and n as 2, and the high-order discrete matrix expression using the Padé approximation is obtained: Further simplifying the expression, we can get A d2 .

[0077] Figure 2 is the relationship between the various coordinate systems involved in the back-EMF observer, the two-phase stationary coordinate system αβ, the two-phase synchronous rotating coordinate system d real q real and estimate the two-phase synchronous rotating coordinate system dq,d real q real The angle between the coordinate system and the αβ coordinate system is θ e , is the real rotor position of the motor, and the angle between the dq coordinate system and the αβ coordinate system is is the estimated rotor position, and the estimated error of the rotor position is

[0078] The improved back-EMF observer transforms the current signal i in the dq coordinate system d 、i q , the compensated voltage reference signal and the compensated voltage reference signal Converted to observed back EMF

[0079] The improved back-EMF observer includes a dq axis current estimation link and a dq axis back-EMF estimation link. The dq axis current estimation link is used to obtain the observed current signal matrix in the dq coordinate system at the next moment. The dq axis back EMF estimation link is used to obtain the observed back EMF signal matrix in the dq coordinate system at the next moment

[0080] Specifically, the compensated voltage reference signal Compensated voltage reference signal Estimate the current signal i in the two-phase synchronous rotating coordinate system d , estimate the current signal i in the two-phase synchronous rotating coordinate system q , estimate the observed back EMF in the two-phase synchronous rotating coordinate system Estimation of observed back EMF in two-phase synchronously rotating coordinate system Estimation of observed current signal in two-phase synchronous rotating coordinate system And estimate the observed current signal in the two-phase synchronous rotating coordinate system Input to the dq axis current estimation link, the input estimated current signal matrix i in the two-phase synchronous rotating coordinate system dq (k) is multiplied by the improved coefficient matrix A, and the input compensated voltage reference signal matrix is Multiplying with the coefficient matrix B, the input estimated current signal matrix i in the two-phase synchronous rotating coordinate system dq (k) The observed current signal matrix in the estimated two-phase synchronous rotating coordinate system with the input After making the difference, multiply the difference by the gain matrix G1, add the above results and add the input estimated two-phase synchronous rotating coordinate system observation back EMF matrix Get the observed current signal matrix of the next estimated two-phase synchronous rotating coordinate system After a delay of one shot, we get

[0081] The current signal i in the two-phase synchronous rotating coordinate system is estimated d , estimate the current signal i in the two-phase synchronous rotating coordinate system q , estimate the observed back EMF in the two-phase synchronous rotating coordinate system Estimation of observed back EMF in two-phase synchronously rotating coordinate system Estimation of observed current signal in two-phase synchronous rotating coordinate system And estimate the observed current signal in the two-phase synchronous rotating coordinate system Input to the dq axis back EMF estimation link, the input estimated current signal matrix i in the two-phase synchronous rotating coordinate system dq (k) The observed current signal matrix in the estimated two-phase synchronous rotating coordinate system with the input After the difference is made, the difference is multiplied by the gain matrix G2, and the above result is combined with the input estimated observed back-EMF matrix in the two-phase synchronous rotating coordinate system Add together and finally get the observed back-EMF signal matrix in the next moment estimated two-phase synchronous rotating coordinate system After a delay of one shot, we get

[0082] The improved back-EMF observer is expressed as:

[0083]

[0084] in, is the observed dq axis current value at time k+1, is the observed dq axis current value at time k, is the observed value of the dq-axis back EMF at time k+1, is the value of the dq-axis back EMF at the observed time k, is the reference value of the dq axis voltage at time k, i dq (k) is the dq axis current value obtained by the current sensor collecting the three-phase current at time k after Park transformation. Matrices G1 and G2 are feedback gain matrices adaptive to the speed. A and B are coefficient matrices, I is the unit matrix, and O is the zero matrix.

[0085] The improved coefficient matrix A is expressed as:

[0086] A=A d1 +Ad2

[0087] Among them, A d1 is the constant coefficient matrix on the diagonal of the original coefficient matrix After first-order Euler discretization, we get A d2 is the speed coupling coefficient matrix on the anti-diagonal line of the original coefficient matrix After the Padre approximation,

[0088]

[0089] in,

[0090]

[0091]

[0092]

[0093]

[0094] The coefficient matrix B is expressed as

[0095] Furthermore, the feedback gain coefficient matrix is ​​set by directly configuring the poles in the z domain to ensure the convergence of the observer. By using the algebraic properties of the block matrix, the characteristic polynomial det(zI-A1+GL) of the back-EMF observer is improved to be expressed as:

[0096]

[0097] Let det(zI-A1+GL)=det((z-p1)(z-p2)I), and the expression of the feedback gain matrix is ​​obtained according to the coefficient matching principle:

[0098] G1=(1-p1-p2)I+A, G2=(1-p1)(1-p2)I,

[0099] Among them, matrix A is the coefficient matrix related to the speed, and the stability of the observer can be guaranteed by configuring the poles within the unit circle.

[0100] Reference Figure 3 As shown, in one embodiment, a high-power permanent magnet synchronous motor control device is proposed, which includes a Park converter, the above-mentioned improved back-EMF observer, a phase-locked loop, a current PI controller, a speed PI controller and an anti-Park converter with delay compensation; wherein,

[0101] The current signal input end of the Park converter is connected to the output signal of the UVW three-phase current sensor of the permanent magnet synchronous motor; the angle signal of the Park converter Angle between input and phase-locked loop The output end is connected; the current signal output of the Park converter is divided into the d-axis current signal i d With the q-axis current signal i q , the dq axis current signal is connected to the input terminal of the improved back-EMF observer;

[0102] The input end of the back-EMF observer is connected to the output end of the current PI controller and the output end of the Park converter respectively, and the output end of the back-EMF observer is connected to the input end of the phase-locked loop;

[0103] The output of the phase-locked loop estimates the speed With reference speed The output of the phase-locked loop estimates the speed. The output of the phase-locked loop estimates the rotor position angle and is connected to the complex vector decoupling current PI controller and the inverse Park transform input with delay compensation. connected to the input terminals of the complex vector decoupling current PI controller, the inverse Park transform with delay compensation, and the Park transform;

[0104] Current PI controller output connected to the input terminal of the inverse Park transformation with delay compensation;

[0105] Inverse Park transform output with delay compensation Connected to the input of the SVPWM inverter.

[0106] It should be noted that the description based on the back-EMF observer refers to the above embodiment and will not be repeated here.

[0107] Furthermore, the current PI controller is a complex vector decoupling current PI controller. Figure 4 As shown, the implementation method of the complex vector decoupling current PI controller is as follows, which consists of a d-axis PI, a d-axis integrator, a q-axis integrator, and a q-axis PI, and the proportional coefficient of the d-axis PI is k p_id , the integration coefficient is k i_id , the proportional coefficient of the q-axis PI is k p_iq , the integration coefficient is k i_iq , the integration coefficient of the d-axis integrator is k p_id and The coefficient of the q-axis integrator is k p_iq and The product of Equal to the difference between the output of the d-axis PI and the d-axis integrator, the output =Q-axis PI output plus Q-axis integrator output plus The product of ψ and ψ, where ψ is the magnetic flux of the permanent magnet, is a constant.

[0108] k p_id =L d *2π*f c , k i_id =R s *2π*f c , where L d is the d-axis inductance of the permanent magnet synchronous motor, R s is the stator resistance of the permanent magnet synchronous motor, f c Sets the cutoff frequency for the current loop bandwidth.

[0109] Furthermore, the compensation function of the anti-Park converter with delay compensation is expressed as:

[0110] Including angle delay compensation and amplitude compensation, is the angle delay compensation, For amplitude compensation, T s is the program control period in the present invention, which is numerically the same as the PWM switching period. is the electrical angular velocity of the motor rotor.

[0111] In an alternative embodiment, continue to refer to Figure 3 As shown, the high-power permanent magnet synchronous motor control device also includes: a current mean error compensation module, the input end of the current mean error compensation module is respectively connected to the output end of the complex vector decoupling current PI controller and the output of the phase-locked loop to estimate the speed connected, its output terminal i d 、i q and dq axis current signal i d 、i q The sum is calculated and the sum is compared with the reference current After making the difference, the difference is connected to the input of the complex vector decoupling current PI controller. The structural block diagram of the current mean error compensation module is shown in Figure 2. Figure 5 Show.

[0112] Furthermore, the current mean error compensation module uses the dq axis voltage reference value at the input end and the estimated rotor speed The current value of dq axis compensation is obtained through calculation; the current mean error compensation calculation formula is:

[0113]

[0114] Among them, i dq_com is the current value of dq axis compensation, is the reference value of dq axis voltage, T sTo control the cycle, is the estimated value of the motor rotor speed,

[0115] where k p_id =L d *2π*f c , k i_id =R s *2π*f c , where L d is the d-axis inductance of the permanent magnet synchronous motor, R s is the stator resistance of the permanent magnet synchronous motor, the k p_iq =L q *2π*f c , k i_iq =R s *2π*f c , f c Sets the cutoff frequency for the current loop bandwidth.

[0116] Reference Figure 6 As shown, the phase-locked loop includes a four-quadrant inverse tangent function, a PI controller and an integrator. The four-quadrant inverse tangent function atan2 is used to calculate the angle error value. Used to drive the phase-locked loop, Calculated by PI controller Further Integrate to get the motor rotor position angle The proportional coefficient in the PI controller is k p_PLL =2ξω0, the integral coefficient is The integration time constant in the integrator is T s Where ξ is the damping coefficient and ω0 is the phase-locked loop bandwidth.

[0117] In one embodiment, a control method based on the above-mentioned high-power permanent magnet synchronous motor control device is proposed, and the method includes:

[0118] Collect the UVW three-phase current signal of the permanent magnet synchronous motor and obtain the dq axis current value i through the Park converter dq ; Improve the dq axis current estimation link in the back EMF observer according to the dq axis current value i dq , voltage reference value Estimated value of back EMF at time k And the predicted dq axis current at time k Calculate the predicted value of the dq axis current at time k+1 Improve the dq axis back EMF estimation link in the back EMF observer according to the predicted value of the dq axis current at the next moment Predicted dq-axis current at time k And the reference value of voltage Calculate the estimated value of the back EMF at time k+1 Observations Obtaining the estimated motor rotor position through a phase-locked loop and speed The reference value of the dq axis voltage and the observed rotor speed information As the input of the current mean error compensation module, the current value i that needs to be compensated for the current regulation link is calculated dq_com ; The estimated speed With reference speed The q-axis reference current is obtained by using the speed PI controller. The dq axis current signal and the current value i compensated by the dq axis current signal dq_com The sum is calculated and then the difference is made with the reference current as the input of the complex vector decoupling current PI controller. The reference voltage of the dq axis is obtained by calculation. According to the dq axis reference voltage Speed ​​estimate and the rotor position estimate Inverse Park calculation with delay compensation to obtain the reference voltage of αβ axis SVPWM inverter based on Output three-phase voltage to the motor to achieve stable operation of the high-power permanent magnet synchronous motor in the full speed range.

[0119] It should be noted that the description of the high-power permanent magnet synchronous motor control device refers to the above embodiment and will not be repeated here.

[0120] Table 1 Motor parameters and experimental conditions settings

[0121] parameter Numeric parameter Numeric Pole pairs 4 Rated speed (rpm) 2000 Stator resistance Rs(Ω) 1.344 Permanent magnet flux (Wb) 0.267 D-axis inductance (mH) 5.838 Q-axis inductance (mH) 7.380 Switching frequency (kHz) 1 Sampling frequency (kHz) 1

[0122] Said Figure 7 This is a comparison chart of simulation results. At this time, the ratio of the switching frequency to the carrier of the motor working at the rated speed is 7.5, the starting given speed is set to 1500rpm, the starting load is 4N·m, it accelerates to 2000rpm in 1s, loads to 7N·m in 2s, and decelerates to 1500rpm in 3s. Figure 7 (a) is the result diagram of the proposed improved back-EMF observer with current mean error compensation link. Figure 7(b) is the result diagram of the proposed improved back-EMF observer without the current mean error compensation link. In each sub-diagram, from top to bottom, it is the motor speed, d-axis sampled current and estimated current, q-axis sampled current and estimated current, dq-axis estimated back-EMF and rotor position error angle. It can be seen from the experimental results that the proposed improved back-EMF observer can make the motor run stably under low carrier ratio conditions. Figure 7 (a) and Figure 7 (b) The error angle of the proposed current mean error compensation link effectively reduces the rotor position observation error.

[0123] In one embodiment, a high-power permanent magnet synchronous motor control system is proposed. The system includes: the high-power permanent magnet synchronous motor control device described above.

[0124] The description of the high-power permanent magnet synchronous motor control device is based on the above embodiment and will not be repeated here.

[0125] In the description provided herein, a large number of specific details are described. However, it is understood that embodiments of the present invention can be practiced without these specific details. In some instances, well-known methods, structures and techniques are not shown in detail so as not to obscure the understanding of this description.

[0126] In addition, those skilled in the art will understand that, although some embodiments described herein include certain features included in other embodiments but not other features, the combination of features of different embodiments also means being within the scope of protection of the present invention and forming different embodiments. For example, in the above embodiments, those skilled in the art can use them in combination according to the known technical solutions and the technical problems to be solved by this application.

[0127] The above is only a preferred embodiment of the present invention, and does not limit the present invention in any form. Although the present invention has been disclosed as a preferred embodiment as above, it is not used to limit the present invention. Any technician familiar with this patent can make some changes or modify the technical contents suggested above into equivalent embodiments without departing from the scope of the technical solution of the present invention. However, any simple modification, equivalent change and modification made to the above embodiments according to the technical essence of the present invention without departing from the content of the technical solution of the present invention still fall within the scope of the solution of the present invention.

Claims

1. A back electromotive force observer, characterized in that: The back-EMF observer improves the coefficient matrix of the permanent magnet synchronous motor back-EMF observer by using the Padé approximation method; The improved back-EMF observer converts the current signal in the dq coordinate system, the compensated voltage reference signal and the compensated voltage reference signal into the observed back-EMF; The improved back-electromotive force observer includes a dq axis current estimation link and a dq axis back-electromotive force estimation link, wherein the dq axis current estimation link is used to obtain an observation current signal matrix in a dq coordinate system at the next moment, and the dq axis back-electromotive force estimation link is used to obtain an observation back-electromotive force signal matrix in a dq coordinate system at the next moment; The improved back-EMF observer is expressed as: in, is the observed dq axis current value at time k+1, is the observed dq axis current value at time k, is the observed value of the dq-axis back EMF at time k+1, is the value of the dq-axis back EMF at the observed time k, is the reference value of the dq axis voltage at time k, i dq (k) is the dq axis current value obtained by the current sensor collecting the three-phase current at time k after Park transformation. The matrices G1 and G2 are feedback gain matrices adaptive to the speed. A and B are coefficient matrices. I is the unit matrix and O is the zero matrix. The coefficient matrix A is expressed as: A=A d1 +A d2 Among them, A d1 is the constant coefficient matrix on the diagonal of the original coefficient matrix After first-order Euler discretization, we get A d2 is the speed coupling coefficient matrix on the anti-diagonal line of the original coefficient matrix After the Padre approximation, in, The coefficient matrix B is expressed as: Among them, L d , L q is the dq axis inductance of the permanent magnet synchronous motor, R s is the stator resistance of the permanent magnet synchronous motor, ω e is the motor electrical angular velocity, T s is the control cycle of the control program.

2. A back electromotive force observer according to claim 1, characterized in that: The feedback gain coefficient matrix is ​​set by directly configuring the poles in the z domain to ensure the convergence of the observer. The characteristic polynomial det(zI-A1+GL) of the improved back-EMF observer is expressed as follows: Let det(zI-A1+GL)=det((z-p1)(z-p2)I), and the expression of the feedback gain matrix is ​​obtained according to the coefficient matching principle: G1=(1-p1-p2)I+A, G2=(1-p1)(1-p2)I, Among them, matrix A is the coefficient matrix related to the speed, and the stability of the observer can be guaranteed by configuring the poles within the unit circle.

3. A high-power permanent magnet synchronous motor control device, characterized in that: The device comprises: a Park converter, a back-EMF observer according to any one of claims 1 to 2, a phase-locked loop module, a current PI controller, a speed PI controller and an anti-Park converter with delay compensation; The input end of the Park converter is connected to the output signal of the permanent magnet synchronous motor UVW three-phase current sensor respectively, and the angle signal input end of the Park converter is connected to the angle output end of the phase-locked loop; the current signal output of the Park converter is divided into a d-axis current signal i d With the q-axis current signal i q , the dq axis current signal is connected to the input terminal of the improved back-EMF observer; The back-EMF observer input terminal is connected to the current PI controller output terminal and the Park converter output terminal respectively, and the back-EMF observer output terminal is connected to the phase-locked loop input terminal; The output end of the phase-locked loop estimates the rotational speed and the reference rotational speed, and then connects to the input end of the speed PI controller. The output end of the phase-locked loop estimates the rotational speed and connects to the input end of the current PI controller and the inverse Park transformation with delay compensation. The output end of the phase-locked loop estimates the rotor position angle and connects to the input end of the current PI controller, the inverse Park transformation with delay compensation, and the Park transformation. The output end of the current PI controller is connected to the input end of the inverse Park transformation with delay compensation; The output end of the inverse Park transformation with delay compensation is connected to the input end of the SVPWM inverter.

4. A high-power permanent magnet synchronous motor control device according to claim 3, characterized in that: The current PI controller is a complex vector decoupling current PI controller; the complex vector decoupling current PI controller includes a d-axis PI, a d-axis integrator, a q-axis integrator and a q-axis PI; the proportional coefficient of the d-axis PI is k p_id , the integration coefficient is k i_id , the proportional coefficient of the q-axis PI is k p_iq , the integration coefficient is k i_iq , the integration coefficient of the d-axis integrator is k p_id and The coefficient of the q-axis integrator is k p_iq and The product of Equal to the difference between the output of the d-axis PI and the d-axis integrator, the output =Q-axis PI output plus Q-axis integrator output plus The product of ψ and ψ, where ψ is the magnetic flux of the permanent magnet, which is a constant; The k p_id =L d *2π*f c , k i_id =R s *2π*f c , where L d is the d-axis inductance of the permanent magnet synchronous motor, R s is the stator resistance of the permanent magnet synchronous motor, f c Sets the cutoff frequency for the current loop bandwidth.

5. A high-power permanent magnet synchronous motor control device according to claim 3, characterized in that: The compensation function of the anti-Park converter with delay compensation is expressed as: Contains compensation for angle delay and amplitude compensation; is the angle delay compensation, For amplitude compensation, T s is the program control period in the present invention, which is numerically the same as the PWM switching period. is the electrical angular velocity of the motor rotor.

6. A high-power permanent magnet synchronous motor control device according to claim 4, characterized in that: The high-power permanent magnet synchronous motor control device also includes: a current mean error compensation module, whose input end is respectively connected to the output end of the complex vector decoupling current PI controller and the output end of the phase-locked loop to estimate the speed, and whose output end is summed with the dq axis current signal, and the summation result is subtracted from the reference current and the difference is connected to the input end of the complex vector decoupling current PI controller.

7. A high-power permanent magnet synchronous motor control device according to claim 6, characterized in that: The current mean error compensation module uses the dq axis voltage reference value at the input end and the estimated rotor speed The current value of dq axis compensation is obtained through calculation; the current mean error compensation calculation formula is: Among them, i dq_com is the current value of dq axis compensation, is the reference value of dq axis voltage, T s To control the cycle, is the estimated value of the motor rotor speed, 8. A high-power permanent magnet synchronous motor control device according to claim 3, characterized in that: The phase-locked loop includes a four-quadrant inverse tangent function, a PI controller and an integrator; the four-quadrant inverse tangent function atan2 is used to calculate the angle error value Angle error value Used to drive the phase-locked loop, angle error value Calculated by PI controller right Integrate to get the motor rotor position angle The proportional coefficient in the PI controller is k p_PLL =2ξω0, the integral coefficient is The integration time constant in the integrator is T s , where ξ is the damping coefficient and ω is the phase-locked loop bandwidth.

9. A control method for a high-power permanent magnet synchronous motor control device according to any one of claims 3 to 8, characterized in that: The method comprises: Collect the UVW three-phase current signal of the permanent magnet synchronous motor and obtain the dq axis current value i through the Park converter dq ; Improve the dq axis current estimation link in the back EMF observer according to the dq axis current value i dq , voltage reference value Estimated value of back EMF at time k And the predicted dq axis current at time k Calculate the predicted value of the dq axis current at time k+1 Improve the dq axis back EMF estimation link in the back EMF observer according to the predicted value of the dq axis current at the next moment Predicted dq-axis current at time k And the reference value of voltage Calculate the estimated value of the back EMF at time k+1 Observations Obtaining the estimated motor rotor position through a phase-locked loop and speed The reference value of the dq axis voltage and the observed rotor speed information As the input of the current mean error compensation module, the current value i that needs to be compensated for the current regulation link is calculated dq_com ; The estimated speed With reference speed The q-axis reference current is obtained by using the speed PI controller. The current value i of the dq axis current signal and the dq axis current signal compensation dq_com The sum is calculated and then the difference is made with the reference current as the input of the complex vector decoupling current PI controller. The reference voltage of the dq axis is obtained by calculation. According to the dq axis reference voltage Speed ​​estimate and the rotor position estimate Inverse Park calculation with delay compensation to obtain the reference voltage of αβ axis The SVPWM inverter is based on the reference voltage of the αβ axis Output three-phase voltage to the motor to achieve stable operation of the high-power permanent magnet synchronous motor in the full speed range.

10. A high-power permanent magnet synchronous motor control system, characterized in that: The system comprises: a high-power permanent magnet synchronous motor control device as described in any one of claims 3 to 8.

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