Vehicle-mounted bidirectional power converter

Abstract

The invention discloses a vehicle-mounted bidirectional power converter which comprises a pre-stage voltage stabilization module, a pre-stage driving circuit, a post-stage driving circuit, a post-stage filtering module and an electric vehicle battery module, the pre-stage voltage stabilization module is used for balancing circuit voltage and supporting reactive power, the pre-stage driving circuit is of a single-phase full-bridge AD-DC topological structure, and the post-stage filtering module is of a single-phase full-bridge AD-DC topological structure. The front-stage driving circuit is of a half-bridge type bidirectional DC-DC topological structure, the rear-stage driving circuit is of a half-bridge type bidirectional DC-DC topological structure, the front-stage driving circuit and the rear-stage driving circuit are connected together through a direct-current voltage-stabilizing bus, the size is small, the weight is light, a front stage and a rear stage do not interfere with each other, control is easy, the rear-stage filtering module is used for restraining direct-current side second harmonics, the harmonic quantity is prevented from influencing the stability of output direct-current voltage, and the output direct-current voltage is stable. According to the converter, a corresponding control method is designed, non-inductive decoupling control and a single-phase phase-locked loop of an improved loop filter are adopted, the phase of a power grid is accurately and efficiently obtained, and the converter works stably.

Description

Technical Field

[0001] The invention relates to the technical field of electric vehicle power control, in particular to a vehicle-mounted bidirectional power converter. Background Art

[0002] With the number of electric vehicles increasing year by year, a large number of electric vehicles have become a form of mobile energy storage. Electric vehicles can be used as virtual power plants to regulate the power grid. During peak periods, they can send electricity back to the grid, and during low periods, they can obtain electricity from the grid for storage. When a large number of electric vehicles are effectively dispatched, they can adjust the peak and valley differences of the grid and maintain the stability of the grid. However, the electric vehicles currently on the market can only transmit electricity to the grid through charging piles. Due to the high cost and large area of ​​charging piles, it is difficult for ordinary consumers to install them, so it is difficult to build them on a large scale and match them with vehicles. Large electric vehicles cannot effectively participate in grid dispatching. Therefore, how to achieve low-cost, convenient and fast transmission of electricity between electric vehicles and the grid has become a research direction.

[0003] In the process of electric vehicles participating in the energy interaction of the power grid, the core component is the bidirectional power converter. As a channel connecting the power grid and the energy flow of electric vehicles, it must not only meet the bidirectional power flow, but also be able to quickly respond to the grid dispatching commands and output in accordance with the power quality requirements. Summary of the invention

[0004] In order to overcome the above problems, the purpose of the present invention is to provide a vehicle-mounted bidirectional power converter, which has a corresponding control method for its overall topological structure design. The front stage adopts an inductorless decoupling control strategy and a single-phase phase-locked loop with an improved loop filter to accurately and efficiently obtain the grid phase, and can make the converter work stably under the condition of grid distortion. The front-stage drive circuit and the rear-stage drive circuit are connected together through a DC voltage-stabilizing bus, eliminating the heavy transformer, and have a small size and light weight. The front and rear stages do not interfere with each other and the control is simple.

[0005] The technical solution adopted by the present invention is: comprising a front-stage voltage stabilizing module, a front-stage driving circuit, a rear-stage driving circuit, a rear-stage filtering module, and an electric vehicle battery module, wherein the front-stage voltage stabilizing module is used to balance the circuit voltage and support reactive power, the front-stage driving circuit is a single-phase full-bridge AD-DC topology structure, the rear-stage driving circuit is a half-bridge bidirectional DC-DC topology structure, the front-stage driving circuit and the rear-stage driving circuit are connected together through a DC voltage stabilizing bus, and the rear-stage filtering module is used to suppress the second harmonic on the DC side to prevent the harmonic amount from affecting the stability of the output DC voltage and causing low-frequency voltage pulsation on the DC side.

[0006] Furthermore, the front-stage driving circuit includes power tubes IGBT1, IGBT2, IGBT3, IGBT4, inductors L1, L2, resistors R1, R2, and capacitor C1. The IGBT1 and IGBT4 form a pair of bridge arms, and the IGBT2 and IGBT3 form another pair of bridge arms. The triangular wave pulse width modulation method is used to control the opening and closing of each pair of bridge arms to achieve forward rectification and reverse inversion.

[0007] Furthermore, the triangular wave pulse width modulation method adopts unipolar triangular wave pulse width modulation, in which only two switch tubes are in high-frequency switch working state in half a cycle, and the other two are in normally closed or normally open state, and the modulation voltage is from U d Change to 0, and then change from 0 to -U d .

[0008] Furthermore, the unipolar triangular wave pulse width modulation uses an isosceles triangle wave as a carrier and a sine wave as a modulation signal wave. By comparing the two waveforms, in the positive half cycle, when the amplitude of the sine wave modulation signal is greater than the amplitude of the triangular wave carrier, the pulse signal outputs a high level, and when the amplitude of the sine wave modulation signal is less than the amplitude of the triangular wave carrier, the pulse signal outputs a low level; in the negative half cycle, when the amplitude of the triangular wave carrier is greater than the amplitude of the sine wave modulation signal, the pulse signal outputs a high level, and when the amplitude of the triangular wave carrier is less than the amplitude of the sine wave modulation signal, the pulse signal outputs a low level, and a wide and narrow pulse with the same impulse as the sine wave but different shape is obtained, which is a triangular wave pulse width modulation waveform.

[0009] Furthermore, the front-stage driving circuit obtains three AC side voltages by controlling four switch tubes IGBT respectively, and the specific steps are as follows:

[0010] Step 1: Simplify the rectifier circuit and obtain its mathematical model, where E s is the electromotive force on the AC side, i s is the AC side current, L is the grid side inductance, R is the AC side resistance, u AB is the AC side voltage, i dc is the DC side current, u dc is the DC side voltage, R L is the load resistance, E L is the load electromotive force;

[0011] Step 2: Define the switch function as:

[0012]

[0013] Step 3: Control the on and off of the switch tube IGBT. The switch tubes of the same bridge arm cannot be turned on at the same time, otherwise it will cause a short circuit. Therefore, the on state of the IGBT switch tube is:

[0014]

[0015] Step 4: Get the function expression of three voltages and switch tubes:

[0016] u AB =(S A -S B ) dc

[0017] Furthermore, the post-stage drive circuit includes power tubes IGBT5 and IGBT6, inductors L3 and L4, and capacitor C2. The post-stage drive circuit operates in Buck and Boost states, and realizes bidirectional energy flow by controlling the alternating conduction of the switch tubes. When the switch tube IGBT5 is turned on, the switch tube IGBT6 is in the off state and is in the Buck working mode; when the switch tube IGBT6 is turned on, IGBT5 is in the off state and is in the Boost working mode. The waveforms of the power tubes IGBT5 and IGBT6 are complementary, and a dead time is set to prevent short circuit.

[0018] Furthermore, there are two loops in the Buck working mode:

[0019] In the first state of the circuit, the pulse width modulation drive signal turns on the upper bridge arm switch tube IGBT5 and turns off the lower bridge arm switch tube IGBT6. The rectified current on the AC side is supplied to the inductors L3, L4 and the battery V through the switch tube IGBT5. bat To charge, the inductor stores energy in the form of a magnetic field;

[0020] In the second loop state: the lower bridge arm switch tube IGBT6 is turned on, the upper bridge arm IGBT5 is turned off, the current flows through the synchronous diode of IGBT6, the inductor L3, L4, the synchronous diode of IGBT6 and the battery V bat A circuit is formed, the inductor provides energy to the battery, the energy stored in the inductor decreases, and the charging process is completed.

[0021] Furthermore, there are two loops in the Boost working mode:

[0022] In the third state of the circuit: the battery output voltage is increased to the DC bus voltage through the boost chopper, and then fed back to the AC side through the inverter, the driving signal turns on the lower bridge arm IGBT6 and turns off the upper bridge arm IGBT5, and the battery V bat Provide energy to inductors L3 and L4, which store magnetic field energy;

[0023] In the fourth state of the circuit: the upper bridge arm IGBT5 is turned on, the lower bridge arm IGBT6 is turned off, the current is transmitted through the synchronous diode of IGBT5, and the battery V bat Together with inductors L3, L4 and the synchronous diode of IGBT5, the battery Vbat And inductors L3 and L4 simultaneously supply power to the DC bus V dc The energy is fed back, the energy stored in the inductor decreases, and the discharge process begins.

[0024] Furthermore, the bidirectional power converter adopts an inductorless decoupling control method based on SOGI, which realizes orthogonal signal construction through a second-order generalized integrator, adopts a continuous integration inductorless decoupling method, and uses a single-phase phase-locked loop with an improved loop filter for grid-connected control.

[0025] Beneficial effects of the present invention:

[0026] 1. It can complete the process of charging the car power battery from the household power grid to realize the G2V function, and it can also complete the process of feeding the electric vehicle power battery to the residential power grid to realize the V2G function, so that electric vehicles can participate in the energy management and dispatch of the power system through the residential power grid, and realize the construction of a new power system.

[0027] 2. The front-stage drive circuit adopts a single-phase full-bridge AD-DC topology structure to achieve high power factor operation and generate fewer harmonics. The rear-stage drive circuit adopts a half-bridge bidirectional DC-DC topology structure. Through energy storage inductors and filter capacitors, the load current ripple amplitude is reduced, so that the output voltage can be higher or lower than the input voltage, which reduces the voltage stress on the switch tube, the switch tube has high safety performance, a simple circuit structure, a small number of components used, and low cost.

[0028] 3. The bidirectional power converter adopts an inductorless decoupling control method based on SOGI and realizes orthogonal signal construction through a second-order generalized integrator. Compared with the traditional delay construction method, the system response rate is improved. The continuous integration inductorless decoupling method is adopted to avoid insufficient feedforward decoupling caused by inductor saturation, thereby effectively improving the pulse width modulation control accuracy. A single-phase phase-locked loop with an improved loop filter is used for grid-connected control to synchronize the access system voltage with the grid voltage.

[0029] 4. The front-stage drive circuit and the rear-stage drive circuit are connected together through a DC voltage-stabilizing bus to achieve a wider output voltage range and higher power quality, while achieving light weight, small size, and high efficiency, thereby minimizing the weight of the vehicle and reducing the cost of the vehicle. BRIEF DESCRIPTION OF THE DRAWINGS

[0030] Figure 1 A circuit diagram of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0031] Figure 2 A structural diagram of a front-stage driving circuit of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0032] Figure 3A structural diagram of a rear-stage driving circuit of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0033] Figure 4 A schematic diagram of unipolar triangular wave pulse width modulation of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0034] Figure 5 A simplified rectifier circuit model diagram of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0035] Figure 6 A schematic diagram of a rear-stage drive circuit loop of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0036] Figure 7 This is a schematic diagram of a second circuit of a rear-stage driving circuit of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0037] Figure 8 The third schematic diagram of the rear-stage drive circuit loop of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0038] Fig. 9 A fourth schematic diagram of a rear-stage drive circuit loop of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0039] Fig.10 This is a unipolar triangle wave pulse width engineering modulation diagram of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0040] Fig.11 This is a working principle diagram of a front-stage driving circuit of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0041] Fig.12 This is the simulation result of a voltage-frequency mutation phase-locked loop of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0042] Fig.13 This is the simulation result of a voltage phase mutation phase-locked loop of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0043] Fig.14 This is a simulation result of a voltage amplitude mutation phase-locked circuit for a vehicle-mounted bidirectional power converter proposed by the present invention;

[0044] Fig.15 The interference phase-locking result of a vehicle-mounted bidirectional power converter containing direct current and harmonics proposed by the present invention;

[0045] Fig.16 This is a simulation result diagram of the inductor decoupling of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0046] Fig.17This is a unidirectional charging and discharging simulation diagram of a vehicle-mounted bidirectional power converter proposed by the present invention;

[0047] Fig.18 This is a bidirectional charging and discharging simulation diagram of a vehicle-mounted bidirectional power converter proposed by the present invention. DETAILED DESCRIPTION

[0048] The specific implementation of the present invention is described below in conjunction with the accompanying drawings and embodiments:

[0049] It should be noted that the structures, proportions, sizes, etc. illustrated in the drawings of this specification are only used to match the contents disclosed in the specification for people familiar with this technology to understand and read, and are not used to limit the conditions under which the present invention can be implemented. Any structural modification, change in proportional relationship or adjustment of size should still fall within the scope of the technical contents disclosed in the present invention without affecting the effects and purposes that can be achieved by the present invention.

[0050] At the same time, the terms such as "upper", "lower", "left", "right", "middle" and "one" cited in this specification are only for the convenience of description and are not used to limit the scope of implementation of the present invention. Changes or adjustments to their relative relationships should be regarded as the scope of implementation of the present invention without substantially changing the technical content.

[0051] like Figures 1 to 18 As shown, it shows a specific implementation of the present invention: a vehicle-mounted bidirectional power converter disclosed in the present invention includes a front-stage voltage stabilization module, a front-stage drive circuit, a rear-stage drive circuit, a rear-stage filter module, and an electric vehicle battery module. The front-stage voltage stabilization module is used to balance the circuit voltage and support reactive power. The front-stage drive circuit is a single-phase full-bridge AD-DC topology structure, and the rear-stage drive circuit is a half-bridge bidirectional DC-DC topology structure. The front-stage drive circuit and the rear-stage drive circuit are connected together through a DC voltage stabilization bus. The rear-stage filter module is used to suppress the second harmonic on the DC side to prevent the harmonic amount from affecting the stability of the output DC voltage and causing low-frequency voltage pulsation on the DC side.

[0052] In the present invention, Figure 1 As shown, the front-stage drive circuit adopts a single-phase full-bridge AD-DC topology structure to achieve high power factor operation and generate fewer harmonics. The rear-stage drive circuit adopts a half-bridge bidirectional DC-DC topology structure. Through energy storage inductors and filter capacitors, the load current ripple amplitude is reduced, so that the output voltage can be higher or lower than the input voltage, so that the voltage stress on the switch tube is smaller, the switch tube has high safety performance, simple circuit structure, small number of devices used, and low cost. The front-stage drive circuit and the rear-stage drive circuit are connected together through a DC voltage-stabilizing bus, eliminating the heavy transformer, small size, light weight, front and rear stages do not interfere with each other, and the control is simple.

[0053] Specifically, the front-stage driving circuit includes power tubes IGBT1, IGBT2, IGBT3, IGBT4, inductors L1, L2, resistors R1, R2, and capacitor C1. The IGBT1 and IGBT4 form a pair of bridge arms, and the IGBT2 and IGBT3 form another pair of bridge arms. The triangular wave pulse width modulation method is used to control the opening and closing of each pair of bridge arms to achieve forward rectification and reverse inversion.

[0054] Specifically, the triangular wave pulse width modulation method adopts unipolar triangular wave pulse width modulation. In half a cycle, only two switch tubes are in the high-frequency switch working state, and the other two are in the normally closed or normally open state. The modulation voltage is from U d Change to 0, and then change from 0 to -U d .

[0055] Specifically, the unipolar triangular wave pulse width modulation uses an isosceles triangle wave as a carrier and a sine wave as a modulation signal wave. By comparing the two waveforms, in the positive half cycle, when the amplitude of the sine wave modulation signal is greater than the amplitude of the triangular wave carrier, the pulse signal outputs a high level, and when the amplitude of the sine wave modulation signal is less than the amplitude of the triangular wave carrier, the pulse signal outputs a low level; in the negative half cycle, when the amplitude of the triangular wave carrier is greater than the amplitude of the sine wave modulation signal, the pulse signal outputs a high level, and when the amplitude of the triangular wave carrier is less than the amplitude of the sine wave modulation signal, the pulse signal outputs a low level, and a wide and narrow pulse with the same impulse as the sine wave but different shape is obtained, which is a triangular wave pulse width modulation waveform.

[0056] In the present invention, Figure 2 , Figure 4 As shown, in the unipolar modulation half cycle, only two switch tubes are in the high-frequency switching working state, and the other two are in the normally closed or normally open state. The switching loss is relatively small, and the level jump from U d Change to 0, and then change from 0 to -U d , the level jump is small, the leakage current is also small, and the harmonic content and voltage pulsation can be reduced to the greatest extent.

[0057] In actual use, the unipolar triangle wave pulse width modulation process cannot change the polarity of the triangle wave according to the polarity of the sine wave to keep it consistent. Therefore, a sine wave u is introduced in the modulation process. r Auxiliary modulation wave with a phase difference of 180°-u r , compare the two with the triangular carrier, and finally get the triangular pulse width modulation wave.

[0058] The four switch tubes are controlled by four drive signals respectively, wherein the same-side bridge arm drive signal is a set of complementary triangular pulse width modulation waves, such as Fig.10 As shown, the sinusoidal modulation wave ur and the triangular carrier u c The obtained triangular pulse width modulation wave is used to drive IGBT1 and IGBT3. r Greater than u c When the IGBT1 drive signal is high, it is in the on state, and the IGBT3 drive signal is low, it is in the off state. Fig.10 The voltage at point A is u A =U dc ; when u r Less than u c When the IGBT1 drive signal is low, it is in the off state, and the IGBT3 drive signal is high, it is in the on state. At this time, the voltage u A =0;

[0059] Use auxiliary modulation wave -u r and the triangular carrier u c The obtained triangular pulse width modulation wave is used to drive IGBT2 and IGBT4. r Greater than u c When the IGBT2 drive signal is high, it is in the on state, and the IGBT4 drive signal is low, it is in the off state. Fig.10 The voltage at point B is u B =U dc ; when -u r Less than u c When the IGBT2 drive signal is low, it is in the off state, and the IGBT4 drive signal is high, it is in the on state. At this time, the voltage at point B is u B =0, so the voltage between points AB on the AC side of the single-phase rectifier bridge is u AB =u A -u B , the voltage value is U dc ,0,-U dc .

[0060] Specifically, the front-stage driving circuit obtains three AC side voltages by controlling four switch tubes IGBT respectively. The specific steps are as follows:

[0061] Step 1: Simplify the rectifier circuit and obtain its mathematical model, such as Figure 5 As shown, where E s is the electromotive force on the AC side, i s is the AC side current, L is the grid side inductance, R is the AC side resistance, u AB is the AC side voltage, i dc is the DC side current, u dc is the DC side voltage, R L is the load resistance, E Lis the load electromotive force;

[0062] Step 2: Define the switch function as:

[0063]

[0064] Step 3: Control the on and off of the switch tube IGBT. The switch tubes of the same bridge arm cannot be turned on at the same time, otherwise it will cause a short circuit. Therefore, the on state of the IGBT switch tube is:

[0065]

[0066] Step 4: Get the function expression of three voltages and switch tubes:

[0067] u AB =(S A -S B ) dc

[0068] In the present invention, the front-stage driving circuit controls the four switch tubes to turn on and off, and there are four working states. Taking the positive half axis of the AC voltage source as an example, the current reference direction is set to be the same as the direction from the positive pole to the negative pole of the power supply. When the actual current I ac When it is consistent with the reference direction, it is positive, otherwise it is negative. The eight circuits corresponding to the four working states are as follows:

[0069] Working state 1: S A =1,S B =0, that is, IGBT1 and IGBT4 are turned on;

[0070] Circuit 1: If Fig.11 As shown in (a), the current is conducted through the freewheeling diodes of the switch tubes IGBT1 and IGBT4. The current direction on the AC side is positive. The AC side circuit (V, L, R), the freewheeling diodes (D1, D4), and the DC side circuit (C1, R) form a loop. The current I ac It flows through the inductor L1, the resistor R1, and then through the freewheeling diode D1 of IGBT1 to charge the capacitor C1. At the same time, it flows through the load, and then passes through the freewheeling diode D4 of IGBT4, flows through the resistor R2 and the inductor L2 back to the grid, completing the rectification process. In this process, the power supply provides energy to the inductor, capacitor and load, and the inductor energy increases.

[0071] Circuit 2: If Fig.11 As shown in (b), the current is conducted through the switch tubes IGBT1 and IGBT4, and the current direction is negative. The AC side circuit, the switch tubes IGBT1, IGBT4, and the DC side circuit form a loop. The current I acThe current passes through the inductor L2, the resistor R2, and then flows through the capacitor C1 through the IGBT4. The bus terminal capacitor C1 discharges and feeds the current back to the grid through the IGBT1, completing the energy feedback process. In this process, the power supply and the capacitor provide energy to the inductor, the inductor energy decreases, and the capacitor provides energy to the load at the same time.

[0072] Working state 2: S A =1,S B =1, that is, IGBT1 and IGBT2 are turned on;

[0073] Circuit 3: If Fig.11 As shown in (c), when the current passes through the freewheeling diode of the switch tube IGBT1 and IGBT2 to be turned on, the current direction is positive, the AC side circuit, the freewheeling diode D1, and the switch tube IGBT2 form a loop, the AC power supply is in a short-circuited state, the current does not flow through the branch where the capacitor is located, and there is no energy exchange between the AC side and the DC side. In this process, the inductor energy increases and is stored in the form of a magnetic field. The energy required by the load is provided by the capacitor.

[0074] Circuit 4: If Fig.11 As shown in (d), when the current is turned on through the freewheeling diodes of the switch tubes IGBT1 and IGBT2, the current direction is negative. The AC side circuit, the switch tube IGBT1, and the freewheeling diode D2 form a loop. The capacitor C1 and the load form a loop. There is no energy exchange between the AC side and the DC side. The inductor energy is reduced, and the capacitor provides energy to the load.

[0075] Working state 3: S A =0,S B =0, that is, IGBT3 and IGBT4 are turned on;

[0076] Circuit 5: If Fig.11 As shown in (e), when the current passes through the freewheeling diodes of the switch tubes IGBT3 and IGBT4 and is turned on, the current direction is positive, and the AC side circuit, the switch tube IGBT3, and the freewheeling diode D4 form a loop. There is no energy exchange between the AC side and the DC side, the AC power supply is short-circuited, the inductor energy increases, and the load is provided with energy by the capacitor C1.

[0077] Circuit 6: If Fig.11 As shown in (f), when the current passes through the freewheeling diode of the switch tube IGBT3 and IGBT4 to be turned on, the current direction is negative, and the AC side circuit, the freewheeling diode D4, and the switch tube IGBT4 form a loop. The inductor energy is reduced, and the load is supplied with energy by the capacitor. There is no energy exchange between the AC side and the DC side.

[0078] Working status 4: S A =0,S B =1, that is, IGBT2 and IGBT3 are turned on;

[0079] Circuit 7: If Fig.11 As shown in (g), the current is turned on through the switch tubes IGBT2 and IGBT3, and the actual current direction is positive. The AC side circuit, IGBT2 and IGBT3, and the DC side circuit form a loop. The current I ac The current flows through the inductor L1, the resistor R1, and then flows to the capacitor branch through IGBT3. The DC side capacitor C1 is discharged. The current then flows through the switch tube IGBT2, flows through the resistor R2, and flows back to the power supply through the inductor L2. The energy of the inductor magnetic field increases, the power supply and the capacitor provide energy to the inductor, and the capacitor provides energy to the load.

[0080] Circuit 8: If Fig.11 As shown in (h), the current is turned on through the freewheeling diode D2 of IGBT2 and the freewheeling diode D3 of IGBT3. The actual current direction is negative. The AC side circuit, the freewheeling diode D2, the freewheeling diode D3, and the DC side circuit form a loop. The magnetic field energy stored in the inductor is reduced. The power supply and the inductor provide energy to the energy storage capacitor C1 and the load at the same time.

[0081] Specifically, Figure 3 As shown, the post-stage driving circuit includes power tubes IGBT5 and IGBT6, inductors L3 and L4, and capacitor C2. The post-stage driving circuit works in Buck and Boost states, and realizes bidirectional energy flow by controlling the alternating conduction of the switch tubes. When the switch tube IGBT5 is turned on, the switch tube IGBT6 is in the off state and is in the Buck working mode; when the switch tube IGBT6 is turned on, IGBT5 is in the off state and is in the Boost working mode. The waveforms of the power tubes IGBT5 and IGBT6 are complementary, and a dead time is set to prevent short circuit.

[0082] Specifically, there are two loops in the Buck working mode, such as Figure 6 , Figure 7 As shown:

[0083] In the first state of the circuit, the pulse width modulation drive signal turns on the upper bridge arm switch tube IGBT5 and turns off the lower bridge arm switch tube IGBT6. The rectified current on the AC side is supplied to the inductors L3, L4 and the battery V through the switch tube IGBT5. bat To charge, the inductor stores energy in the form of a magnetic field;

[0084] In the second loop state: the lower bridge arm switch tube IGBT6 is turned on, the upper bridge arm IGBT5 is turned off, the current flows through the synchronous diode of IGBT6, the inductor L3, L4, the synchronous diode of IGBT6 and the battery V bat A circuit is formed, the inductor provides energy to the battery, the energy stored in the inductor decreases, and the charging process is completed.

[0085] Specifically, there are two loops in the Boost working mode, such as Figure 8 , Fig. 9 As shown:

[0086] In the third state of the circuit: the battery output voltage is increased to the DC bus voltage through the boost chopper, and then fed back to the AC side through the inverter, the driving signal turns on the lower bridge arm IGBT6 and turns off the upper bridge arm IGBT5, and the battery V bat Provide energy to inductors L3 and L4, which store magnetic field energy;

[0087] In the fourth state of the circuit: the upper bridge arm IGBT5 is turned on, the lower bridge arm IGBT6 is turned off, the current is transmitted through the synchronous diode of IGBT5, and the battery V bat Together with inductors L3, L4 and the synchronous diode of IGBT5, the battery V bat And inductors L3 and L4 simultaneously supply power to the DC bus V dc The energy is fed back, the energy stored in the inductor decreases, and the discharge process begins.

[0088] Specifically, the bidirectional power converter adopts an inductorless decoupling control method based on SOGI. The strategy realizes orthogonal signal construction through a second-order generalized integrator, adopts a continuous integration inductorless decoupling method, and uses a single-phase phase-locked loop with an improved loop filter for grid-connected control.

[0089] In the present invention, the bidirectional power converter adopts an inductorless decoupling control method based on SOGI, and realizes orthogonal signal construction through a second-order generalized integrator. Compared with the traditional delay construction method, the system response rate is improved. The continuous integration inductorless decoupling method is adopted to avoid insufficient feedforward decoupling caused by inductor saturation, thereby effectively improving the pulse width modulation control accuracy. The single-phase phase-locked loop with an improved loop filter is used for grid-connected control, so that the access system voltage is synchronized with the grid voltage.

[0090] Example

[0091] On the Matlab / Simulink simulation platform, the phase-locked effect of the improved single-phase phase-locked loop is simulated and verified under the conditions of sudden changes in voltage frequency, phase, amplitude, and simultaneous injection of DC components and harmonic components.

[0092] Grid distortion phase-lock verification

[0093] Simulation environment settings: grid input voltage amplitude is 100V, frequency is 50Hz, and initial phase is 0rad. In order to observe the relationship between the phase-locked phase and input voltage, the phase-locked phase is enlarged 10 times during simulation. When the grid voltage frequency changes suddenly, the phase-locked loop simulation results are as follows: Fig.12As shown in the figure, when t = 0.03s, the voltage frequency suddenly changes from 50Hz to 53Hz. It can be seen from the simulation diagram that after about 0.06s, the phase-locked loop tracks the phase of the upper grid, and the output voltage waveform is as follows Fig.12 As shown in (a), Fig.12 (b) is the phase-locked loop phase and grid voltage waveform results. When the mutation occurs, the voltage waveform and phase waveform of the next cycle deviate. After about three cycles, the phase waveform and the voltage waveform zero point coincide, and the phase lock is completed. Fig.12 (c) shows the error of the phase of the traditional single-phase phase-locked loop (O-PLL) and the modified single-phase phase-locked loop (M-PLL). Under the distorted working condition, the phase of the traditional phase-locked loop and the modified phase-locked loop is compared with the grid phase to obtain the phase-locked error diagram. Fig.12 (c) From the error comparison, it can be seen that the error waveform of the improved single-phase phase-locked loop is smooth, close to zero, and the phase error is small; the error waveform of the traditional single-phase phase-locked loop fluctuates greatly, and compared with the improved phase-locked loop, it is a certain distance away from the zero point of the vertical coordinate, and the error is slightly larger.

[0094] When the voltage phase changes suddenly, the phase-locked loop simulation results are as follows Fig.13 As shown in the figure, similarly, when time t = 0.03s, the phase suddenly changes by 90 degrees, and the voltage waveform changes significantly. After 0.06s, the output waveform of the phase-locked loop is consistent with the grid. In terms of error, when the sudden change occurs, the phase of the phase-locked loop cannot immediately track the grid phase, and the waveform shows a significant protrusion. After phase-locked adjustment, it gradually returns to near zero. When the phase of the improved phase-locked loop jumps, the error waveform is relatively gentle, and the phase-locked error is still small, which is slightly better than the traditional phase-locked loop. Fig.13 (c).

[0095] When the voltage amplitude suddenly changes, the phase-locked simulation results are as follows: Fig.14 As shown in the figure, it can be seen that the influence of the sudden change of the grid amplitude on the phase is not very obvious, and accurate phase locking can be completed in a short time. The error of the improved phase-locked loop is not much different from that of the traditional phase-locked loop, which is basically the same as shown in the figure. Fig.14 As shown in (c), by comparison, it can be concluded that the improved phase-locked loop has higher phase-locking accuracy when common distortion in the power grid occurs.

[0096] Phase-locked verification of simultaneous DC and harmonic injection into the power grid

[0097] Simulation settings, when t = 0.03s, the power grid simultaneously injects 20% of the fifth harmonic and 10% of the DC component. The simulation results are as follows: Figure 3-4As shown in the figure. After about 0.03s of simultaneous injection of DC and harmonic components, the phase-locked output voltage waveform of the traditional single-phase phase-locked loop deviates from the grid voltage, and the offset becomes larger and larger, eventually leading to phase-lock failure. After the DC and harmonic components are injected, the improved single-phase phase-locked loop can still track the grid voltage and phase, and phase-lock is successful.

[0098] like Fig.15 As shown in the figure, the error comparison between the traditional single-phase phase-locked loop and the improved single-phase phase-locked loop shows that the traditional single-phase phase-locked loop has a large phase error and a certain volatility under the interference of DC and harmonics; the improved single-phase phase-locked loop has a phase error close to zero, which is significantly smaller than the traditional phase-locked loop, with small volatility and is relatively stable. It can be seen that the single-phase phase-locked loop proposed in this paper is significantly better than the traditional single-phase phase-locked loop when the DC harmonic component of the power grid interferes.

[0099] Through the above analysis and comparison, it can be seen that the single-phase phase-locked loop of the improved loop filter is not only slightly better than the traditional single-phase phase-locked loop in terms of phase, amplitude and frequency distortion of the grid voltage, but also can achieve the tracking effect; and it can still achieve the ideal phase-locked accuracy when the grid contains both DC and harmonic disturbances, which shows that it has excellent adaptability and good dynamic performance. It can provide accurate phase information and grid-connected conditions for the bidirectional power converter of electric vehicles. It can enable the bidirectional power converter of electric vehicles to have a wider range of application scenarios, effectively improve the anti-interference ability of the converter, and make it stable.

[0100] Electric Vehicle Bidirectional Power Converter Charging and Discharging Simulation

[0101] The electric vehicle bidirectional power converter model is built on the Matlab / Simulink simulation platform, and the traditional inductor decoupling control and the improved inductorless decoupling control are simulated respectively. The improved control strategy is used for three different charging modes of electric vehicles: unidirectional charging and discharging, bidirectional constant current charging and discharging, and bidirectional variable current charging and discharging, to verify the feasibility of the inductorless decoupling control strategy proposed in this paper. The simulation system parameters are shown in Table 1:

[0102] Table 1 System parameters

[0103]

[0104] Inductor Decoupling Simulation

[0105] Firstly, the traditional inductor decoupling control and the improved inductorless decoupling control are simulated. By comparing the current after decoupling with the reference current before decoupling, the influence of inductor saturation on decoupling is analyzed.

[0106] like Fig.16(a) shows the output waveform of the improved inductorless decoupling control. In order to facilitate the comparative analysis of the waveforms before and after decoupling, the amplitude of the output waveform after decoupling is multiplied by a constant at 0.04s to distinguish the waveforms before and after decoupling. Taking the q-axis coordinate current as an example, through comparative analysis, it can be seen that under the inductorless decoupling control, the frequency and phase of the current waveform iq after decoupling are the same as the reference current iq* before decoupling, and it is not affected by the saturation of the inductor coil.

[0107] Fig.16 (b) is the output waveform with inductor decoupling. It can be seen that after about 0.06s, the inductor reaches saturation, and the current waveform iq after decoupling deviates from the reference current iq* before decoupling. The inductor tends to saturation due to excessive current, causing a delay in decoupling and failure to fully decouple, resulting in a phase deviation.

[0108] Fig.16 (c) is the comparison of the decoupling errors of the inductor coils. The error waveform of the non-inductor decoupling is relatively stable and is always near the zero point. It can be seen that the error of the non-inductor decoupling control strategy is small and approaches zero, achieving complete decoupling. Due to the saturation of the inductor coil, the error of the inductor decoupling control strategy gradually increases, and the error waveform gradually moves away from the zero point. The decoupling fails, which will affect the subsequent PWM control signal and cause control imbalance.

[0109] In summary, it can be seen that when the current is too large, the inductor will be saturated, resulting in insufficient decoupling. Compared with traditional inductor decoupling, the improved inductorless decoupling control does not contain the inductor term, and the use of the integral module avoids the influence of inductor saturation, which can effectively improve the control accuracy and control precision.

[0110] Electric vehicle unidirectional charging and discharging simulation

[0111] In order to verify the feasibility of the improved inductorless control strategy, the electric vehicle unidirectional charging and discharging and bidirectional charging and discharging are simulated and verified. The rated voltage of the electric vehicle battery is set to 150V, the charging and discharging current is 10A, and the charging and discharging time is 20s to achieve the constant current charging and discharging process. The battery charging and discharging current is as follows: Fig.17 As shown in (a), the current value is negative when the battery is charged, and positive otherwise. During the charging process, the bidirectional power converter first performs rectification. From the simulation results, it can be seen that after a 1s overshoot, the battery is charged with a current of 10A; during the discharging process, the battery is discharged directly, there is no overshoot at the battery end, and it is discharged with a constant current of 10A.

[0112] When the battery is charging or discharging, the voltage across the battery is Fig.17As shown in (b), in the charging state, after 20s, the battery voltage rises from 149V to 154V and gradually reaches a fixed voltage; in the discharging state, the load current produces a certain voltage drop on the battery internal resistance, and the battery voltage drops from 149V to 146V and gradually tends to be stable.

[0113] The initial SOC of the battery is set to 50%. The battery state of charge (SOC) changes during charging and discharging as follows: Fig.17 As shown in (c), after 20s of charging process, the battery SOC increases from 50% to 50.4%, and the charge increases linearly with the charging time; during the discharge process, the battery state of charge decreases from 50% to 49.6%, and the charge decreases linearly, completing the entire discharge process.

[0114] Fig.17 (d) is the DC bus voltage during unidirectional charging and discharging. In the charging state, after 1.5s, the DC bus voltage reaches 300V and the bus voltage stabilizes at 300V. During the discharge process, since the battery voltage directly reaches the DC bus voltage of 300V through the boost chopper, the overshoot time is significantly shorter than the charging process.

[0115] From the above analysis, it can be seen that in the unidirectional charging and discharging mode, this control strategy can realize the energy exchange between electric vehicles and the power grid, control the battery charging and discharging current, ensure that the DC bus voltage is in a stable state, maintain the power balance between the AC side and the DC side, and improve the transmission efficiency.

[0116] Electric vehicle bidirectional charging and discharging simulation

[0117] Two modes are used in the bidirectional charging and discharging simulation of electric vehicles: Mode 1 is the bidirectional constant current mode; Mode 2 is the bidirectional variable current mode. The effectiveness of the inductorless decoupling control strategy is verified by using the two bidirectional charging and discharging modes. Fig.18 (a) and (b) show a bidirectional constant current charge and discharge mode. During the time of 0-7s, the bidirectional power converter charges with a current of 10A, and the voltage across the battery rises from 150V to 152V, completing the charging process. During the time of 7-13s, the bidirectional power converter does not perform charging and discharging, and the charging and discharging current is 0. Due to the internal resistance of the battery, when charging stops, the voltage across the battery begins to drop gradually. During the time of 13-20s, the bidirectional power converter works in the discharge state, and the battery discharges with a current of 10A and feeds back to the power grid, and the voltage across the battery begins to drop from 151V.

[0118] Fig.18(c) and (d) are bidirectional current conversion charging and discharging modes. In 0-3s, the bidirectional power converter charges the battery with a current of 10A, and the voltage across the battery increases from 150V to 151V. Then the converter charging current increases to 20A. After 4s, the voltage across the battery increases to 152V, with a faster growth rate and a larger slope. In 7-13s, the charging current is 0, and no energy flows in the bidirectional power converter. The internal resistance of the battery causes the voltage across the battery to gradually decrease. In 13-15s, the battery discharges with a current of 10A, and the energy at the battery end is fed to the grid by the converter. In 15-20s, the battery discharge current increases to 20A, and the discharge continues for 5s, and the voltage across the battery drops to 150V.

[0119] Fig.18 (e) is the change of battery SOC during bidirectional charging and discharging. It can be seen from the figure that the battery charge growth rate is different due to different current sizes during charging and discharging. During charging, the charge increases, and during discharging, the charge decreases. Due to the short simulation time, the charge change is small.

[0120] Fig.18 (f) is the DC bus voltage during the bidirectional charging and discharging process. During the charging and discharging process, the DC bus voltage is always maintained at 300V, the voltage ripple is controlled within 2%, and the error is 2V.

[0121] From the above analysis, it can be concluded that under this control method, the power converter can realize constant current and variable current charging and discharging. By adjusting the charging current, the fast charging and slow charging modes can be switched. Under the condition of stabilizing the DC bus voltage, the battery can be deeply charged and discharged, thereby extending the battery life.

[0122] Through the simulation of single-phase phase-locked loop and converter charging and discharging, it can be seen that the electric vehicle bidirectional power converter can realize bidirectional energy flow under the cooperation of improved phase-locked loop strategy and inductorless decoupling control strategy, improve the anti-interference ability of the converter through the phase-locked loop, realize stable operation under complex power grid conditions, and increase the application space of the converter.

[0123] The preferred embodiments of the present invention are described in detail above in conjunction with the accompanying drawings, but the present invention is not limited to the above embodiments, and various changes can be made within the knowledge scope of ordinary technicians in this field without departing from the purpose of the present invention.

[0124] Many other changes and modifications may be made without departing from the concept and scope of the present invention.It should be understood that the present invention is not limited to the specific embodiments, and the scope of the present invention is defined by the appended claims.

Claims

1. A vehicle-mounted bidirectional power converter, characterized in that: It includes a front-stage voltage stabilization module, a front-stage drive circuit, a rear-stage drive circuit, a rear-stage filter module, and an electric vehicle battery module. The front-stage voltage stabilization module is used to balance the circuit voltage and support reactive power. The front-stage drive circuit is a single-phase full-bridge AD-DC topology structure, and the rear-stage drive circuit is a half-bridge bidirectional DC-DC topology structure. The front-stage drive circuit and the rear-stage drive circuit are connected together through a DC voltage stabilization bus. The rear-stage filter module is used to suppress the second harmonic on the DC side to prevent the harmonic amount from affecting the stability of the output DC voltage and causing low-frequency voltage pulsation on the DC side.

2. The vehicle-mounted bidirectional power converter according to claim 1, characterized in that: The front-stage driving circuit includes power tubes IGBT1, IGBT2, IGBT3, IGBT4, inductors L1, L2, resistors R1, R2, and capacitor C1. The IGBT1 and IGBT4 form a pair of bridge arms, and the IGBT2 and IGBT3 form another pair of bridge arms. The triangular wave pulse width modulation method is used to control the opening and closing of each pair of bridge arms to achieve forward rectification and reverse inversion.

3. The vehicle-mounted bidirectional power converter according to claim 2, characterized in that: The triangular wave pulse width modulation method adopts unipolar triangular wave pulse width modulation. In half a cycle, only two switch tubes are in high-frequency switching working state, and the other two are in normally closed or normally open state. The modulation voltage is from U d Change to 0, and then change from 0 to -U d .

4. The vehicle-mounted bidirectional power converter according to claim 3, characterized in that: The unipolar triangular wave pulse width modulation adopts an isosceles triangular wave as a carrier wave and a sine wave as a modulation signal wave. By comparing the two waveforms, in the positive half cycle, when the amplitude of the sine wave modulation signal is greater than the amplitude of the triangular wave carrier wave, the pulse signal outputs a high level, and when the amplitude of the sine wave modulation signal is less than the amplitude of the triangular wave carrier wave, the pulse signal outputs a low level; in the negative half cycle, when the amplitude of the triangular wave carrier wave is greater than the amplitude of the sine wave modulation signal, the pulse signal outputs a high level, and when the amplitude of the triangular wave carrier wave is less than the amplitude of the sine wave modulation signal, the pulse signal outputs a low level, and a wide and narrow pulse with the same impulse as the sine wave but different shape is obtained, which is a triangular wave pulse width modulation waveform.

5. The vehicle-mounted bidirectional power converter according to claim 2, characterized in that: The front-stage driving circuit obtains three AC side voltages by controlling four switch tubes IGBT respectively. The specific steps are as follows: Step 1: Simplify the rectifier circuit and obtain its mathematical model, where E s is the electromotive force on the AC side, i s is the AC side current, L is the grid side inductance, R is the AC side resistance, u AB is the AC side voltage, i dc is the DC side current, u dc is the DC side voltage, R L is the load resistance, E L is the load electromotive force; Step 2: Define the switch function as: Step 3: Control the on and off of the switch tube IGBT. The switch tubes of the same bridge arm cannot be turned on at the same time, otherwise it will cause a short circuit. Therefore, the on state of the IGBT switch tube is: Step 4: Get the function expression of three voltages and switch tubes: in AB =(S A -WITH B )in dc 。 6. The vehicle-mounted bidirectional power converter according to claim 1, characterized in that: The post-stage drive circuit includes power tubes IGBT5 and IGBT6, inductors L3 and L4, and capacitor C2. The post-stage drive circuit works in Buck and Boost states, and realizes bidirectional energy flow by controlling the alternating conduction of the switch tubes. When the switch tube IGBT5 is turned on, the switch tube IGBT6 is in the off state and is in the Buck working mode; when the switch tube IGBT6 is turned on, IGBT5 is in the off state and is in the Boost working mode. The waveforms of the power tubes IGBT5 and IGBT6 are complementary, and a dead time is set to prevent short circuit.

7. The vehicle-mounted bidirectional power converter according to claim 6, characterized in that: There are two loops in the Buck working mode: In the first state of the circuit, the pulse width modulation drive signal turns on the upper bridge arm switch tube IGBT5 and turns off the lower bridge arm switch tube IGBT6. The rectified current on the AC side is supplied to the inductors L3, L4 and the battery V through the switch tube IGBT5. bat To charge, the inductor stores energy in the form of a magnetic field; In the second loop state: the lower bridge arm switch tube IGBT6 is turned on, the upper bridge arm IGBT5 is turned off, the current flows through the synchronous diode of IGBT6, the inductor L3, L4, the synchronous diode of IGBT6 and the battery V bat A circuit is formed, the inductor provides energy to the battery, the energy stored in the inductor decreases, and the charging process is completed.

8. The vehicle-mounted bidirectional power converter according to claim 6, characterized in that: There are two loops in the Boost working mode: In the third state of the circuit: the battery output voltage is increased to the DC bus voltage through the boost chopper, and then fed back to the AC side through the inverter, the driving signal turns on the lower bridge arm IGBT6 and turns off the upper bridge arm IGBT5, and the battery V bat Provide energy to inductors L3 and L4, which store magnetic field energy; In the fourth state of the circuit: the upper bridge arm IGBT5 is turned on, the lower bridge arm IGBT6 is turned off, the current is transmitted through the synchronous diode of IGBT5, and the battery V bat Together with inductors L3, L4 and the synchronous diode of IGBT5, the battery V bat And inductors L3 and L4 simultaneously supply power to the DC bus V dc The energy is fed back, the energy stored in the inductor decreases, and the discharge process begins.

9. The vehicle-mounted bidirectional power converter according to claim 1, characterized in that: The bidirectional power converter adopts an inductorless decoupling control method based on SOGI. This strategy realizes orthogonal signal construction through a second-order generalized integrator, adopts a continuous integration inductorless decoupling method, and uses a single-phase phase-locked loop with an improved loop filter for grid-connected control.

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